Compensation band-gap reference circuit for sigma-delta ADC circuit

Abstract

A compensation band-gap reference circuit used for a sigma-delta ADC circuit comprises a self-starting circuit, a positive temperature coefficient current circuit used for generating current positively correlated with absolute temperature, a negative temperature coefficient current circuit used for generating current negatively correlated with the absolute temperature, a compensation circuit used for generating compensation voltage and a reference voltage generating circuit. The reference voltage generation circuit generates positive temperature coefficient voltage from positive temperature coefficient current through a resistor, generates negative temperature coefficient voltage from negative temperature coefficient current through the resistor, and then superposes the generated positive temperature coefficient voltage and negative temperature coefficient voltage to generate first-order temperature compensation band-gap reference voltage with convex curvature; and then a VCC voltage is generated by a compensation circuit, and the VCC voltage is subtracted from the first-order temperature compensation band-gap reference voltage with the convex curvature to obtain a high-precision band-gap reference voltage. According to the invention, the influence of the band-gap reference voltage source on the sigma-delta ADC circuit is reduced, so that the output precision of the sigma-delta ADC circuit is improved.

Description

Technical Field

[0001] The present invention relates to the technical field of integrated circuits, and in particular to a compensation bandgap reference circuit for a Σ-ΔADC circuit. Background Art

[0002] The bandgap reference circuit is a basic module in the field of analog integrated circuit design. Its function is to provide a reference voltage for the overall circuit that varies very little with external factors. The voltage should have good temperature stability and a high power supply rejection ratio, that is, it should vary little with temperature and power supply voltage.

[0003] In the Σ-ΔADC circuit, the performance of the bandgap reference circuit directly determines the output accuracy of the Σ-ΔADC circuit. The output accuracy of the bandgap reference circuit depends largely on the temperature coefficient. The traditional bandgap reference is formed by superimposing voltages with negative temperature coefficients and positive temperature coefficients in a suitable ratio. The negative temperature coefficient voltage is generated by the emitter-base voltage VEB of the transistor, and the positive temperature coefficient voltage is generated by the difference of VEB of multiple transistors working at different current densities. At this time, the generated voltage is a first-order temperature-compensated bandgap reference source. Due to the inherent nonlinear component of the transistor itself, the output accuracy of the bandgap quasi-source is greatly limited. Under normal circumstances, the temperature coefficient of the first-order temperature-compensated bandgap reference source is about 10ppm / ℃ to 30ppm / ℃. The use of such a large temperature coefficient bandgap reference source in the Σ-ΔADC circuit will not only cause the change of the quantization step size of the Σ-ΔADC and introduce too much linear error, but also cause the integrator output in the Σ-ΔADC circuit to drift, resulting in inaccurate signal processing; the above-mentioned influence will reduce the output accuracy of the Σ-ΔADC. Summary of the invention

[0004] In view of the technical problem that the output accuracy of the traditional bandgap reference circuit is reduced when used in the Σ-ΔADC circuit due to its large temperature coefficient, the present technical solution provides a compensated bandgap reference circuit for the Σ-ΔADC circuit, which effectively reduces the temperature coefficient of the bandgap reference circuit and the influence of the bandgap reference voltage source on the Σ-ΔADC circuit while slightly increasing the power consumption, thereby improving the output accuracy of the Σ-ΔADC circuit; it can effectively solve the above-mentioned problem.

[0005] The present invention is achieved through the following technical solutions:

[0006] A compensated bandgap reference circuit for a Σ-ΔADC circuit comprises: a self-starting circuit, a positive temperature coefficient current circuit, a negative temperature coefficient current circuit and a reference voltage generating circuit; the compensated bandgap reference circuit also comprises a compensation circuit, and the compensation circuit is respectively connected to the positive temperature coefficient current circuit, the negative temperature coefficient current circuit and the reference voltage generating circuit; the positive temperature coefficient current circuit generates a current that is positively correlated with the absolute temperature, the negative temperature coefficient current circuit generates a current that is negatively correlated with the absolute temperature, the compensation circuit generates a compensation voltage, and the reference voltage generating circuit generates a positive temperature coefficient voltage by passing the positive temperature coefficient current through a resistor, and generates a negative temperature coefficient voltage by passing the negative temperature coefficient current through a resistor, and then superimposes the generated positive temperature coefficient voltage and negative temperature coefficient voltage on each other to generate a first-order temperature compensated bandgap reference voltage with a convex curvature, and then generates a VCC voltage by using the compensation circuit, and subtracts the VCC voltage from the first-order temperature compensated bandgap reference voltage with a convex curvature to obtain a final bandgap reference voltage.

[0007] Furthermore, the positive temperature coefficient current circuit includes an operational amplifier OPA1, four MOS tubes MP6, MP7, MP8, MP9, a resistor R1, and two PNP transistors Q1, Q2; the gate of the MOS tube MP6 is connected to the gates of the MOS tubes MP10 and MP12 in the self-starting circuit, the operational amplifier OPA1, and the compensation circuit.

[0008] Furthermore, the negative temperature coefficient current circuit includes an operational amplifier OPA2, a MOS tube MP14, and a resistor R2; the gate of the MOS tube MP14 is connected to the operational amplifier OPA2, the gate of the MOS tube MP13 in the reference voltage generating circuit, and the gate of the MOS tube MP11 in the compensation circuit.

[0009] Further, the reference voltage generating circuit includes two MOS tubes MP12 and MP13 and two resistors R3 and R4; the compensation circuit includes seven MOS tubes MN5, MN6, MN7, MN8, MN9, MP10 and MP11;

[0010] The MN5 and MP10 are in the same branch, the gate of MN5 is connected to its drain, the gate of MN5 is connected to the gates of MN7 and MN9, the drain of MN5 is connected to the drain of MP10, and the source of MN5 is connected to the ground;

[0011] The gate of MP10 is connected to the gate of MP12, and the source of MP10 is connected to the power supply VDD;

[0012] The MP11, MN6 and MN7 are in the same branch, the gate of MN6 is connected to its drain, the gate of MN6 is connected to the gate of MN8, the drain of MN6 is connected to the drain of MP11, the source of MN6 is connected to the drain of MN7, and the source of MN7 is connected to the ground;

[0013] The source of MP11 is connected to the power supply VDD;

[0014] The MN8 and MN9 are in the same branch, the source of MN9 is connected to the ground, and the drain of MN9 is connected to the source of MN8;

[0015] The drain of the MN8 is connected to the resistors R3 and R4.

[0016] Furthermore, the self-starting circuit includes 7 MOS tubes MP1, MP2, MP3, MP4, MN1, MN2, and MN3; at the moment the circuit is powered on, the MP1, MP2, and MP3 tubes are turned on, and a small current is injected to increase the gate voltage of the MN2 tube, so that the MN2 tube is turned on, and the drain of MN2 is also powered on. Through the negative feedback of the operational amplifier OPA1, the core circuit of the entire bandgap reference starts to work; after the circuit works normally, the current mirrors MN1 and MN3 are turned on, the gate voltage of MN2 is pulled down, and the MN2 tube is turned off.

[0017] Furthermore, when the temperature of the compensation circuit is low, the negative temperature coefficient current I IPTAT Greater than the positive temperature coefficient current I PTAT , then the mirror current a×I obtained by MP11 tube IPTAT Greater than the mirror current b×I obtained by MP10 tube PTAT At this time, the gate-source voltage of MP11 tube |V GSP11 |Very big,|V GSP11 |-|V THP11 | is greater than its drain-source voltage |V DSP11 |, the MP11 tube works in the linear region, which is equivalent to a linear resistor, and its expression is shown in Formula 4:

[0018]

[0019] where u p represents the hole mobility, c ox Represents the gate oxide capacitance per unit area, W P11 / L P11 Represents the width-to-length ratio of the MP11 tube, V THP11 Represents the threshold voltage of the MP11 tube.

[0020] Then the MN7 tube obtains the mirror current b×I through the current mirror copy PTATSince MP11 works in the linear region and MN6 is connected by diode, the main current flowing through MP11, MN6 and MN7 is I N7 , which is expressed as shown in Formula 5:

[0021]

[0022] where u n represents the electron mobility, c ox Represents the gate oxide capacitance per unit area, W N7 / L N7 Represents the width-to-length ratio of the MN7 tube, V THN7 Represents the threshold voltage of the MN7 tube.

[0023] Set the proportional coefficient c of MN8 tube so that it works in the linear region at this time. MN9 tube obtains the mirror current b×d×I through current mirror replication. PTAT Since MN8 operates in the linear region, the main current flowing through MN8 and MN9 is I N9 , at this time the compensation current I X1 =I N9 , which is expressed as shown in Formula 6:

[0024] I X1 =b×d×I PTAT (Formula 6)

[0025] The compensation current generates a voltage V that is positively correlated with the absolute temperature through resistor R4. PT2 , which is expressed as shown in Formula 7:

[0026] V PT2 =I X1 ×R4 (Formula 7)

[0027] As the temperature increases, the positive temperature coefficient current I PTAT Slowly increases, negative temperature coefficient current I IPTAT Slowly decreases, when the temperature reaches a certain value, the positive temperature coefficient current I PTAT Greater than the negative temperature coefficient current I IPTAT , from formula 4, we can know that the positive temperature coefficient current I PTAT The increase in current I N7 The gate-source voltage V of MN7 tube increases without changing the width-to-length ratio of MN7 tube. GSN7 becomes larger, making V GSN7 -V THN7 Greater than its drain-source voltage V DSN7 , its working area changes from the saturation area to the linear area, which is equivalent to a linear resistor, and its expression is shown in Formula 8:

[0028]

[0029] With I IPTAT The gate potential of MP14 tube increases. Since the gates of MP11 tube and MP14 tube are connected, the gate potential of MP11 tube also increases, making the gate-source voltage of MP11 tube |V GSP11 | is decreasing, |V GSP11 |-|V THP11 | is less than its drain-source voltage |V DSP11 |, so that it changes from the linear region to the saturation region; at this time, the main current flowing through MP11, MN6, and MN7 tubes is I P11 , its expression is shown in Formula 9.

[0030] I P11 =a×I IPTAT (Formula 9)

[0031] Set the proportional coefficient d of the MN9 tube so that it works in the linear region at this time. The MN8 tube obtains the mirror current c×I through the current mirror copy. P11 Since the MN9 tube works in the linear region, the main current flowing through the MN8 and MN9 tubes is I N8 , at this time the compensation current I X2 =I N8 , which is expressed as shown in formula 10:

[0032] I X2 =c×I P11 =c×a×I IPTAT (Formula 10)

[0033] The compensation current generates a voltage V that is negatively correlated with the absolute temperature through resistor R4. NT2 , which is expressed as shown in formula 11:

[0034] V NT2 =I X2 ×R4(Formula 11)

[0035] The voltage V generated at this time is positively correlated with the absolute temperature through the compensation circuit and the reference voltage generation circuit. PT2 and the voltage V which is negatively correlated with the absolute temperature NT2 The compensation voltage V CC , using a first-order compensated bandgap reference voltage V REF1 Subtract the compensation voltage V generated by the compensation circuit CC The compensated bandgap reference voltage V REF2 , V REF2 The expression is shown in formula 12:

[0036] V REF2 =VREF1 -(V PT2 +V NT2 )=V REF1 -V CC (Formula 12)

[0037] The expression of the temperature coefficient formula is shown in Formula 13:

[0038]

[0039] According to the temperature coefficient formula, V MAX V REF2 Maximum voltage, V MIN V REF2 The minimum voltage, using the first-order temperature compensation voltage V REF1 Subtract the compensation voltage V CC , we can find that the compensated circuit V MAX and V MIN becomes closer, that is, V MAX -V MIN Becomes smaller, thereby reducing the temperature coefficient and achieving a compensation effect.

[0040] Beneficial Effects

[0041] The present invention proposes a compensation bandgap reference circuit for a Σ-Δ ADC circuit, which has the following beneficial effects compared with the prior art:

[0042] (1) This technical solution adds a compensation circuit, in terms of temperature coefficient, at low temperatures, the negative temperature coefficient current I IPTAT Greater than the positive temperature coefficient current I PTAT , causing the related tube to work in the linear region, and its value is equivalent to a linear resistor. The compensation current I generated by the current mirror X1 Used to generate a positive temperature coefficient voltage V PT2 At high temperatures, the positive temperature coefficient current I PTAT Greater than the negative temperature coefficient current I IPTAT , the relevant tube switches from the saturation region to the linear region, and at the same time, the compensation current I X2 Generates a negative temperature coefficient voltage V NT2 The compensation circuit converts the positive temperature coefficient voltage V PT2 and negative temperature coefficient voltage V PT2 The compensation voltage V is generated by superposition CC , and through the first-order compensation reference voltage V REF1 Subtract the compensation voltage V CC The compensated voltage V REF2 . V REF2 V MAX and V MINGetting closer, through the simulation results and the temperature coefficient formula, it can be obtained that the temperature coefficient is reduced. The reduction of the temperature coefficient can reduce the influence of the Σ-Δ ADC circuit on temperature and improve its working accuracy.

[0043] (2) Compared with the existing second-order temperature-compensated bandgap reference voltage circuit, the compensation circuit in the present technical solution is composed of only 7 MOS tubes. Regardless of whether it is working at low temperature or high temperature, there will always be a part of the MOS tubes that will work in the linear region, so that the power consumption will be reduced to a certain extent. Therefore, the power consumption has certain advantages compared with the existing second-order temperature-compensated bandgap reference voltage circuit. BRIEF DESCRIPTION OF THE DRAWINGS

[0044] Figure 1 It is a schematic diagram of the structure of the compensated bandgap reference circuit of the present invention.

[0045] Figure 2 Generates a schematic for a first-order temperature compensated bandgap reference voltage.

[0046] Figure 3 This is a schematic diagram of the bandgap reference voltage generated by the compensation circuit of the present invention.

[0047] Figure 4 This is a first-order temperature compensated bandgap reference voltage curve.

[0048] Figure 5 This is a bandgap reference voltage curve generated by the compensation circuit of the present invention. DETAILED DESCRIPTION

[0049] The following will be combined with the accompanying drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. The described embodiments are only part of the embodiments of the present invention, rather than all the embodiments. Under the premise of not departing from the design concept of the present invention, various modifications and improvements made by ordinary persons in the art to the technical solutions of the present invention should all fall within the protection scope of the present invention.

[0050] Embodiment 1:

[0051] like Figure 1 As shown, a compensated bandgap reference circuit for a Σ-Δ ADC circuit comprises: a self-starting circuit, a positive temperature coefficient current circuit, a negative temperature coefficient current circuit, a compensation circuit and a reference voltage generating circuit; the compensation circuit is respectively connected to the positive temperature coefficient current circuit, the negative temperature coefficient current circuit and the reference voltage generating circuit.

[0052] The conventional bandgap reference circuit has a large temperature coefficient, so its output accuracy is reduced when used in a Σ-ΔADC circuit. The present invention aims to provide a compensated bandgap reference circuit for a Σ-ΔADC circuit, which can effectively reduce the temperature coefficient of the bandgap reference circuit while slightly increasing power consumption, thereby improving the output accuracy of the Σ-ΔADC circuit.

[0053] The self-starting circuit eliminates the degeneracy point of the circuit, that is, prevents the circuit from entering an undesirable steady state, such as a zero current steady state, when powered on, so that the circuit enters a correct working state. The self-starting circuit includes MOS tubes MP1, MP2, MP3, MP4, MN1, MN2, and MN3.

[0054] The positive temperature coefficient current circuit generates a current that is positively correlated with the absolute temperature; the positive temperature coefficient current circuit includes an operational amplifier OPA1, four MOS tubes MP6, MP7, MP8, MP9, a resistor R1, and two PNP transistors Q1, Q2; the gate of the MOS tube MP6 is connected to the gates of the MOS tubes MP10 and MP12 in the self-starting circuit, the operational amplifier OPA1, and the compensation circuit.

[0055] The negative temperature coefficient current circuit generates a current that is negatively correlated with the absolute temperature; the negative temperature coefficient current circuit includes an operational amplifier OPA2, a MOS tube MP14, and a resistor R2; the gate of the MOS tube MP14 is connected to the operational amplifier OPA2, the gate of the MOS tube MP13 in the reference voltage generating circuit, and the gate of the MOS tube MP11 in the compensation circuit.

[0056] The compensation circuit generates a compensation voltage, which effectively reduces the temperature coefficient of the bandgap reference source; the compensation circuit includes 7 MOS tubes MN5, MN6, MN7, MN8, MN9, MP10, and MP11; MN5 and MP10 are in the same branch, the gate of MN5 is connected to its drain, and the gate of MN5 is connected to the gates of MN7 and MN9, the drain of MN5 is connected to the drain of MP10, and the source of MN5 is connected to the ground; the gate of MP10 is connected to the gate of MP12, and the source of MP10 is connected to the power supply. VDD is connected; the MP11, MN6 and MN7 are in the same branch, the gate of MN6 is connected to its drain, and the gate of MN6 is connected to the gate of MN8, the drain of MN6 is connected to the drain of MP11, the source of MN6 is connected to the drain of MN7, and the source of MN7 is connected to the ground; the source of MP11 is connected to the power supply VDD; the MN8 and MN9 are in the same branch, the source of MN9 is connected to the ground, and the drain of MN9 is connected to the source of MN8; the drain of MN8 is connected to resistors R3 and R4.

[0057] The reference voltage generating circuit comprises two MOS tubes MP12 and MP13 and two resistors R3 and R4. The reference voltage generating circuit generates a positive temperature coefficient voltage by passing the positive temperature coefficient current through the resistor, and generates a negative temperature coefficient voltage by passing the negative temperature coefficient current through the resistor, and then superimposes the generated positive temperature coefficient voltage and negative temperature coefficient voltage to generate a first-order temperature compensated bandgap reference voltage with a convex curvature, and then generates a VCC voltage by using a compensation circuit, and subtracts the VCC voltage from the first-order temperature compensated bandgap reference voltage with a convex curvature to obtain a final bandgap reference voltage.

[0058] At the moment the circuit is powered on, MP1, MP2, and MP3 tubes are turned on, injecting a small current to increase the gate voltage of MN2 tube, causing MN2 tube to be turned on, and the drain of MN2 is also powered on. Through the negative feedback of op amp OPA1, the core circuit of the entire bandgap reference starts to work; after the circuit works normally, current mirrors MN1 and MN3 are turned on, lowering the gate voltage of MN2 and turning off MN2 tube to prevent it from affecting the circuit operation.

[0059] Due to the negative feedback of the operational amplifier OPA1, the potentials of points A and B in the circuit are equal. Considering the central symmetry to improve the matching degree, the area of ​​PNP transistors Q1 and Q2 is 1:8, and the emitter-base voltage of Q1 is V EB1 , Q2 emitter-base voltage is V EB2 , the voltage difference between them △V EB It is a voltage that is positively correlated with the absolute temperature, and generates a current I that is positively correlated with the absolute temperature through the resistor R1. PTAT , whose expression is shown in Formula 1.

[0060]

[0061] Then, the MP12 tube obtains the mirror current B×I by replicating it through the common source and common gate current mirror. PTAT .

[0062] Due to the negative feedback of the operational amplifier OPA2, the potentials of points B and C in the circuit are equal, and the emitter-base voltage of Q1 is V EB1 It is a voltage that is negatively correlated with the absolute temperature, and generates a current I that is negatively correlated with the absolute temperature through the resistor R2. IPTAT , whose expression is shown in Formula 2.

[0063]

[0064] Through the P tube current mirror, the MP13 tube obtains the mirror current A×I IPTAT .

[0065] The reference voltage generation circuit converts the above current B×IPTAT A positive temperature coefficient voltage V is generated by resistors R3 and R4 PT1 , negatively correlated with temperature current A×I IPTAT A negative temperature coefficient voltage V is generated by resistors R3 and R4 NT1 , the positive temperature coefficient voltage V PT1 and negative temperature coefficient voltage V NT1 Derivative the temperature T, and we can find that when the temperature is low, It will show positive temperature characteristics. When the temperature is high, It will show negative temperature characteristics, so in the whole temperature range, V PT1 and V NT1 Superimposed on each other, a first-order temperature compensated bandgap reference voltage V with a convex curvature is presented. REF1 The schematic diagram of the whole process is as follows Figure 2 As shown, V REF1 The expression of is shown in formula 3.

[0066] V REF1 =V PT1 +V NT1 (Formula 3)

[0067] At this time, the temperature coefficient of the first-order bandgap reference voltage is very large, which is difficult to meet the requirements of the Σ-Δ ADC circuit.

[0068] The compensation circuit proposed in this embodiment can effectively reduce the temperature coefficient of the bandgap reference voltage, and its working principle is as follows:

[0069] When the temperature is low, the negative temperature coefficient current I IPTAT Greater than the positive temperature coefficient current I PTAT , then the mirror current a×I obtained by MP11 tube IPTAT Greater than the mirror current b×I obtained by MP10 tube PTAT At this time, the gate-source voltage of MP11 tube |V GSP11 |Very big,|V GSP11 |-|V THP11 | is greater than its drain-source voltage |V DSP11 |, it works in the linear region, equivalent to a linear resistor, and its expression is shown in Formula 4.

[0070]

[0071] where u p represents the hole mobility, c ox Represents the gate oxide capacitance per unit area, W P11 / L P11 Represents the width-to-length ratio of the MP11 tube, V THP11Represents the threshold voltage of the MP11 tube.

[0072] Then the MN7 tube obtains the mirror current b×I through the current mirror copy PTAT Since MP11 works in the linear region and MN6 is connected by diode, the main current flowing through MP11, MN6 and MN7 is I N7 , whose expression is shown in Formula 5.

[0073]

[0074] where u n represents the electron mobility, c ox Represents the gate oxide capacitance per unit area, W N7 / L N7 Represents the width-to-length ratio of the MN7 tube, V THN7 Represents the threshold voltage of the MN7 tube.

[0075] Set the proportional coefficient c of MN8 tube so that it works in the linear region at this time. MN9 tube obtains the mirror current b×d×I through current mirror replication. PTAT Since MN8 operates in the linear region, the main current flowing through MN8 and MN9 is I N9 , at this time the compensation current I X1 =I N9 , whose expression is shown in Formula 6.

[0076] I X1 =b×d×I PTAT (Formula 6)

[0077] The compensation current generates a voltage V that is positively correlated with the absolute temperature through resistor R4. PT2 , whose expression is shown in Formula 7.

[0078] V PT2 =I X1 ×R4 (Formula 7)

[0079] As the temperature increases, the positive temperature coefficient current I PTAT Slowly increases, negative temperature coefficient current I IPTAT Slowly decreases, when the temperature reaches a certain value, the positive temperature coefficient current I PTAT Greater than the negative temperature coefficient current I IPTAT , from formula 4, we can know that the positive temperature coefficient current I PTAT The increase in current I N7 The gate-source voltage V of MN7 tube increases without changing the width-to-length ratio of MN7 tube. GSN7 becomes larger, making V GSN7 -V THN7 Greater than its drain-source voltage VDSN7 , its working area changes from the saturation area to the linear area, which is equivalent to a linear resistor, and its expression is shown in Formula 8.

[0080]

[0081] With I IPTAT The gate potential of MP14 tube increases. Since the gates of MP11 tube and MP14 tube are connected, the gate potential of MP11 tube also increases, making the gate-source voltage of MP11 tube |V GSP11 | is decreasing, |V GSP11 |-|V THP11 | is less than its drain-source voltage |V DSP11 |, so that it changes from the linear region to the saturation region. At this time, the main current flowing through MP11, MN6, and MN7 is I P11 , its expression is shown in Formula 9.

[0082] I P11 =a×I IPTAT (Formula 9)

[0083] The proportional coefficient d of the MN9 tube is set so that it works in the linear region at this time. The MN8 tube obtains the mirror current c×IP11 through the current mirror replication. Since the MN9 tube works in the linear region, the main current flowing through the MN8 and MN9 tubes is IN8. At this time, the compensation current IX2=IN8, and its expression is shown in Formula 10.

[0084] I X2 =c×I P11 =c×a×I IPTAT (Formula 10)

[0085] The compensation current generates a voltage VNT2 which is negatively correlated with the absolute temperature through the resistor R4, and its expression is shown in Formula 11.

[0086] V NT2 =I X2 ×R4(Formula 11)

[0087] By means of the compensation circuit and the reference voltage generating circuit, the voltage VPT2 positively correlated with the absolute temperature and the voltage VNT2 negatively correlated with the absolute temperature are superimposed on each other to generate the compensation voltage VCC. The compensation voltage VCC generated by the compensation circuit is subtracted from the first-order compensated bandgap reference voltage VREF1 to obtain the compensated bandgap reference voltage VREF2. The compensation circuit of this embodiment has a compensation principle as shown below: Figure 3 As shown, the expression of VREF2 is shown in formula 12.

[0088] V REF2 =V REF1 -(VPT2 +V NT2 )=V REF1 -V CC (Formula 12)

[0089] The expression of the temperature coefficient formula is shown in Formula 13.

[0090]

[0091] According to the temperature coefficient formula, VMAX represents the maximum voltage of VREF2, and VMIN represents the minimum voltage of VREF2. By subtracting the compensation voltage VCC from the first-order temperature compensation voltage VREF1, we can find that the VMAX and VMIN of the compensated circuit become closer, that is, VMAX-VMIN becomes smaller, thereby reducing the temperature coefficient and achieving the compensation effect.

[0092] According to the experimental results, the simulation results of the first-order temperature compensated bandgap reference voltage are as follows Figure 4 As shown, its reference voltage V MAX is 900.92mV, V MIN The output fluctuation is about 1.51mV, and the temperature coefficient is about 9.32ppm / ℃. The simulation results of the compensation circuit provided by the present invention to generate the bandgap reference voltage are as follows: Figure 5 As shown, its reference voltage V MAX is 899.28mV, V MIN It is 898.86mV, the output fluctuation is about 0.42mV, and the temperature coefficient is about 2.53ppm / ℃, which is about 3.6 times lower than the temperature coefficient of the first-order temperature compensated bandgap reference voltage. The experimental results are shown in Table 1.

[0093]

[0094] In summary, according to the table data and simulation results, the compensation circuit proposed in this design can effectively reduce the temperature coefficient and can be applied in the Σ-Δ ADC circuit.

Claims

1. A compensated bandgap reference circuit for a Σ-Δ ADC circuit, comprising: A self-starting circuit, a positive temperature coefficient current circuit, a negative temperature coefficient current circuit and a reference voltage generating circuit; characterized in that: the compensated bandgap reference circuit also includes a compensation circuit, and the compensation circuit is respectively connected to the positive temperature coefficient current circuit, the negative temperature coefficient current circuit and the reference voltage generating circuit; the positive temperature coefficient current circuit generates a current that is positively correlated with the absolute temperature, the negative temperature coefficient current circuit generates a current that is negatively correlated with the absolute temperature, the compensation circuit generates a compensation voltage, and the reference voltage generating circuit generates a positive temperature coefficient voltage by passing the positive temperature coefficient current through a resistor, and generates a negative temperature coefficient voltage by passing the negative temperature coefficient current through a resistor, and then superimposes the generated positive temperature coefficient voltage and negative temperature coefficient voltage on each other to generate a first-order temperature compensated bandgap reference voltage with a convex curvature, and then uses the compensation circuit to generate a VCC voltage, and subtracts the VCC voltage from the first-order temperature compensated bandgap reference voltage with a convex curvature to obtain a final bandgap reference voltage.

2. The compensated bandgap reference circuit for a Σ-Δ ADC circuit according to claim 1, characterized in that: The positive temperature coefficient current circuit includes an operational amplifier OPA1, four MOS tubes MP6, MP7, MP8, MP9, a resistor R1, and two PNP transistors Q1, Q2; the gate of the MOS tube MP6 is connected to the gates of the self-starting circuit, the operational amplifier OPA1 and the MOS tubes MP10 and MP12 in the compensation circuit.

3. The compensated bandgap reference circuit for a Σ-Δ ADC circuit according to claim 2, characterized in that: The negative temperature coefficient current circuit includes an operational amplifier OPA2, a MOS tube MP14, and a resistor R2; the gate of the MOS tube MP14 is connected to the operational amplifier OPA2, the gate of the MOS tube MP13 in the reference voltage generation circuit, and the gate of the MOS tube MP11 in the compensation circuit.

4. The compensated bandgap reference circuit for a Σ-Δ ADC circuit according to claim 3, characterized in that: The reference voltage generating circuit includes two MOS tubes MP12 and MP13 and two resistors R3 and R4; the compensation circuit includes seven MOS tubes MN5, MN6, MN7, MN8, MN9, MP10 and MP11; The MN5 and MP10 are in the same branch, the gate of MN5 is connected to its drain, the gate of MN5 is connected to the gates of MN7 and MN9, the drain of MN5 is connected to the drain of MP10, and the source of MN5 is connected to the ground; The gate of MP10 is connected to the gate of MP12, and the source of MP10 is connected to the power supply VDD; The MP11, MN6 and MN7 are in the same branch, the gate of MN6 is connected to its drain, the gate of MN6 is connected to the gate of MN8, the drain of MN6 is connected to the drain of MP11, the source of MN6 is connected to the drain of MN7, and the source of MN7 is connected to the ground; The source of MP11 is connected to the power supply VDD; The MN8 and MN9 are in the same branch, the source of MN9 is connected to the ground, and the drain of MN9 is connected to the source of MN8; The drain of the MN8 is connected to the resistors R3 and R4.

5. The compensated bandgap reference circuit for a Σ-Δ ADC circuit according to claim 1, characterized in that: The self-starting circuit includes 7 MOS tubes MP1, MP2, MP3, MP4, MN1, MN2, and MN3; at the moment the circuit is powered on, the MP1, MP2, and MP3 tubes are turned on, and a small current is injected to increase the gate voltage of the MN2 tube, so that the MN2 tube is turned on, and the drain of MN2 is also powered on. Through the negative feedback of the operational amplifier OPA1, the core circuit of the entire bandgap reference starts to work; after the circuit works normally, the current mirrors MN1 and MN3 are turned on, the gate voltage of MN2 is pulled down, and the MN2 tube is turned off.

6. The compensated bandgap reference circuit for a Σ-Δ ADC circuit according to claim 1, characterized in that: When the temperature of the compensation circuit is low, the negative temperature coefficient current I IPTAT Greater than the positive temperature coefficient current I PTAT , then the mirror current a×I obtained by MP11 tube IPTAT Greater than the mirror current b×I obtained by MP10 tube PTAT At this time, the gate-source voltage of MP11 tube |V GSP11 |Very big,|V GSP11 |-|V THP11 | is greater than its drain-source voltage |V DSP11 |, the MP11 tube works in the linear region, which is equivalent to a linear resistor. Its resistance expression is shown in Formula 4: where u p represents the hole mobility, c ox Represents the gate oxide capacitance per unit area, W P11 / L P11 Represents the width-to-length ratio of the MP11 tube, V THP11 Represents the threshold voltage of MP11 tube; Then the MN7 tube obtains the mirror current b×I through the current mirror replication. PTAT Since MP11 works in the linear region and MN6 is connected by diode, the main current flowing through MP11, MN6 and MN7 is I N7 , which is expressed as shown in Formula 5: where u n represents the electron mobility, c ox Represents the gate oxide capacitance per unit area, W N7 / L N7 Represents the width-to-length ratio of the MN7 tube, V THN7 Represents the threshold voltage of MN7 tube; Set the proportional coefficient c of MN8 tube so that it works in the linear region at this time. MN9 tube obtains the mirror current b×d×I through current mirror replication. PTAT Since MN8 operates in the linear region, the main current flowing through MN8 and MN9 is I N9 , at this time the compensation current I X1 =I N9 , which is expressed as shown in Formula 6: I X1 =b×d×I PTAT (Formula 6) The compensation current generates a voltage V that is positively correlated with the absolute temperature through resistor R4. PT2 , which is expressed as shown in Formula 7: V PT2 =I X1 ×R4 (Formula 7) As the temperature increases, the positive temperature coefficient current I PTAT Slowly increases, negative temperature coefficient current I IPTAT Slowly decreases, when the temperature reaches a certain value, the positive temperature coefficient current I PTAT Greater than the negative temperature coefficient current I IPTAT , from formula 4, we can know that the positive temperature coefficient current I PTAT The increase in current I N7 The gate-source voltage V of MN7 tube increases without changing the width-to-length ratio of MN7 tube. GSN7 becomes larger, making V GSN7 -V THN7 Greater than its drain-source voltage V DSN7 , its working area changes from the saturation area to the linear area, which is equivalent to a linear resistor. Its resistance expression is shown in Formula 8: With I IPTAT The gate potential of MP14 tube increases. Since the gates of MP11 tube and MP14 tube are connected, the gate potential of MP11 tube also increases, making the gate-source voltage of MP11 tube |V GSP11 | is decreasing, |V GSP11 |-|V THP11 | is less than its drain-source voltage |V DSP11 |, so that it changes from the linear region to the saturation region; at this time, the main current flowing through MP11, MN6, and MN7 tubes is I P11 , its expression is shown in Formula 9. I P11 =a×I IPTAT (Formula 9) Set the proportional coefficient d of the MN9 tube so that it works in the linear region at this time. The MN8 tube obtains the mirror current c×I through the current mirror copy. P11 Since the MN9 tube works in the linear region, the main current flowing through the MN8 and MN9 tubes is I N8 , at this time the compensation current I X2 =I N8 , which is expressed as shown in formula 10: I X2 =c×I P11 =c×a×I IPTAT (Formula 10) The compensation current generates a voltage V that is negatively correlated with the absolute temperature through resistor R4. NT2 , which is expressed as shown in formula 11: V NT2 =I X2 ×R4 (Formula 11) The voltage V generated at this time is positively correlated with the absolute temperature through the compensation circuit and the reference voltage generation circuit. PT2 and the voltage V which is negatively correlated with the absolute temperature NT2 The compensation voltage V CC , using a first-order compensated bandgap reference voltage V REF1 Subtract the compensation voltage V generated by the compensation circuit CC The compensated bandgap reference voltage V REF2 , V REF2 The expression is shown in formula 12: V REF2 = V REF1 - (V PT2 + V NT2 ) = V REF1 - V CC (Equation 12) The expression of the temperature coefficient formula is shown in Formula 13: According to the temperature coefficient formula, V MAX V REF2 Maximum voltage, V MIN V REF2 The minimum voltage, using the first-order temperature compensation voltage V REF1 Subtract the compensation voltage V CC , then the compensated circuit V MAX and V MIN becomes closer, that is, V MAX -V MIN Becomes smaller, thereby reducing the temperature coefficient and achieving a compensation effect.

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