Spatial decoupling TBM cutterhead multiphase motor position-sensorless field weakening control method

Abstract

The invention discloses a space decoupling TBM cutterhead multi-phase motor sensorless field weakening control method, which comprises the following steps of: converting five-phase current of a TBM cutterhead FP-FIPM motor into alpha-beta axis and dq axis fundamental wave current and third harmonic current, calculating an estimated electrical angle and an estimated rotating speed by adopting a sensorless space decoupling strategy for estimating a rotor position, and calculating the position of the TBM cutterhead FP-FIPM according to the estimated electrical angle and the estimated rotating speed. Weak magnetic control is distributed to a fundamental wave space, sensorless control is distributed to a third harmonic space, a self-adaptive third harmonic current injection weak magnetic control method is adopted, and the torque output capacity of a weak magnetic area is improved through the weak magnetic control of the fundamental wave space and third harmonic current with a controllable injection ratio. The problem of space coupling brought by existing position-sensorless flux-weakening control in the fundamental wave space implementation process is solved, the estimation precision of position-sensorless control under flux-weakening operation is effectively improved, and the motor keeps low coupling, high precision, high robustness and high reliability in a complex multi-disturbance environment. And long-term stable high-efficiency operation of the TBM cutterhead driving system is ensured.

Description

Technical Field

[0001] The present invention belongs to the technical field of high-performance control of multi-phase motors, and relates to a spatially decoupled position sensorless control for a five-phase magnetic field enhancement type permanent magnet motor, and specifically to a position sensorless magnetic weakening control method for a cutterhead multi-phase motor of a TBM (Tunnel Boring Machine) full-face tunnel boring machine. Background Art

[0002] TBM (Tunnel Boring Machine) is a full-face tunnel boring machine. It is a large-scale tunnel construction equipment that integrates mechanical, electrical, hydraulic, optical, and gas systems, including a series of continuous operations such as tunnel excavation, support, and slag discharge. It has a wide range of applications in underground rail transit and tunnel excavation. In specific applications, TBM uses a cutterhead for cutting operations, and the cutterhead is directly driven by the cutterhead drive system. Therefore, as the core component of TBM, the performance of the cutterhead drive system directly affects the excavation efficiency and quality of TBM. At present, motor drive, as a driving mode of TBM, has the advantages of fast response and high efficiency, and is widely used in the TBM field. The special working occasions of TBM require its cutterhead motor drive system to have high power, high efficiency, strong robustness, high reliability, low-speed high torque, high fault tolerance and other performance. For example, in different working conditions such as rapid excavation and low-speed escape in soft rock and broken geological belts, TBM puts forward high requirements on the speed regulation range and torque output capacity of the cutterhead drive motor. Therefore, a high-performance cutterhead motor drive system is an important prerequisite for ensuring high-efficiency operation of the TBM.

[0003] The five-phase flux-intensifying permanent magnet (FP-FIPM) motor has the advantages of high efficiency, high power density, wide speed regulation range, low torque ripple, strong low-speed torque output ability and strong fault tolerance ability, and is very suitable as the drive motor of the TBM cutterhead drive system. In addition, in order to meet the requirements of rapid tunneling of TBM under complex geological conditions such as hard rock, soft rock and fracture zone, the five-phase FP-FIPM motor in its cutterhead motor drive system needs to operate above the rated speed. The literature "Fuzzy Logic Speed Control of Permanent Magnet Synchronous Machine and Feedback Voltage Ripple Reduction in Flux-Weakening Operation Region" (IEEE Transactions on Industry Applications, 2020) adopts a field-weakening control based on voltage feedforward, which can effectively broaden the speed range of the motor and avoid the serious dependence on motor parameters and complex calculations. At present, some technologies have extended the application scope of field-weakening control to five-phase permanent magnet motors. However, it should be noted that different from traditional five-phase permanent magnet motors, the FP-FIPM motor has a reduced torque output ability in the field-weakening operation region due to its inverse salient pole characteristics. And there has not been any relevant technological breakthrough in the field-weakening control of FP-FIPM motors yet.

[0004] In addition, the working environment of the TBM cutterhead drive system is complex and harsh. The system often operates in high-temperature, high-humidity, dusty, high-pressure environments for a long time and faces complex and changeable geological environments, resulting in the position sensors in the cutterhead motor drive system being extremely prone to failure, seriously affecting the working efficiency of the TBM cutterhead drive system. Moreover, due to the special working conditions of the TBM, it is relatively difficult to repair or even replace faulty components. The sensorless control strategy based on back electromotive force, as an effective solution, can ensure the stable operation of the TBM cutterhead drive system after the position sensor fails. For this reason, the literature with Chinese Patent No. CN202011424018.6 proposes "a sensorless control method for a five-phase permanent magnet synchronous motor", which is a sensorless control method for a five-phase permanent magnet synchronous motor based on the third harmonic back electromotive force, and explores a sensorless control strategy based on the fundamental back electromotive force under weak magnetic operation, but ignores the complex coupling relationship between weak magnetic control and sensorless control, greatly increasing the weak magnetic current error, estimated rotor position error and torque ripple. Therefore, there is an urgent need in this field to develop a sensorless weak magnetic control strategy for the TBM cutterhead FP-FIPM motor that can meet low coupling, high precision, strong robustness, high reliability and easy application in a complex multi-disturbance environment, so as to ensure the long-term stable and high-efficiency operation of the TBM cutterhead drive system. Summary of the Invention

[0005] Object of the Invention: Aiming at the problems existing in the prior art, the present invention proposes a sensorless weak magnetic control method for a multi-phase motor of a TBM cutterhead with spatial decoupling based on an Adaptive Frequency Complex Coefficient Filter Sliding Mode Observer (hereinafter referred to as AFCCF-SMO). By introducing the dual-space design concept of weak magnetic control and sensorless control, an adaptive third harmonic current injection weak magnetic strategy is proposed in the fundamental space, and an AFCCF-SMO sensorless control strategy is proposed in the third harmonic space, which can not only cleverly avoid the rotor position estimation error caused by spatial coupling and complex interference, improve the performance of sensorless control, but also improve the torque output ability of the TBM cutterhead FP-FIPM motor under weak magnetic control.

[0006] Technical Solution: To achieve the above object of the invention, the technical solution adopted by the sensorless weak magnetic control method for a multi-phase motor of a TBM cutterhead with spatial decoupling of the present invention includes the following steps:

[0007] Step 1): Transform the five-phase current of the TBM cutterhead FP-FIPM motor into fundamental current and third harmonic current on the αβ axis and dq axis;

[0008] Step 2): Use the αβ-axis third harmonic current, the αβ-axis third harmonic voltage at the previous moment feedback by the inverse Park transformation, the equivalent back electromotive force h α3 (t), h β3 (t) and the estimated third harmonic back electromotive force at the previous moment feedback as inputs, output the estimated third harmonic current error to the sigmoid function module, and output the synchronous frequency to the adaptive frequency complex coefficient filter module; the sigmoid function module calculates the equivalent back electromotive force h α3 (t + 1), h β3 (t + 1); use the equivalent back electromotive force h α3 (t + 1), h β3 (t + 1) and the third harmonic back electromotive force to calculate the back electromotive force feedback error. The adaptive frequency complex coefficient filter module calculates the third harmonic back electromotive force at the current moment with the back electromotive force feedback error and the synchronous frequency as inputs

[0009] Step 3): The third harmonic back electromotive force obtains the estimated electrical angle through a phase-locked loop, and the estimated electrical angle is transformed into the estimated mechanical angular velocity;

[0010] Step 4): The speed error after comparing the estimated mechanical angular velocity with the given speed is adjusted to obtain the current amplitude i s * , and calculate the voltage limit u lim according to the dq-axis voltage feedback by the current loop PI regulation, and calculate the difference Δu s from the voltage amplitude; if Δu s is greater than or equal to 0, use the i s * and Δu s as inputs to output the dq-axis current reference value. If Δu s is less than 0, first generate the leading angle β, then calculate the dq-axis current reference value, and then calculate the given value of the third harmonic current through the third harmonic injection ratio calculation;

[0011] Step 5): After comparing the fundamental wave current and the third harmonic current of the dq-axis with the dq-axis current reference value and the given value of the third harmonic current respectively, output the dq-axis voltage through the current loop PI regulation, then obtain the αβ-axis voltage and the third harmonic space voltage through the inverse Park transformation, and generate a switching sequence to form an equivalent voltage through a five-phase inverter.

[0012] The present invention has the following beneficial effects after adopting the above technical solutions:

[0013] 1) This invention introduces the concept of spatial decoupling into the sensorless weak magnetic drive system of the TBM cutter disc FP-FIPM motor for the first time, and innovatively proposes a spatial decoupling strategy of dual-space design. The weak magnetic control is cleverly allocated to the fundamental wave space, while the sensorless control is allocated to the third harmonic space, which solves the spatial coupling problem caused by the existing sensorless weak magnetic control in the fundamental wave space, and effectively improves the estimation accuracy of the sensorless control under weak magnetic operation. The proposed spatial decoupling strategy enables the TBM cutter disc FP-FIPM motor to maintain low coupling, high precision, strong robustness and high reliability in a complex multi-disturbance environment, thereby ensuring the long-term stable and efficient operation of the TBM cutter disc drive system.

[0014] 2) This invention innovatively proposes a spatial decoupling position sensorless control strategy based on AFCCF-SMO, and distributes the position sensorless control to the third harmonic space, overcoming the problem of weak magnetic current error, estimated rotor position error and torque pulsation caused by the coupling of traditional sliding mode observer (SMO) control and weak magnetic control. The accuracy of rotor position estimation is improved, thereby ensuring that the TBM cutterhead drive system can still operate stably and accurately after a position sensor failure occurs.

[0015] 3) The present invention utilizes the TBM cutter head FP-FIPM motor L d3 Almost equal to L q3 The characteristics of the sensorless weak magnetic drive system are simplified, and the extended third harmonic back-EMF equation is simplified, thereby simplifying the observer design process and further reducing the difficulty of application, debugging and improvement of the sensorless weak magnetic drive system in actual engineering.

[0016] 4) The AFCCF-SMO proposed in the present invention significantly simplifies the parameter setting process in the traditional sliding mode observer due to its adaptive frequency characteristics, which is conducive to practical application and engineering.

[0017] 5) The present invention overcomes the difficult problems of phase delay and amplitude attenuation of the low-pass filter in traditional SMO control through the adaptive frequency and bandpass filtering characteristics of the proposed AFCCF-SMO, and further improves the accuracy of estimated position.

[0018] 6) The present invention proposes an adaptive third harmonic injection weak magnetic method to reasonably distribute the current in the fundamental plane and the third harmonic plane in the electromagnetic torque, thereby improving the current utilization rate and increasing the speed operating range of the TBM cutterhead FP-FIPM motor, which is beneficial for the TBM cutterhead drive system to be suitable for different working conditions such as rapid excavation and low-speed escape.

[0019] 7) The self - adaptive third - harmonic injection field - weakening method proposed by the present invention can make full use of the third - harmonic torque in the third - harmonic plane of the TBM cutterhead FP - FIPM motor, improve the torque output ability of the motor in the field - weakening region, overcome the adverse effect of the reluctance torque of the TBM cutterhead FP - FIPM motor on the torque output ability in the field - weakening region, and meet the requirements of the TBM for rapid tunneling under complex geological conditions such as hard rock, soft rock, and fracture zones. Description of the Drawings

[0020] Figure 1 It is a schematic structural diagram of the TBM cutterhead FP - FIPM motor controlled by the control method of the present invention;

[0021] Figure 2 It is a block diagram of the composition principle of the space - decoupled position - less control system of the TBM cutterhead FP - FIPM motor based on AFCCF - SMO of the present invention;

[0022] Figure 3 For Figure 2 It is a block diagram of the composition principle of the AFCCF - SMO module based on the third - harmonic back - electromotive force in

[0023] Figure 4 For Figure 2 It is a block diagram of the composition principle of the field - weakening module with self - adaptive third - harmonic current injection in

[0024] Figure 5 They are the operating waveforms with traditional field - weakening control at different speeds;

[0025] Figure 6 They are the operating waveforms with the self - adaptive third - harmonic injection field - weakening control of the present invention;

[0026] Figure 7 They are the comparison waveforms of the estimated speeds with traditional SMO and AFCCF - SMO of the present invention;

[0027] Figure 8 They are the comparison waveforms of the estimated rotor positions with traditional SMO and AFCCF - SMO of the present invention;

[0028] Figure 9 They are the operating waveforms of sensorless control during field - weakening operation with the traditional strategy;

[0029] Figure 10 They are the operating waveforms with the space - decoupled position - less control based on AFCCF - SMO of the present invention.

[0030] In the figure: 1. Stator; 2. Rotor; 3. Armature teeth; 4. Fault-tolerant teeth; 5. Armature winding; 6. Permanent magnet; 7. Magnetic barrier; 8. Current Clark transformation module; 9. Current Park transformation module; 10. AFCCF-SMO module based on third-harmonic back electromotive force; 11. Phase-locked loop module; 12. Electrical angle differential module; 13. Speed transformation module; 14. Speed loop PI module; 15. Field weakening module with adaptive third-harmonic current injection; 16. Current loop PI module; 17. Inverse Park transformation module; 18. SVPWM module; 19. Five-phase inverter module; 20. TBM cutterhead FP-FIPM motor; 21. Observer αβ-axis current estimation module; 22. Sigmoid function module; 23. Adaptive frequency complex coefficient filter module; 24. u lim Limiting module; 25. Maximum Torque Per Ampere (hereinafter referred to as MTPA) and field weakening control module; 26. Third-harmonic injection ratio calculation module. Specific implementation manner

[0031] In order to make the objectives, technical solutions and advantages of the present invention clearer and more understandable, the present invention will be further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the present invention and are not used to limit the present invention.

[0032] As Figure 1 shown, the structural schematic diagram of the TBM cutterhead FP-FIPM motor controlled by the control method of the present invention includes: a stator 1, a rotor 2, armature teeth 3, fault-tolerant teeth 4, an armature winding 5, a permanent magnet 6 and a magnetic barrier 7. The stator 1 is sleeved outside the rotor 2. Armature teeth 3 and fault-tolerant teeth 4 are evenly distributed at intervals in the circumferential direction of the inner circle of the stator 1. The tooth widths of the armature teeth 3 and the fault-tolerant teeth 4 are not equal; an armature winding coil 5 is wound around the armature teeth 3, which is a single-layer concentrated winding, and two adjacent single-layer concentrated windings are isolated by the fault-tolerant teeth 4; the total number of teeth of the armature teeth 3 and the fault-tolerant teeth 4 is 20. The number of pole pairs of the permanent magnet 6 is 6. In combination with the slot-pole matching, the permanent magnet 6 adopts a spoke-type arrangement with a flux concentration effect. The rotor 2 realizes the magnetic field enhancement effect of the motor by designing two magnetic barriers 7, which shows the inverse salient pole effect that the fundamental wave space direct-axis inductance L d1 is greater than the fundamental wave space quadrature-axis inductance L q1 Compared with traditional permanent magnet synchronous motors, the TBM cutterhead FP-FIPM motor has the advantages of high efficiency, high power density, wide speed regulation range, low torque ripple, strong low-speed torque output ability and strong fault tolerance ability, and is very suitable as the drive motor of the TBM cutterhead drive system.

[0033] Figure 2The following is the structural block diagram of the sensorless control system for the spatial decoupling of the FP-FIPM motor of the TBM cutterhead based on AFCCF-SMO of the present invention. Comparing with the structural diagram of the sensorless control system for the five-phase permanent magnet synchronous motor based on the traditional sliding mode observer, the present invention adopts the adaptive third harmonic injection field weakening method to replace the traditional field weakening method, and alleviates the adverse effect of the reluctance torque of the FP-FIPM motor of the TBM cutterhead by injecting a third harmonic component with a variable ratio to the fundamental wave, improves the torque output capacity of the motor in the field weakening region, and meets the requirements of the high-performance TBM cutterhead drive system for the speed regulation range and torque output capacity. At the same time, compared with the traditional SMO control, the present invention is based on the spatial decoupling sensorless control strategy of AFCCF-SMO, and distributes the sensorless control to the third harmonic space, overcoming the problems of field weakening current error, estimated rotor position error and torque ripple caused by the coupling of the traditional SMO control and the field weakening control. In addition, the characteristics of AFCCF-SMO for adaptive frequency and band-pass filtering further improve the accuracy of the estimated position. This ensures that the TBM cutterhead drive system still operates stably and with high precision after the position sensor fails.

[0034] The sensorless control system for the spatial decoupling of the FP-FIPM motor of the TBM cutterhead includes: a current Clark transformation module 8, a current Park transformation module 9, an AFCCF-SMO module 10 based on the back electromotive force of the third harmonic, a phase-locked loop (PPL) module 11, an electrical angle differential module 12, a speed transformation module 13, a speed loop PI module 14, an FW (field weakening) module 15 for adaptive third harmonic current injection, a current loop PI module 16, an inverse Park transformation module 17, an SVPWM module 18, and a five-phase inverter module 19. This control system is connected to the FP-FIPM motor 20 of the TBM cutterhead to achieve control. The specific control steps are as follows:

[0035] Step 1): Use the Clark transformation and Park transformation to decouple the five-phase current i ABCDE output by the FP-FIPM motor 20 of the TBM cutterhead into the fundamental wave current and the third harmonic current of the αβ axis and the dq axis.

[0036] In order to decouple the five-phase current i ABCDE into the fundamental wave current and the third harmonic current of the αβ axis, connect the output end of the FP-FIPM motor 20 of the TBM cutterhead to the current Clark transformation module 8 and the current Park transformation module 9 in sequence,

[0037] collect the five-phase current i ABCDE and the five-phase voltage u ABCDE output by the FP-FIPM motor 20 of the TBM cutterhead. The five-phase current i ABCDE is namely five current signals i A , i B, i C , i D , i E , five-phase voltage u ABCDE That is, five voltage signals u A , u B , u C , u D , u E . Five-phase current i ABCDE is input into the current Clark transformation module 8, and the corresponding currents i α1 , i β1 , i α3 , i β3 , i0:

[0038]

[0039] Among them, i0 is the zero-sequence current, and the Clark transformation matrix is:

[0040]

[0041] In the formula, T Clark is the Clark transformation matrix, α is the angle between the axes of adjacent two-phase windings, and for the FI-IPM motor, α is 2π / 5.

[0042] The output terminals of the current Clark transformation module 8 are respectively connected to the current Park transformation module 9 and the AFCCF-SMO module 10 based on the third-harmonic back electromotive force. The αβ-axis currents i α1 , i β1 , i α3 , i β3 are input into the current Park transformation module 9, and the αβ-axis currents i α3 , i β3 are input into the AFCCF-SMO module 10 based on the third-harmonic back electromotive force.

[0043] The current Park transformation module 9 uses the αβ-axis currents i α1 , i β1 , i α3 , i β3 and the estimated electrical angle output by the phase-locked loop module 11 as input signals, and transforms them into the corresponding dq-axis currents i d1 , i q1 , i d3 , i q3 :

[0044]

[0045] Among them, \(i_0\) is the zero-sequence current, and the extended Park transformation matrix is:

[0046]

[0047] In the formula, \(T\) Park_ex is the extended Park matrix, is the estimated electrical angle. Compared with the traditional control method, since the sensorless control adopted in the present invention uses the estimated electrical angle to replace the true rotor position, that is, the true electrical angle \(\theta\) e for coordinate transformation.

[0048] Step 2): Based on the currents and voltages \(i\) α3 , \(i\) β3 , \(u\) α3 , \(u\) β3 in the third-harmonic space, a sensorless space decoupling strategy for estimating the rotor position is used to calculate the estimated electrical angle and the estimated rotational speed

[0049] The working environment of the TBM cutterhead drive system is complex and harsh. The system often operates in high-temperature, high-humidity, dusty, high-voltage and other environments for a long time, and faces a complex and changeable geological environment, resulting in the position sensor in the cutterhead motor drive system being extremely prone to failure, seriously affecting the working efficiency of the TBM cutterhead drive system. For this reason, the sensorless control strategy is an effective solution. However, the complex coupling relationship between the field-weakening control and the sensorless control greatly increases the field-weakening current error, the estimated rotor position error and the torque ripple. Therefore, a sensorless field-weakening control strategy that meets low coupling, high precision, strong robustness, high reliability and easy application in a complex multi-disturbance environment is needed. For this reason, the present invention adopts the AFCCF-SMO module 10 based on the third-harmonic back electromotive force, as Figure 3 shown. This module is composed of an observer \(\alpha\beta\)-axis current estimation module 21, a sigmoid function module 22 and an adaptive frequency complex coefficient filter module 23. The output ends of the observer \(\alpha\beta\)-axis current estimation module 21 are respectively connected to the sigmoid function module 22 and the adaptive frequency complex coefficient filter module 23, and the outputs of the sigmoid function module 22 and the adaptive frequency complex coefficient filter module 23 are also fed back to the observer \(\alpha\beta\)-axis current estimation module 21 to form a closed loop.

[0050] By realizing sensorless control in the third-harmonic space, the influence of the coupling between the field-weakening control and the sensorless control on the position estimation accuracy is suppressed.

[0051] The third-harmonic space currents \(i\) α3 , \(i\) β3, the third harmonic space voltage u at the previous moment t output by the inverse Park transformation module 17 α3 (t), u β3 (t), the equivalent back electromotive force h α3 (t), h β3 (t) and the estimated third harmonic back electromotive force at time t feedback by the adaptive frequency complex coefficient filter module 23 are used as the inputs of the observer αβ-axis current estimation module 21. The observer αβ-axis current estimation module 21 outputs the estimated third harmonic current error to the sigmoid function module 22, and outputs the synchronous frequency of the AFCCF-SMO to the adaptive frequency complex coefficient filter module 23.

[0052] According to the extended third harmonic back electromotive force model of the TBM cutter head FP-FIPM motor, with the third harmonic space current i α3 , i β3 , the third harmonic space voltage u at the previous moment output by the inverse Park transformation module 17 α3 (t), u β3 (t), the equivalent back electromotive force h at the previous moment feedback by the sigmoid function module 22 α3 (t), h β3 (t) and the estimated third harmonic back electromotive force at time t feedback by the adaptive frequency complex coefficient filter module 23 are used as the inputs of the observer αβ-axis current estimation module 21. The observer αβ-axis current estimation module 21 obtains the estimated third harmonic current error The calculation method is as follows:

[0053]

[0054] Where, L d3 is the third harmonic space d-axis inductance, R s is the stator resistance, and p is the differential operator.

[0055] In addition, the observer αβ-axis current estimation module 21 also needs to calculate the synchronous frequency of the AFCCF-SMO and output it to the adaptive frequency complex coefficient filter module 23. The observer αβ-axis current estimation module 21 first calculates the extended third harmonic back electromotive force e α3 , e β3 , the method is as follows:

[0056]

[0057] The αβ-axis current estimation module 21 of the observer further calculates the synchronous frequency of the AFCCF-SMO based on the extended third harmonic back electromotive force e α3 ,e β3 ,according to the following method

[0058]

[0059] The obtained synchronous frequency of the AFCCF-SMO is used as the input of the adaptive frequency complex coefficient filter module 23

[0060] The sigmoid function module 22 takes the estimated third harmonic current error output by the αβ-axis current estimation module 21 of the observer as the input, and calculates the equivalent back electromotive force h α3 (t + 1), h β3 (t + 1), and the calculation method is as follows

[0061]

[0062] where σ is the sliding mode gain

[0063] The equivalent back electromotive force h α3 (t + 1), h β3 (t + 1) is the output value of the sigmoid function module 22 at the current moment, and the equivalent back electromotive force h α3 (t), h β3 (t) is the feedback value fed back to the αβ-axis current estimation module 21 of the observer in the previous sampling period

[0064] The sigmoid function is continuous within the boundary layer and continuously transitions at a lower frequency on the sliding mode surface, effectively suppressing the high-frequency switching brought by the sign function, reducing the chattering of the predicted value, and improving the robustness of the observer. The expression of the sigmoid function is as follows

[0065]

[0066] where x is the input of the function, which is the estimated third harmonic current error in the present invention e is the natural constant

[0067] The sensorless space decoupling strategy of the present invention not only improves the accuracy of rotor position estimation, but also simplifies the extended third harmonic back electromotive force equation, thereby simplifying the observer design process and reducing the difficulty of application, debugging and improvement of the present sensorless field weakening drive system in practical engineering

[0068] An adaptive frequency complex coefficient filter is used to replace the low-pass filter, and the accuracy of the estimated position is further improved through the characteristics of adaptive frequency and band-pass filtering.

[0069] The equivalent back electromotive force h α3 (t + 1), h β3 (t + 1), at the current moment output by the sigmoid function module 22, and the estimated third harmonic back electromotive force at the previous moment fed back by the adaptive frequency complex coefficient filter module 23 are subtracted to obtain the back electromotive force feedback error at the previous moment. The adaptive frequency complex coefficient filter module 23 uses the back electromotive force feedback error at the previous moment and the synchronous frequency of AFCCF-SMO output by the observer αβ-axis current estimation module 21 as inputs and performs the following calculations:

[0070]

[0071] where k c is a variable coefficient. The adaptive frequency complex coefficient filter module 23 then performs the following calculations on the intermediate variable z α3 (t), z β3 (t):

[0072]

[0073] where the estimated third harmonic back electromotive force is the output value of the adaptive frequency complex coefficient filter module 23 at the current moment, and the estimated third harmonic back electromotive force is the feedback value fed back to the observer αβ-axis current estimation module 21 in the previous sampling period.

[0074] The estimated third harmonic back electromotive force output by the adaptive frequency complex coefficient filter module 23 at the current moment is also used as the output of the AFCCF-SMO module 10 based on the third harmonic back electromotive force.

[0075] Step 3): The output end of the AFCCF-SMO module 10 based on the third harmonic back electromotive force is sequentially connected to a phase-locked loop module 11, an electrical angle differential module 12, and a rotational speed conversion module 13.

[0076] The phase-locked loop module 11 uses the estimated third harmonic back electromotive force at the current moment output by the AFCCF-SMO module 10 based on the third harmonic back electromotive force and the estimated electrical angle at the previous moment fed back by the phase-locked loop module 11 As input, the normalized error signal ε3 is calculated through the phase detector. The calculation method is as follows:

[0077]

[0078] Based on the normalized error signal ε3, the estimated electrical angle at the current moment is obtained through the PI link and the integral link as a loop filter.

[0079]

[0080] Among them, s represents the differential, k p , k i is the proportional coefficient and integral coefficient of the phase-locked loop. Estimated electrical angle is the output value of the phase-locked loop module 11 at the current time, used in coordinate transformations throughout the system; estimates the electrical angle It is the feedback value fed back to the phase-locked loop module 11 in the previous sampling period.

[0081] Estimated electrical angle The estimated electrical angular velocity is calculated by the electrical angle differential module 12 Then the speed conversion module 13 is divided by the pole pair number to obtain the estimated mechanical angular velocity With the given speed ω m * After comparison, a speed closed loop is formed.

[0082] The AFCCF-SMO proposed in the present invention improves the rotor position estimation accuracy and can ensure that the TBM cutter head drive system can still operate stably and with high precision after a position sensor failure occurs.

[0083] Step 4): The present invention adopts a magnetic weakening method of adaptive third harmonic current injection to improve the torque output capacity of the magnetic weakening area by controlling the magnetic weakening of the fundamental wave space and injecting a third harmonic current with a controllable ratio.

[0084] Due to the reverse salient pole characteristics, the reluctance torque of the TBM cutterhead FP-FIPM motor in the weak magnetic field area has an adverse effect on the torque output capacity. The adaptive third harmonic current injection weak magnetic field method improves the torque output capacity of the TBM cutterhead FP-FIPM motor in the weak magnetic field area, which can meet the needs of TBM rapid excavation under complex geological conditions such as hard rock, soft rock, and broken belts. The specific method is:

[0085] 4.1) Set the given speed ω m * and estimated mechanical angular velocity For comparison, given speed ω m * Subtract the estimated mechanical angular velocity The speed error Δω ism , is input to the speed loop PI module 14. In the speed loop PI module 14, the speed error Δω m is adjusted by the conventional PI to obtain the current amplitude i s * , the current amplitude i s * is output from the speed loop PI module 14 to the field weakening module 15 with adaptive third harmonic current injection. The dq-axis voltages u d1 (t), u q1 (t) feedback by the current loop PI module 16 are also input to the field weakening module 15 with adaptive third harmonic current injection.

[0086] As Figure 4 shown in the block diagram of the composition principle of the field weakening module 15 with adaptive third harmonic current injection, this module consists of a u lim limiting module 24, an MTPA and field weakening control module 25, and a third harmonic injection ratio calculation module 26. The output end of the u lim limiting module 24 is respectively connected to the MTPA and field weakening control module 25 and the third harmonic injection ratio calculation module 26 to respectively judge whether to enter the field weakening control stage and whether to adopt the field weakening control with adaptive third harmonic current injection proposed by the present invention. The output end of the MTPA and field weakening control module 25 is connected to the third harmonic injection ratio calculation module 26. lim The limiting module 24 calculates the difference Δu d1 between the voltage limit u q1 and the voltage amplitude with the dq-axis voltages u lim (t), u s (t) feedback by the current loop PI module 16 as the input, and outputs to the MTPA and field weakening control module 25 and the third harmonic injection ratio calculation module 26. The difference Δu s of the voltage amplitude is:

[0087]

[0088] Among them, the voltage limit u lim is the amplitude of the maximum output voltage of the inverter to the motor terminal, and is set to a certain value according to the bus supply voltage.

[0089] 4.2) According to the positive and negative of the voltage difference Δu lim output by the limiting module 24, it is divided into two cases: s If the voltage difference Δu

[0090] of the voltage amplitude sGreater than or equal to 0, indicating that the dq-axis current vector is in the first quadrant. At this time, the voltage amplitude difference is input to the MTPA and field weakening control module 25, and the MTPA and field weakening control module 25 adopts MTPA control. The MTPA and field weakening control module 25 uses the phase current amplitude i output by the speed loop PI module 14 s * and u lim The voltage difference Δu output by the limiter module 24 s as the input, and outputs the dq-axis current reference values i d1 * , i q1 * , and the calculation method is as follows:

[0091]

[0092] where L d1 , L q1 are the dq-axis inductances in the fundamental wave plane, and the output of the third harmonic injection ratio calculation module 26 is set to i q3 * =0. ψ f1 is the permanent magnet flux linkage in the fundamental wave space of the motor.

[0093] If the voltage amplitude difference Δu s is less than 0, indicating that the dq-axis current vector is in the second quadrant. At this time, the voltage amplitude difference is input to the MTPA and field weakening control module 25, and the adaptive third harmonic current injection field weakening method proposed by the present invention is adopted:

[0094] First, the MTPA and field weakening control module 25 uses u lim The voltage amplitude difference Δu output by the limiter module 24 s generates the leading angle β through a PI regulator. The leading angle β is the angle between the dq-axis current vector and the q-axis, and the calculation method is:

[0095] β=(k ii / s + k pp )Δu s (16)

[0096] where k ii , k pp are the PI regulator parameters, and the magnitudes of k ii , k pp are obtained through repeated experiments. 1 / s represents the integral operation.

[0097] Then calculate the dq-axis current reference values i d1 * , i q1 * The method is as follows:

[0098]

[0099] Then, the third harmonic injection ratio calculation module 26 calculates the dq axis current reference value i based on formula (16): d1 * ,i q1 * , calculate the third harmonic current given value i q3 * .i q3 * The calculation method is:

[0100] i q3 * =ri q1 * (18)

[0101] Among them, r is defined as the third harmonic injection ratio. Under the limitation of inverter capacity and with the maximum value of electromagnetic torque as the target, the specific expression of the optimal third harmonic current injection ratio r is:

[0102]

[0103] Among them, ψ f1 is the permanent magnet flux linkage in the motor fundamental wave space, ψ f3 It is the permanent magnet flux linkage in the third harmonic space of the motor.

[0104] The dq axis current reference value i output by the MTPA and the weak magnetic control module 25 d1 * ,i q1 * , the third harmonic q-axis current given value i output by the third harmonic injection ratio calculation module 26 q3 * All are outputs of the field weakening module 15 for adaptive third harmonic current injection.

[0105] TBM needs to work in different working conditions such as fast excavation in soft rock and broken geological zones and low-speed escape. The strategy of using MTPA control below the base speed and adaptive third harmonic current injection weak magnetic field above the base speed can meet the requirements of complex working conditions for the speed regulation range and torque output capacity of the TBM cutterhead drive motor.

[0106] Step 5): After the currents of the dq axes of the fundamental wave space and the third harmonic wave space are compared with the given values, the voltages of the fundamental wave space and the third harmonic wave space are calculated through the current loop PI module 16 and input into the SVPWM module to generate a switching sequence.

[0107] The actual dq axis current i output by the current Park conversion module 9 is d1 ,i q1 ,id3 , i q3 and the dq-axis current reference values i d1 * , i q1 * , i d3 * , i q3 * (where i d3 * = 0) are compared one by one to obtain the corresponding differential currents Δi d1 , Δi q1 , Δi d3 , Δi q3 , which are used as the inputs of the current-loop PI module 16. The output of the current-loop PI module 16 is sequentially connected to the inverse Park transformation module 17, the SVPWM module 18, the five-phase inverter module 19, and the TBM cutter head FP-FIPM motor 20. The current-loop PI module 16 performs PI-link calculations on the differential currents Δi d1 , Δi q1 , Δi d3 , Δi q3 and outputs the dq-axis voltages u d1 , u q1 , u d3 , u q3 . The inverse Park transformation module 17 uses the dq-axis voltages u d1 , u q1 , u d3 , u q3 and the estimated electrical angle at the current moment output by the phase-locked loop module 11 as inputs, and obtains the fundamental voltages u α1 , u β1 on the αβ axis and the third-harmonic space voltages u α3 , u β3 through inverse Park transformation. The inverse Park transformation matrix is:

[0108]

[0109] where T anti_Park is the inverse Park transformation matrix, is the estimated electrical angle.

[0110] The inverse Park transformation module 17 converts the dq-axis voltages u d1 , u q1 , u d3 , u q3 into the αβ-axis voltages u α1 , u β1 , u α3 , uβ3 , the calculation formula is:

[0111]

[0112] In the formula, u0 is the zero-sequence component, whose main function is to make the matrix calculation formula (19) hold and is not used as the output of the inverse Park transformation module 17.

[0113] The inverse Park transformation module 17 outputs the αβ-axis voltages u α1 , u β1 , u α3 , u β3 to the SVPWM module 18. The SVPWM module 18 generates the switching sequence S α1 , u β1 , u α3 , u β3 according to the input reference voltages u ABCDE in the fundamental wave and third harmonic spaces of the αβ-axis. The SVPWM module 18 outputs the switching sequence S ABCDE to the five-phase inverter module 19. The five-phase inverter module 19 reproduces the equivalent five-phase voltage u ABCDE in the form of PWM and outputs it to the TBM cutter head FP-FIPM motor 20.

[0114] Figure 5 shows the operating waveforms of the traditional field-weakening control method at different speeds, Figure 6 shows the operating waveforms of the sensorless field-weakening control method for the multi-phase motor of the TBM cutter head with spatial decoupling of the present invention at different speeds. Figure 5 and Figure 6 in which n is the speed, 400 rpm / div indicates that the amplitude per vertical grid is 400 rpm. i d , i q are the dq-axis currents, 3 A / div indicates that the amplitude per vertical grid is 3 A, T e is the torque, 5 N·m / div indicates that the amplitude per vertical grid is 5 N·m; Time is the abscissa time, 2 s / div indicates that the amplitude per horizontal grid is 2 s. It can be seen from Figure 5 that as the speed increases, the pulsations of the current and torque increase rapidly. It can be seen from Figure 6 that after the speed is increased to 1000 rpm, the stabilization time is shortened from 4 s to 1.6 s. The current and torque pulsations of the adaptive third-harmonic injection field-weakening control of the present invention are well suppressed under the entire operating conditions, and the dynamic performance of the system is also significantly improved, indicating that the adaptive third-harmonic injection field-weakening control of the present invention can improve the load capacity of the TBM cutter head FP-FIPM motor.

[0115] Figure 7The comparison waveforms of the estimated speed using the traditional SMO and the AFCCF-SMO of the present invention are given. Figure 8 The comparison waveforms of the estimated rotor position using the traditional SMO and the AFCCF-SMO of the present invention are given. Figure 7 and Figure 8 where n in [[ ]] is the rotational speed, is the estimated rotational speed, and 400 rpm / div indicates that the amplitude of each grid in the longitudinal direction is 400 rpm. is the rotational speed estimation error, and 200 rpm / div indicates that the amplitude of each grid in the longitudinal direction is 200 rpm. θ is the rotor position, is the estimated rotor position, and π rad / div indicates that the amplitude of each grid in the longitudinal direction is π rad. is the rotor position estimation error, and 0.5 rad / div indicates that the amplitude of each grid in the longitudinal direction is 0.5 rad. T e is the torque, and 5 N·m / div indicates that the amplitude of each grid in the longitudinal direction is 5 N·m. Time is the abscissa time, and 200 ms / div indicates that the amplitude of each grid in the transverse direction is 200 ms. Figure 7 , Figure 8 Each figure is divided into two versions of 200 ms / div and 5 ms / div from top to bottom. It can be found from the comparison that when operating under the same working conditions, the estimated rotational speed error and rotor position error of the AFCCF-SMO of the present invention are significantly reduced, indicating that the AFCCF-SMO can significantly improve the estimation performance of sensorless control.

[0116] Figure 9 The operating waveforms of the traditional strategy are given. Figure 10 The operating waveforms of the space decoupled sensorless control based on the AFCCF-SMO of the present invention under field weakening operation are given. Figure 9 and Figure 10 in [[ ]] is the rotational speed estimation error, and 100 rpm / div indicates that the amplitude of each grid in the longitudinal direction is 100 rpm. is the rotor position estimation error, and 0.5 rad / div indicates that the amplitude of each grid in the longitudinal direction is 0.5 rad. T e is the torque, and 5 N·m / div indicates that the amplitude of each grid in the longitudinal direction is 5 N·m. Time is the abscissa time, and 2 s / div indicates that the amplitude of each grid in the transverse direction is 2 s. As Figure 9 shown, due to the influence of negative magnetoresistance torque and spatial coupling, the estimated rotational speed error and rotor position error fluctuate violently with the changes of speed and load. As Figure 10 shown, the estimated rotational speed error and rotor position error are significantly reduced. At the same time, due to the adaptive frequency and band-pass filtering characteristics of the proposed AFCCF-SMO, the error fluctuations caused by variable speed and variable load are significantly reduced.

[0117] In summary, the present invention first uses Clark transformation and Park transformation to transform the five-phase current i ABCDE Decoupled into the fundamental current and third harmonic current of αβ axis and dq axis; the current voltage i in the third harmonic space α3 ,i β3 ,u α3 ,u β3 As input, a sensorless spatial decoupling strategy is used to estimate the rotor position and calculate the estimated electrical angle and estimated speed The performance of position sensorless control is improved by comprehensively considering the coupling of position sensorless control and weak magnetic control; a weak magnetic control method with adaptive third harmonic current injection is adopted to improve the torque output capacity of the weak magnetic area through weak magnetic control in the fundamental space and injection of controllable third harmonic current; finally, after comparing the currents of the dq axes in the fundamental space and the third harmonic space with the given values, the voltages in the fundamental space and the third harmonic space are calculated by the current loop PI module and input into the SVPWM module to generate a switching sequence. The AFCCF-SMO in the present invention overcomes the problem of weak magnetic current error, estimated rotor position error and torque pulsation caused by the coupling of traditional SMO control and weak magnetic control, and improves the estimation accuracy of position sensorless control under weak magnetic operation; the adaptive third harmonic injection weak magnetic method adopted improves the current utilization rate and the speed operating range of the TBM cutter head FP-FIPM motor, which is beneficial for the TBM cutter head drive system to be suitable for different working conditions such as rapid excavation and low-speed escape; compared with the existing research on position sensorless control, the present invention comprehensively considers the spatial coupling problem brought about by the position sensorless weak magnetic control in the fundamental wave space, and innovatively proposes a spatial decoupling strategy of dual-space design, which greatly improves the overall performance of the drive system and provides a feasible solution for the TBM cutter head FP-FIPM motor to maintain low coupling, high precision, strong robustness, high reliability, long-term stable high-efficiency operation in a complex multi-disturbance environment.

Claims

1. A spatially decoupled TBM cutterhead multiphase motor position sensorless weak magnetic control method, characterized in that The following steps are involved: Step 1): transform the five-phase current of the TBM cutterhead FP-FIPM motor into the fundamental current and third harmonic current of the αβ axis and dq axis; Step 2): The αβ axis third harmonic current, the αβ axis third harmonic voltage at the last moment of the inverse Park transformation feedback, and the equivalent back electromotive force h of the last moment of the feedback α3 (t), h β3 (t) and the estimated third harmonic back EMF of the previous moment of feedback As input, the estimated third harmonic current error is output to the sigmoid function module, and the synchronization frequency is output to the adaptive frequency complex coefficient filter module; The sigmoid function module calculates the equivalent back electromotive force h at the current moment α3 (t+1), h β3 (t+1); the equivalent back electromotive force h α3 (t+1), h β3 (t+1) and the third harmonic back electromotive force The back-EMF feedback error is obtained by subtraction. The adaptive frequency complex coefficient filter module uses the back-EMF feedback error and the synchronization frequency as input to calculate the third harmonic back-EMF at the current moment. Step 3): The third harmonic back electromotive force The estimated electrical angle is obtained through a phase-locked loop, and the estimated electrical angle is converted into an estimated mechanical angular velocity; Step 4): The speed error after comparing the estimated mechanical angular velocity with the given speed is adjusted to obtain the current amplitude i s * , calculate the voltage limit u according to the dq axis voltage feedback of the current loop PI regulation lim The difference between the voltage amplitude and s ; If Δu s is greater than or equal to 0, with the i s * and Δu s As input, output dq axis current reference value, if Δu s If it is less than 0, the lead angle β is generated first, and then the dq axis current reference value is calculated, and then the third harmonic current given value is calculated by the third harmonic injection ratio; Step 5): After comparing the fundamental current and third harmonic current of the dq axis with the dq axis current reference value and the third harmonic current given value, the output dq axis voltage is adjusted by the current loop PI, and then the αβ axis voltage and the third harmonic space voltage are obtained through the inverse Park transformation, and the switching sequence is generated to form an equivalent voltage through the five-phase inverter.

2. The method for position sensorless weak magnetic control of a TBM cutterhead multiphase motor with spatial decoupling according to claim 1 is characterized by: The estimated third harmonic current error described in step 2) u α3 (t),u β3 (t) is the αβ axis third harmonic voltage at the last moment of the anti-Park transformation feedback, L d3 is the third harmonic space d-axis inductance, R s is the stator resistance, and p is the differential operator.

3. The method for position sensorless weak magnetic control of a TBM cutterhead multiphase motor with spatial decoupling according to claim 2 is characterized in that: Synchronous frequency Among them, e α3 =u α3 (t)-i α3 (pL d3 +R s ), e β3 =u β3 (t)-i β3 (pL d3 +R s ).

4. The method for position sensorless weak magnetic control of a spatially decoupled TBM cutterhead multiphase motor according to claim 3 is characterized in that: Equivalent back electromotive force h α3 (t), h β3 (t) is: The expression of the sigmoid function is, σ is the sliding mode gain, x is the input of the function, and e is a natural constant.

5. The method for position sensorless weak magnetic control of a spatially decoupled TBM cutterhead multiphase motor according to claim 4 is characterized in that: The adaptive frequency complex coefficient filter module first calculates the intermediate variables: k c is the coefficient; Then calculate the third harmonic back electromotive force at the current moment 6. The method for position sensorless weak magnetic control of a spatially decoupled TBM cutterhead multiphase motor according to claim 1, characterized in that: In step 3), the phase-locked loop calculates the normalized error signal ε3 through a phase detector: Then calculate the estimated electrical angle at the current moment s represents differential, k p , k i are the proportional coefficient and integral coefficient of the phase-locked loop.

7. The method for position sensorless weak magnetic control of a spatially decoupled TBM cutterhead multiphase motor according to claim 6 is characterized in that: In step 4), the voltage limit u lim The difference between the voltage amplitude Voltage limit u lim is the maximum output voltage of the inverter to the motor terminal, u d1 (t),u q1 (t)u d1 (t) is the dq axis voltage of the current loop PI regulation feedback, voltage limit u lim It is the maximum voltage amplitude of the inverter output to the motor terminal.

8. The method for position sensorless weak magnetic control of a spatially decoupled TBM cutterhead multiphase motor according to claim 7 is characterized in that: If the voltage amplitude difference Δu s is greater than or equal to 0, then the dq axis current reference value i d1 * ,i q1 * for: L d1 , L q1 is the dq-axis inductance of the fundamental wave plane, ψ f1 It is the permanent magnet flux linkage in the fundamental wave space of the motor; If the voltage amplitude difference Δu s Less than 0, the lead angle β=(k ii / s+k pp )△u s , k ii , k pp is the PI regulator parameter, then the dq axis current reference value i d1 * =i s * sinβ,i q1 * =i s * cosβ, third harmonic current given value i q3 * =ri q1 * , where the third harmonic injection ratio ψ f1 is the permanent magnet flux linkage in the motor fundamental wave space, ψ f3 It is the permanent magnet flux linkage in the third harmonic space of the motor.

9. The method for position sensorless weak magnetic control of a TBM cutterhead multiphase motor with spatial decoupling according to claim 8 is characterized in that: In step 5), the αβ axis fundamental voltage and the third harmonic voltage are obtained by inverse Park transformation: u0 is the zero-sequence component.

10. The method for position sensorless weak magnetic control of a TBM cutterhead multiphase motor with spatial decoupling according to claim 1, characterized in that: In step 1), the five-phase current i ABCDE Clark transformation matrix T Clark The transformation results in the fundamental current of αβ axis and the third harmonic current i α1 ,i β1 ,i α3 ,i β3 : Then through the extended Park transformation matrix T Park_ex Get the dq axis fundamental current and the third harmonic current dq axis fundamental current and the third harmonic current i d1 ,i q1 ,i d3 ,i q3 : α is the angle between the axes of two adjacent phase windings, i0 is the zero-sequence current, It is the estimated electrical angle at the current moment obtained by the phase-locked loop.

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