A multi-stage filter amplifier suppressor

Abstract

The present application discloses a multi-stage filtering and amplifying suppressor, comprising: a signal acquisition module, a spatial attenuation and calculation module, a multi-stage filtering and amplifying module, and a simulation verification module; the signal acquisition module obtains the frequency of the received signal and collects the interference signal, and calculates the oscillator distribution spacing according to the frequency of the received signal; the spatial attenuation and calculation module calculates the spatial attenuation according to the spacing between the transmitting and receiving antennas; the multi-stage filtering and amplifying module performs multi-stage filtering and amplification on the received signal according to the interference signal and the spatial attenuation, and at the same time processes the insertion loss generated by the multi-stage filtering and amplification; through structural isolation and multi-stage filtering and amplifying circuits, the transmitting signal A1 is gradually suppressed from 40dBm to ‑75dBm, and the receiving signal B1 is increased from ‑100dBm to ‑56dBm, ultimately making the A1 signal lower than the B1 signal, thereby solving the interference problem of the short message system transmitting signal on the receiving channel.

Description

Technical Field

[0001] The present invention relates to the technical field of navigation communication equipment, and in particular to a multi-stage filtering amplification suppressor. Background Art

[0002] The airborne short message transceiver system belongs to the navigation and communication system, which consists of a transceiver antenna, a short message transceiver, a short message control handle, a locking cross-linking box, a mounting bracket, a handle bracket, a radio frequency cable and other components. It is mainly used to receive, send and display short message information, obtain the current position through the on-board navigation system, and realize point-to-point communication.

[0003] Due to size limitations and installation requirements, the short message system and receiving antenna are typically designed as an integrated whole. The short message system receives signals at the B1 frequency point in BD3 at 1575.42 MHz ± 16.368 MHz, while the transmitting antenna receives signals at a frequency point between 1615 MHz and 1627 MHz. Due to the close frequency difference between the two, the short message system's transmitted signal can directly interfere with the receiving antenna's signal path, causing saturation and distortion of the receiving antenna's signal channel, rendering the receiver unable to perform positioning calculations.

[0004] Traditional interference suppression methods primarily rely on filtering technology, designing filters with specific frequency responses to remove or attenuate interfering signals. However, this approach has significant limitations. On the one hand, while suppressing interference, filters often also attenuate the target signal to some extent. This is particularly true when the interfering and target signal spectra overlap, significantly reducing the filtering effectiveness. On the other hand, in complex and changing interference environments, a single filter cannot achieve comprehensive interference suppression, requiring the design of a more complex filtering network or the adoption of a multi-stage filtering architecture, which undoubtedly increases system complexity and cost. Summary of the Invention

[0005] The present application provides a multi-stage filtering and amplifying suppressor, which gradually suppresses the transmitting signal A1 from 40dBm to -75dBm through structural isolation and multi-stage filtering and amplifying circuits, and at the same time increases the receiving signal B1 from -100dBm to -56dBm, ultimately making the A1 signal lower than the B1 signal, solving the problem of interference of the transmitting signal of the short message system on the receiving channel.

[0006] The present application provides a multi-stage filtering and amplifying suppressor, comprising: a signal acquisition module, a spatial attenuation and calculation module, a multi-stage filtering and amplifying module, and a simulation verification module; the signal acquisition module obtains the frequency of a received signal and collects an interference signal, calculates the vibrator distribution spacing according to the frequency of the received signal, and converts the vibrator distribution spacing into the spacing between transmitting and receiving antennas; the spatial attenuation and calculation module calculates the spatial attenuation amount according to the spacing between transmitting and receiving antennas; the multi-stage filtering and amplifying module performs multi-stage filtering and amplification on the received signal according to the interference signal and the spatial attenuation amount, and simultaneously processes the insertion loss generated by the multi-stage filtering and amplification; the simulation verification module constructs a simulation model based on the results of the multi-stage filtering and amplification, and verifies the results of the multi-stage filtering and amplification.

[0007] Preferably, the vibrator distribution spacing is calculated according to the obtained frequency of the received signal, and the formula is: , where L is the spacing between the oscillators; f is the frequency of the received signal; C is the speed of light, which is 3×10 8 m / s.

[0008] Preferably, the unit of the vibrator distribution spacing is converted into kilometers and used as the spacing between the transmitting and receiving antennas. The spatial attenuation is calculated based on the spacing between the transmitting and receiving antennas. The formula is: , where is the spatial attenuation; 32.45 is a constant term, which represents the parameter of the automatic spatial path loss model, that is, the power attenuation of the signal from the transmitter to the receiver due to the propagation distance and frequency; F is the frequency; and D is the distance between the transmitting and receiving antennas.

[0009] Preferably, the interference signal is subjected to multi-stage filtering and amplification through spatial attenuation. First, the interference signal is subjected to primary filtering using a filter, and the signal at the transmitting end is attenuated at the same time; then, the interference signal that has passed the primary filtering is subjected to primary amplification, and the primary amplification is amplified using a low-noise amplifier; then, the interference signal that has passed the primary amplification is subjected to secondary filtering; then, the interference signal that has passed the secondary filtering is subjected to secondary amplification, and the secondary amplification is also amplified using a low-noise amplifier; finally, the interference signal that has passed the secondary amplification is subjected to tertiary filtering.

[0010] Preferably, the multi-stage filtering and amplifying module is used for processing insertion loss, specifically:

[0011] S201, using a power splitter to divide the receiving end signal into multiple paths to form multiple receiving channels, calculating the noise figure within a single channel based on the insertion loss of the filter, and calculating the delay difference;

[0012] S202, multiple receiving channels share the same local oscillator;

[0013] S203 , using a signal merging algorithm to calculate the insertion loss of the multiple channels, and calculating the noise coefficient of the multiple channels based on the noise coefficient within a single channel.

[0014] Preferably, the noise coefficient of a single channel in a multi-channel is calculated using the following formula: ,in, is the noise figure of a single channel; is the noise coefficient of the power divider; is the noise factor of the filter; is the gain of the power divider; is the gain of the filter; is the noise figure of the primary low noise amplifier; is the gain of the primary low noise amplifier; is the noise figure of the two-stage low-noise amplifier. The formula for calculating the noise figure of multiple channels is: ,in, is the noise figure of the multi-channel, is the noise coefficient of a single channel, N is the number of channels, and the noise coefficient of a single channel is the noise coefficient of the first receiving channel or the noise coefficient of the second receiving channel.

[0015] Preferably, the calculation formula for the multi-channel insertion loss is: ,in, is the equivalent insertion loss after multi-channel merging, is the insertion loss of a single channel, is the signal-to-noise ratio gain brought by MRC merging, and N is the number of channels.

[0016] Preferably, the step of calculating the time delay difference by the spatial attenuation and calculation module is:

[0017] S301, dividing the signal from the transmitting end into real-time data and calibration data, and classifying the interference level according to the power and signal-to-noise ratio of the received signal at the receiving end;

[0018] S302, performing delay matching on multiple channels, and obtaining delay differences of the multiple channels according to the delay matching;

[0019] S303: Compensate for multiple channels according to the obtained delay difference.

[0020] Preferably, the real-time data is communication data, and the transmitting end is equipped with a pseudo-random sequence generation module, which generates a pseudo-random sequence according to a preset algorithm and parameters, and the pseudo-random sequence is used as calibration data.

[0021] One or more technical solutions provided in this application have at least the following technical effects or advantages: through structural isolation (10cm spacing) and a multi-stage filtering and amplification circuit (three-stage filtering + two-stage amplification), the transmit signal A1 is gradually suppressed from 40dBm to -75dBm, while the receive signal B1 is simultaneously increased from -100dBm to -56dBm, ultimately making the A1 signal lower than the B1 signal. This solves the problem of interference from the transmit signal of the short message system on the receive channel. Simulation and measured data verify the effectiveness of the design, achieving efficient suppression of strong interference signals and precise amplification of weak receive signals.

[0022] The system significantly improves B1 signal reception performance through a dual-channel diversity reception architecture and maximum ratio combining technology. Spatial / polarization diversity reduces signal correlation, enhancing multipath fading mitigation. MRC combining in the dual-channel system reduces the 3.5 dB insertion loss of a single channel to 0.5 dB. After MRC combining, the noise figure is reduced from 3.92 dB for a single channel to 2.42 dB.

[0023] By dynamically adjusting the insertion density of calibration data, accurate dual-channel delay estimation and compensation are achieved in different interference scenarios, thereby ensuring correct signal reception and processing, improving the performance and stability of the communication system. Data is classified, calibration data is dynamically scheduled, and delay differences are calculated. Channel signals are calibrated based on the delay differences, improving delay compensation accuracy and reducing pseudorange errors.

[0024] By actively injecting reverse interference signals, the system can achieve precise suppression and effective control of specific interference signals, improving signal quality, optimizing spectral characteristics, and reducing the impact of high-frequency leakage and low-frequency background noise on the communication system, thereby significantly enhancing the performance and reliability of the communication system. The generated reverse interference signal is divided into four sub-bands based on the signal's frequency domain characteristics: the core suppression band, the edge suppression band, the high-frequency leakage suppression band, and the low-frequency leakage suppression band. Each sub-band is designed for different interference characteristics, achieving precise control of the interference signal. BRIEF DESCRIPTION OF THE DRAWINGS

[0025] Figure 1 This is a schematic diagram of a multi-stage filter amplifier suppressor according to the present invention;

[0026] Figure 2 This is a simulation result diagram of the multi-stage filtering and amplification suppression of the present invention;

[0027] Figure 3 This is a diagram showing actual test results of the multi-stage filtering and amplification suppression of the present invention;

[0028] Figure 4 A block diagram of a multi-stage filter amplifier suppressor according to the present invention;

[0029] Figure 5 A schematic diagram of a process for reducing insertion loss using multiple receiving channels according to an embodiment of the present invention;

[0030] Figure 6 A schematic diagram of a process for calculating delay inequality according to an embodiment of the present invention;

[0031] Figure 7 The figure is a flow chart of controlling original interference by injecting reverse interference signals according to an embodiment of the present invention. DETAILED DESCRIPTION

[0032] To facilitate understanding of the present invention, the present application will be described more comprehensively below with reference to the relevant drawings; the drawings show preferred embodiments of the present invention, but the present invention can be implemented in many different forms and is not limited to the embodiments described herein; on the contrary, the purpose of providing these embodiments is to enable a more thorough and comprehensive understanding of the disclosed content of the present invention.

[0033] It should be noted that the terms “vertical”, “horizontal”, “up”, “down”, “left”, “right” and similar expressions used in this document are for illustrative purposes only and do not represent the only implementation method.

[0034] Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by those skilled in the art to which this invention pertains; the terms used herein in the specification of the present invention are for the purpose of describing specific embodiments only and are not intended to limit the present invention; the term "and / or" used herein includes any and all combinations of one or more of the associated listed items.

[0035] Example 1: Figure 4 This is a flow chart of a multi-stage filtering and amplifying suppressor according to an embodiment of the present invention, comprising: a signal acquisition module, a spatial attenuation and calculation module, a multi-stage filtering and amplifying module, and a simulation verification module. The signal acquisition module is electrically connected to the spatial attenuation and calculation module, the spatial attenuation and calculation module is electrically connected to the multi-stage filtering and amplifying module, and the multi-stage filtering and amplifying module is electrically connected to the simulation verification module.

[0036] The signal acquisition module obtains the frequency of the received signal and the interference signal, calculates the vibrator distribution spacing according to the frequency of the received signal, and converts the vibrator distribution spacing into the spacing between the transmitting and receiving antennas. The spatial attenuation and calculation module calculates the spatial attenuation according to the spacing between the transmitting and receiving antennas. The frequency of the received signal is obtained using a spectrum analyzer, and the signal of the standard signal generator and the received signal are simultaneously input into the spectrum analyzer. The frequency of the standard signal is adjusted so that its spectrum line coincides with the spectrum line of the received signal. At this time, the frequency of the standard signal is the frequency of the received signal. The vibrator distribution spacing is calculated according to the obtained frequency of the received signal. The formula is: , where L is the spacing between the oscillators; f is the frequency of the received signal; C is the speed of light, which is 3×10 8 m / s, convert the vibrator distribution spacing into km as the spacing between the transmitting and receiving antennas, and calculate the spatial attenuation based on the spacing between the transmitting and receiving antennas. The formula is: ,in, is the spatial attenuation; 32.45 is a constant term, which represents the parameter of the automatic spatial path loss model, that is, the power attenuation of the signal from the transmitter to the receiver due to the propagation distance and frequency; F is the frequency (the signal transmission frequency is the same as the reception frequency); D is the distance between the transmitting and receiving antennas.

[0037] For example, an airborne short message system uses an integrated design. The transmitting signal is A1 and the receiving antenna receives the signal B1, both at 1575.42MHz. To enhance anti-interference capability, the system uses a multi-element antenna. The spacing between the elements needs to be optimized based on the signal frequency and spatial attenuation. C is 3×10 8 m / s, f is 1575.42MHz, substitute it into the formula: , D=0.0001km, F is 1575.42MHz, substitute F and D into the formula: =16.53dB. In this embodiment, the spatial attenuation is simplified to 15dB.

[0038] The multi-stage filtering and amplification module performs multi-stage filtering and amplification on the received signal according to the interference signal and spatial attenuation; connects to the antenna to be tested through a spectrum analyzer, sets scanning parameters covering the target frequency band, locates the interference direction in combination with a directional antenna, and uses high-resolution bandwidth and detection mode to capture transient or interference characteristics, and obtains the interference signal through time-frequency analysis software. Because the obtained interference signal will cause distortion of the received signal, the interference signal is filtered out.

[0039] The interference signal is filtered and amplified in multiple stages by the spatial attenuation, such as Figure 1As shown, first, a high-power tolerant filter is used to perform primary filtering on the interference signal. The filter filters out the interference signal outside the target frequency band through its highly selective suppression characteristics, and at the same time implements controllable attenuation on the signal in the transmission channel to balance the signal strength and system security; then the interference signal after the primary filtering is primary amplified, and the primary amplification is amplified by a low-noise amplifier. The low-noise amplifier controls the noise contribution while providing moderate gain to avoid nonlinear distortion of the amplifier due to signal overload. The amplifier accurately matches the input signal strength to ensure that the transmitted signal does not cause saturation, and performs basic enhancement on the weak received signal; then, the interference signal after the primary amplification is secondary filtered, and a high-power tolerant surface acoustic wave filter is used for secondary filtering. The filter uses surface acoustic wave technology to achieve higher-precision spectrum shaping, and its out-of-band suppression capability can further eliminate the residual out-of-band interference components while maintaining continuous attenuation of the transmitted signal. At this stage, a steeper The roll-off characteristic forms a clearer spectral isolation between the transmitted signal and the received signal; then, the interference signal that has passed the secondary filtering is amplified in the second stage. The secondary amplification is also amplified by a low-noise amplifier. This amplifier provides higher gain while maintaining low noise characteristics, and gradually improves the received signal strength through the multi-stage amplification cascade effect. At this stage, while continuously attenuating the transmit signal strength, the received signal is step-by-step enhanced to ensure that the two always maintain a controllable level difference in the processing path; finally, the interference signal that has passed the secondary amplification is filtered in the third stage. The filter is a cavity filter. The cavity filter uses the high Q value characteristics of the metal resonant cavity to achieve ultra-wideband out-of-band suppression, and implements ultimate attenuation on the residual transmit signal to ensure the purity of the receiving channel. While eliminating all out-of-band interference, it minimizes signal distortion through sophisticated impedance matching design; the high-power tolerance filter, low-noise amplifier, high-power tolerance surface acoustic wave filter and cavity filter are connected in series to form a circuit.

[0040] A specific example is: a satellite communication system needs to suppress the interference of the transmitting signal (marked as A1 signal) on the receiving channel (marked as B1 channel) to ensure that the B1 satellite signal (initial strength of -100dBm) is not overwhelmed at the receiving end. The system adopts a series architecture of three-stage filtering and two-stage amplification. The specific implementation is as follows: a high-power ceramic dielectric filter is selected in the primary filtering stage. The out-of-band suppression of the ceramic dielectric filter is ≥ 40dB. The initial strength of the A1 signal is 40dBm, and the attenuation of the antenna coupling path is 15dB. The strength of the A1 signal after coupling = the initial A1 signal strength - the attenuation of the antenna coupling path, P 耦 =40dBm-15dB=25dBm, A1 signal strength after primary filtering = A1 signal strength after coupling - filter out-of-band suppression, P 处滤=25dBm-40dB=-15dBm, the A1 signal strength drops to -15dBm, and the B1 signal strength remains at -100dBm because it does not enter the filter frequency band; a gallium arsenide low noise amplifier (LNA) is selected for the primary amplification stage. The gain of the gallium arsenide low noise amplifier is 20dB and the noise figure is 0.6dB. The A1 signal strength after the first stage of amplification = the A1 signal strength after filtering + the amplifier gain, PA 一放 =-15dBm+20dB=5dBm, the B1 signal strength after the first stage amplification = the initial B1 signal strength + amplifier gain, PB 一放 =-100dBm+20dB=-80dBm, A1 signal strength is 5dBm, B1 signal strength is increased to -80dBm; High-power surface acoustic wave filter is selected in the secondary filtering stage, the out-of-band suppression of high-power surface acoustic wave filter is ≥ 40dB, filter insertion loss (in-band): 3dB, A1 signal strength after secondary filtering = A1 signal strength after primary amplification - filter out-of-band suppression, PA 二滤 =5dBm-40dB=-35dBm, B1 signal strength after secondary filtering = B1 signal strength after primary amplification - filter insertion loss, PB 二滤 =-80dBm-3dB=-83dBm, the signal strength of A1 drops to -35dBm, and the signal strength of B1 drops to -83dBm; a gallium nitride low-noise amplifier is used in the secondary amplification stage. The gain of the gallium nitride low-noise amplifier is 30dB, and the noise figure is 1dB. The signal strength of A1 after secondary amplification = the signal strength of A1 after secondary filtering + the amplifier gain, PA 二放 =-35dBm+30dB=-5dBm, B1 signal strength after amplification = B1 signal strength after secondary filtering + amplifier gain, PB 二放 =-83dBm+30dB=-53dBm, A1 signal strength is -5dBm, B1 signal strength is increased to -53dBm; metal cavity filter is selected in the three-stage filtering stage, the metal cavity filter has an out-of-band suppression of ≥ 70dB, and the insertion loss (in-band) is 3dB. The A1 signal strength after the three-stage filtering = L signal strength after the second-stage amplification - filter out-of-band suppression, PA 三滤 =-5dBm-70dB=-75dBm, B1 signal strength after three-stage filtering = B1 signal strength after two-stage amplification - filter insertion loss, PB 三滤 =-53dBm-3dB=-56dBm, the A1 signal strength drops to -75dBm, and the B1 signal strength drops to -56dBm.

[0041] The simulation verification module constructs a simulation model based on the results of multi-stage filtering and amplification, verifies the results of multi-stage filtering and amplification, uses ADS to construct the simulation model, uses the constructed simulation model to simulate the parameters of each stage filter and the gain / phase curve of the amplifier respectively, ensures that the single-stage performance meets the standard, connects the filter and the amplifier according to the actual circuit topology, analyzes the frequency response characteristics after cascading, inputs A1 signal and B1 signal, observes the output signal strength of B1 channel and the interference strength of A1 signal, calculates the interference suppression ratio, and generates Figure 2 The simulation result diagram is as follows. According to the above steps S101 to S102, actual tests are performed on it. A finished antenna including multi-stage filtering and amplification suppression design is used. The frequency and power acquisition results are as follows Figure 3 As shown, compare the measured data with the simulation results. If the error is large, check whether the actual parameters of the filter are consistent with the simulation model.

[0042] The technical solution in the above-mentioned embodiment of the present application has at least the following technical effects or advantages: through structural isolation (10cm spacing) and a multi-stage filtering and amplification circuit (three-stage filtering + two-stage amplification), the transmitting signal A1 is gradually suppressed from 40dBm to -75dBm, and the receiving signal B1 is increased from -100dBm to -56dBm, ultimately making the A1 signal lower than the B1 signal, solving the problem of interference of the short message system transmitting signal on the receiving channel. The simulation and measured data verify the effectiveness of the design, achieving efficient suppression of strong interference signals and precise amplification of weak received signals.

[0043] Example 2: Based on the filter insertion loss in Example 1, each filter level will cause a certain insertion loss to the target signal (B1 signal) while suppressing the interference signal. For example, the B1 signal has a 3dB loss after the second-stage filtering. If the cumulative loss of the multi-stage filtering is too large, the B1 signal strength will be too low, affecting the receiving sensitivity and reducing the overall performance of the system. In particular, when receiving weak signals, the signal-to-noise ratio required by the system cannot be met. In this embodiment, multiple receiving channels (hierarchical reception) are used, each channel is filtered separately and the signal is merged, and the influence of the single-channel insertion loss is offset by statistical gain, such as Figure 5 shown.

[0044] S201, using a power splitter to divide the receiving end signal into multiple paths to form multiple receiving channels, and calculating the noise figure in a single channel based on the insertion loss of the filter;

[0045] Furthermore, the received signal is input to the input port of the Wilkinson power divider through the antenna. The Wilkinson power divider is a passive power distribution device used to divide the input signal into two outputs while maintaining isolation between the output ports. The two output ports of the Wilkinson power divider are respectively connected to two independent RF receiving channels, namely the first receiving channel and the second receiving channel. The first receiving channel and the second receiving channel both include the same filter, low-noise amplifier and ADC.

[0046] Calculate the insertion loss of the first receiving channel. The insertion loss in the first receiving channel mainly comes from the power splitter loss and the filter insertion loss. The noise figure of the first receiving channel is calculated based on the noise figures of the power splitter, filter, and low-noise amplifier. The formula is: ,in, is the noise figure of the first receiving channel; is the noise coefficient of the power divider; is the noise factor of the filter; is the gain of the power divider (linear value); is the gain of the filter; is the noise figure of the primary low noise amplifier; is the gain of the primary low noise amplifier; is the noise figure of the second-stage low-noise amplifier.

[0047] S202, the first receiving channel and the second receiving channel share the same local oscillator;

[0048] Specifically, the first receiving channel and the second receiving channel share the same local oscillator (LO) signal, so that the phases of the signals after frequency conversion of the first receiving channel and the second receiving channel are consistent. The local oscillator signal provides a unified carrier frequency and phase reference for the first receiving channel and the second receiving channel. Pilot signals with known frequency and phase are superimposed on the first receiving channel and the second receiving channel. Fourier transform is performed on the pilot signals of the two channels to extract their phase information, and the phase difference of the second receiving channel relative to the first receiving channel is calculated, and phase rotation compensation is performed on the second receiving channel.

[0049] S203, using a signal merging algorithm to calculate the insertion loss of multiple channels, and calculating the noise figures of multiple channels based on the noise figure within a single channel;

[0050] Specifically, in this embodiment, the multi-channel is a dual-channel. In the dual-channel receiving system, the insertion loss of the dual channels is calculated by using a merging algorithm. The merging algorithm is the maximum ratio combining (MRC) technology. MRC is a linear merging technology that maximizes the signal-to-noise ratio of the combined signal by weighted summing of the signals of each channel. If the signal-to-noise ratios of the two channels are the same, the weights are both 0.5. For two independent channels, the signal-to-noise ratio after MRC merging is the sum of the signal-to-noise ratios of each channel: SNR 合并 =SNR1+SNR2, where SNR 合并 is the combined signal-to-noise ratio, SNR1 is the signal-to-noise ratio of the first receiving channel, and SNR2 is the signal-to-noise ratio of the second receiving channel. Since SNR1=SNR2=SNR, SNR 合并 =2SNR, the formula for improving the multi-channel equivalent insertion loss is: ,in, is the equivalent insertion loss after multi-channel merging, here refers to dual channels, is the insertion loss of a single channel, is the signal-to-noise ratio gain brought by MRC combining, N=2.

[0051] After the two channels are combined by MRC, the formula for calculating the combined noise coefficient is: ,in, is the combined dual-channel noise figure, is the noise figure of the first receiving channel, and N is the number of channels.

[0052] A specific example is: the insertion loss of a single channel mainly comes from the power divider and the filter. The power divider loss is 0.5dB, and the filter loss is 3 dB. =0.5dB+3dB=3.5dB, =3.5dB-3.0dB=0.5dB, for the calculation of single-channel noise figure: ,in, =1, (power divider), =10 3 / 10 ≈2.0, (filter), =10 0.6 / 10 ≈1.15, =10 20 / 10 =100 (primary LNA), =10 1 / 10 ≈1.26, =10 30 / 10 =1000 (secondary LNA), substituting into: =1+1.122+0.336+0.0058=2.4638, which is converted into dB: NF单 =10lg(2.4638)≈3.92dB, ≈1.232, converted to dB: NF MRC =10lg (1.232)≈0.9dB.

[0053] The technical solutions in the above-mentioned embodiments of the present application have at least the following technical effects or advantages: through the dual-channel diversity reception architecture and maximum ratio combining technology, the system can significantly improve the reception performance of the B1 signal: through spatial / polarization diversity, the signal correlation is reduced and the anti-multipath fading capability is improved. The dual-channel system optimizes the single-channel 3.5 dB insertion loss to 0.5 dB through MRC combining. After the dual-channel MRC combining, the noise figure is reduced from 3.92 dB of the single channel to 0.9 dB.

[0054] Example 3: Based on the inter-symbol interference caused by the delay mismatch generated in the technical solutions of Example 1 and Example 2 above, this embodiment dynamically calibrates the delay difference to achieve signal phase alignment and improve efficiency, such as Figure 6 shown.

[0055] S301, dividing the signal from the transmitting end into real-time data and calibration data, and classifying the interference level according to the power and signal-to-noise ratio of the received signal at the receiving end;

[0056] Furthermore, real-time data is normal communication data, which contains information that needs to be transmitted in actual business. The transmitter is equipped with a special pseudo-random sequence generation module, which generates a pseudo-random sequence (PRN code) according to preset algorithms and parameters. The pseudo-random sequence is used as calibration data. The pseudo-random sequence has good autocorrelation and cross-correlation characteristics and can meet the requirements of delay calibration.

[0057] The acquisition module at the receiving end is used to measure the power of the received signal, separate the signal component and the noise component from the collected signal, and use the signal processing algorithm to obtain the signal-to-noise ratio. The power threshold is determined based on factors such as the system's performance requirements, signal transmission characteristics, and actual application scenarios. The power of the received signal measured in real time is compared with the set power threshold. If the received signal power is less than the set power threshold, the current communication environment is determined to be a low-interference scenario. In the low-interference scenario, the system inserts calibration data every 10 ms and only transmits basic timestamps to reduce power consumption while still meeting basic delay calibration requirements. If the received signal power is greater than or equal to the set power threshold, the current communication environment is determined to be a high-interference scenario. At this time, the system inserts calibration data every 2 ms. Calibration data is inserted once, and the insertion density can be dynamically adjusted according to the interference intensity to track and compensate for delay changes more promptly and ensure signal quality. The system pre-sets the signal-to-noise ratio threshold based on the actual system performance and communication environment requirements. The signal-to-noise ratio threshold is set through a large number of experiments and simulation analyses. It is used to measure the interference intensity and guide the adjustment of the calibration data insertion frequency. When the monitored signal-to-noise ratio is lower than the signal-to-noise ratio threshold, it indicates that the interference intensity has increased. The calibration data insertion control module will adjust the calibration data insertion frequency from once every 2ms to once every 1.5ms, which can track delay changes more promptly and improve the accuracy of delay calibration.

[0058] S302, performing delay matching on the first receiving channel and the second receiving channel, and obtaining a delay difference between the two channels according to the delay matching;

[0059] Specifically, delay matching is a mathematical operation used to measure the similarity between two signals. In delay estimation, by performing delay matching on the signals of the first and second receiving channels, the optimal matching position between the two signals can be found, thereby determining the delay difference. The calculation formula for delay matching is: ,in, is the delay difference; To maximize the parameter estimate, the optimal delay estimate is selected from all possible candidate delay values ​​τ, which maximizes the cross-correlation function. N is the length of the delay matching sequence, which determines the accuracy and computational complexity of the delay matching. The longer the sequence length, the more accurate the delay matching result, but the greater the computational complexity. represents the signal sequence of the first receiving channel, that is, the received PRN code sequence; τ is the delay variable, which represents the delay offset of the second receiving channel signal relative to the first receiving channel signal.

[0060] S303, compensating the second receiving channel according to the obtained delay difference;

[0061] Furthermore, according to the calculated delay difference , the delay difference is a continuous physical quantity, while digital signal processing is performed in the discrete time domain. Therefore, Δτ needs to be converted into discrete sampling points. The sampling period Ts is an important parameter in digital signal processing. It represents the time interval between two adjacent sampling points. By dividing the time delay difference Δτ by the sampling period Ts, the number of sampling points that need to be compensated can be obtained. The compensation formula for the second receiving channel is: ,in, To compensate the second receiving channel, Ts is the sampling period; is the delay difference; Indicates floor operation. The second receiving channel is aligned with the signal of the first receiving channel by performing digital delay line compensation.

[0062] The technical solutions in the above-mentioned embodiments of the present application have at least the following technical effects or advantages: by dynamically adjusting the insertion density of the calibration data, accurate dual-channel delay estimation and compensation are achieved in different interference scenarios, thereby ensuring the correct reception and processing of the signal, improving the performance and stability of the communication system, classifying the data, dynamically scheduling the calibration data, calculating the delay difference, calibrating the channel signal according to the delay difference, improving the delay compensation accuracy, and reducing the pseudorange error.

[0063] Example 4: According to the above-mentioned example 1, the interference is passively suppressed by multi-stage filtering, but the target signal is attenuated. In this example, the reverse interference signal is actively injected to actively suppress the original interference, thereby reducing the interference at the expense of the target signal strength. Figure 7 shown.

[0064] S401, generating a reverse interference signal using a frequency conversion module according to the interference signal;

[0065] Furthermore, the system uses an NVMe SSD as a storage module, which pre-stores the spectral characteristic parameters of the A1 signal. The center frequency, modulation mode and bandwidth of the A1 signal are read from the NVMe SSD, and the DPUCVD accelerator on the frequency conversion module Xilinx RFSoC ZCU111 is used to generate baseband I / Q data. The baseband generation algorithm set in the DPUCVD accelerator is used to generate I / Q data, and the generated baseband I / Q data is conjugated to generate a reverse interference signal.

[0066] S402, dividing the reverse interference signal into multiple sub-bands according to the signal frequency domain characteristics;

[0067] Specifically, the generated reverse interference signal is divided into four sub-bands according to the signal frequency domain characteristics, namely the core suppression band, the edge suppression band, the high-frequency leakage suppression band and the low-frequency leakage suppression band. For the core suppression band, the frequency band range is 1571.34MHz-1579.50MHz, the bandwidth is 8.16MHz, and the core suppression band covers the center frequency of the original interference (1575.42MHz) and ±4 MHz range, suppressing the main energy; for the edge suppression band, the frequency band range is 1563.25MHz-1571.34MHz and 1579.50MHz-1587.59MHz, with a bandwidth of 16.32MHz. The edge suppression band suppresses the transition frequency band where the main energy spreads to both sides to prevent interference tailing; for the high-frequency leakage suppression band, the frequency band range is 1587.59MHz-1595.68MHz, with a bandwidth of 8.09MHz. The high-frequency leakage suppression band targets the higher harmonics and out-of-band spurious of the A1 signal; for the low-frequency leakage suppression band, the frequency band range is 1559.16MHz-1563.25MHz, with a bandwidth of 4.09MHz. The low-frequency leakage suppression band suppresses low-frequency base noise to avoid interference lower sideband rise.

[0068] S403, obtaining the power of the core suppression band, the edge suppression band, the high-frequency leakage suppression band, and the low-frequency leakage suppression band, and calculating the reverse interference amplitude of each sub-band according to the power of each sub-band;

[0069] Furthermore, for the core suppression band, the center frequency f is determined according to the frequency range of the core suppression band. c =1575.42 MHz, use spectrum analyzer to measure the power at the center frequency to obtain the core frequency power. Based on the core frequency power, calculate the reverse interference amplitude of the core suppression band. The formula is: ,in, is the reverse interference amplitude of the core suppression band, The power of the core frequency point; for the edge suppression band, according to the frequency range of the edge suppression band, the frequency range is close to the low-frequency and high-frequency boundaries of the core suppression band, and the average power of the two edge frequency bands is monitored. ,in, It represents the average power of the low-frequency and high-frequency bands of the edge suppression band. The reverse interference amplitude of the edge suppression band is calculated based on the average power of the edge band. The formula is: ,in, is the reverse interference amplitude of the edge suppression band, is the average power of the edge band; for the high-frequency leakage suppression band, the center frequency point is determined as the high-frequency leakage peak power according to the frequency range of the high-frequency leakage suppression band, and the reverse interference amplitude of the high-frequency leakage suppression band is calculated according to the high-frequency leakage peak power: ,in, is the reverse interference amplitude of the high-frequency leakage suppression band, is the high-frequency leakage peak power; for the low-frequency leakage suppression band, the center frequency point is determined as the low-frequency base noise power according to the frequency range of the low-frequency leakage suppression band, and the reverse interference amplitude of the low-frequency leakage suppression band is calculated according to the low-frequency base noise power: ,in, is the reverse interference amplitude of the low-frequency leakage suppression band, is the low-frequency base noise power, in the above formula 、 、 These are the maximum allowable ranges.

[0070] S404, controlling the original interference according to the obtained reverse interference amplitude of each sub-band;

[0071] Specifically, for the core suppression band, the generated high-amplitude core suppression band reverse interference signal is superimposed on the main lobe of the original interference signal in anti-phase. Anti-phase superposition means that the phase of the reverse interference signal is 180 degrees different from the phase of the main lobe of the original interference signal. When the two signals meet in space, due to their opposite phases, destructive interference will occur. The result of destructive interference is the formation of a "spectral pit" on the spectrum of the core suppression band. This "spectral pit" greatly reduces the power of the main lobe of the original interference signal. By precisely controlling the amplitude and phase of the reverse interference signal, the main lobe power can be forced to be reduced by 20 dB, thereby effectively suppressing the interference energy in the core suppression band and ensuring A1 The signal main lobe energy is significantly suppressed. For the edge suppression band, a reverse interference signal is generated and applied to the transition region of the edge suppression band. Unlike the high-amplitude reverse interference of the core suppression band, the reverse interference amplitude of the edge suppression band is relatively low. The purpose is to gently suppress the interference energy in the transition region. This gentle suppression can prevent the interference energy from excessively spreading to both sides of the core suppression band, thereby maintaining the relative stability of the signal spectrum. By gently suppressing the transition region, the transition band power of the edge suppression band is attenuated by ≥15dB. This attenuation effect effectively reduces the distribution of interference energy in the transition region, avoids the occurrence of interference "shoulder" rise, and further optimizes the spectral characteristics of the signal. For the high-frequency leakage suppression band, the generated reverse interference signal is a narrowband signal targeting high-frequency harmonics. This narrowband reverse signal is superimposed with the high-frequency harmonics in the original interference signal. Since the frequency and phase of the reverse interference signal match the high-frequency harmonics, destructive interference will occur between them. Through the destructive interference between the narrowband reverse signal and the high-frequency harmonics, the high-frequency harmonics can be effectively eliminated, making the spurious suppression ratio ≥30 dBc. This means that the spurious signal power within the high-frequency leakage suppression band is reduced by at least 30 dB relative to the carrier signal power, significantly improving the signal quality and purity. For the low-frequency leakage suppression band, the generated low-amplitude reverse interference signal is applied to the low-frequency leakage suppression band. Due to the low amplitude of the reverse interference signal, it can interact with the base noise in the original interference signal in a gentle manner. Through the interaction between the low-amplitude reverse interference and the base noise, the base noise power is reduced by ≥10 dB. This reduction in base noise directly improves the sensitivity of the receiver, enabling the receiver to detect weaker signals. For example, the receiver sensitivity can be improved to -130 dBm, which means that the receiver can operate normally at lower signal power levels, improving the performance and reliability of the communication system.

[0072] The technical solution in the above-mentioned embodiment of the present application has at least the following technical effects or advantages: by actively injecting a reverse interference signal, the system can achieve precise suppression and effective control of a specific interference signal, improve signal quality, optimize spectrum characteristics, reduce the impact of high-frequency leakage and low-frequency base noise on the communication system, thereby significantly improving the performance and reliability of the communication system, and dividing the generated reverse interference signal into four sub-bands: a core suppression band, an edge suppression band, a high-frequency leakage suppression band, and a low-frequency leakage suppression band according to the signal frequency domain characteristics. Each sub-band is designed for different interference characteristics to achieve precise control of the interference signal.

[0073] The foregoing description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Various modifications and variations are readily apparent to those skilled in the art. Any modifications, equivalent substitutions, or improvements made within the spirit and principles of the present invention shall be included within the scope of protection of the present invention.

Claims

1. A multi-stage filter amplifier suppressor, characterized in that: include: Signal acquisition module, spatial attenuation and calculation module, multi-stage filtering and amplification module and simulation verification module; The signal acquisition module obtains the frequency of the received signal and collects the interference signal, calculates the vibrator distribution spacing according to the frequency of the received signal, and converts the vibrator distribution spacing into the spacing between the transmitting and receiving antennas. The spatial attenuation and calculation module calculates the spatial attenuation according to the spacing between the transmitting and receiving antennas; the multi-stage filtering and amplification module performs multi-stage filtering and amplification on the received signal according to the interference signal and the spatial attenuation, and processes the insertion loss generated by the multi-stage filtering and amplification; the simulation verification module constructs a simulation model based on the results of the multi-stage filtering and amplification, and verifies the results of the multi-stage filtering and amplification; S401, generating a reverse interference signal using a frequency conversion module according to the interference signal; S402, dividing the generated reverse interference signal into four sub-bands according to the signal frequency domain characteristics, namely a core suppression band, an edge suppression band, a high-frequency leakage suppression band, and a low-frequency leakage suppression band; S403: Obtain the power of the four sub-bands, calculate the reverse interference amplitude of each sub-band according to the power of each sub-band, and for the core suppression band, determine its center frequency f according to the frequency range of the core suppression band. c =1575.42 MHz. Use spectrum analyzer to measure the power at the center frequency to obtain the core frequency power. Calculate the reverse interference amplitude of the core suppression band based on the core frequency power. The formula is: ,in, is the reverse interference amplitude of the core suppression band, The power of the core frequency point; for the edge suppression band, according to the frequency range of the edge suppression band, the frequency range is close to the low-frequency and high-frequency boundaries of the core suppression band, and the average power of the two edge frequency bands is monitored. ,in, is the average power of the edge band, and It represents the average power of the low-frequency and high-frequency bands of the edge suppression band. The reverse interference amplitude of the edge suppression band is calculated based on the average power of the edge band. The formula is: ,in, is the reverse interference amplitude of the edge suppression band, is the average power of the edge band; for the high-frequency leakage suppression band, the center frequency point is determined as the high-frequency leakage peak power according to the frequency range of the high-frequency leakage suppression band, and the reverse interference amplitude of the high-frequency leakage suppression band is calculated according to the high-frequency leakage peak power: ,in, is the reverse interference amplitude of the high-frequency leakage suppression band, is the high-frequency leakage peak power; for the low-frequency leakage suppression band, the center frequency point is determined as the low-frequency base noise power according to the frequency range of the low-frequency leakage suppression band, and the reverse interference amplitude of the low-frequency leakage suppression band is calculated according to the low-frequency base noise power: ,in, is the reverse interference amplitude of the low-frequency leakage suppression band, is the low-frequency base noise power, in the above formula All are the maximum allowable ranges; S404: Control the original interference according to the obtained reverse interference amplitude of each sub-band.

2. A multi-stage filter amplifier suppressor according to claim 1, characterized in that: The vibrator distribution spacing is calculated based on the frequency of the received signal. The formula is: , where L is the spacing between the oscillators; f is the frequency of the received signal; and C is the speed of light, which is 3×108 m / s.

3. A multi-stage filter amplifier suppressor as claimed in claim 2, characterized in that: The unit of the vibrator distribution spacing is converted into kilometers and used as the spacing between the transmitting and receiving antennas. The spatial attenuation is calculated based on the spacing between the transmitting and receiving antennas. The formula is: ,in, is the spatial attenuation; 32.45 is a constant term, which represents the parameter of the automatic spatial path loss model, that is, the power attenuation of the signal from the transmitter to the receiver due to the propagation distance and frequency; F is the frequency; D is the distance between the transmitting and receiving antennas.

4. The multi-stage filter amplifier suppressor according to claim 1, characterized in that: The interference signal is subjected to multi-stage filtering and amplification through spatial attenuation. First, a filter is used to perform primary filtering on the interference signal, and the signal at the transmitting end is attenuated at the same time; then, the interference signal that has passed the primary filtering is subjected to primary amplification, and the primary amplification is amplified by a low-noise amplifier; then, the interference signal that has passed the primary amplification is subjected to secondary filtering; then, the interference signal that has passed the secondary filtering is subjected to secondary amplification, and the secondary amplification is also amplified by a low-noise amplifier; finally, the interference signal that has passed the secondary amplification is subjected to tertiary filtering.

5. The multi-stage filter amplifier suppressor according to claim 1, characterized in that: The multi-stage filtering and amplification module is used to process insertion loss, specifically: S201, using a power splitter to divide the receiving end signal into multiple paths to form multiple receiving channels, calculating the noise figure within a single channel based on the insertion loss of the filter, and calculating the delay difference; S202, multiple receiving channels share the same local oscillator; S203 , using a signal merging algorithm to calculate the insertion loss of the multiple channels, and calculating the noise coefficient of the multiple channels based on the noise coefficient within a single channel.

6. The multi-stage filter amplifier suppressor according to claim 5, characterized in that: The noise figure of a single channel in a multi-channel system is calculated using the following formula: ,in, is the noise figure of a single channel; is the noise coefficient of the power divider; is the noise factor of the filter; is the gain of the power divider; is the gain of the filter; is the noise figure of the primary low noise amplifier; is the gain of the primary low noise amplifier; is the noise figure of the two-stage low-noise amplifier. The formula for calculating the noise figure of multiple channels is: ,in, is the noise figure of the multi-channel, is the noise coefficient of a single channel, N is the number of channels, and the noise coefficient of a single channel is the noise coefficient of the first receiving channel or the noise coefficient of the second receiving channel.

7. The multi-stage filter amplifier suppressor according to claim 5, characterized in that: The calculation formula for multi-channel insertion loss is: ,in, is the equivalent insertion loss after multi-channel merging, is the insertion loss of a single channel, is the signal-to-noise ratio gain brought by MRC merging, and N is the number of channels.

8. The multi-stage filter amplifier suppressor according to claim 1, characterized in that: The steps of calculating the time delay difference by the spatial attenuation and calculation module are as follows: S301, dividing the signal from the transmitting end into real-time data and calibration data, and classifying the interference level according to the power and signal-to-noise ratio of the received signal at the receiving end; S302, performing delay matching on multiple channels, and obtaining delay differences of the multiple channels according to the delay matching; S303: Compensate for multiple channels according to the obtained delay difference.

9. The multi-stage filter amplifier suppressor according to claim 8, characterized in that: The real-time data is the communication data. The transmitter is equipped with a pseudo-random sequence generation module. The pseudo-random sequence generation module generates a pseudo-random sequence according to a preset algorithm and parameters. The pseudo-random sequence is used as calibration data.

Images