Active power decoupling type power converter and control method thereof

Abstract

The invention discloses an active power decoupling type power converter which is provided with an active power decoupling branch, one end of the active power decoupling branch is connected to the negative end of a direct current bus, and the other end of the active power decoupling branch is connected with a load neutral point and a filtering neutral point; the invention also discloses a control method of the power converter. The control method comprises the following steps: obtaining a first input quantity according to a difference value between an input side power actual value and an input side power reference value; obtaining a second input quantity according to the harmonic component of the load side power of the power converter; taking the sum of the first input quantity and the second input quantity as a reference value of an active power decoupling branch current inner loop, obtaining a decoupling inductance voltage reference value according to a difference value between the reference value and a decoupling inductance current actual value, taking the decoupling inductance voltage reference value as a common-mode value of first to fourth bridge arm modulation waves in an inverter circuit at the output side of the power converter, adding the common-mode value with the modulation waves, and performing modulation to obtain a decoupling inductance voltage reference value; and switching signals of four bridge arms of the inverter circuit are obtained. According to the invention, the problem that the pulsating power generated by the load side is coupled to the input side when the power converter is under the unbalanced load condition can be solved.

Description

Technical Field

[0001] The present invention belongs to the technical field of power conversion, and relates to an active power decoupling technology for power converters, and particularly to an active power decoupling type power converter and its control method. Background Art

[0002] In recent years, with the rapid development of power electronics technology and the continuous expansion of application fields, power converters play an increasingly important role in new energy, electric vehicles, smart grids, industrial automation and other fields. Traditional power converters mostly adopt passive or single control strategies to achieve energy conversion and regulation. However, with the improvement of system integration and the increasing influence of external disturbances such as grid fluctuations and load changes, traditional solutions can no longer meet the high requirements of modern applications in terms of dynamic response, energy transfer efficiency and system stability. Especially in multi-port systems or composite energy flow management systems, the interaction between multiple sources and multiple loads often leads to the mutual coupling between different power branches, resulting in a decline in the dynamic performance of the system and making the design of control strategies more complex. Therefore, it is particularly urgent to develop a new type of power converter with high efficiency, good robustness and capable of realizing power decoupling control.

[0003] Traditional matrix converters can achieve bidirectional energy conversion between direct current and alternating current without using a traditional DC intermediate link; matrix converters can generate almost perfect sine wave input and output current and voltage waveforms with low harmonic content, thus improving the power quality; they can adjust the input power factor within a wide range, enabling them to flexibly adjust the magnitude and phase of the power factor according to different loads and operating conditions. Due to the many advantages of matrix converters, they have wide applicability in various application scenarios. However, traditional matrix converters also have some problems that limit their applications, mainly reflected in the following aspects: First, the number of switching devices is large. Both direct matrix converters and bipolar matrix converters require 18 bidirectional switches, while back-to-back converters only require 12. Therefore, some scholars have proposed sparse matrix converters and ultra-sparse matrix converters, etc., but at the expense of the ability of four-quadrant operation.

[0004] Second, input-output power coupling. Under unbalanced load conditions, the three-phase output voltage should be sinusoidal and symmetric as the control target. At this time, the total three-phase load power will contain second-harmonic fluctuating power, and this fluctuating power will be directly reflected in the input power. Given that the three-phase input voltage is sinusoidal and symmetric, the fluctuating power will cause the input current to distort and the harmonic content to exceed the grid standard. The input-output power coupling problem is caused by the characteristic of matrix converters without DC energy storage elements, and there is currently no effective solution.

[0005] Active power decoupling technology handles power fluctuations by adding active switches and small-capacity energy storage elements to reduce or replace large-capacity passive power decoupling elements. Conventional active power decoupling usually requires the addition of active switching devices and still needs to be improved. Summary of the Invention

[0006] The object of the present invention is to provide an active power decoupling type power converter and a control method thereof, so as to solve the problem that the pulsating power generated on the load side of the power converter is coupled to the input side under unbalanced load conditions.

[0007] In order to achieve the above object, the solution of the present invention is: An active power decoupling power converter includes an input-side rectifier circuit, a capacitor parallel circuit, an output-side inverter circuit, a three-phase load circuit, and a three-phase filter circuit. The input-side rectifier circuit is a three-phase three-bridge-arm structure, and the output-side inverter circuit is a three-phase four-bridge-arm structure. The DC positive terminal of the input-side rectifier circuit, the DC positive terminal of the output-side inverter circuit, and the positive terminal of the capacitor parallel circuit are all connected to the DC bus positive terminal. The DC negative terminal of the input-side rectifier circuit, the DC negative terminal of the output-side inverter circuit, and the negative terminal of the capacitor parallel circuit are all connected to the DC bus negative terminal. The first to third bridge arms in the output-side inverter circuit are respectively connected to one end of the three-phase load circuit and the three-phase filter circuit, the other end of the three-phase load circuit is short-circuited to the load neutral point, the other end of the three-phase filter circuit is short-circuited to the filter neutral point, and the fourth bridge arm in the output-side inverter circuit is connected to the load neutral point and the filter neutral point; and an active power decoupling branch is also included, one end of the active power decoupling branch is connected to the negative end of the DC bus, and the other end of the active power decoupling branch is respectively connected to the load neutral point and the filter neutral point.

[0008] The active power decoupling branch includes a filter inductor and a filter capacitor connected in series.

[0009] As mentioned above, a control method of an active power decoupling type power converter includes: Get the actual power value of the power converter input side P in , according to the actual value of the input side power P in and input side power reference value P in *The difference is the first input i Lp *; Get the load side power of the power converter P load , according to the load side power of the power converter P load The harmonic component of the feedforward compensation is used as the second inputi Le *; Take the sum of the first input quantity i Lp * and the second input quantity i Le * as the reference value of the inner loop of the active power decoupling branch current, i Ln *; according to the reference value i Ln * and the actual value of the decoupling inductor current i Ln to obtain the reference value of the decoupling inductor voltage u Ln *; Use the reference value of the decoupling inductor voltage u Ln * as the common mode value of the modulation waves of the first to fourth bridge arms in the inverter circuit on the output side of the power converter, and modulate the sum of the common mode value and the modulation wave to obtain the switching signals of the first to fourth bridge arms.

[0010] Among them, obtaining the actual value of the power on the input side of the power converter P in , including, Obtain the voltage amplitude V x and current amplitude I x of each phase on the input side, x = a, b, c; According to the voltage amplitude and current amplitude, calculate the actual value of the active power on the input side P in ,

[0011] Among them, θ x is the phase difference between the voltage and current of each phase on the input side.

[0012] Among them, according to the actual value of the power on the input side P in and the reference value of the power on the input side P in * to obtain the first input quantity i Lp , including, Send the difference between the actual value of the power on the input side P in and the reference value of the power on the input side P in * into the parallel resonance controller to obtain the first input quantity i Lp*; among them, the transfer function of the parallel resonance controller is,

[0013] where G R_2ω is the transfer function of the second harmonic component with a resonance frequency of 2 ω n ; G R_4ω is the transfer function of the fourth harmonic component with a resonance frequency of 4 ω n ; K r is the resonance gain, ω n is the fundamental angular frequency.

[0014] Among them, obtaining the power on the load side of the power converter P load , includes, obtaining the voltage amplitude V o_x and current amplitude I o_x of each phase on the load side, x = a, b, c; calculating the power on the load side P load based on the voltage amplitude and current amplitude,

[0015] where, θ o_x is the phase difference between the voltage and current of each phase on the load side.

[0016] Among them, obtaining the feedforward compensation amount based on the harmonic components of the power P load on the load side of the power converter as the second input quantity i Le *, includes, using a notch filter to extract the harmonic components of the power P load on the load side of the power converter to obtain the DC component of the load side power P load_dc , and the transfer function of the notch filter is:

[0017] where, ω n is the fundamental angular frequency; subtracting the load side power P load from the DC component of the load side power P load_dc to obtain the second harmonic component of the load side powerP load_2ω ; According to the second harmonic component of the load-side power P load_2ω and the input-side voltage amplitude u sm calculate the feedforward compensation amount, which is the second input quantity i Le *,

[0018] wherein, the input-side voltage amplitude u sm is calculated according to the following formula wherein, V α , V β are the α-axis and β-axis components of the three-phase input voltage on the input side respectively, U sa , U sb , U sc are the amplitudes of the phase voltages of the three-phase input voltages respectively.

[0019] wherein, according to the reference value i Ln * and the actual value of the decoupling inductor current i Ln get the reference value of the decoupling inductor voltage u Ln *, including, send the difference between the reference value i Ln * and the actual value of the decoupling inductor current i Ln into a proportional-integral multi-resonant controller to obtain the reference value of the decoupling inductor voltage u Ln * ; wherein, the transfer function of the proportional-integral multi-resonant controller is

[0020] wherein, K p , K i , K r are the proportional gain, integral gain and resonant gain respectively, ω r is the cut-off frequency, ω n is the fundamental angular frequency.

[0021] wherein, taking the decoupled inductor voltage reference value u Ln * as the common mode value of the modulation waves of the first to fourth bridge arms in the inverter circuit on the output side of the power converter, modulating the sum of the common mode value and the modulation waves to obtain the switching signals of the first to fourth bridge arms, including acquiring the three-phase voltage on the load side u o and the three-phase current of the filter inductor i L , and obtaining the α, β, and 0-axis components u o of the three-phase load voltage u o_α 、 u o_β 、 u o_0 and the α, β, and 0-axis components i L of the three-phase current of the filter inductor i L_α 、 i L_β 、 i L_0 ; setting the three-phase load voltage reference value as u a 、 u b 、 u c , obtaining the α-axis and β-axis components of the three-phase load voltage reference value through Clark transformation u α 、 u β , and setting the reference value of the zero-axis component of the load voltage u 0 to be 0; subtracting the α-axis, β-axis, and 0-axis components u α 、 u β 、 u 0 of the three-phase load voltage reference value from the α-axis, β-axis, and 0-axis components u o_α 、 u o_β 、 u o_0 of the actual value of the load voltage respectively, and inputting the difference into the load voltage outer-loop QPR controller for regulation to output the reference values i L_α *、 i L_β *、i L_0 *; Among them, the transfer function of the load voltage outer-loop QPR controller is

[0022] Among them, K up and K ur are the proportional gain and resonance gain of the load voltage outer-loop QPR controller respectively; ω uc is the cut-off frequency, ω n is the fundamental angular frequency; Subtract the α-axis, β-axis, and 0-axis components of the actual value of the filter inductor current from the reference values i L_α *, i L_β *, i L_0 * of the α-axis, β-axis, and 0-axis filter inductor current inner-loop respectively, and input the differences into the filter inductor current inner-loop QPR controller for adjustment. The outputs corresponding to the α-axis and β-axis filter inductor current inner-loops are subjected to Clark inverse transformation to obtain the modulation voltages i L_α and i L_β and i L_0 of the three-phase load. The output of the 0-axis filter inductor current inner-loop is used as the second component of the modulation voltage of the fourth bridge arm u am and u bm and u cm ; Among them, the transfer function of the filter inductor current inner-loop QPR controller is u 0m ; Among them,

[0023] Among them, K ip and K ir are the proportional gain and resonance gain of the filter inductor current inner-loop QPR controller respectively; ω ic is the cut-off frequency; Add the common-mode voltage 0.75 to the three-phase load modulation voltages u am and u bm and u cm simultaneously respectively usm 1. The first component of the modulation voltage of the fourth bridge arm u Ln * , to obtain the modulation voltages of the three-phase bridge arms u AO . u BO . u CO , add the first component of the modulation voltage of the fourth bridge arm u Ln * and the second component of the modulation voltage of the fourth bridge arm u 0m , then add the common-mode voltage 0.75 u sm , to obtain the modulation voltage of the fourth bridge arm u NO .

[0024] Each of the above modulation voltages is modulated to obtain the switching signals of the first to fourth bridge arms of the output-side inverter circuit.

[0025] After adopting the above scheme, the beneficial effects of the present invention are as follows: (1) In the power converter of the present invention, a multiplexed fourth bridge arm is adopted, reducing the number of switches, cost and losses; (2) For the power coupling phenomenon that the pulsating power caused by the unbalanced load is coupled to the input side, the present invention adopts an active power decoupling control method combining the double-loop control of the fourth bridge arm multiplexed topology structure with the input power outer loop - decoupling branch current inner loop and feed-forward control to suppress it, and fully suppresses the second-harmonic and fourth-harmonic pulsating power coupled to the input side; (3) For the fourth bridge arm of the inverter stage, the present invention adopts an active power decoupling control method combining the double-loop control of the input power outer loop - decoupling branch current inner loop and feed-forward control, and uses the decoupling branch to generate pulsating power to compensate for the pulsating power generated under unbalanced conditions; for the other three bridge arms of the inverter stage, the present invention adopts a voltage-current double-loop control strategy based on the stationary coordinate system to achieve the control goal of three-phase sinusoidal balance of the output voltage; for the rectifier stage, the present invention adopts a control strategy of uncontrolled voltage maximum bridge arm and harmonic injection of the voltage intermediate value bridge arm to achieve the control goal of three-phase sinusoidal balance of the input current; the present invention solves the problem of waveform distortion of the power converter under unbalanced conditions and improves the working performance of the power converter under unbalanced conditions. Description of the Drawings

[0026] Figure 1 is the topology structure of the active power decoupling type power converter of the present invention; Figure 2 is the active power decoupling control block diagram of the present invention; Figure 3 is the input-side power waveform diagram before active power decoupling of the present invention; Figure 4 is the input-side power waveform diagram after active power decoupling of the present invention; Figure 5 is the input-side power outer-loop control block diagram of the present invention; Figure 6 is the control block diagram of the load fluctuation power feed-forward compensation method of the present invention; Figure 7 is the power decoupling branch current inner-loop control block diagram of the present invention; Figure 8 is the control block diagram of the voltage-current double-loop control strategy in the stationary coordinate system of the present invention; Figure 9 is the control block diagram of the uncontrolled voltage extreme bridge arm and the harmonic injection of the voltage intermediate bridge arm of the present invention. Detailed implementation manners

[0027] To make the objectives, technical solutions and advantages of the embodiments of the present invention clearer, the technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are some, but not all, of the embodiments of the present invention. All other embodiments obtained by those skilled in the art based on the embodiments of the present invention without creative efforts shall fall within the protection scope of the present invention.

[0028] This embodiment provides an active power decoupling type power converter, the topological structure of which is as Figure 1 shown. The input side is a common three-phase rectifier bridge structure, and a DC bus capacitor is connected in parallel between the input side and the load side C dc . The topological structure of the inverter stage is based on the traditional three-phase four-leg inverter stage topological structure. The midpoint of the fourth leg is connected to the neutral point of the three-phase filter capacitor and the neutral point N of the load together. Through this branch, the fourth leg participates in controlling the neutral point potential. A branch composed of a filter inductor L n and a filter capacitor C n connected in series has its two ends respectively connected to the negative bus end of the fourth leg and the load neutral point. This branch is the decoupling branch for active power decoupling.

[0029] The present invention also provides a control method for an active power decoupling type power converter, the control block diagram of which is as Figure 2 shown, and includes the following steps: Step S1.1 Input power outer loop. Calculate the actual value of the input-side power according to the formula P in , and subtract the actual value from the set input-side power reference value P in . The difference is used as the input of the power outer loop, and the output of the power outer loop P in * is one of the input quantities of the active power decoupling branch current inner loop. i Lp

[0030] Step S1.2 Load fluctuation power feedforward compensation. Calculate the load power using a mathematical formula P load , extract the harmonic components through a notch filter, and then calculate the feedforward compensation amount according to the harmonic components i Le *, which is used as another input quantity of the active power decoupling branch current inner loop.

[0031] Step S1.3 Active power decoupling branch current inner loop. Add the output of the input-side power outer loop obtained in Step S1.1 i Lp * to the feedforward compensation amount obtained in Step S1.2 i Le *. The sum obtained is used as the reference value of the active power decoupling branch current inner loop i Ln *. Sample the actual value of the decoupling inductor current i Ln , subtract the actual value from the reference value i Ln * i Ln . After the combined action of the decoupling branch current inner loop and the sequence component control, the reference value of the decoupling inductor voltage is obtained u Ln *.

[0032] Step S1.4 Switching signal modulation. Adopt the voltage-current double-loop control strategy in the stationary coordinate system, and use the reference value of the decoupling inductor voltage u Ln * as the common-mode value of the modulation waves of the four bridge arms. Add it to the modulation waves of the four bridge arms obtained by the unbalanced load control method respectively, and then through the modulation action, the switching signals of the four bridge arms are obtained. The waveforms of the input-side power before and after implementing this active power decoupling control method are shown in Figure 3 and Figure 4 .

[0033] The control block diagram of the input-side power outer loop control disclosed in the present invention is shown in Figure 5 and includes the following steps: ​Step S2.1 Calculate the actual value of the active power on the input side P in :

[0034] where x =a, b, c. V x and I x respectively represent the voltage amplitude and current amplitude of a certain phase on the input side, θ x represents the phase difference between the voltage and current of the certain phase.

[0035] Step S2.2 Subtract the actual value of the input-side power calculated in Step S1.1 P in * from the set input-side power reference value P in to obtain the difference between the two.

[0036] Step S2.3 Use the difference obtained in Step S2.2 as the input of the parallel resonance controller G r (s), and the transfer function of the parallel resonance controller is:[[]]

[0037] where G R_2ω is the transfer function of the double-frequency component with a resonance frequency of 2 ω n , and G R_4ω is the transfer function of the quadruple-frequency component with a resonance frequency of 4 ω n ; K r is the resonance gain, ω n is the fundamental angular frequency.

[0038] Step S2.4 Use the output of the parallel resonance controller i Lp * as the first input quantity of the inner loop of the decoupling branch current.

[0039] The control block diagram of the load fluctuation power feedforward compensation method provided by the present invention is as shown in Figure 6 and includes the following steps: Step S3.1 Calculate the load-side power P load :

[0040] where x = a, b, c. Vo_x and I o_x respectively represent the voltage amplitude and current amplitude of a certain phase on the load side, and cos θ o_x represents the phase difference between the voltage and current of a certain phase.

[0041] Step S3.2 Use the load-side power P load as the input of the notch filter, and the output of the notch filter is the DC component of the load-side power P load_dc , and the transfer function of the notch filter is:

[0042] where ω n is the fundamental angular frequency.

[0043] Step S3.3 Subtract the load-side power P load obtained in Step S3.1 from the DC component of the load-side power P load_dc obtained in Step S3.2 to get the second harmonic component of the load-side power P load_2ω .

[0044] Step S3.4 Sample the three-phase input voltages u sa , u sb , u sc , and calculate the input-side voltage amplitude u sm :

[0045] In the formula, U sa , U sb , U sc respectively represent the amplitudes of the voltages of each phase of the three-phase input voltage, V α , V β respectively represent the α-axis and β-axis components of the three-phase input voltage, u sm represents the amplitude of the input voltage.

[0046] Step S3.5 According to the second harmonic component of the load-side power P load_2ωand the input voltage amplitude obtained in step S3.4 u sm Calculate the second input quantity of the current inner loop i Le *:

[0047] The control block diagram of the current inner loop of the power decoupling branch disclosed in the present invention is as Figure 7 shown, and includes the following steps: In step S4.1, add the first input quantity obtained in step S1.4 i Lp * and the second input quantity obtained in step S3.5 i Le * to be used as the reference value of the current inner loop of the active power decoupling branch i Ln *.

[0048] In step S4.2, sample the actual value of the current inner loop of the active power decoupling branch i Ln , and subtract the actual value from the reference value of the current inner loop of the active power decoupling branch obtained in step S4.1 i Ln * as the input of the proportional-integral multi-resonant controller G i Ln (s). The transfer function of the controller is: c wherein,

[0049] where, K p 、 K i 、 K r are the proportional gain, integral gain and resonant gain respectively, ω r is the cut-off frequency.

[0050] The output of the proportional-integral multi-resonant controller in step S4.3 is the reference value of the decoupling inductor voltage and serves as the first component of the modulation voltage of the fourth leg u Ln * .

[0051] The control block diagram of the voltage-current double-loop control strategy in the stationary coordinate system provided by the present invention is as Figure 8 shown, and includes the following steps: In step S5.1, sample the three-phase voltage of the load u o and the three-phase current of the filter inductor iL , the three-phase load voltage is obtained by Clark transformation u o for the α, β, and 0-axis components u o_α and u o_β and u o_0 and the three-phase current of the filter inductor i L for the α, β, and 0-axis components i L_α and i L_β and i L_0 .

[0052] Step S5.2 Set the reference value of the three-phase load voltage as u a and u b and u c , and obtain the α-axis and β-axis components of the reference value of the three-phase load voltage through Clark transformation u α and u β , and set the reference value of the zero-axis component of the load voltage u 0 to be 0.

[0053] Step S5.3 Use the α-axis, β-axis, and 0-axis components of the reference value of the three-phase load voltage u α and u β and u 0 to subtract the α-axis, β-axis, and 0-axis components of the actual value of the load voltage u o_α and u o_β and u o_0 respectively. The difference is used as the input of the outer-loop QPR controller of the load voltage. After being adjusted by the controller, the output quantities are used as the reference values of the inner loops of the α-axis, β-axis, and 0-axis filter inductor currents i L_α *, i L_β *, i L_0 *. The transfer function of the outer-loop QPR controller G V is:

[0054] In the formula, K up and Kur are the proportional gain and the resonant gain of the load voltage outer-loop QPR controller, respectively; ω uc is the cut-off frequency.

[0055] Step S5.4 Subtract the α-axis, β-axis, and 0-axis components of the actual filter inductor current from the reference values of the α-axis, β-axis, and 0-axis filter inductor current inner loops obtained in step S5.3 i L_α *, i L_β *, i L_0 *, respectively, and use the differences as the inputs of the filter inductor current inner-loop QPR controller. The transfer function of the filter inductor current inner-loop QPR controller G i L_α , i L_β , i L_0 , respectively. The difference is used as the input of the filter inductor current inner-loop QPR controller. The transfer function of the filter inductor current inner-loop QPR controller G i is:

[0056] In the formula, K ip , K ir are the proportional gain and the resonant gain of the filter inductor current inner-loop QPR controller, respectively; ω ic is the cut-off frequency.

[0057] Step S5.5 The outputs of the α-axis and β-axis filter inductor current inner loops are subjected to Clark inverse transformation to obtain the modulation voltages of the three-phase load u am , u bm , u cm . The output of the 0-axis filter inductor current inner loop is used as the second component of the modulation voltage of the fourth bridge arm u 0m .

[0058] Step S5.6 Add the common-mode voltage 0.75 u am , u bm , u cm to the modulation voltages of the three-phase load u sm and the first component of the modulation voltage of the fourth bridge arm obtained in step S1.3 u Ln * , respectively, to obtain the modulation voltages of the three-phase bridge arms in the inverteru AO and u BO and u CO , add the first component of the modulation voltage of the fourth bridge arm u Ln * and the second component of the modulation voltage of the fourth bridge arm u 0m , then add the common-mode voltage 0.75 u sm , to obtain the modulation voltage of the fourth bridge arm u NO .

[0059]

[0060] The modulation voltage can be modulated to obtain the switching signals of the four bridge arms of the inverter.

[0061] The control block diagram of the control strategy of uncontrolled voltage extreme value bridge arm and harmonic injection control of voltage intermediate value bridge arm disclosed by the present invention is as shown in Figure 9 and includes the following steps: Step S6.1 Sample the three-phase voltage on the input side u s and the three-phase current i s .

[0062] Step S6.2 Distinguish the voltage extreme value bridge arm and the voltage intermediate value bridge arm through the angular frequency of the three-phase voltage on the input side. Among them, the voltage extreme value bridge arm is the power frequency bridge arm, the phases with the maximum and minimum instantaneous values of the voltage intermediate value input voltage are uncontrolled, the voltage intermediate value bridge arm injects quasi-third harmonic current, and the switching signals of the two voltage extreme value bridge arms are obtained according to the look-up table method.

[0063] Step S6.3 Calculate the reference value of the quasi-third harmonic current i y *, subtract the actual value of the current of the phase connected to the voltage intermediate value bridge arm i y , through the action of the PI regulator, obtain the switching signal of the voltage intermediate value bridge arm (i.e., the high-frequency bridge arm) by the PWM method, and realize the control method of the rectifier stage.

[0064] Those skilled in the art should understand that the embodiments of the present invention can be provided as a method, a system, or a computer program product. Therefore, the present invention can take the form of a complete hardware embodiment, a complete software embodiment, or an embodiment combining software and hardware aspects. Moreover, the present invention can take the form of a computer program product implemented on one or more computer-usable storage media (including but not limited to disk memory, CD-ROM, optical memory, etc.) that contain computer-usable program code. The solutions in the embodiments of the present invention can be implemented in various computer languages. For example, object-oriented programming languages such as Java and interpreted scripting languages such as JavaScript, etc.

[0065] The present invention is described with reference to the flowcharts and / or block diagrams of methods, apparatuses (systems), and computer program products according to the embodiments of the present invention. It should be understood that each flow and / or block in the flowchart and / or block diagram, as well as the combination of flows and / or blocks in the flowchart and / or block diagram, can be implemented by computer program instructions. These computer program instructions can be provided to the processor of a general-purpose computer, a special-purpose computer, an embedded processor, or other programmable data processing devices to generate a machine, such that the instructions executed by the processor of the computer or other programmable data processing devices generate means for implementing the functions specified in Figure 1 one flow or multiple flows and / or blocks Figure 1 one block or multiple blocks.

[0066] These computer program instructions can also be stored in a computer-readable memory that can direct a computer or other programmable data processing devices to work in a specific manner, such that the instructions stored in the computer-readable memory generate a manufactured article including instruction means that implement the functions specified in Figure 1 one flow or multiple flows and / or blocks Figure 1 one block or multiple blocks.

[0067] These computer program instructions can also be loaded onto a computer or other programmable data processing devices, such that a series of operation steps are executed on the computer or other programmable devices to generate a computer-implemented process. Thus, the instructions executed on the computer or other programmable devices provide steps for implementing the functions specified in Figure 1 one flow or multiple flows and / or blocks Figure 1 one block or multiple blocks.

[0068] Although the preferred embodiments of the present invention have been described, those skilled in the art can make additional changes and modifications once they learn the basic creative concepts. Therefore, the appended claims are intended to be construed as including the preferred embodiments and all changes and modifications that fall within the scope of the present invention.

[0069] Obviously, those skilled in the art can make various changes and modifications to the present invention without departing from the spirit and scope of the present invention. Thus, if these modifications and variations of the present invention fall within the scope of the claims of the present invention and their equivalent technologies, the present invention is also intended to include these modifications and variations.

Claims

1. An active power decoupling type power converter, comprising an input side rectifying circuit, a capacitor parallel circuit, an output side inverting circuit, a three-phase load circuit and a three-phase filtering circuit, wherein, The input - side rectifier circuit has a three - phase three - leg structure, and the output - side inverter circuit has a three - phase four - leg structure. The positive DC terminal of the input - side rectifier circuit, the positive DC terminal of the output - side inverter circuit, and the positive terminal of the capacitor parallel circuit are all connected to the positive terminal of the DC bus. The negative DC terminal of the input - side rectifier circuit, the negative DC terminal of the output - side inverter circuit, and the negative terminal of the capacitor parallel circuit are all connected to the negative terminal of the DC bus. The first to third legs in the output - side inverter circuit are respectively connected to one end of the three - phase load circuit and the three - phase filter circuit. The other end of the three - phase load circuit is short - circuited to form a load neutral point, and the other end of the three - phase filter circuit is short - circuited to form a filter neutral point. The fourth leg in the output - side inverter circuit, the load neutral point, and the filter neutral point are connected. It is characterized in that: an active power decoupling branch is further included. One end of the active power decoupling branch is connected to the negative terminal of the DC bus, and the other end of the active power decoupling branch is respectively connected to the load neutral point and the filter neutral point.

2. The active power decoupling type power converter according to claim 1, wherein: The active power decoupling branch includes a filter inductor and a filter capacitor connected in series.

3. The control method of an active power decoupling type power converter as described in claim 1, characterized in that: Including, Obtain the actual value of the input - side power of the power converter P in , according to the actual value of the input - side power P in and the input - side power reference value P in *, obtain the first input quantity i Lp *; Obtain the power on the load side of the power converter P load , and obtain a feedforward compensation amount according to the harmonic components of the power on the load side of the power converter P load as the second input quantity i Le *; Take the sum of the first input quantity i Lp * and the second input quantity i Le * as the reference value of the inner loop of the active power decoupling branch current, i Ln * and based on the difference between the reference value i Ln * and the actual value of the decoupling inductor current i Ln obtain the reference value of the decoupling inductor voltage u Ln *; Taking the decoupled inductor voltage reference value u Ln *as the common mode value of the modulation waves of the first to fourth bridge arms in the inverter circuit on the output side of the power converter, modulating the sum of the common mode value and the modulation waves to obtain the switching signals of the first to fourth bridge arms.

4. The control method according to claim 3, characterized in that: Obtain the actual value of the input-side power of the power converter P in , including, Obtain the voltage amplitudes of each phase on the input side V x and the current amplitudes I x , where x = a, b, c; Calculate the actual value of the active power on the input side based on the voltage amplitude and the current amplitude P in , ， wherein, θ x is the phase difference between the phase voltages and currents on the input side.

5. The control method according to claim 3, wherein: According to the actual value of the input side power P in and the reference value of the input side power P in to obtain the first input quantity i Lp *, including, Send the actual value of the input - side power P in and the reference value of the input - side power P in * into the parallel - resonance controller to obtain the first input quantity i Lp *; where the transfer function of the parallel - resonance controller is ， Among them, G R_2ω is the transfer function of the second harmonic component with a resonant frequency of 2 ω n , and G R_4ω is the transfer function of the fourth harmonic component with a resonant frequency of 4 ω n ; K r is the resonant gain, ω n and is the fundamental angular frequency.

6. The control method according to claim 3, characterized in that: Obtaining the power on the load side of the power converter P load , including, Obtain the voltage amplitude of each phase on the load side V o_x and the current amplitude I o_x , where x = a, b, c; Calculate the power on the load side based on the voltage amplitude and current amplitude P load , ， where, cos θ o_x is the phase difference between the phase voltages and currents on the load side.

7. The control method according to claim 3, wherein: According to the harmonic components of the power on the load side of the power converter P load to obtain a feedforward compensation amount as the second input quantity i Le *, including, The notch filter is adopted to extract the harmonic components of the power on the load side of the power converter, so as to obtain the DC component of the power on the load side P load , and the transfer function of the notch filter is as follows: P load_dc ​ ， Among them, ω n is the fundamental angular frequency; The load-side power P load is subtracted from the DC component of the load-side power P load_dc to obtain the second harmonic component of the load-side power P load_2ω ; According to the second harmonic component of the load-side power P load_2ω and the input-side voltage amplitude u sm calculate the feedforward compensation amount, which is the second input quantity i Le *, ， wherein, the input-side voltage amplitude u sm is calculated according to the following formula ， Among them, V α , V β are the α-axis and β-axis components of the three-phase voltage on the input side respectively, U sa , U sb , U sc are the amplitudes of the phase voltages of the three-phase input voltages respectively.

8. The control method according to claim 3, characterized in that: According to the reference value i Ln * the difference from the actual value of the decoupling inductor current i Ln to obtain the reference value of the decoupling inductor voltage u Ln *, including, Send the reference value i Ln * and the actual value of the decoupling inductor current i Ln into a proportional-integral multi-resonant controller to obtain the reference value of the decoupling inductor voltage u Ln *; where the transfer function of the proportional-integral multi-resonant controller is ， Among them, K p , K i , K r are the proportional gain, integral gain, and resonant gain respectively, ω r is the cut-off frequency, ω n is the fundamental angular frequency.

9. The control method according to claim 3, characterized in that: Taking the decoupled inductor voltage reference value u Ln * as the common-mode value of the modulation waves of the first to fourth bridge arms in the inverter circuit on the output side of the power converter, modulating the sum of the common-mode value and the modulation waves to obtain the switching signals of the first to fourth bridge arms, including Obtain the three-phase voltage on the load side u o and the three-phase current of the filter inductor i L , and use the Clark transformation to obtain the α, β, and 0-axis components of the three-phase voltage on the load u o , u o_α , u o_β , u o_0 and the α, β, and 0-axis components of the three-phase current of the filter inductor i L , i L_α , i L_β , i L_0 ; Set the three-phase load voltage reference value to u a , u b , u c . Obtain the α-axis and β-axis components of the three-phase load voltage reference value through Clark transformation u α , u β , and set the reference value of the zero-axis component of the load voltage u 0 to 0; Subtract the α-axis, β-axis, and 0-axis components of the three-phase load voltage reference value u α , u β , u 0 from the α-axis, β-axis, and 0-axis components of the actual load voltage respectively u o_α , u o_β , u o_0 . Input the difference into the load voltage outer-loop QPR controller for adjustment, and output the reference values of the α-axis, β-axis, and 0-axis filter inductor currents for the inner loop i L_α *, i L_β *, i L_0 *; where the transfer function of the load voltage outer-loop QPR controller is ， Among them, K up and K ur are the proportional gain and resonant gain of the load voltage outer-loop QPR controller respectively; ω uc is the cut-off frequency, ω n is the fundamental angular frequency; Subtract the α-axis, β-axis, and 0-axis components of the actual filter inductor current from the reference values of the α-axis, β-axis, and 0-axis filter inductor current inner loops respectively i L_α *, i L_β *, i L_0 * respectively, and input the differences into the QPR controller of the filter inductor current inner loop for adjustment. The outputs corresponding to the α-axis and β-axis filter inductor current inner loops are subjected to Clark inverse transformation to obtain the modulation voltages of the three-phase load i L_α , i L_β , i L_0 , and the output of the 0-axis filter inductor current inner loop is used as the second component of the modulation voltage of the fourth bridge arm u am , u bm , u cm . Among them, the transfer function of the QPR controller of the filter inductor current inner loop is u 0m ; where ， Among them, K ip , K ir are the proportional gain and resonant gain of the QPR controller for the inner loop of the filter inductor current, respectively; ω ic is the cut-off frequency; Modulate the three-phase load voltage u am 、 u bm 、 u cm Add the common-mode voltage 0.75 u sm and the first component of the modulation voltage of the fourth bridge arm u Ln * simultaneously to obtain the modulation voltages of the three-phase bridge arms u AO 、 u BO 、 u CO Add the first component of the modulation voltage of the fourth bridge arm u Ln * and the second component of the modulation voltage of the fourth bridge arm u 0m and then add the common-mode voltage 0.75 u sm to obtain the modulation voltage of the fourth bridge arm u NO , ， Each of the modulation voltages is modulated to obtain the switching signals of the first to fourth legs of the output - side inverter circuit.

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