Totem-pole power factor correction converter and control method thereof

Abstract

The invention provides a totem-pole power factor correction converter and a control method thereof. The totem-pole power factor correction converter includes: a plurality of phase legs connected in parallel with each other, the midpoint of each phase leg being connected to a corresponding input terminal; a plurality of boost inductors, each boost inductor being connected to a midpoint between the corresponding input terminal and the corresponding phase pin; a plurality of capacitors, one end of each capacitor is connected between the corresponding boost inductor and the input end, and the other end of each capacitor is connected to the neutral end; the diode bridge arm is connected with the plurality of phase pins in parallel, and the midpoint of the diode bridge arm is connected to the neutral end; the output filter is connected with the diode bridge arm in parallel; the T-shaped pin is connected between the midpoint of the diode bridge arm and the midpoint of the output filter and comprises two active switches which are connected in series; and a control system configured to turn on the two active switches of the T-pin near the AC voltage zero crossing during non-unit power factor operation.

Description

Technical Field

[0001] This case involves the field of on-board charging of electric vehicles, and specifically, involves a totem pole power factor correction converter for an on-board charger of an electric vehicle and a control method thereof. Background Art

[0002] Electric vehicles (EVs) are powered by electric motors rather than internal combustion engines. Because electric motors do not emit greenhouse gases, as internal combustion engines do, demand for EVs has surged in an era of global concern regarding environmental protection.

[0003] When connected to the AC utility system, state-of-the-art power electronic rectifiers must comply with input current and harmonic standards. Limits on low-frequency harmonic distortion are very strict and are typically met by applying a sinusoidal input current to the power converter that interfaces with the AC power system. A power converter that interfaces with the AC system and achieves low harmonic distortion simulates a resistive load, forcing its input current to follow the input voltage waveform. One of the most common solutions for providing a sinusoidal input current is through a so-called conventional boost converter. Figure 1 An exemplary embodiment is disclosed. The boost converter 10 includes an input diode bridge with rectifiers (D1, D2, D3, D4), a boost inductor (L1), a controlled switch device (S1), a boost diode (D1), a filter capacitor (C o ) and load (R). Figure 1 In FIG, the load (R) is illustrated as a resistor, but may also be another device including a downstream converter (e.g., an isolated DC-DC converter for regulating the DC voltage provided to the actual end-user load or the DC current for battery charging). The boost converter 10 can be appropriately controlled to extract a nearly sinusoidal AC input current, thereby producing a near-unity power factor.

[0004] Most electric vehicles are equipped with an on-board charger (OBC), which can be used to charge the vehicle battery at work, home, or other locations. Typically, an electric vehicle draws power from the AC grid via the OBC, which converts the power into electricity that can be used to charge the battery. For example, a 100kWh battery pack takes approximately 14 hours to charge using a 7kW OBC. The input power to the OBC may consist of single-phase AC voltage or, in some cases, three-phase AC voltage. Figure 2A single-phase electric vehicle onboard charger system 12 is disclosed. In this exemplary embodiment, the system 12 includes an AC / DC rectifier 14, a DC / DC converter 16, and a high voltage (HV) battery 18. In power factor correction (PFC) mode, the AC / DC rectifier 14 shapes the input current so that the DC / DC converter 16 acts as a resistor. That is, the input current has the same sinusoidal shape as the grid voltage, and there is no phase shift between the voltage and current. Unlike PFC mode, in non-unity power factor operation, such as Figure 3 The input current and grid phase diagram 20 is shown as an example. The phase current and grid voltage shown therein have a predetermined displacement angle θ_vi, so that the power factor becomes PF=cos(θ vi As shown by the superimposed labels in Figure 20, the waveforms represent the phase currents at unity PF, leading power factor, and lagging power factor. The waveforms with dashed lines represent the scaled phase voltage waveforms.

[0005] For distributed energy resources (DERs), including wind turbines and solar power plants, a typical power factor requirement or specification is approximately 0.95 (leading and lagging) (see A. Ellis, R. Nelson, E. Von Engeln, R. Walling, J. MacDowell, L. Casey, E. Seymour, W. Peter, C. Barker, B. Kirby, et al., “Reactive power performance requirements for wind and solar plants,” in 2012 IEEE Power and Energy Society General Meeting, pp. 1–8, IEEE, 2012). In an Australian study, a power factor of 0.8 was also used for both capacitive and inductive power factors (see D. Condon and D. McPhail, “Voltage regulation of distribution networks using inverter reactive power functionality - Australian utility experience,” in 2015 IEEE PES Asia-Pacific Power and Energy Engineering Conference (APPEEC), pp. 1–5, 2015). Until around 2022, electric vehicle onboard chargers (OBCs) were not required to provide reactive power capabilities. However, as the penetration of renewable energy sources across the grid continues to increase, some reactive power compensation requirements or specifications are emerging for relatively small devices (e.g., 22 kW OBCs). Reactive power compensation provides or absorbs reactive current, thereby regulating the voltage at the grid connection point. This evolution is not much different from the grid requirements for wind turbines and solar power plants, which began with simple active power injection and then moved on to frequency control and weak grid support (see "Inverter-based resource performance and analysis, technical workshop, NERC IRPT meeting, February 2019." https: / / www.nerc.com / comm / PC / IRPTF%20Workshops / IRPTF_Workshop_Presentations.pdf. Accessed: 2022-07-12). Summary of the Invention

[0006] According to one aspect disclosed in the present invention, a method for operating a totem pole power factor correction (PFC) converter is provided. The totem pole power factor correction converter includes: three input terminals, a neutral terminal, and two output terminals; a plurality of phase legs connected in parallel with each other, wherein each phase leg includes an upper switch and a lower switch connected in series, and there is a midpoint between the upper switch and the lower switch, and the midpoint of each phase leg is connected to a corresponding input terminal among the three input terminals; a plurality of boost inductors, each boost inductor is connected to a corresponding input terminal among the three input terminals and a midpoint of a corresponding phase leg among the plurality of phase legs; a plurality of capacitors, one end of each capacitor is connected between a corresponding boost inductor among the plurality of boost inductors and the input terminal, and the other end of each capacitor is connected to the midpoint. a neutral terminal; a diode bridge arm connected in parallel with the plurality of phase legs and comprising two diodes connected in series, wherein a midpoint is defined between the two diodes, and the midpoint of the diode bridge arm is connected to the neutral terminal; an output filter connected in parallel with the diode bridge arm and comprising two capacitors connected in series, wherein a midpoint is defined between the two capacitors; a T-leg connected between the midpoint of the diode bridge arm and the midpoint of the output filter and comprising two active switches connected in series; and a control system configured to execute a method, the method comprising: during non-unity power factor operation, turning on the two active switches of the T-leg near an AC voltage zero crossing.

[0007] According to another aspect disclosed herein, a totem pole power factor correction (PFC) converter is provided. The totem pole power factor correction (PFC) converter includes: three input terminals, a neutral terminal, and two output terminals, wherein the three input terminals are connected to an AC power source; a first switch leg, a second switch leg, and a third switch leg connected to the three input terminals and electrically connected in parallel with each other, wherein each of the first, second, and third switch legs includes an upper switch and a lower switch connected in series, and has a midpoint electrically connected to a corresponding phase of the AC power source via an inductor; a diode leg electrically connected in parallel with the first, second, and third switch legs, and including two diodes connected in series, wherein the diode leg has a midpoint electrically connected to the neutral terminal; an output filter connected in parallel with the diode leg, and including two capacitors connected in series, wherein the two capacitors have a midpoint between them; and a T-leg connected between the midpoint of the diode leg and the midpoint of the output filter, and including two active switches connected in series.

[0008] These and other aspects of the present invention will be apparent from and elucidated with reference to the embodiments described hereinafter. BRIEF DESCRIPTION OF THE DRAWINGS

[0009] Many aspects of the present embodiments may be better understood with reference to the following drawings. The components in the drawings are not necessarily drawn to scale; emphasis is placed on clearly illustrating the principles of the present embodiments. In the drawings, like reference numerals designate corresponding parts throughout the various views.

[0010] Figure 1 is a schematic diagram illustrating a boost converter.

[0011] Figure 2 The present invention is a schematic diagram showing a single-phase electric vehicle on-board charger system.

[0012] Figure 3 The present invention discloses an input current and grid phase diagram, and an example of reflecting different phase currents according to unity power factor and non-unity power factor calculation.

[0013] Figures 4A to 4B The three-phase / single-phase combination circuit adopts three-phase input according to the relay position ( Figure 4A ) or single-phase input ( Figure 4B ) operation example.

[0014] Figure 5 A demonstration of a single-phase bridgeless totem-pole rectifier for an on-board charger with staggered three legs is disclosed.

[0015] Figures 6A to 6B It is revealed separately Figure 5 Single-phase bridgeless totem pole rectifier for positive input, input voltage VAC>0, input current IAC>0( Figure 6A ) and for negative input, input voltage VAC<0, input current IAC<0( Figure 6B ) schematic diagram.

[0016] 7A to 7B Schematic diagram containing selected waveforms showing how the phase leg reconstruction of a bridgeless totem-pole rectifier works at unity power factor operation.

[0017] Figures 8A to 8B Contains diagrams of selected waveforms showing Figures 6A to 6B How does the bridgeless totem pole rectifier perform phase leg reconstruction under non-unity power factor operation?

[0018] Figure 9 is a schematic diagram illustrating a bridgeless totem-pole converter with a T-type leg, which is capable of achieving non-unity power factor operation by switching to T-type operation near the voltage zero crossing point.

[0019] Figure 10 Is to reveal the spreader and T-shaped foot ( Figure 9 ) Schematic diagram of the gating signal near the zero voltage crossing.

[0020] Figure 11 is a schematic diagram illustrating an exemplary three-phase / single-phase combined AC / DC converter of the present invention, which is capable of implementing a control / modulation method for non-unity power factor operation.

[0021] Figure 12 is revealed Figure 11 A three-phase / single-phase combination AC / DC converter is provided based on a bridgeless totem-pole converter derived from relay activation / deactivation, configured to enable a control / modulation method for non-unity power factor operation.

[0022] Figure 13 This invention discloses an embodiment of the present invention for Figure 12 Schematic diagram showing the interleaved gate control signals and boost choke current ripple waveforms for the three PWM-modulated phase legs of a totem-pole converter.

[0023] Figure 14 Schematic diagram of some selected waveforms of the control / modulation method of the embodiment of the present invention to achieve operation under non-unity power factor by actively modulating the expanded phase leg.

[0024] Figure 15 This is to reveal the embodiment of the present case Figure 14 Further expand the duty cycle of the PWM modulation phase leg.

[0025] Figure 16 It is an example of selecting the line cycle frequency waveform in the embodiment of the present case, which shows the input voltage VAC, the input current reference (for each phase), the duty cycle of the PWM modulated phase leg, the voltage at the midpoint of the expanded phase leg, the current of the boost inductor L1, L2, L3 and the total current input IAC.

[0026] Figure 17 The switch S in the phase leg of the embodiment of this case is disclosed. N and S P Example gating.

[0027] Figure 18A and Figure 18B This is a schematic diagram showing another control / modulation method in this case, which realizes reactive power control by switching the slow leg so that the voltage between the midpoints of the two phase legs is VDC, 0 and -VDC respectively within one switching cycle, which is called bipolar switching.

[0028] Figure 19 An exemplary three-phase / single-phase combination AC / DC converter with an expander leg is disclosed. The expander leg includes two diodes and a T-leg. The T-leg has two active switches and a relay for switching from a DC bus midpoint to the expander leg midpoint according to three-phase and single-phase configurations.

[0029] Figure 20The present invention discloses that the relay configuration is based on the embodiment of the present invention. Figure 19 Schematic diagram of a totem-pole converter derived from a three-phase / single-phase combination AC / DC converter, with its expanded legs including two diodes and a T-leg with two active switches.

[0030] Figure 21 1 is a timing diagram illustrating the generation of a T-type leg gate signal under the condition of leading power factor according to an embodiment of the present invention.

[0031] Figure 22 The embodiment of the present invention discloses a change in the phase angle (ωT1) of the T-type gate signal activation, wherein the phase shift thereof changes by about 0.8 to 0.95.

[0032] Figure 23 The embodiment of the present invention discloses a change in the phase angle (ωT2) of the T-type gate signal activation, wherein the change in the phase shift lags by approximately 0.8 to 0.95.

[0033] FIG. 24A to FIG. 24B Schematic diagram illustrating some selected waveforms during non-unity power factor operation under leading operation with a power factor of 0.9 when the AC current and voltage are in opposite directions in an embodiment of the present invention, including an amplification.

[0034] FIG. 25A to FIG. 25B Schematic diagrams, including magnification, illustrating some selected waveforms during non-unity power factor operation at a power factor lagging 0.9 when the AC current and voltage are in opposite directions in an embodiment of the present invention.

[0035] FIG. 26A to FIG. 26B 1 and 2 are simulation results showing some selected waveforms during non-unity power factor operation when the power factors lead 0.796 and 0.955, respectively, in the embodiments of the present invention.

[0036] FIG. 27A to FIG. 27B The figures are simulation results showing some selected waveforms during non-unity power factor operation when the power factors lag by 0.796 and 0.955, respectively, in the embodiments of the present invention.

[0037] Figure 28 The present invention discloses an embodiment of a three-phase / single-phase AC / DC converter and a control system with fractional switching closed-loop control.

[0038] Figure 29 The present invention discloses an embodiment of a three-phase / single-phase AC / DC converter and a control system with T-type leg switch closed-loop control. DETAILED DESCRIPTION

[0039] This application discloses, in certain embodiments, a three-phase / single-phase combination AC / DC converter capable of achieving non-unity power factor operation. In one embodiment, the three-phase / single-phase combination AC / DC converter is configured to operate as a single-phase bridgeless totem-pole PFC circuit for single-phase input to achieve non-unity power factor operation. Existing combination circuits can also function as totem-pole PFC circuits when operating in single-phase operation, but cannot achieve non-unity power factor operation due to the reconfiguration of the spread phase legs.

[0040] In some embodiments of a three-phase / single-phase combination AC / DC converter, when the input phase voltage is zero or near zero, the line-frequency voltage is temporarily switched at a high frequency to spread the phase legs, providing a degree of freedom and thus controllability of non-zero phase currents. By doing so, the converter (rectifier) ​​can operate at non-unity power factors, making it compatible with future grid requirements.

[0041] In some embodiments of a three-phase / single-phase combination AC / DC converter, at least two switches are added between the diode spreader leg and the DC bus midpoint, thereby forming a T-type totem-pole bridgeless PFC circuit or device. The T-leg activates near the AC voltage zero crossing and when the AC voltage and current are of opposite polarity. This provides the appropriate voltage across the inductor to establish current near the zero voltage crossing. Additionally, when the AC current and voltage are of opposite polarity, the T-leg turns on and conducts current in the reverse direction. This enables the converter (rectifier) ​​to operate at non-unity power factor, making it compatible with future grid requirements. Furthermore, non-unity power factor operation can be achieved using only low-cost diodes in the spreader leg, reducing cost and control complexity.

[0042] To summarize certain features of the three-phase / single-phase combination AC / DC converter of the present invention, a description will now be made in detail with reference to the three-phase / single-phase combination AC / DC converter illustrated in the accompanying drawings. Furthermore, although details of one or more embodiments are defined or described herein, such details are not necessarily integral to every embodiment, and not all stated advantages are necessarily related to a single embodiment. Rather, the intent is to cover alternatives, modifications, and equivalents within the spirit and scope of the present invention as defined by the appended claims. While the focus is on three-phase / single-phase combination circuits, specific portions of some embodiments may represent solutions for existing non-unity power factor applications. For example, a single-phase topology utilizing one or more combinations of a T-type configuration, a diode in the spreader leg, or a temporary high switch in the spreader leg may be used in certain electric vehicle implementations. Furthermore, while the focus of this disclosure is on electric vehicle onboard charger applications, certain embodiments described herein may also be used in other applications, such as power conversion in data center applications. It is important to note that two or more embodiments of this disclosure may be interchangeable or combined in any combination. Furthermore, the terms "rectifier" and "converter" are used interchangeably herein, and the term "converter" should be understood to have a broader scope. It is worth noting that the waveforms referred to in this case are analog waveforms. Furthermore, it should be understood that within the context of this disclosure, the claims are not necessarily limited to the specific embodiments listed in the specification.

[0043] Before that, some additional explanations are given on some of the challenges of operating under non-unit power factors. As mentioned above, the input power supply of the on-board charger can include single-phase AC voltage or three-phase AC voltage. Currently, many on-board chargers operate under single-phase input conditions, with low output power and long charging time. With the increase in the capacity of power batteries of new electric vehicles and the demand for fast charging, the demand for three-phase input and higher power on-board chargers is also increasing. The three-phase / single-phase topology or architecture usually includes a front-stage circuit and a rear-stage isolated DC-DC circuit. The front-stage PFC circuit can reduce harmonic distortion while achieving PFC and provide a small ripple DC bus voltage for the rear-stage DC-DC circuit. The rear-stage DC-DC circuit can also output stable and adjustable high-voltage DC power to the power battery. Reference Figures 4A to 4B, which discloses an exemplary three-phase / single-phase combination circuit 22 that connects the utility grid to a DC / DC converter for the implementation of an on-board charger for electric vehicles. The three-phase / single-phase combination circuit 22 includes three input terminals A, B, C, a neutral terminal N, two output terminals p, n, an input filter, multiple boost inductors L1, L2, L3, L4, L5, L6, multiple capacitors, multiple phase legs (or switch phase legs, bridge arms, PWM modulation phase legs), a diode bridge arm (or slow leg), an output filter (or capacitor leg) and multiple relays. The three input terminals A, B, C are connected to the AC input of the utility grid to receive a three-phase AC input or a single-phase AC input. The input filter is electrically connected between the three input terminals A, B, C and the multiple phase legs to filter the aforementioned AC input. The multiple phase legs are connected in parallel to each other, and each phase leg includes two switches connected in series, such as an upper switch (S1, S3, S5, S7, S9, S 11 ) and the following switches (S2, S4, S6, S8, S 10, S 12 ). The aforementioned switches (S1, S2, S3, S4, S5, S6, S7, S8, S9, S 10 ,S 11 ,S 12 ) can be a metal oxide semiconductor field effect transistor with an antiparallel diode, but is not limited to this. The upper switch (S1, S3, S5, S7, S9, S 11 ) and the lower switch (S2, S4, S6, S8, S 10, S 12 ) have a midpoint between them, which is electrically connected to the corresponding input terminals A, B, and C. Each boost inductor L1, L2, L3, L4, L5, and L6 is electrically connected between the corresponding input terminals A, B, and C and the midpoint of the corresponding phase leg. One end of each capacitor is connected between the corresponding boost inductor L1, L2, L3, L4, L5, and L6 and the input terminals A, B, and C, and the other end of each capacitor is connected to the neutral terminal N. The diode bridge arm is connected in parallel with multiple phase legs and includes two diodes D N ,D P Connected in series, two diodes D N ,D PThere is a midpoint between them. The output filter is connected in parallel with the diode bridge leg and includes two capacitors connected in series, with a midpoint between the two capacitors. The multiple relays include a first relay RL1 and a second relay RL2. One end of the first relay RL1 is connected to one end of one of the multiple boost inductors L1, L2, L3, L4, L5, and L6, and the other end of the first relay RL1 is connected to the midpoint of a corresponding phase leg in the multiple phase legs. One end of the second relay RL2 is connected to the neutral terminal N, and the other end of the second relay RL2 is selectively connected to the midpoint of the diode bridge leg or the midpoint of the output filter. The illustrated three-phase / single-phase combination circuit 22 (or three-phase / single-phase combination AC / DC converter) can operate with three-phase (combination circuit 22A) and single-phase (combination circuit 22B) inputs depending on the positions (activated / deactivated) of the first relay RL1 and the second relay RL2. In the case of three-phase input, the second relay RL2 is open (ie, the other end of the second relay RL2 is connected to the midpoint of the output filter), and the first relay RL1 is closed. Figure 4A The configuration of the combined circuit 22A is shown. When the single-phase input is applied, the first relay RL1 is open and the second relay RL2 is closed (ie, the other end of the second relay RL2 is connected to the midpoint of the diode bridge arm). Figure 4B The configuration of the combinational circuit 22B is shown.

[0044] For single-phase input, the combined circuit 22B operates as a (bridgeless) totem-pole PFC rectifier 22B-1, as shown in FIG. Figure 5 As shown. The combination circuit 22B-1 can interconnect the on-board charger and the single-phase power grid. In this embodiment, the combination circuit 22B-1 includes three input terminals A, B, C, a neutral terminal N, two output terminals p, n, an input filter, three boost inductors L1, L2, L3, three capacitors, three phase legs (or switch phase legs, bridge arms, PWM modulation phase legs), a diode bridge arm (or slow leg or slow bridge arm) and an output filter (or capacitor leg). The three input terminals A, B, C are configured to receive single-phase input. The input filter is electrically connected between the three input terminals A, B, C and the three phase legs. The three phase legs are connected in parallel to each other, and each phase leg includes two switches connected in series. There is a midpoint between the upper switch (S1, S3, S5) and the lower switch (S2, S4, S6) of each phase leg, and the aforementioned midpoint is electrically connected to the corresponding input terminal A, B, C. Each boost inductor L1, L2, L3 is electrically connected between the corresponding input terminal A, B, C and the midpoint of the corresponding phase leg. One end of each capacitor is connected between the corresponding boost inductor L1, L2, L3 and the input terminal A, B, C, and the other end of each capacitor is connected to the neutral terminal N. The diode bridge arm is connected in parallel with the three phase legs and includes two diodes D N ,DP Connected in series, two diodes D N ,D P There is a midpoint between them. The output filter is connected in parallel with the diode bridge arm and includes two capacitors connected in series. The midpoint of the diode bridge arm is electrically connected to the neutral terminal N and connected to the common node of the grid neutral line. In some embodiments, the three phase legs (i.e., Figure 5 From left to right in the figure, the first phase leg has complementary switches S1 and S2, the second phase leg has complementary switches S3 and S4, and the third phase leg has complementary switches S5 and S6) carrying equal amplitude currents. These three phase legs operate at high frequency through boost inductors (chokes) L1, L2, and L3 and are interleaved with each other to reduce high-frequency ripple in the grid AC current. However, the single-phase bridgeless totem pole rectifier 22B-1 itself does not support reactive power flow. This is because the slow leg 24 (i.e., Figure 5 、 Figure 6A and Figure 6B Including diode D N and D P Due to the nature of the legs), it is also called the expanded (or expanded) phase leg 24 in the power conversion industry, and it is connected to the neutral line of the grid, which will be further explained below. Figure 6A (for positive input, input voltage VAC>0, input current IAC>0) and Figure 6B (For negative input, input voltage VAC < 0, input current IAC < 0) The example circuit represents 26A, 26B, the phase leg 24 is expanded to periodically reconnect the neutral terminal (N) of the grid voltage to the negative terminal (n) of the DC link (such as Figure 6A The circuit in FIG26A represents the circuit in FIG26A) and the positive (p) rail (as shown in FIG26A). Figure 6B The circuit in Figure 26B shows that the grid voltage is always positive relative to the negative potential of the DC link. More generally, for unity power factor, the input voltage VAC and the input current IAC are in phase. When the input voltage VAC and the input current IAC are positive ( Figure 6A ) P When the input voltage VAC and input current IAC are negative ( Figure 6B ) N This reconfiguration allows the PWM modulated phase leg to regulate the grid current. Note that for simplicity, only a single PWM modulated phase leg is shown.

[0045] Reference 7A to 7B , which reveals how a bridgeless totem-pole rectifier operating at unity power factor performs phase leg reconstruction. 7A to 7B Several waveforms are shown, including Figure 7B The chart in the figure has the suffix B. Figure 7AA close-up view of the corresponding numbered chart (with suffix A) in the figure is shown. 7A to 7B From top to bottom, the input AC voltage waveform graph 28 (i.e., 28A, 28B), the phase leg (PWM modulation) duty cycle waveform graph 30 (i.e., 30A, 30B), the PWM active waveform graph 32 (i.e., 32A, 32B), and the input AC current (+reference) waveform graph 32 (i.e., 34A, 34B) are shown. Note that the input AC current refers to the actual measured current, while the reference refers to the controller reference. Figure 7A As shown in the input AC voltage waveform diagram 28A and the phase leg duty cycle waveform diagram 30A, when the input AC voltage, and therefore the AC current (see input AC current + reference waveform diagram 34A), changes polarity, the spread phase leg reconstruction occurs. If the power factor is 1 (i.e., the shapes of the voltage and current waveforms are equal, but not necessarily the amplitudes), then during the reconfiguration of the spread phase leg, the modulation of the PWM phase leg (i.e., Figures 6A to 6B The pin with switches S1 and S2 in the circuit) will be closed (ie, as shown in the PWM active waveform diagram 32A), which is typically as shown in FIG. Figure 7B The PWM active waveform shown in FIG32B shows multiple PWM cycles. Figure 5 The diode D P and D N By being replaced by a controllable switch, the midpoint voltage of the phase leg can be actively controlled instead of being passively driven by the input AC current (i.e., the positive input current causes the diode D P Forward bias, negative input current makes diode D N Forward bias). However, while spreading the phase legs theoretically allows for reconfiguration using control signals within a single switching cycle, measurement errors in input voltage sensing, filter delays, and controller delays make this approach impractical. Therefore, gating signals are typically not applied near the zero AC voltage crossing point. The absence of modulation near zero AC voltage results in zero phase current. However, due to the relatively small ideal input current, the impact on the phase current total harmonic distortion (THD) is relatively small.

[0046] Figures 8A to 8B Revealed and 7A to 7BSimilar waveform graphs, but for non-unity power factor operation, include (from top to bottom) input AC voltage waveform graph 36 (i.e., 36A, 36B), phase leg (PWM modulation) duty cycle waveform graph 38 (i.e., 38A, 38B), PWM active waveform graph 40 (i.e., 40A, 40B), and input AC current (+reference) or total input current waveform graph 42 (i.e., 42A, 42B). When the power factor is non-unity, turning off the PWM phase leg near the zero voltage crossing may cause significant perturbations in the non-zero phase current, as shown in input AC current + reference waveform graph 42A ( Figure 8A ) and 42B( Figure 8B ) as shown.

[0047] from Figures 8A to 8B From the observation, it is worth noting that Figures 6A to 6B In the figure, 26A and 26B are represented. When the input voltage VAC and the input current IAC are out of phase, the diode D P and diode D N Unable to conduct current, an additional path is required for current to flow. In the reference ("Single-Phase GaN-Based T-Type Totem-Pole Rectifier With Full-Range ZVS Control and Reactive Power Regulation," in IEEE Transactions on Power Electronics, vol. 38, no. 2, pp. 2191-2201, Feb. 2023, doi: 10.1109 / TPEL.2022.3215969 by J. Sun, L. Zhu, R. Qin, DJ Costinett and LMT Olbert.), a method for non-unity power factor operation of a single-phase totem pole PFC circuit or device (i.e., rectifier) ​​operating in critical conduction mode (CRM) is discussed. In the reference, and with reference to Figure 9 The circuit 44 in FIG. 4 proposes a solution to connect two bidirectional current-conducting MOSFETs (S3 and S4) in the expander leg. However, as discussed in the document, even with these switches, when the input voltage VAC is approximately equal to 0, the voltage is insufficient to generate current at a non-unity power factor. The document also proposes a bridgeless totem pole converter with a T-type configuration, which includes two additional active switches (S3 and S4) connected between the expander leg AC point 46 and the DC bus midpoint 48. 5a and S 5b ). Two active switches S in T-type configuration 5a and S5b Starts only near the zero voltage crossing. This arrangement may help to establish the required voltage across the inductor near the zero crossing of the AC voltage and further assist in current shaping (see Figure 10 ), thereby achieving the required current. In practice, circuit 44 comprises a bridgeless totem pole converter with a T-type leg, capable of achieving non-unity power factor operation by switching to T-type operation near the zero voltage crossing. When the voltage amplitude is higher than the boundary voltage |V bound |, the operation remains unchanged, and switches S1 and S2 operate at high frequency, while the active switches S3 and S4 of the expansion leg (slow leg) are continuously turned on or off according to the voltage direction, such as Figure 10 The device switching sequence diagram 50 is shown. For example, the device switching sequence diagram 50 shows the gate control signals of the expander and the T-leg near the zero voltage crossing.

[0048] See also Figure 11 , which discloses an exemplary three-phase / single-phase combined AC / DC converter 52 connecting the utility grid to a DC / DC converter 54 for implementation of an electric vehicle on-board charger while being capable of operating at non-unity power factor operation. The three-phase / single-phase combined AC / DC converter 52 is topologically similar to Figure 4A and Figure 4B The three-phase / single-phase combination circuit 22 shown has some similarities, wherein the same components and numbers represent similar components and functions, and are not described again here. Figure 4A and Figure 4B The embodiment of the three-phase / single-phase combination circuit 22 shown in FIG. 5 further includes a third relay RL3 connected to the fast switch pin (ie, including the switch S 11 and S 12 The last fast switching foot) and the slow foot (ie, including the diode D N and D P The illustrated three-phase / single-phase combination AC / DC converter 52 can operate with both three-phase and single-phase inputs depending on the positions (activated / deactivated) of the first, second, and third relays RL1, RL2, and RL3.

[0049] In the case of unidirectional input, by selecting the first relay RL1 to be open, the second relay RL2 to be closed (i.e., the other end of the second relay RL2 is connected to the midpoint of the diode bridge arm), and the third relay RL3 to be closed, the three-phase / single-phase combined AC / DC converter 52 operates as a totem pole PFC converter 56 for single-phase, non-unity power factor operation. The circuit configuration is as follows: Figure 12As shown. The totem pole PFC converter 56 connects the utility grid to the DC / DC converter 54. The totem pole PFC converter 56 receives the utility grid input (input voltage VAC, input current IAC) at the input filter 58 and allows the current in the boost chokes (inductors) L1 to L3 to be staggered in single-phase operation. It is worth noting that in some embodiments, a different number of chokes (e.g., less than or more than three) can be used. Figure 12 In the exemplary topology shown in FIG, a totem pole PFC converter 56 includes a first phase leg 60 (also called a first switching leg, and having complementary switches S1 and S2), a second phase leg 62 (also called a second switching leg, and having complementary switches S3 and S4), a third phase leg 64 (also called a third switching leg, and having complementary switches S5 and S6), a fourth phase leg (also called a slow leg or an expander or an expanded phase leg, having a switch S N and S P ), and the diode phase leg (with diode D N and D P , also known as diode bridge arm, diode leg, expander or expander leg diode). It should be noted that the terms "expander" and "expanding" are used interchangeably in this article. To simplify the operation analysis, it is assumed that the output capacitance C O1 and C O2 High enough so that the output voltage V O can be considered to be reasonably constant over the entire AC line cycle.

[0050] The midpoint 70 of the fourth phase leg 66 is connected to the midpoint 72 of the diode phase leg 68, and the two midpoints 70 and 72 are further connected to the neutral terminal of the filter capacitor through a point 74. N and S P The internal diode of the diode phase leg 68 is compared to the external diode D N and D P , has a higher voltage drop. Therefore, the diode D N and D P With switch S N and S P Parallel connection can reduce the total conduction loss.

[0051] Figure 13 1 shows the gate control signals (top graph 76 ) and the associated boost choke currents (current ripple waveforms, bottom graph 78 ) of the three PWM modulated phase legs 60 , 62 , and 64 of the totem pole PFC converter 56 for single-phase operation.

[0052] In an exemplary operation at unity power factor, when both the input voltage VAC and the input current IAC are positive, and the switch S of the fourth phase leg 66 is N and SP When not conducting, the diode D of the diode phase leg 68 P Similarly, when the input voltage VAC and the input current IAC are both negative, and the switch S of the fourth phase leg 66 is N and S P Not conducting, diode D at diode phase leg 68 N For standard operation with unity power factor, the switch S in the fourth phase leg 66 is forward biased and conducts current IAC. N and S P No operation is necessary because the diode D in the diode phase leg 68 P and D N The input voltage VAC will be automatically expanded. However, when the diode D P When forward biased, it may be beneficial for the fourth phase leg 66 switch S P Similarly, when the diode D of the diode phase leg 68 is turned on, an additional path is provided for the input current IAC, thereby reducing the conduction loss. N When the forward bias is on, the switch S of the fourth phase leg 66 N In this case, the switch S of the fourth phase leg 66 N and S P Can be used as a synchronous rectifier.

[0053] In operation where the power factor is not 1, the input current IAC may be negative or positive and is independent of the polarity of the input voltage VAC. Figure 3 As shown in Figure 2. Since the polarity of the input voltage VAC and the input current IAC is not the same in the entire half-line cycle, the diode D P and D N The automatic input voltage expansion will no longer work.

[0054] One possible solution to force the input voltage VAC is to use Figure 12 The switch S of the fourth phase leg 66 N and S P That is, when the input voltage VAC is positive, the switch S P When the input voltage VAC is negative, the switch S N This behavior is significantly different from the operation of synchronous rectifiers, which do not allow for gating intervals (which are required when synchronous rectifiers operate at 50 / 60Hz). Synchronous rectifiers used to spread the phase legs are usually turned on with a delay of several switching cycles to prevent issues with input voltage V AC polarity sensing.

[0055] In order to achieve continuous modulation of the input current, the switch S N and S PThe fourth phase leg 66 should be reconfigured within a single switching cycle. This raises two issues:

[0056] 1) The aforementioned input voltage polarity sensing issue - sensor, filter, sampling, and computational delays need to be considered. If the voltage polarity is incorrect, even temporarily, the rectifier will lose its current regulation capability.

[0057] 2) Even if the input voltage VAC is correctly developed, the PWM modulated phase legs 60, 62, 64 (with switches S1...S6) do not have enough voltage margin to fully control the input current to the desired set point. Consider the following scenario, where the input voltage VAC is only slightly positive (e.g., VAC=0.01V), and the switches S P is activated and commands the input current IAC to be negative (i.e., Figure 3 Leading PF, grid angle = 180 degrees). Due to the switch S P is turned on, the midpoint of the expanded phase is clamped to the negative rail of the DC link, so its voltage is 0V. To force the input current IAC to be negative, the voltage across the boost chokes L1…L3 needs to be negative. However, when the top switch S1 is turned on, the voltage across the boost choke L1 is (V O –VAC)~V O When the bottom switch S2 is on, the voltage across the boost choke L1 is (–VAC) ~ -0.01 V. Therefore, there is not enough negative voltage in the system to successfully control the input current IAC.

[0058] In one embodiment, in order to achieve continuous modulation and maintain current control near the zero AC voltage crossing, the PWM modulation phase legs 60, 62, 64 and 66 should be avoided from operating at extreme duty cycles (i.e., <1% and >99% in one embodiment, or other than 0% or 100%). In one embodiment, the phase legs 60, 62, 64 and 66 are operated in such a manner that the unrolled phase average voltage is maintained between 0V and V O non-extreme values ​​between .

[0059] Please note Figure 14 and Figure 15 . Figure 14 Some selected waveform diagrams are disclosed that illustrate an embodiment of a modulation method for implementing a totem pole PFC converter 56 by actively modulating the fourth or spread phase leg 66. Figure 12 ) Operation at non-unity power factor. Figure 14 The waveform diagram includes input voltage VAC 80, input current reference 82, top switch gating signal 84, switch S NThe gate control signal 86 (i.e., originating from the controller), the extended phase leg voltage 88 (i.e., measured from the converter circuit), and the boost choke L1...L3 current 90. Figure 15 include the Figure 14 same waveform diagrams, and further expand these diagrams by showing the duty cycles of the PWM modulated phase leg waveforms 92 (i.e., phase legs 60, 62, and 64). At Figure 14 (and Figure 15 ), referring to the waveform diagram 88, the extended phase leg voltage starts at 0V (when VAC > +25V), then transitions to an average value of approximately V O / 2 (during the period when -25V < VAC < 2�V), and finally increases to V O (when VAC < -25V). In addition, it can be seen from the waveform diagram 84 that the top switch gate control signals of the three PWM modulated phase legs also change from a low duty cycle (when VAC > 25V) to approximately 50% + VAC / V O duty cycle (when -25V < VAC < 25V), and then change to a high duty cycle (when VAC < -25V). However, Figure 15 [[ID=十六]]the duty cycles shown in the waveform diagram 92 of [[ID=十七]] Figure 12 (the duty cycles of the three PWM modulated phase legs 60, 62, and 64 are substantially equal) are not extreme, i.e., not 0% or 100% (i.e., not <1% or >99%). For example, the minimum duty cycle is 4% and the maximum duty cycle is 96%. Therefore, the current controller of the totem pole PFC converter 56 (

[0060] ) will not lose regulation.

[0061] Table 1 reveals the approximate duty cycles of the PWM modulation and extended phase legs during the entire line cycle. Overall, Table 1 reflects the operating regions selected based on the VAC phase angle. It should be noted that θ TR1 and θ<& TR2 values depend on the circuit implementation and its limitations. For example, in one embodiment, these values are within the range where VAC is less than approximately 8% - 10% VAC MAX . θ<& VAC is the angle of the grid, in degrees. In this specific example, θ<& TR1 = θ<& TR2 = 5 degrees. In a physically non-ideal system, the duty cycles of the PWM modulated phase legs can be further adjusted by additional feedforward terms and feedback controllers.

[0062]

[0063] Table 1

[0064] In addition, if Figure 14 As shown, when the fourth phase leg 66 (i.e., the spread phase leg) is modulated and the rectifier operates in the N+1 region, the switching frequency is reduced to approximately half. While current ripple would typically be expected to increase, the current ripple actually remains relatively low because both the PWM-modulated phase legs 60, 62, and 64 and the spread phase leg 66 are modulated, increasing the number of voltage transitions the boost choke experiences during each switching cycle. One reason for the reduced switching frequency in the N+1 region is that the fourth phase leg 66 carries the combined current of the first three phase legs 60, 62, and 64, thereby increasing conduction and switching losses. The reduction in switching frequency in this region subsequently reduces switching losses. Furthermore, in the N+1 region, the gate signals for the PWM-modulated phase legs 60, 62, and 64 are not phase-shifted relative to each other, which helps minimize overall current ripple because the carriers of the PWM-modulated phase legs 60, 62, and 64 and the spread phase leg 66 are synchronized.

[0065] When the synchronous rectifier is operated at 50 / 60Hz, a gated interval is used. For reactive power control, this interval should not occur even in the extended leg. However, when the fast leg and slow leg are at the switching frequency and follow the same Figures 14 and 15 When operating with a modulation control similar to that shown in , no spacing is required.

[0066] Figure 16 Some selected line cycle waveforms are disclosed, including input voltage VAC 94, input current reference 96, duty cycle 98 of the phase legs (i.e., phase legs 60, 62, and 64), voltage 100 of the expanded phase leg (i.e., phase leg 66), boost choke L1...L3 102, and input current 104. The total input current is shown as Figure 16 As shown in waveform graph 104, no significant distortion occurs in the N+1 region (where expanded phase leg 66 is modulating), despite the low switching frequency and the absence of phase shift in the carriers of PWM modulated phase legs 60, 62, and 64. The two vertical lines (n waveform graph 98) represent the zero angle of input voltage VAC and the zero angle of input current IAC. The power factor in this example is 0.9 leading.

[0067] Furthermore, it is worth noting that despite the fact that there is a displacement of about 25.7 degrees between the input voltage VAC and the input current IAC at a power factor of 0.9, the N+1 region is actually quite narrow, spanning about 10 degrees in the simulation. The N+1 region may be more or less appropriate for the application and the hardware used. One purpose of the N+1 region is to avoid extreme duty cycles of the PWM modulated phase legs, which is independent of the desired power factor. However, as mentioned above, when the power factor is 1, the current near the zero voltage crossing cannot be regulated, so the PWM modulated phase legs are simply turned off. Since the commanded current amplitude is very low, this does not affect the input current THD, as Figure 7A shown, for unity power factor operation.

[0068] Furthermore, it should be noted that although the duration of the N+1 region may be very short, when the input voltage AC is negative and the input current IAC is positive, the switch S of the phase leg 66 is N Should be turned on with a 100% duty cycle. Figure 17 As shown, in addition to showing Figure 16 In addition to the waveform diagrams 94 and 96, the switch S N The gate control signal 106, switch S P The waveform diagram of the gate control signal 108 and the expanded phase leg voltage 110. If the switch S N Not open, diode D N will become a blocking voltage, and the diode D P will carry current. However, this will result in the expanded phase voltage being 0V instead of V O Since the input AC voltage is also negative, current regulation will be lost.

[0069] On the contrary, when the input voltage AC is positive and the input current IAC is negative, the switch S P In the remaining cases outside the N+1 region, the polarity of the input voltage VAC and the input current IAC are the same, and both switches SN and SP can be turned off without loss of function.

[0070] It is worth noting that the above combined Figures 12 to 17 The control method described works with arbitrary power factors and arbitrary current shapes.

[0071] 18A to 18B Another method of generating reactive current is disclosed in . Figure 18A The waveform diagrams of VAC / IAC waveform chart 112A, VAB waveform chart 114A and IL waveform chart 116A are disclosed. Figure 18B An enlarged view of the same waveform is disclosed and is labeled using the same figure number, but with the suffix B instead of the suffix A. 18A to 18BThe transition from unipolar operation to bipolar operation near the zero crossing of the input voltage VAC is shown for reactive power control, such that the voltage between the fast leg and the slow leg changes from VDC to 0 and then to negative VDC within one switching cycle (i.e., the phase leg 66 is switched so that the voltage between the two midpoints (VAB) is VDC, 0, and -VDC within one cycle, which is also called bipolar switching). It is worth noting that one switching cycle refers to one PWM time period (typically >5kHz). Figure 18B Revealed Figure 18A The inductor current is generated using three switching states, which are similar to the Figures 14 to 16 In contrast to the control method shown in , the voltage only changes from 0 to VDC and from 0 to negative VDC in one switching cycle.

[0072] Having described what is generally referred to as a fractional switching control approach, attention is now turned to certain embodiments of variations on the totem-pole converter architecture. Figure 19 An embodiment of a three-phase / single-phase combined AC / DC converter 118 is disclosed. In this embodiment, the three-phase / single-phase combined AC / DC converter 118 connects the utility grid to a DC / DC converter for use in an electric vehicle onboard charger, while being capable of operating at non-unity power factor operation. The three-phase / single-phase combined AC / DC converter 118 is topologically similar to the Figure 11 The three-phase / single-phase combined AC / DC converter 52 shown has some similarities, where the same elements and numbers represent similar elements and functions, and are not described again here. Figure 11 The embodiment of the three-phase / single-phase combined AC / DC converter 52 is shown in FIG. 1 . The three-phase / single-phase combined AC / DC converter 118 of this embodiment omits the first relay RL1, the second relay RL2, and the third relay RL3 of the three-phase / single-phase combined AC / DC converter 52. The three-phase / single-phase combined AC / DC converter 118 includes a diode D N and D P The expander (diode) foot 120 has two active switches S A1 and S A2 The T-leg 122 and the relay RL1 are connected. One end of the relay RL1 is connected to the neutral terminal N, and the other end of the relay RL1 is selectively connected to the midpoint of the expander leg 120 or the DC bus midpoint (i.e., the midpoint of the output filter or the midpoint of the capacitor leg). The relay RL1 is configured to switch from the DC bus midpoint to the midpoint of the expander leg 120 depending on whether the three-phase / single-phase combination AC / DC converter 118 is operating in a three-phase configuration or a single-phase configuration. The two active switches S A1 and S A2(or T-leg switches) are connected in series and connected between the midpoint of the expander leg 120 and the DC bus midpoint. This architecture / topology enables the three-phase / single-phase combined AC / DC converter 118 to operate according to non-unity power factor operation. In unidirectional input operation, by selecting relay RL1 to switch between the midpoint connected to the diode bridge leg and the neutral terminal N, the three-phase / single-phase combined AC / DC converter 118 can be used as Figure 20 The totem pole PFC converter 128 is shown in operation. The totem pole PFC converter 128 connects the utility grid to the DC / DC converter 54. The totem pole PFC converter 128 receives the utility grid input (input voltage VAC, input current IAC) at the input filter 58, and the currents generated in the boost chokes (inductors) L1 to L3 can be interleaved in single-phase operation. It is worth noting that in some embodiments, a different number of chokes (e.g., less than / more than three) can be used. Figure 20 In the exemplary architecture or topology shown in FIG, the totem pole PFC converter 128 includes a first phase leg 130 (or a first switch leg, having complementary switches S1 and S2), a second phase leg 132 (or a second switch leg, having complementary switches S3 and S4), and a third phase leg 134 (or a third switch leg, having complementary switches S5 and S6), an expansion leg (or a diode bridge leg) 120, a T-leg 122, and a capacitor leg, as described above. To simplify the operation analysis, it is assumed that the output capacitance C of the capacitor leg is O1 and C O2 High enough so that the output voltage V O can be considered to be reasonably constant over the entire AC line cycle.

[0073] When the input voltage VAC and the input current IAC are both positive, and the active switch S of the T-type leg 122 is A1 、S A2 When closed, the diode D of pin 120 is expanded P Similarly, when both the input voltage VAC and the input current IAC are negative, the active switch S of the T-leg 122 is A1 、S A2 Close, expand the diode D of pin 120 N For standard operation with unity power factor, the active switch S of the T-leg 122 is forward biased and conducts current IAC. A1 and S A2 No need to run, because the diode D of pin 120 is expanded P and D N The input voltage VAC will be automatically expanded.

[0074] When the power factor is not 1, the input current IAC may be negative or positive regardless of the polarity of the input voltage VAC, such as Figure 3 As shown in Figure 2. Since the polarity of voltage VAC and current IAC is not the same in the entire half-line cycle, the diode D P and D N The automatic input voltage expansion will no longer work.

[0075] Since the diode D P and D N When forward biased, current can only be conducted in one direction. Therefore, when the input voltage VAC and the input current IAC are of opposite polarity, a path is required for the current to flow. This path is achieved by adding two active switches (i.e., S A1 and S A2 ) is provided. T-leg 122 provides a flow path for the boost inductor current and provides voltages VAC-VDC / 2 and VAC+VDC / 2 across the boost inductor when the input voltage VAC is close to zero and the current has a high amplitude (i.e., see Figure 3 ).

[0076] In addition, briefly explain Figure 10 When the AC voltage amplitude is lower than V bound When the AC voltage exceeds V bound When , switches S3 and S4 will be turned on.

[0077] However, in certain embodiments disclosed herein, when the input voltage VAC and the input current IAC (with ripple) are of opposite polarity, non-unity power factor operation is achieved using only the diode and the conducting T-leg switch in the expander leg 120. In addition, the T-leg switch remains on for a short period of time after the zero crossing of the input voltage VAC to prevent the expander leg 120 from jumping too quickly between VDC and 0V. The T-leg switch operates at line frequency, so there is little or no switching loss.

[0078] refer to Figure 21 , when there is only a diode in the slow leg, T1 and T2 need to be extended so that the T-leg switch will conduct as long as ILripple is opposite in sign to the input voltage VAC, because the diode can only conduct current in one direction. Therefore, for the leading power factor, T1 can be calculated by the following formula: acmax *sin(ωT1+φ)+|V acmax *sin(ωt)|*DT s / 2L>0, where Iacmax is the peak AC current, φ is the phase shift between the input voltage VAC and the input current IAC, V acmax is the peak AC current amplitude, D is the duty cycle, Ts is the switching cycle.

[0079] When T2 is selected, the T-type foot switch is turned on when the duty cycle is less than 0.05.

[0080] For the lagging power factor, T2 can be calculated by the following formula: acmax *sin(ωT1+φ)-|V acmax *sin(ωt)|*DT s / 2L<0.

[0081] When T1 is selected, the T-type foot switch is turned on when the duty cycle is less than 0.05.

[0082] Now refer to Figure 21 , which reveals the timing of the gate signal generation of the T-leg 122 under the leading power factor condition. Specifically, the waveform chart includes VAC / IAC, REF 136, duty cycle (chopper + expander) 138, midpoint voltage (chopper + expander) and SN / SP gate enable / disable 140, T-leg switch gate signal 142, boost choke current 144 and boost choke current-Ref / filter 146. It is worth noting that Figure 21 Three regions are shown (Region 1, Region 2, and Region 3). Specifically, T-mode modulation (non-PWM, line frequency only) is active in Regions 1, 2, and 3. In Region 1, even though both the input voltage and input current polarity are negative, as shown in waveform diagram 136, waveform diagram 144 shows that the boost choke current begins to reach 0A due to ripple. At this point, oscillation on expander leg 120 can be avoided by activating T-mode. In Region 2, the input voltage and input current polarity are inconsistent, also indicating that T-mode mode needs to be activated. In Region 3, operation cannot switch from T-mode to DP because the PWM modulation phase leg duty cycle saturates. Therefore, operation waits a short time to deactivate the T-type leg switch gate signal. The PWM duty cycle drops to approximately 5%. Figure 21 The waveforms in Figure 1 also reveal that interval T1 is calculated from the time the boost choke current ripple is greater than 0 until the input voltage VAC crosses 0 V. Interval T2 is calculated from the time the duty cycle becomes > 0.05 or the time the minimum threshold value is reached.

[0083] Figures 22 to 23 The changes in region T1 and region T2 are shown for a range of leading and lagging power factors, respectively. For example, Figure 22 The figure reveals the change in the phase angle (ωT1) when the T-gate signal is enabled. The phase shift range is approximately 0.8-0.95 lead, and its values ​​are also shown in the table on the right side of the figure. Figure 23The figure reveals the change in the phase angle (ωT2) when the T-gate signal is enabled. The phase shift ranges from about 0.8 to 0.95 hysteresis, and its values ​​are also shown in the table on the right side of the figure.

[0084] Figure 24A and Figure 24B It is disclosed that the totem pole PFC converter 128 ( Figure 20 ) Some selected waveforms during non-unity power factor operation at a power factor of 0.9 leading operation, and their enlarged views ( Figure 24B ). Figure 24A and Figure 24B The waveform diagram includes VAC / IAC, REF 148 (148A, 148B), T-type leg switch gate signal 150A (and T-type leg switch gate signal Sag signal 150B), Vmunf voltage 152 (152A, 152B), i.e., the midpoint voltage of the expander leg 120, iL1 inductor current 154 (154A, 154B), fast leg gate signal Sg signal 156, and fast leg switch Vds voltage 158. Figure 24A In the figure, the gate control signal 150A of the T-type foot switch is disclosed, and when the active switch S A1 and S A2 When conducting, the midpoint voltage of the expander pin 152A is at VDC / 2. Figure 24B In FIG, an enlarged view of the gate control signal in the high-frequency leg (ie, switches S1 and S2 of the first phase leg) is revealed. As can be seen from the figure, when the AC voltage is positive, a negative average current is generated.

[0085] Figure 25A and Figure 25B It is disclosed that the totem pole PFC converter 128 ( Figure 20 ) Some selected waveforms during non-unity power factor operation with 0.9 lagging operation and their enlarged views ( Figure 25B ). Waveform chart 160~170 and Figure 24A and Figure 24B The waveform diagrams 148 to 158 are similar, so for the sake of brevity, each individual description is omitted here. Figure 25A and Figure 25B It can be observed that the input current ripple can be reduced by interleaving multiple phase legs.

[0086] Figure 26A and Figure 26B Targeted respectively Figure 24A and Figure 24B The waveform diagrams of some types shown in reveal simulation results for leading power factors of 0.796 and 0.955. As the power factor increases, the T-leg 122 is enabled for a longer time.

[0087] Figure 27A and Figure 27B The simulation results reveal that the lagging power factors are 0.796 and 0.955 respectively.

[0088] It is worth noting that, combined with Figures 20 to 25B The control method described will work at any power factor and any current shape.

[0089] Figure 28 and Figure 29 An exemplary control system 172 (ie, Figure 28 172A and Figure 29172B) is used to control the three-phase / single-phase combination AC / DC converters 52 and 118 (including the totem-pole PFC converters 56 and 128). For example, the control system 172A can be used to control the three-phase / single-phase AC / DC converter 52 to achieve non-unity power factor operation and unity power factor operation. Similarly, the control system 172B can be used to control the three-phase / single-phase combination AC / DC converter 118 to achieve non-unity power factor operation and unity power factor operation. In some embodiments, the control system 172A can be incorporated into a separate device from the control system 172B collaborative device, or in some embodiments, the functionality of both can be combined into a single device. The control system 172 can be embodied in a custom or commercially available processor, including a single-core or multi-core central processing unit (CPU), a tensor processing unit (TPU), a graphics processing unit (GPU), a vector processing unit (VPU), or a co-processor among multiple processors, a semiconductor-based microprocessor (in the form of a microchip), a macroprocessor, one or more application-specific integrated circuits (ASICs), a field programmable gate array (FPGA), a plurality of appropriately configured digital logic gates, and / or other conventional electrical configurations, comprising discrete components, both individually and in various combinations, to coordinate the overall operation of the control system 172. When a particular embodiment of the control system 172 is implemented at least in part by software (including firmware), it should be noted that the software can be stored on various non-transitory computer-readable (storage) media for use by or in conjunction with various computer-related systems or methods. In the context of the present disclosure, a computer-readable medium can include an electronic, magnetic, optical, or other physical device or means that can contain or store a computer program (i.e., executable program code or instructions) for use by or in conjunction with the control system 172. When a particular embodiment of the control system 172 is implemented at least in part by hardware, this functionality may be implemented using any one or a combination of the following technologies, all of which are already available in the art: discrete logic circuits having logic gates for implementing logic functions based on data signals, application-specific integrated circuits (ASICs) having appropriate combinational logic gates, programmable gate arrays (PGAs), field-programmable gate arrays (FPGAs), TPUs, GPUs, and / or other accelerators / coprocessors, etc.

[0090] Special reference Figure 28 , which discloses a three-phase / single-phase combined AC / DC converter 52 and a control system 172A for fractional switching loop control. The control system 172A includes various functions that can be implemented in hardware, software, or a combination of both as described above. The control system 172A includes a multiplier 174, a voltage controller 176, a current controller 178, a duty cycle calculator 180A, a counter 182, a first sensor, a second sensor, and a third sensor. Although the operation of the control system 178 is based on Figure 28 The structure is self-explanatory, but is briefly described below. The first sensor is configured to sense the input voltage, the second sensor is configured to sense the input current, and the third sensor is configured to sense the output voltage. Multiplier 174 is connected to the first sensor. Voltage controller 176 is connected between the third sensor and multiplier 174. The measured output voltage and the reference output voltage Voref are input to voltage controller 176, and the magnitude of the reference input current is generated by comparing their errors. Multiplier 174 multiplies the aforementioned magnitude by the unit sine wave generated by the input voltage and by the power factor angle to generate a reference current signal I Lref The current controller 178 is connected to the multiplier 174 and the second sensor, wherein the measured input current and the reference current signal I Lref The input current controller 178 compares the error and provides it to the duty cycle calculator 180A. Duty cycle calculator 180A generates a duty cycle based on the error and the AC voltage amplitude. When the AC voltage is below 8% of the peak AC voltage, fractional switching is enabled. This duty cycle is compared with the sawtooth waveform in counter 182, and gate signals are generated for all active components. In this embodiment, the operation of the three-phase / single-phase AC / DC converter 52 is as previously described and will not be further elaborated here.

[0091] Special reference Figure 29 , which discloses a three-phase / single-phase combination AC / DC converter 118 and a control system 172B for closed-loop control with a T-leg switch. Figure 28 The same numbers are used to mark similar functions in the figure, where the T1 / T2 calculation and the duty cycle calculation are marked as 180B. The error output by the voltage controller 176 is multiplied by the power factor angle through the multiplier 174 to generate the reference current signal I Lref When the T-switch is disabled, the error in the current controller 178 generates a duty cycle. T1 and T2 are calculated based on the AC voltage and reference current. In this embodiment, the operation of the three-phase / single-phase AC / DC converter 118 is as previously described and will not be further elaborated upon.

[0092] Although the present case has been shown and described in detail in the drawings and the foregoing description, these illustrations and descriptions should be regarded as illustrative or exemplary, not restrictive. Therefore, it should be understood that the features mentioned in the appended claims (or the converter and the first and second modulation methods of the preceding paragraphs) are followed by reference symbols, which are only for the purpose of enhancing the intelligibility of the claims and do not limit the scope of the claims or the description in any way. The present case is not limited to the disclosed embodiments. By studying the drawings, the disclosure and the appended claims, those skilled in the art can understand and implement other variations of the disclosed embodiments when practicing the claimed invention. It is worth noting that various combinations of the disclosed embodiments can be used, and therefore, reference to one embodiment or one embodiment does not mean that features in that embodiment are excluded from being used together with features in other embodiments. In the appended claims, the word "comprising" does not exclude other elements or steps, and the indefinite article "a" or "an" does not exclude a plurality.

[0093] The present invention may be modified in various ways by those skilled in the art, but all modifications are within the protection of the appended claims.

Claims

1. A method of operating a totem pole power factor correction converter, wherein the totem pole power factor correction converter comprises: Three input terminals, one neutral terminal, and two output terminals; a plurality of phase legs connected in parallel with each other, wherein each phase leg includes an upper switch and a lower switch connected in series, and a midpoint is defined between the upper switch and the lower switch, and the midpoint of each phase leg is connected to a corresponding input terminal among the three input terminals; a plurality of boost inductors, each of the boost inductors being connected between a corresponding input terminal among the three input terminals and the midpoint of a corresponding phase leg among the plurality of phase legs; a plurality of capacitors, one end of each capacitor being connected between a corresponding boost inductor among the plurality of boost inductors and the input terminal, and the other end of each capacitor being connected to the neutral terminal; a diode bridge arm connected in parallel with the plurality of phase legs and comprising two diodes connected in series, wherein a midpoint is defined between the two diodes, and the midpoint of the diode bridge arm is connected to the neutral terminal; an output filter connected in parallel with the diode bridge arm and comprising two capacitors connected in series, wherein a midpoint is defined between the two capacitors; a T-shaped leg connected between the midpoint of the diode bridge arm and the midpoint of the output filter and comprising two active switches connected in series; as well as A control system configured to execute the method, the method comprising: When the totem pole power factor correction converter operates in a non-unity power factor state, the two active switches of the T-leg are turned on near an AC voltage zero crossing. 2 . The method of claim 1 , wherein operating the upper switch and the lower switch of the plurality of phase legs comprises operating at a duty cycle greater than 0% and less than 100%.

3. The method of claim 1 , wherein operating the plurality of phase legs comprises operating the plurality of phase legs at different duty cycles, and the plurality of phase legs are not phase-shifted relative to each other.

4. The method of claim 1, wherein during non-unity power factor operation, the two active switches of the T-leg are switched at line frequency. 5 . The method of claim 4 , further comprising turning on the two active switches of the T-leg when an inductor current including ripple has a polarity opposite to that of an AC voltage and during non-unity power factor operation.

6. A totem pole power factor correction converter, comprising: three input terminals, a neutral terminal, and two output terminals, wherein the three input terminals are connected to an AC power source; a first switch leg, a second switch leg, and a third switch leg connected to the three input terminals and electrically connected in parallel with each other, wherein each of the first switch leg, the second switch leg, and the third switch leg includes an upper switch and a lower switch connected in series, and has a midpoint electrically connected to a corresponding phase of the AC power source via an inductor; a diode bridge arm electrically connected in parallel with the first switch leg, the second switch leg, and the third switch leg, and comprising two diodes connected in series, wherein the diode bridge arm has a midpoint electrically connected to the neutral terminal; as well as an output filter connected in parallel with the diode bridge arm and comprising two capacitors connected in series, wherein a midpoint is defined between the two capacitors; A T-shaped leg is connected between the midpoint of the diode bridge arm and the midpoint of the output filter and includes two active switches connected in series.

7. The totem pole power factor correction converter as claimed in claim 6 further comprises three capacitors, wherein first ends of the three capacitors are electrically connected to the three phases of the AC power source respectively, and second ends of the three capacitors are connected to the neutral terminal.

8. The totem pole power factor correction converter as claimed in claim 6, further comprising an input filter electrically connected to the three input terminals to filter the single-phase input.

9. The totem pole power factor correction converter as claimed in claim 6, further comprising a control system configured to turn on the two active switches of the T-leg near a zero crossing of an AC voltage.

10. The totem pole power factor correction converter of claim 9, further comprising a control system configured to turn on the two active switches of the T-leg when an inductor current including a ripple has opposite polarities to an AC voltage and during non-unity power factor operation.

11. The totem pole power factor correction converter as claimed in claim 6, wherein the switching operation of the upper switch and the lower switch of the first switch leg, the second switch leg and the third switch leg operates at a duty cycle greater than 0% and less than 100%.

12. The totem pole power factor correction converter as claimed in claim 6, wherein the first switch leg, the second switch leg and the third switch leg operate at different duty cycles, and the first switch leg, the second switch leg and the third switch leg have no phase shift relative to each other.

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