Dual-beam time modulator design method and dual-beam time modulator

Abstract

The invention provides a design method of a dual-beam time modulator. The method comprises the following steps: expanding a time modulation function into Fourier series; keeping the amplitude weighted value of the target sideband consistent, and setting the amplitude weighted values of the other sidebands as 0; setting the initial phase of the target sideband as a certain value, and setting the initial phases of the other sidebands as 0; a + 1 secondary sideband and a-1 secondary sideband are selected as target sidebands, the initial phase of the + 1 secondary sideband is independently controllable, the amplitudes of the + 1 secondary sideband and the-1 secondary sideband are set to be 1, the initial phases are set to be delta phi and 0 respectively, and a dual-beam time modulation function with the adjustable initial phases is constructed; a programmable time modulator architecture is utilized to realize a dual-beam time modulation function with an adjustable initial phase.

Description

Technical Field

[0001] The present invention belongs to the field of radio frequency microwave integrated circuit design, and in particular relates to a dual-beam time modulator design method and a dual-beam time modulator. Background Art

[0002] With the development of wireless communication systems, phased array systems have been widely used in various fields such as intelligence, automation, communications, navigation, and electronic warfare. Phased array systems have become one of the core technologies in many systems. To meet ever-changing needs and challenges, the design, function, and performance requirements of phased array systems are constantly increasing. Due to the radiation characteristics of traditional phased arrays, the beam pointing is controlled by controlling the feed amplitude and phase. Higher precision often means more complex design and higher cost. Especially in the millimeter wave frequency band, the high precision and low cost of traditional phased array systems are more difficult to coexist.

[0003] A time-modulated array (TMA) is a technology that achieves beam steering by periodically adjusting the on-off state of each antenna element (i.e., controlling the antenna's operating time). Unlike traditional phased array systems that achieve beam steering by controlling phase, a time-modulated antenna array introduces a fourth dimension—time. By periodically enabling and disabling the excitation of individual elements to control their phase and amplitude, the antenna array's beam direction is controlled by combining phase difference and time delay. Because TMAs can control the phase and amplitude by controlling the delay and duration of the time modulation sequence through a field-programmable gate array (FPGA), the FPGA's maximum clock frequency also determines the phase resolution of the time modulation. Compared to traditional phased arrays, where the resolution depends on the accuracy of the phase shifter, TMAs exhibit superior high-resolution performance.

[0004] TMA generates sideband signals that radiate at frequencies offset from the carrier frequency by multiples of the time modulation rate. Sideband radiation is generally considered a harmful phenomenon. As the number of sidebands increases, their energy decreases, so only the +1 sideband is often utilized. However, with TMA, sideband radiation is not necessarily harmful. The generated sidebands can be used for multi-beam scanning, achieving the goal of simultaneously receiving signals from multiple directions. Without increasing hardware costs, the system's processing power can be improved simply by optimizing the switching sequence.

[0005] When performing multi-beam scanning with a limited number of beams, the system's spectral degradation and reduced efficiency are acceptable. In this case, if we can overcome the disadvantage of the traditional TMA, in which beam pointing depends on the number of sidebands and the length of the delay, and add a variable to adjust the pointing of each beam, then we can use the existing beams to achieve combined reception of signals at more angles. Therefore, it is necessary to design a multi-beam time modulator in which the pointing of each beam is independently adjustable. Summary of the Invention

[0006] The purpose of the present invention is to provide a dual-beam time modulator to address the problems existing in the above-mentioned prior art.

[0007] The technical solution to achieve the purpose of the present invention is: a dual-beam time modulator design method, comprising:

[0008] Expand the time modulation function into a Fourier series;

[0009] Keep the amplitude weighting value of the target sideband consistent and set the amplitude weighting values ​​of the other sidebands to 0;

[0010] Set the initial phase of the target sideband to a certain value, and the initial phases of the remaining sidebands to 0;

[0011] Select +1 sideband and -1 sideband as target sidebands, and the initial phase of +1 sideband is independently controllable, then the amplitude of +1 sideband and -1 sideband are both set to 1, and the initial phase is set to and 0, constructing a dual-beam time modulation function with adjustable initial phase;

[0012] A dual-beam time modulation function with adjustable initial phase is realized using a programmable time modulator architecture;

[0013] The programmable time modulator architecture includes a differential orthogonal signal generating unit, an I-channel vector synthesis phase shift unit, a Q-channel vector synthesis phase shift unit, a variable gain amplifier unit, a phase shift digital timing control unit, and an amplification digital timing control unit; the phase function is realized by controlling the I-channel vector synthesis phase shift unit and the Q-channel vector synthesis phase shift unit through the phase shift digital timing control unit, and the amplitude function is realized by controlling the variable gain amplifier unit through the amplification digital timing control unit.

[0014] The present invention also proposes a dual-beam time modulator, comprising: a differential orthogonal signal generating unit, an I-way vector synthesis phase shifting unit, a Q-way vector synthesis phase shifting unit, a variable gain amplifying unit, a phase shifting digital timing control unit, and an amplifying digital timing control unit;

[0015] RF input differential signal RF IN + and RF IN– Input the differential orthogonal signal generating unit. The I-channel signal and the Q-channel signal outputted by the differential orthogonal signal generating unit are inputted into the I-channel vector synthesis phase shifting unit and the Q-channel vector synthesis phase shifting unit respectively. The positive outputs of the I-channel vector synthesis phase shifting unit and the Q-channel vector synthesis phase shifting unit are superimposed to obtain the superimposed signal VM+. The superimposed signal VM+ is inputted into the positive pole of the variable gain amplifier unit. The negative outputs of the I-channel vector synthesis phase shifting unit and the Q-channel vector synthesis phase shifting unit are superimposed to obtain the superimposed signal VM-. The superimposed signal is inputted into the negative pole of the VM-variable gain amplifier unit. The output end of the variable gain amplifier unit is also the output end of the dual-beam time modulator. The RF output differential signal RF OUT + and RF OUT –, the output of the phase-shift digital timing control unit is connected to the I-path vector synthesis phase-shift unit and the Q-path vector synthesis phase-shift unit; the output of the amplification digital timing control unit is connected to the variable gain amplification unit; the phase function is realized by controlling the I-path vector synthesis phase-shift unit and the Q-path vector synthesis phase-shift unit through the phase-shift digital timing control unit, and the amplitude function is realized by controlling the variable gain amplification unit through the amplification digital timing control unit.

[0016] Compared with the prior art, the present invention has the following significant advantages:

[0017] 1) Dual-beam: The present invention can generate ±1 sidebands for dual-beam scanning. Compared with single-beam time modulation, its main advantage is that it allows signals to be received in different directions at the same time, enhancing multi-target detection and achieving more efficient beam switching and coverage, thereby greatly improving the system's scanning speed.

[0018] 2) Adjustable initial phase: The initial phase of the +1 sideband generated by the present invention is independently adjustable. Compared with the multiple harmonics of the traditional time modulator, which can only point to the array normal when the delay is 0, the present invention has significantly improved the control ability of the beam and can flexibly design the angle between the two beams, so that the system can realize the combined reception of signals in multiple directions, optimize the scanning efficiency, and greatly improve the flexibility of the system.

[0019] 3) High precision: The time modulator used in the present invention introduces a time parameter to control the phase. The phase shift accuracy is determined by the timing control accuracy of the FPGA. Compared with traditional phase shifters whose phase shift accuracy is restricted by circuit hardware, structure and cost, the present invention can also maintain high phase adjustment accuracy at a low cost.

[0020] 4) High efficiency: The time modulation function of the present invention is fitted by the variable gain amplifier unit in N steps, which can increase the number of steps of the time amplitude function fitted by the variable gain amplifier unit to achieve 4Nf p Harmonics within the bandwidth are suppressed.

[0021] 5) Low phase error: The vector synthesis phase shift unit and variable gain amplifier unit of the present invention both use an impedance-invariant structure. The equivalent gate width of the transistors in the I and Q branches remains unchanged in different phase shift states and gain states, that is, the output impedance of each branch remains unchanged, thereby achieving extremely low phase error. BRIEF DESCRIPTION OF THE DRAWINGS

[0022] Figure 1 is a circuit block diagram of a dual-beam time modulator of the present invention;

[0023] Figure 2 It is a specific implementation circuit of the dual-beam time modulator of the present invention;

[0024] Figure 3 It is the beam pattern of the ±1st sideband with adjustable initial phase generated by the present invention;

[0025] Figure 4 are the amplitude function and phase function of the time modulation function of the present invention;

[0026] Figure 5 It is the step-wise fitting time series of the amplitude and phase functions when the initial phase of the present invention is 0°;

[0027] Figure 6 It is the step-wise fitting time series of the amplitude and phase functions when the initial phase is 45°;

[0028] Figure 7 It is the nine-way control timing of the variable gain amplifier unit by the amplifying digital timing control unit when the initial phase of the present invention is 0°;

[0029] Figure 8 The digital phase shift control unit of the present invention controls the eight-way multiplexing control timing of the I / Q vector synthesis phase shift unit when the initial phase is 0°;

[0030] Figure 9 It is the RMS phase shift error of the +1 sideband of the present invention when it is delayed in the 1 to 10 GHz frequency band.

[0031] Figure 10 It is the RMS phase shift error of the first sideband of the present invention when it is delayed in the 1 to 10 GHz frequency band. DETAILED DESCRIPTION

[0032] In order to make the purpose, technical solutions and advantages of this application more clear, the following further describes this application in detail with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain this application and are not intended to limit this application.

[0033] In one embodiment, combined Figure 1The circuit structure of the dual-beam time modulator is given, which includes: a differential orthogonal signal generating unit 1, an I-way vector synthesis phase shifting unit 2, a Q-way vector synthesis phase shifting unit 3, a variable gain amplifier unit 4, a phase shift digital timing control unit 5 and an amplification digital timing control unit 6. The differential orthogonal signal generating unit is characterized by converting the RF input differential signal RF IN + and RF IN – Converted into four orthogonal differential signals, whose input ports are connected to the differential input ports of the time modulator, whose I-channel differential signal output ports are connected to the differential input ports of the I-channel vector synthesis phase shift unit, and whose Q-channel differential signal output ports are connected to the differential input ports of the Q-channel vector synthesis phase shift unit.

[0034] The I-path vector synthesis phase shift unit modulates the I-path differential signal in time sequence, and together with the Q-path vector synthesis phase shift unit, implements fitting modulation of the phase function of the dual-beam time function with adjustable initial phase. Its differential input port is connected to the I-path differential output port of the differential orthogonal signal generating unit, its positive output port is connected to the positive output port of the Q-path vector synthesis phase shift unit, and then connected to the positive input port of the variable gain amplifier after superposition, and its negative output port is connected to the negative output port of the Q-path vector synthesis phase shift unit, and then connected to the negative input port of the variable gain amplifier after superposition.

[0035] The Q-path vector synthesis phase shift unit is characterized in that it modulates the Q-path differential signal in time sequence, and together with the I-path vector synthesis phase shift unit, realizes fitting modulation of the phase function of the dual-beam time function with adjustable initial phase. Its differential input port is connected to the Q-path differential output port of the differential orthogonal signal generating unit, its positive output port is connected to the positive output port of the I-path vector synthesis phase shift unit, and then connected to the positive input port of the variable gain amplifier after superposition, and its negative output port is connected to the negative output port of the I-path vector synthesis phase shift unit, and then connected to the negative input port of the variable gain amplifier after superposition.

[0036] The variable gain amplification unit amplifies the differential signal in time sequence to achieve fitting modulation of the amplitude function of the dual-beam time function with adjustable initial phase. Its positive input port is connected to the positive output ports of the I-way vector synthesis phase shift unit and the Q-way vector synthesis phase shift unit, its negative input port is connected to the negative output ports of the I-way vector synthesis phase shift unit and the Q-way vector synthesis phase shift unit, and its output port is connected to the differential output port of the time modulator.

[0037] The phase-shift digital timing control unit is characterized by outputting specific high and low levels that change with time through an FPGA, thereby controlling the phase states of the I-path vector synthesis phase shift unit and the Q-path vector synthesis phase shift unit in a timing sequence. Its input port is connected to the FPGA input, and its output port is connected to the control signal input ports of the I-path vector synthesis phase shift unit and the Q-path vector synthesis phase shift unit.

[0038] The amplifying digital timing control unit is characterized by outputting specific high and low levels that change with time through FPGA, controlling the gain state of the variable gain amplifying unit in a timing manner, and its input port is connected to the FPGA input, and the output port is connected to the control signal input port of the variable gain amplifying unit.

[0039] In other embodiments, the I-path vector synthesis phase shift unit, the Q-path vector synthesis phase shift unit, and the variable gain amplification unit may be replaced by other phase shift structures and variable gain amplification structures.

[0040] Combine Figure 2 , the description of the core circuit module is as follows:

[0041] The I-path vector synthesis phase shift unit 2, the Q-path vector synthesis phase shift unit 3, and the variable gain amplifier unit 4 all use a common source and common gate structure. The amplification formula of the common source and common gate structure is shown in formula (1), and the transconductance formula is shown in formula (2). Since the time modulator of the present invention adopts an impedance-constant structure and other variables are kept constant, variable gain amplification can be achieved by changing only the gate width W of the common gate tube.

[0042] |A v |=G m ·R out (1)

[0043]

[0044] A v is the amplification gain, G m is the transistor transconductance, R out is the transistor output impedance, μ n is the electron mobility, C ox is the capacitance per unit area of ​​the transistor, W and L are the width and length of the transistor respectively, V GS and V TH are the transistor gate-source voltage and threshold voltage, respectively.

[0045] I-way vector synthesis phase shift unit 2 Figure 2 As shown in FIG, it includes four common-gate MOS transistor arrays M1, M2, M3 and M4. Each MOS transistor array contains 8 MOS transistors. The gate control signals of MOS array M1 and MOS array M4 are P I <7:0>, the gate control signals of MOS array M2 and MOS array M3 are the inverted signals P IB <7:0>. Each MOS transistor array is divided into 4 pairs, each pair has two MOS transistors with the same gate width, P I <7:6> The gate width of the MOS tube controlled by is W1, P I <5:4> The gate width of the MOS tube controlled by is W2, P I<3:2> The gate width of the MOS tube controlled by is W3, P I The gate width of the MOS tube controlled by <1:0> is W4. The gate width of each MOS tube is different, and the gate width ratio is W1:W2:W3:W4=1-cos(θ1):cos(θ1)-cos(θ2):cos(θ2)-cos(θ3):cos(θ3)=1:2.8:4.3:5, θ k =k×360 / 2 4 (k=1,2,3) In any phase shift state, the number of MOS transistors connected to the I+ and I- branches in the open state and the closed state is the same. The MOS transistor distribution of the Q-path vector synthesis phase shift unit 3 is the same as that of the I-path vector synthesis phase shift unit 2. The corresponding control signal is determined by P I <7:0> and P IB <7:0> changes to P Q <7:0> and P QB <7:0>, in any phase shift state, the number of MOS tubes connected to the Q+ and Q- branches in the open state and the closed state is the same.

[0046] Since each pair of MOS tubes of the I / Q path vector synthesis phase shift unit is affected by the timing control signal P output by the phase shift digital control unit 5, I / Q <2(n-1)+1> and P I / Q <2(n-1)> (n=1,2,3,4) control, the effective gate width in any phase shift state can be calculated by formula (3), here the effective gate width W of I channel is given I,eff .

[0047]

[0048] In a further embodiment, the variable gain amplifier unit 4 is as follows Figure 2 As shown, it includes four common-gate MOS transistor arrays M5, M6, M7 and M8. The gate control signals of MOS array M5 and MOS array M8 are V G <8:0>, the gate control signals of MOS array M6 and MOS array M7 are the inverted signals V GB <8:0>. Each MOS tube array contains 9 MOS tubes, and the gate width of each MOS tube is different. G <8> The gate width of the controlled MOS tube is W G8 , V G <7> The gate width of the controlled MOS tube is W G7 , V G <6> The gate width of the controlled MOS tube is W G6 , V G <5> The gate width of the controlled MOS tube is W G5 , V G<4> The gate width of the controlled MOS tube is W G4 , V G <3> The gate width of the controlled MOS tube is W G3 , V G <2> The gate width of the controlled MOS tube is W G2 , V G <1> The gate width of the controlled MOS tube is W G1 , V G <0> The gate width of the controlled MOS tube is W G0 , the gate width ratio is W G0 :W G1 :W G2 :W G3 :W G4 :W G5 :W G6 :W G7 :W G8 =1-cos(θ G1 ): cos(θ G1 )-cos(θ G2 ): cos(θ G2 )-cos(θ G3 ): cos(θ G3 )-cos(θ G4 ): cos(θ G4 )-cos(θ G5 ): cos(θ G5 )-cos(θ G6 ): cos(θ G6 )-cos(θ G7 ): cos(θ G7 ): 1=1: 2.96: 4.81: 6.48: 7.89: 9: 9.77: 10.16: 52.07, θ Gk =k×360 / 2 5 (k=1,2,…,7), under any gain state, the number of MOS tubes connected to the VM+ and VM- branches in the on state and the off state is the same.

[0049] Each MOS tube of the variable gain amplifier unit 4 is amplified by the control signal V output by the digital control unit 6. G <n>and V GB <n>(N=0,1,2,…,8) control, the effective gate width W of the common gate stage of the variable gain amplifier unit 4 is given by formula (4): V,eff .

[0050]

[0051] Combine Figure 2 The I-path vector synthesis phase shift unit 2, the Q-path vector synthesis phase shift unit 3, and the variable gain amplifier unit 4 all adopt an impedance-invariant structure. Each unit contains a four-MOS transistor array. Under any phase shift / gain state, the total gate width of the equivalently turned-on / off MOS transistors on each branch (I+, I-, Q+, Q-, VM+, VM-) remains unchanged, being half the sum of the gate widths of all MOS transistors on each branch, thereby maintaining a constant output impedance.

[0052] Combine Figure 3 The time modulator of the present invention needs to generate ±1 sidebands and suppress the other useless sidebands. When no delay is applied, that is, when Δt = 0, it can ensure that the beam direction of the -1 sideband remains unchanged, and the beam of the +1 sideband points to a specific direction as needed. The angle between the two sidebands is The time modulation function at this time is as shown in formula (5).

[0053]

[0054] Observe formula (5), α m is the amplitude weighted value on the m-th sideband, is the initial phase weight of the m-th sideband. To use the ±1st sideband for the dual-beam scanning, the amplitudes of the ±1st sideband need to be kept consistent, while the amplitudes of the other sidebands need to be set to 0. To change the initial phase of the +1st sideband, the initial phase of the +1st sideband needs to be set to The initial phase of the -1 sideband is set to 0.

[0055] Therefore, the dual-beam time modulation function of the present invention that can change the initial phase can be obtained as formula (6):

[0056]

[0057] The dual-beam time modulation function with initial phase can be achieved by using amplitude modulation and phase modulation on the radio frequency signal. Based on formula (6), is the time-modulated phase function, is the time modulation amplitude function, combined with Figure 4 , use MATLAB to draw the initial phase of the +1 sideband The amplitude function and phase function when the amplitude function is 0°, the amplitude function is similar to the period T p / 2 cosine absolute value function, the phase function is the period T p , a square wave function whose phase switches between two phases 180° apart, T p is the modulation function period, f p =1 / T p The generated sideband signal is modulated at the frequency f p The frequency of the -1 sideband and the +1 sideband retained by the present invention are (f0-f p ) and (f0+f p ), f0 is the carrier frequency.

[0058] In the simulation of the present invention, the time amplitude function is fitted by 9 steps, and the gains are normalized to sin(kπ / 16), k=0, 1, 2, ..., 8, combined with Figure 7 , the amplified digital timing control unit 6 provides 9-way control timing for the variable gain amplifier unit 4, so that the control timing of the variable gain amplifier unit 4 is approximately Figure 5 The 9-level step-shaped cosine absolute value function has a pulse width corresponding to each gain step of τ k (k=0,1,2,…,8), for example, the normalized gain is sin(8π / 16) and the corresponding pulse width is τ8. When the pulse width satisfies τ8=T p / 32, τ7=3τ8=3T p / 32,τ6=5τ8=5T p / 32, τ5=7τ8=7T p / 32, τ4=9τ8=9T p / 32, τ3=11τ8=11T p / 32,τ2=13τ8=13T p / 32, τ1=15τ8=15T p / 32, τ0=17τ8=17T p / 32, thus completing the fitting of the amplitude function of the time modulation function.

[0059] There are 4 MOS tube arrays in the variable gain amplifier unit 4. Each MOS tube array has 9 MOS tubes with different gate widths. Each MOS tube is affected by the timing control signal V G <n>and its reverse signal V GB <n>(N=0,1,2,…,8) control, the amplification digital timing control unit 6 provides 9 gain control timings for the variable gain amplification unit 4 as shown in Table 1, with a period of T p / 2, where V G <8> The gate width of the controlled MOS tube is W G8 =52.07, V G <8> is fixed to high level "1" to keep the output polarity of variable gain amplifier unit 4 unchanged when the switch states of other MOS tubes change, because when V G <8> When set to a low level "0", the absolute gain state is the same as the existing gain state, but the polarity is changed.

[0060] Table 1

[0061] state Normalized amplitude <![CDATA[V G <0>]]> <![CDATA[V G <1>]]> <![CDATA[V G <2>]]> <![CDATA[V G <3>]]> <![CDATA[V G <4>]]> <![CDATA[V G <5>]]> <![CDATA[V G <6>]]> <![CDATA[V G <7>]]> <![CDATA[V G <8>]]> 1 sin(8×π / 16) 1 1 1 1 1 1 1 1 1 2 sin(7×π / 16) 0 1 1 1 1 1 1 1 1 3 sin(6×π / 16) 0 0 1 1 1 1 1 1 1 4 sin(5×π / 16) 0 0 0 1 1 1 1 1 1 5 sin(4×π / 16) 0 0 0 0 1 1 1 1 1 6 sin(3×π / 16) 0 0 0 0 0 1 1 1 1 7 sin(2×π / 16) 0 0 0 0 0 0 1 1 1 8 sin(1×π / 16) 0 0 0 0 0 0 0 1 1 9 sin(0×π / 16) 0 0 0 0 0 0 0 0 1

[0062] In the simulation of the present invention, the superposition of the I-way vector synthesis phase shift unit 2 and the Q-way vector synthesis phase shift unit 3 can achieve a 4-bit phase shift number, and the phase shift accuracy is 360° / 2 4 =22.5°, combined Figure 8 The phase shift digital timing control unit 5 provides eight-way multiplexing control timing for the I-way vector synthesis phase shift unit 2 and the Q-way vector synthesis phase shift unit 3, so that the I and Q-way control timings are approximately superimposed. Figure 5 The square wave function in has two phases that differ by 180°, thus completing the fitting of the time phase function.

[0063] The I-path vector synthesis phase shift unit 2 and the Q-path vector synthesis phase shift unit 3 each have 4 MOS transistor arrays, each MOS transistor array is divided into 4 pairs, each pair has two MOS transistors with the same gate width, and each pair of MOS transistors with the same gate width is controlled by the timing control signal P I / Q <2(n-1)+1> and P I / Q <2(n-1)> (n=1,2,3,4) and its reverse signal control, I-way vector synthesis phase shift unit 2 is controlled by the phase shift digital timing control unit 5 to generate the control timing P I <7:0> and its inverted signal P IB <7:0> control, Q-way vector synthesis phase shift unit 3 is controlled by the phase shift digital timing control unit 5 to generate the control timing P Q <7:0> and its inverted signal P QB <7:0> control, in order to reduce the number of control signals, the control signals of I and Q paths are multiplexed, that is, P I <7> and P Q <1> Reverse Merger, P I <6> and P Q <0> The merger of P I <5> and P Q <3> The reverse phase merger, P I <4> and P Q <2> Merge, P I <3> and P Q <4> The reverse phase merger, P I <2> and P Q <5> Merge, P I <1> and P Q <6> The merger of P I <0> and P Q <7> Reverse merger, similarly merge their reverse signals.

[0064] The I-channel and Q-channel level states for the phase control timing of 0 to 360 degrees with an accuracy of 22.5 degrees are given in Table 2 and Table 3 respectively. When confirmed, apply and Phase corresponding I, Q path control timing. For example Figure 8 , given when When the phase shift digital timing control unit is 0°, it realizes the control timing of 0° and 180°, and the phase shift digital timing control unit is 0° to 17T. p / 64 and 49T p / 4~T p Time P I <7:0> are all high to achieve a 0° phase. p / 4~49T p / 4 is all low level to achieve a 180° phase.

[0065] Table 2

[0066] state Phase / ° Normalized amplitude <![CDATA[P I <7>]]> <![CDATA[P I <6>]]> <![CDATA[P I <5>]]> <![CDATA[P I <4>]]> <![CDATA[P I <3>]]> <![CDATA[P I <2>]]> <![CDATA[P I <1>]]> <![CDATA[P I <0>]]> 1 0 sin(4×π / 8) 1 1 1 1 1 1 1 1 2 22.5 sin(3×π / 8) 0 1 1 1 1 1 1 1 3 45 sin(2×π / 8) 0 1 0 1 1 1 1 1 4 67.5 sin(1×π / 8) 0 1 0 1 0 1 1 1 5 90 sin(0×π / 8) 0 1 0 1 0 1 1 0 6 112.5 -sin(1×π / 8) 0 1 0 1 0 1 0 0 7 135 -sin(2×π / 8) 0 1 0 1 0 0 0 0 8 157.5 -sin(3×π / 8) 0 1 0 0 0 0 0 0 9 180 -sin(4×π / 8) 0 0 0 0 0 0 0 0 10 202.5 -sin(3×π / 8) 1 0 0 0 0 0 0 0 11 225 -sin(2×π / 8) 1 0 1 0 0 0 0 0 12 247.5 -sin(1×π / 8) 1 0 1 0 1 0 0 0 13 270 sin(0×π / 8) 1 0 1 0 1 0 0 1 14 292.5 sin(1×π / 8) 1 0 1 0 1 0 1 1 15 315 sin(2×π / 8) 1 0 1 0 1 1 1 1 16 337.5 sin(3×π / 8) 1 0 1 1 1 1 1 1

[0067] Table 3

[0068]

[0069]

[0070] Compared to Figure 5 Initial phase The time modulation function at 0° is When the initial phase changes, the amplitude of the time modulation amplitude function remains unchanged, and the overall timing is advanced. The value of the time-modulated phase function is and Switching between two phases, the overall timing is advanced Combine Figure 5 and Figure 6 To observe the initial phase of the +1 sideband The change in the timing of the time modulation amplitude and phase function when changes.

[0071] Figure 5 is the initial phase of the +1 sideband The time modulation amplitude function and phase function of the time modulation amplitude function are fitted in one cycle. The time modulation amplitude function is in the range of 0~T p The peak value is obtained at 8T p / 32~9T p / 32 to obtain the valley value; the time phase function is 0~17T p / 64 and 49T p / 4~T p The phase is 0 at 17T p / 4~49T p / 4, the phase is 180°. Figure 6 The initial phase of the +1 sideband is The time modulation amplitude function and phase function of the time modulation amplitude function are fitted in one cycle. The time modulation amplitude function is at 15T. p / 32~T p / 2, the peak value is obtained at 7T p / 32~8T p / 32 to obtain the valley value; the time phase function is between 0 and 15T p / 64 and 47T p / 4~T p The sum phase is 22.5° at 15T p / 64~47T p When / 64, the phase is 202.5°. Figure 6 Compared to Figure 5 , the timing is advanced by 22.5T p / 4π is T p / 32.

[0072] Delay the time modulation function and get Figure 9 and Figure 10 ±1st sideband RMS phase error diagram.

[0073] When the amplitude function and phase function of the time modulation function are as follows Figure 5 As shown, for the modulation function U(t) of the dual-beam time modulation chip that can change the initial phase, in order to facilitate the calculation of the power suppression ratio, the amplitude weighting value α in formula (5) is m and phase weighted values The replacement is shown in formula (7):

[0074]

[0075] Among them A k is the kth sideband (kf p ) complex amplitude (k=0,±1,…,±∞), under the timing requirements above, derive A k The formula is shown in formula (8):

[0076]

[0077] It can be seen from the above formula that at the required -1st and +1st sidebands, the complex amplitude A k The modulus value is constant. In the simulation of the present invention, the useless sidebands except the (32l±1) sidebands (l=0, ±1, ±2,…, ±∞) are effectively suppressed, and the power suppression ratio of the maximum useless sideband (+33 sideband) reaches -29.83dBc.

[0078] The above description is only a specific embodiment of the present invention. Any feature disclosed in this specification, unless otherwise stated, can be replaced by other equivalent or alternative features with similar purposes; all disclosed features, or all steps in the methods or processes, except for mutually exclusive features and / or steps, can be combined in any way.< / n> < / n> < / n> < / n>

Claims

1. A dual-beam temporal modulator design method, characterized in that: include: Expand the time modulation function into a Fourier series; Keep the amplitude weighting value of the target sideband consistent and set the amplitude weighting values ​​of the other sidebands to 0; Set the initial phase of the target sideband to a certain value, and the initial phases of the remaining sidebands to 0; Select +1 sideband and -1 sideband as target sidebands, and the initial phase of +1 sideband is independently controllable, then the amplitude of +1 sideband and -1 sideband are both set to 1, and the initial phase is set to and 0, constructing a dual-beam time modulation function with adjustable initial phase; A dual-beam time modulation function with adjustable initial phase is realized using a programmable time modulator architecture; The programmable time modulator architecture includes a differential orthogonal signal generating unit, an I-channel vector synthesis phase shift unit, a Q-channel vector synthesis phase shift unit, a variable gain amplifier unit, a phase shift digital timing control unit, and an amplification digital timing control unit; the phase function is realized by controlling the I-channel vector synthesis phase shift unit and the Q-channel vector synthesis phase shift unit through the phase shift digital timing control unit, and the amplitude function is realized by controlling the variable gain amplifier unit through the amplification digital timing control unit.

2. The dual-beam temporal modulator design method according to claim 1, wherein: The specific method of expanding the time modulation function into a Fourier series is: Among them, U(t) is the time modulation function, m is the number of sidebands, α m is the amplitude weighted value on the mth sideband, e is a natural constant, is the initial phase weight value of the m-th sideband, f p is the time modulation frequency, and t is the time modulation delay.

3. The dual-beam temporal modulator design method according to claim 1, wherein: The dual-beam time modulation function with adjustable initial phase is specifically: Where, is the initial phase value of the +1 sideband, is the time-modulated phase function, is the time modulation amplitude function, f p is the time modulation frequency.

4. A dual-beam temporal modulator, characterized in that: include: Differential orthogonal signal generating unit, I-way vector synthesis phase shifting unit, Q-way vector synthesis phase shifting unit, variable gain amplifying unit, phase shifting digital timing control unit and amplifying digital timing control unit; RF input differential signal RF IN + and RF IN – Input the differential orthogonal signal generating unit. The I-channel signal and the Q-channel signal outputted by the differential orthogonal signal generating unit are inputted into the I-channel vector synthesis phase shifting unit and the Q-channel vector synthesis phase shifting unit respectively. The positive outputs of the I-channel vector synthesis phase shifting unit and the Q-channel vector synthesis phase shifting unit are superimposed to obtain the superimposed signal VM+. The superimposed signal VM+ is inputted into the positive pole of the variable gain amplifier unit. The negative outputs of the I-channel vector synthesis phase shifting unit and the Q-channel vector synthesis phase shifting unit are superimposed to obtain the superimposed signal VM-. The superimposed signal is inputted into the negative pole of the VM-variable gain amplifier unit. The output end of the variable gain amplifier unit is also the output end of the dual-beam time modulator. The RF output differential signal RF OUT + and RF OUT –, the output of the phase-shift digital timing control unit is connected to the I-path vector synthesis phase-shift unit and the Q-path vector synthesis phase-shift unit; the output of the amplification digital timing control unit is connected to the variable gain amplification unit; the phase function is realized by controlling the I-path vector synthesis phase-shift unit and the Q-path vector synthesis phase-shift unit through the phase-shift digital timing control unit, and the amplitude function is realized by controlling the variable gain amplification unit through the amplification digital timing control unit.

5. The dual-beam temporal modulator according to claim 4, characterized in that: The I-way vector synthesis phase shift unit includes four common-gate MOS transistor arrays M1, M2, M3 and M4. Each common-gate MOS transistor array includes eight MOS transistors. The gate control signals of the common-gate MOS transistor array M1 and the common-gate MOS transistor array M4 are P I <7:0>, the gate control signal of the common-gate MOS transistor array M2 and the common-gate MOS transistor array M3 is the inverted signal P IB <7:0>; Each common-gate MOS transistor array is divided into 4 pairs, each pair has two MOS transistors with the same gate width, P I <7:6> The gate width of the MOS tube controlled by is W1, P I <5:4> The gate width of the MOS tube controlled by is W2, P I <3:2> The gate width of the MOS tube controlled by is W3, P I The gate width of the MOS tube controlled by <1:0> is W4. The gate width of each pair of MOS tubes is different, and the gate width ratio is W1:W2:W3:W4=1-cos(θ1):cos(θ1)-cos(θ2):cos(θ2)-cos(θ3):cos(θ3)=1:2.8:4.3:5, θ k =k×360 / 2 4 , k = 1, 2, 3, in any phase shift state, the number of MOS tubes on the I+ and I- branches that are in the open state and the number of MOS tubes that are in the closed state are the same.

6. The dual-beam temporal modulator according to claim 5, characterized in that: The structure of the Q-path vector synthesis phase shift unit is the same as that of the I-path vector synthesis phase shift unit.

7. The dual-beam temporal modulator according to claim 4, wherein: The variable gain amplifier unit includes four common-gate MOS transistor arrays M5, M6, M7 and M8. The gate control signals of the MOS array M5 and the MOS array M8 are V G <8:0>, the gate control signals of MOS array M6 and MOS array M7 are the inverted signals V GB <8:0>. Each MOS tube array contains 9 MOS tubes, and the gate width of each MOS tube is different. G <8> The gate width of the controlled MOS tube is W G8 , V G <7> The gate width of the controlled MOS tube is W G7 , V G <6> The gate width of the controlled MOS tube is W G6 , V G <5> The gate width of the controlled MOS tube is W G5 , V G <4> The gate width of the controlled MOS tube is W G4 , V G <3> The gate width of the controlled MOS tube is W G3 , V G <2> The gate width of the controlled MOS tube is W G2 , V G <1> The gate width of the controlled MOS tube is W G1 , V G <0> The gate width of the controlled MOS tube is W G0 , the gate width ratio is W G0 :W G1 :W G2 :W G3 :W G4 :W G5 :W G6 :W G7 :W G8 =1-cos(θ G1 ): cos(θ G1 )-cos(θ G2 ): cos(θ G2 )-cos(θ G3 ): cos(θ G3 )-cos(θ G4 ): cos(θ G4 )-cos(θ G5 ): cos(θ G5 )-cos(θ G6 ): cos(θ G6 )-cos(θ G7 ): cos(θ G7 ): 1=1: 2.96: 4.81: 6.48: 7.89: 9: 9.77: 10.16: 52.07, θ Gk =k×360 / 2 5 , k=1,2,…,7, under any gain state, the number of MOS tubes connected to the VM+ and VM- branches in the on state and the off state is the same.

Images