DC-DC converter with bypass connection

Abstract

The invention relates to a DC-DC converter with a bypass connection. A direct current-direct current (DC-DC) converter includes an output node connectable to a battery pack, a first set of power switches, an isolation circuit, and a second set of power switches connected to the isolation circuit. The second set of power switches is arranged in three switch pairs. A second pair of switches and a third pair of switches are connected in parallel, the third pair of switches being connected to the output node. A boost capacitor is arranged in parallel with the second pair of switches and the third pair of switches. A bypass connection directly connects a power switch in the first switch pair to the output node.

Description

TECHNICAL FIELD

[0001] The present disclosure relates to circuit topologies and control methods for performing direct current-direct current conversion processes.

[0002] INTRODUCTION

[0003] Electric vehicles, backup power sources, power generation stations, and other mobile and stationary battery power systems utilize rechargeable battery packs as direct current (DC) energy storage devices. As constituent electrochemical battery cells of the battery pack are depleted, the cells can be recharged using off-board charging stations. For battery packs having a relatively high voltage capability, such as 400 volt (V) to 800 volt (V) traction battery packs used to energize one or more alternating current (AC) traction motors on a mobile system, fast battery charging can be achieved via a direct current fast charging (DCFC) process. The relatively high charging power during the DCFC phase significantly shortens the completion time of the battery charging event relative to AC-based "Level One" or "Level Two" charging.

[0004] A typical DCFC charging station uses a voltage rectifier connected to AC grid power to convert the AC input waveform to a DC output waveform and provide the necessary power factor correction. A direct current-direct current (DC-DC) converter receives the input voltage from the voltage rectifier and outputs the required charging current to the connected battery pack. A given charging station can be used to charge a fleet of traction battery packs having different voltage capabilities. As such, high voltage charging solutions of 50 kilowatts to 300 kilowatts or higher are typically required to effectively charge battery packs having different battery voltage capabilities. SUMMARY

[0005] Described herein are direct current-direct current (DC-DC) converters, as well as circuits and charging stations using the DC-DC converters. The need to effectively charge battery packs having different battery voltage capabilities can be met by constructing a DC-DC converter according to the present disclosure. In particular, the contemplated converter topology incorporates a switching hardware circuit having multiple pairs of switches and a bypass connection. The bypass connection directly connects a semiconductor-based power switch in one of the pairs of switches to the charging battery pack.

[0006] DC-DC converters can be constructed from multiple pairs of switches, which in some implementations can be packaged in separate power modules. The converters can include a dual active bridge (DAB) configuration (i.e., a power side bridge and a battery side bridge) and an inductor-capacitor (LC) circuit. While the disclosed converters can be bidirectional in their design, in certain implementations, such as when charging a propulsion battery pack of an electric vehicle, circuit components of the converter can be used to step up the charging voltage to the always higher voltage level required to charge the connected battery pack, as the case can be.

[0007] To reduce losses associated with the step-up, the converters in the embodiments described herein include the bypass connection described above, i.e., a conductive wire or trace. In one or more implementations, the bypass connection directly connects a switch of the battery side bridge to the battery pack. Thus, using the bypass connection as described in detail herein enables the step-up while reducing the required charging current of the battery pack.

[0008] Embodiments of a DC-DC converter include an output node connectable to a battery pack, a first set of power switches, an isolation circuit, and a second set of power switches. The power switches in the second set are connected to the isolation circuit and arranged in three pairs of switches. The three pairs of switches. A second pair of switches and a third pair of switches of the three pairs of switches are connected in parallel, the third pair of switches being connected to the output node. A step-up capacitor is arranged in parallel with the second pair of switches and the third pair of switches. The bypass connection directly connects a power switch of the first pair of switches to the output node. The foregoing summary is not intended to represent every implementation or aspect of the present disclosure. Rather, the foregoing summary illustrates certain novel aspects and features that, alone or in combination, can represent an improvement over the prior art. The foregoing summary, as well as the following detailed description of representative implementations and modes of the present disclosure, will be better understood in conjunction with the accompanying drawings and the appended claims. BRIEF DESCRIPTION OF DRAWINGS

[0009] The drawings described herein are for purposes of illustration only and are not intended to limit the scope of the present disclosure. It is to be understood that the drawings are not to scale and that, unless otherwise specifically noted, the drawings are meant as illustrative examples, and not as a defining as to the scope of the disclosure.

[0010] Figure 1 is a schematic diagram of an electrical system having a direct current-direct current (DC-DC) converter with a bypass connection constructed in accordance with the present disclosure.

[0011] Figure 1A is a schematic diagram of a DC-DC converter in Figure 1 in accordance with an alternative construction.

[0012] Figure 2A is a representative time diagram of an intermediate voltage in a first stage of the exemplary DC-DC converter shown in Figure 1 is a representative time diagram of an intermediate voltage in a first stage of the exemplary DC-DC converter shown in

[0013] Figure 2B is Figure 1 a representative time diagram of the output current and the boost current of the DC-DC converter shown in

[0014] Figure 2C is Figure 1 a representative time diagram of the primary transformer current of the first stage of the DC-DC converter shown in

[0015] Figure 3A is Figure 2A a representative time diagram of the intermediate voltage in over a shorter time interval in

[0016] Figure 3B is Figure 2B a representative time diagram of the output current and the boost current in over a shorter time interval in

[0017] Figure 3C is Figure 2C a representative time diagram of the primary transformer current in over a shorter time interval in

[0018] The present disclosure can be modified in alternative forms or embodied in other ways, wherein representative embodiments are shown in the drawings and are described in detail below. The inventive aspects of the present disclosure are not limited to the disclosed embodiments. Rather, the present disclosure is intended to cover alternatives that fall within the scope of the present disclosure as defined by the appended claims. DETAILED DESCRIPTION

[0019] With reference to the drawings, wherein like reference numerals refer to like or similar components throughout the several views, Figure 1 a circuit 10 is shown having a direct current-direct current (DC-DC) converter 12, a voltage rectifier 13, and respective first and second inductors LI and L2. In a non-limiting scenario, the circuit 10 is operable to charge a rechargeable battery pack 16 after first converting an alternating current (AC) source voltage (V PFC ) from an AC power source 18. The converter 12, as described in detail herein, is configured to be connected to the battery pack 16 at an output node N6 via the inductor L2 (shown far right in Figure 1 ) during a direct current fast charging (DCFC) process. The inductor L2 and the output node N6 can be incorporated into a charging cord / charging connector of an electric vehicle supply equipment (EVSE) charging station.

[0020] When charging the connected battery pack 16, the Figure 1The DC-DC converter 12 in the embodiment of the present invention performs the DC-DC conversion process to achieve a reduced current and a wide output voltage range during charging operation with minimal electrical losses. The converter 12 described herein does not use a buck stage to reduce the voltage, but instead incorporates a bypass connection 14 around the circuit components used to perform the boost function. The bypass connection 14 connects one of the three switch pairs (SP1, SP2, SP3) of the converter 12 (specifically, the first switch pair SP1) via the output node N6 (at Figure 1 The bypass connection 14 shown in FIG. 1 is directly connected to the battery pack 16. Using the bypass connection 14 shown also has the benefit of reducing current flow during boost operation, thereby increasing the operating efficiency of the converter 12.

[0021] exist Figure 1 In the representative circuit topology of FIG, battery pack 16 may alternatively embody lithium-ion, nickel metal hydride (NiMH), nickel cadmium (NiCd), or another battery chemistry suitable for the application. For example, battery pack 16 may be configured as a high voltage rechargeable battery pack for powering electric vehicles or stationary power stations, or as a backup energy source for residential or commercial buildings. In one or more specific implementations, "high voltage" refers to, but is not limited to, approximately 400V to 800V or 1000V or higher.

[0022] The DC-DC converter 12 may be connected to an AC power source 18, such as grid power, via a voltage rectifier 13. That is, the voltage rectifier 13 may be connected to the AC power source 18 to receive an AC input waveform therefrom, and the voltage rectifier 13 may output a DC voltage waveform as an input voltage (V PFC Although shown schematically for simplicity of illustration, the voltage rectifier 13 may be configured to provide power factor correction (PFC) as desired.

[0023] In one or more embodiments, Figure 1 The components of the DC-DC converter 12 in can be packaged as a first power module 20 and a second power module 30. In this instance, the "module" may include the hardware components shown and may include a protective housing (not shown) to protect such components from moisture and debris. The second power module 30 can be connected to the first power module 20 and configured to increase its output voltage level. The input side 200 of the first power module 20 is connected to the voltage rectifier 13. In addition, the first power module 20 and the second power module 30 can be isolated from each other by an intermediate n:1 transformer 21, where n is the turns ratio of the primary (P) winding and the secondary (S) winding of the transformer 21. In some specific implementations, the transformer 21 can be constructed as a 1:1 transformer, i.e., n=1, but the present disclosure is not limited to such embodiments.

[0024] Still referring to Figure 1 , the DC-DC converter 12 includes a plurality of semiconductor-based power switches. Specifically, the example first power module 20 can include a first set of power switches (S1, S2, S3, and S4) arranged as a power side bridge, as shown. A second set of power switches arranged in the three switch pairs SP1, SP2, and SP3 described above (i.e., six power switches S5, S6, S7, S8, S9, and S10) are hardware components of the second power module 30, as described below. In various embodiments, the various power switches S1-S6 can be constructed as silicon-based or silicon carbide-based metal oxide semiconductor field effect transistors (MOSFETs), silicon-based insulated gate bipolar transistors (IGBTs), or wide band gap (WBG) gallium nitride (GaN) switches, by way of example and without limitation.

[0025] The present teachings allow for an increased intermediate voltage (V + ) between the positive supply rail 111 mid and the negative supply rail 111-. + Accordingly, the voltage rating of the individual switches in the switch pairs SP2 and SP3 (i.e., the power switches S7, S8, S9, and S10 in the representative circuit topology of Figure 1 ) exceeds the voltage rating of the power switches S5 and S6 in the remaining one of the three switch pairs (i.e., the switch pair SP1). For example, in one possible implementation, the voltage rating of the power switches S7, S8, S9, and S10 in the switch pairs SP2 and SP3 can be at least about 800 V to about 1000 V. However, the actual voltage rating will depend on the particular voltages and currents of the implemented circuit 10. To this end, the power switches S7, S8, S9, and S10, when constructed as power switches, can benefit from being constructed from silicon carbide (SiC) material, e.g., a SiC MOSFET in one possible non-limiting implementation.

[0026] The power switches S1-S4 of the DC-DC converter 12 can be arranged as an H-bridge. As understood in the art, an H-bridge is generally composed of four switching elements, such as MOSFETs, transistors, or other suitable switching elements, arranged in a bridge configuration with a load connected between two center nodes (N1, N2) of the H-bridge, as shown. The power switches S1 and S3 are electrically connected to the positive voltage rail 11 + of the converter 12, and thus function as the nominal “upper” switches in the circuit 10. The power switches S2 and S4 are electrically connected to the negative voltage rail 11 - , i.e., electrical ground, and thus function as the nominal “lower” switches, i.e., “upper” and “lower” describing the positive voltage rail connection and the negative voltage rail connection, respectively.

[0027] Power switches (S1, S2) are connected at node N1. Similarly, power switches (S3, S4) are connected at node N2. Nodes N1 and N2 in turn are connected to opposite ends of a primary winding (P) of transformer 21 to drive isolation circuit 23, which in Figure 1 the illustrated configuration, the LC circuit implementation of isolation circuit 23 is not operated near its resonant frequency. Here, power can be converted via phase-shift operation.

[0028] As understood in the art, phase-shift control (change in the relative time or position of a waveform or signal compared to a reference signal or waveform) and frequency control (change in the switching frequency relative to the resonant frequency of the LC circuit) are used in the control of LC circuits to effect power conversion. That is, when regulating the output voltage or current, the voltage-current waveform phase or frequency relationship can be adjusted. In particular, the LC circuit of isolation circuit 23, as illustrated in Figure 1 the illustrated configuration, the LC circuit implementation of isolation circuit 23 is not operated near its resonant frequency. Here, power can be converted via phase-shift operation. Figure 1 The first capacitor C1 in the non-limiting topology in

[0029] Still referring to Figure 1 , three switch pairs SP1, SP2 and SP3, which collectively include power switches S5 to S10, are connected to a secondary winding (S) of transformer 21 via intervening isolation circuit 23. In the illustrated arrangement, power switches (S5, S6) forming first switch pair SP1 are connected together at node N3. Similarly, power switches (S7, S8) forming second switch pair SP2 are connected at node N4. Power switches (S9, S10) in turn form third switch pair SP3, with power switches S9 and S10 connected together at node N5. In this implementation, node N3 is connected to isolation circuit 23, while node N4 is connected to the secondary winding (S) of transformer 21.

[0030] Figure 1The third switch pair SP3 is arranged in parallel with the second switch pair SP2 in the DC-DC converter 12 to form a boost stage, wherein the switch pairs SP2 and SP3 are connected via respective positive voltage rails 111 + and negative voltage rails 111- to the second capacitor C2 (hereinafter referred to as boost capacitor C2). The boost capacitor C2 arranged in parallel with the switch pairs SP2 and SP3 carries an elevated intermediate voltage (V mid ). The switch pair SP3 is directly connected to the output node N6. During a DCFC charging event, the node N5 provided at the output side 300 of the second power module 30 is able to be connected to the battery pack 16 via the boost inductor L2. The connection of the node N5 to the battery pack 16 is achieved via the output node N6. In one or more implementations, the output node N6 can be a charging connection point of an EVSE charging station using the DC-DC converter 12, e.g. its charging coupler (not shown).

[0031] As shown in Figure 1 , in a representative charging operation, the connected battery pack 16 at a battery voltage level (V at ) receives a charging output current (i out ) from the output node N6. The output current (i out ) in this topology is the sum of a boost current (i L ) through the second inductor L2 and a bypass current (i B ) which bypasses the boost stage (switch pairs SP2 and SP3) via the bypass connection 14. That is, the switch pair SP1 is directly connected to the battery pack 16 via the output node N6 and the bypass connection 14, i.e. instead of connecting the power switch S5 to the positive voltage rail 111 + . Thus, Figure 1 the topology in aims to provide an increased voltage during the boost phase of operation with increased charging efficiency without requiring additional hardware or circuit complexity.

[0032] As mentioned above, the four power switches S1 to S4 and four of the six remaining power switches S5 to S10 together form a dual active bridge (DAB) with a total of eight power switches (i.e. S1 to S8). The power switches S1 to S4 are arranged to form a supply side bridge. The switches S5 to S8 are arranged in this embodiment to form a battery side bridge, wherein the terms “supply side” and “battery side” denote the proximity with respect to the AC power source 18 and the battery pack 16, respectively. In the envisaged embodiment, only one switch pair (i.e. the second switch pair SP2) of the battery side bridge is elevated to a higher voltage. The second switch pair SP2 and the third switch pair SP3 together output an average voltage over the boost capacitor C2, as shown in the figure, thereby forming the intermediate voltage (V mid ).

[0033] The boost operation of the DC-DC converter 12 in Figure 1A involves the controlled operation of the power switches S1-S4, the transformer 21, the isolation circuit 23, and the first switch pair SP1 and the second switch pair SP2 (specifically, the switches S5-S8). The first capacitor C1 of the isolation circuit 23 as described above blocks any DC offset between the two switch pairs SP2 and SP3, essentially averaging the applied voltage to the boosted intermediate voltage (V mid ).

[0034] With brief reference to Figure 1 , the skilled person will appreciate that Figure 1A the bypass connection 14 in Figure 1 may extend between the switch S6 and the output node N6. The battery pack 16 in such an implementation is connected to the common positive voltage rail 111 + and the output node N6. In fact, Figure 1 the solution in + replaces the common ground in Figure 1 with the common positive voltage rail 111 + .

[0035] Control loops: Figure 2A The control of the representative DC-DC converter 12 in may occur in two dynamically decoupled control loops, the operation of which is independently regulated by a specific controller (i.e., the first control processor 40A (CP-A) or the second control processor 40B (CP-B), e.g., a microprocessor, central processing unit, or integrated circuit programmed and thus operable to control the functions of the circuit 10 described herein). The respective first control processor 40A and second control processor 40B or a further plurality of control processors can receive input signals (CC bat , CC PFC ) in the form of, for example, reported battery voltage (V IN-A ), source voltage (V IN-B ), on / off conduction states of various power switches (S1-S10), temperature, etc.

[0036] In response to the input signals (CC IN-A , CC IN-B ), the respective first control processor 40A and second control processor 40B are configured to output corresponding control signals (CC OUT-A , CC OUT-B ) to the various switches to control the power flow on the circuit 10 as needed and to independently control the on / off conduction states of different power switch groups, e.g., the power switches S9 and S10 are controlled by the second control processor 40B to control the boost voltage (intermediate voltage V mid ), and the remaining power switches S1-S8 are controlled via the first control processor 40A to control the power flow.

[0037] Using a corresponding control process, the first control processor 40A is operable to control the corresponding on / off conduction states of the eight power switches S1-S8 and the state of the isolation circuit 23. That is, the first control processor 40A can be used to help control the power flow on the isolated converter via state control of the various power switches S1, S2, S3, S4, S5, S6, S7, and S8. In one or more implementations, the first control processor 40A can control the dual active bridge (DAB), i.e., the power switches forming the above-mentioned power supply side bridge and battery side bridge, via single phase shift control as a possible first process. As understood by those skilled in the art, this entails dynamically changing the phase angle to control the power flow on the transmission line, e.g., using a tap to introduce a controllable voltage into the magnetic circuit of the transformer 21 via the secondary winding.

[0038] The power flow on the isolation circuit 23 can be achieved via variable frequency control as a second process, with other possible methods possibly being used within the scope of the present disclosure. As understood by those skilled in the art, the power flow in an LC circuit is determined by the impedance of the circuit. The impedance, in turn, is a function of the frequency of the applied voltage (or current). As such, the frequency of the applied signal can be changed to change the impedance of the LC circuit. In resonant converters, the switching frequency can be varied around the resonant frequency of the LC circuit to control the power flow and regulate the output voltage. Thus, implementing variable frequency control in the DC-DC converter 12 can include using a feedback control loop to monitor the output voltage and adjust the input frequency.

[0039] The second boost circuit controlled via the second control processor (CP-B) 40B can be used to control the intermediate voltage (V mid ), which can be achieved by varying the duty cycle of the switches (S9, S10) in the third switch pair SP3. As used herein, the duty cycle (expressed as a percentage or ratio between 0 and 1) is the ratio of the time a switch is in the on / conductive state to the time that same switch is off / non-conductive. Thus, a 0% duty cycle is always off, and a 100% duty cycle is always on. In the context of the DC-DC converter 12, the duty cycle variation can be achieved, for example, by changing the ratio of switch S10 on / switch S9 off, or vice versa.

[0040] To this end, the intermediate voltage (V mid ) can be set by the second control processor 40B according to the following equation:

[0041] V mid = 2 / PFC -V bat

[0042] where V PFCis the source voltage from the voltage rectifier 13, and V bat is the voltage capability of the battery pack 16 (“battery voltage”). Thus, the intermediate voltage (V PFC ) can be determined from the supply voltage (V bat ) and the battery voltage (V mid ) of the battery pack 16. If V bat exceeds V PFC , the second control processor 40B can cause the power switch S9 to be in an on state, i.e., a conducting state, such that the intermediate voltage V mid is equal to the battery voltage V bat .

[0043] In the present approach, the intermediate voltage V mid can be set much higher than the usual voltage in a conventional two-stage converter. For example, in some implementations, in a non-limiting example 100 kW charging event from a 700 V input, the intermediate voltage V mid can be set to be about 30% to 35% higher using the 1 : 1 transformer 21 for a battery pack 16 at 450 V and a boost voltage (V mid ) of 950 V. The relatively higher intermediate voltage (V mid ) helps to reduce losses in the converter 12, albeit at the cost of requiring higher rated voltages for the power switches S7, S8, S9, and S10. Due to the 1 : 1 configuration of the transformer 21, a high voltage implementation of the battery pack 16 (e.g., 700 V to 800 V or higher) can be charged with the power switch S9 always on / conducting, at which time only half of the battery current flows through the power switch S9. That is, in the case where the supply voltage (V PFC ) is approximately equal to or exceeds the battery voltage (V bat ), nothing will happen during the boost, with the bypass connection 14 ensuring that only 50% of the charging power is delivered through the inactive switch assembly.

[0044] Reference is now made to Figure 2B , Figure 2C and Figure 1 , showing the operation of the DC-DC converter 12 in Figure 2A over a representative time interval via traces 50, 52, 54, and 56, where time (t) is expressed in milliseconds (ms). In this example, both the DAB control and the boost control start at t = 0 and continue until about t = 1.6 ms, after which the DC-DC converter 12 is controlled to steady state operation. That is, in this non-limiting example case, Figure 2B trace 50 in mid represents the intermediate voltage (V mid)’s possible trajectory.

[0045] Figure 1 The traces 52 and 54 in FIG. 5 show the output current (i out ) and boost current (i L ), which is located in Figure 3B In this example, the output current (i out ) decreases rapidly during the boost phase and eventually settles to a relatively steady-state value of about 15A by completing the boost phase at t=1.6ms. At this point, the boost current (i L ) continues to change with sawtooth / triangle wave (see Figure 1 ), which in this example varies between about 0A and about 10A. Over the same boost interval, Figure 3C The primary current (I p ) gradually decreases to a smooth, well-defined oscillation range, which in this exemplary case is about ±8 A (see Figure 3A Note that the boost current (i L ) is less than the output current (i out ), which means that the bypass connection 14 effectively reduces the current flowing through the corresponding second switch pair SP2 and third switch pair SP3.

[0046] At a representative shorter time interval of 1.6ms to 1.65ms, Figure 3B 、 Figure 3C and Figure 2A Corresponding to Figure 2B 、 Figure 2C and Figures 2A to 2C To better show Figure 1 The traces 50, 52, 54 and 56 in FIG1 and continue beyond the boost phase to further illustrate steady-state operation. After the boost phase is complete, the boosted intermediate voltage (V mid ) oscillates within a narrow window around its average value, which in this representative case is about 860 V, to provide a Figure 3B The continuous output voltage at which the connected battery pack 16 is charged. Similarly, Figure 3C The output current (i out ) steady-state properties, where the boost current (i L ) remains well below the output current (i out ) and the boost current (i L ) provides output current (i out ) Figure 1 Trace 56 will arrive at Figure 1 The primary current (iP ) appear as smooth, approximately sinusoidal waveforms, indicating reduced losses and increased efficiency.

[0047] Thus, use of the bypass connection 14 extending from the power switch pair SP1 to the output node N6 enables boosting to high voltage with reduced current during the boost phase. Possible collateral benefits of the foregoing teachings include increased efficiency of stationary EV chargers required to charge battery packs 16 having different voltage capabilities (e.g., about 400V to 800V or higher), and a wide range of other applications utilizing boost power supplies and power generation stations. Among other possible users, providers of EVSE stations that need to support many different battery voltage levels can benefit from ​ alternative circuit topologies in FIG. 21.

[0048] While several modes for carrying out the present teachings have been described in detail, those familiar with the art to which this teaching relates will recognize various alternative aspects for practicing the present teachings within the scope of the appended claims. The above description and drawings are illustrative and exemplary of the entire range of alternatives that are implicitly and explicitly present in the contents of this disclosure, and are not limiting of the scope of alternatives that can be implemented based on the contents of this disclosure. Furthermore, the concepts described herein expressly include combinations and sub-combinations of the elements and features described. The detailed description and drawings are supportive and descriptive of the present teachings, but the scope of the present teachings is defined solely by the claims.

Claims

1. A direct current (DC-DC) converter, comprising: an output node connectable to a battery pack; The first group of power switches; Isolation circuit; and a second group of power switches connected to the isolation circuit and arranged in three switch pairs, the three switch pairs including a first switch pair, a second switch pair, and a third switch pair, wherein the second switch pair and the third switch pair are connected in parallel, and wherein the third switch pair is connected to the output node; a boost capacitor arranged in parallel with the second switch pair and the third switch pair; and A bypass connection connects the power switch of the first switch pair directly to the output node.

2. The converter of claim 1, wherein the isolation circuit comprises an inductor-capacitor (LC) circuit.

3. The converter according to claim 1, wherein: The first set of power switches includes four power switches arranged to form an H-bridge; and The four power switches in the first group of power switches and the four power switches in the second group of power switches together form a dual active bridge (DAB) having eight power switches.

4. The converter according to claim 3, further comprising: A control processor is operable to control the corresponding conductive states of the eight power switches and the state of the isolation circuit using corresponding control processes.

5. The converter according to claim 4, wherein the corresponding control process includes single phase shift control and variable frequency control. 6 . The converter of claim 1 , wherein voltage ratings of the switches in the second and third switch pairs exceed voltage ratings of the power switches in the first switch pair. 7 . The converter of claim 6 , wherein the voltage rating of the power switches in the second and third switch pairs is at least about 900 volts. 8 . The converter of claim 1 , wherein the power switches in the second switch pair and the third switch pair are silicon carbide (SiC) switches.

9. The converter of claim 8, wherein the SiC switch comprises a SiC metal oxide silicon field effect transistor (MOSFET).

10. The converter according to claim 1, further comprising: A control processor is operable to vary a duty cycle of the power switches in the second and third switch pairs.

11. The converter of claim 10 , wherein the control processor operable to vary the duty cycle of the power switches in the second and third switch pairs is operable to set the boosted intermediate voltage on the boost capacitor based on the supply voltage and the battery voltage of the battery pack.

12. The converter according to claim 11, wherein: The control processor operable to change the duty cycle of the power switch is configured to maintain the power switch of the third switch pair in an on state when the battery voltage exceeds the supply voltage so that the intermediate voltage is equal to the battery voltage.

13. A circuit for charging a battery pack, comprising: a voltage rectifier connectable to an alternating current (AC) voltage source, the voltage rectifier being configured to rectify an AC input waveform from the AC voltage source to generate a direct current (DC) output waveform; and a direct current to direct current (DC-DC) converter, connected to the voltage rectifier, comprising: Output node; a first power module having four power switches arranged as an H-bridge; and a second power module, connected to the first power module and configured to increase an output voltage level of the first power module, the second power module comprising: a first switch pair, a second switch pair, and a third switch pair of power switches, wherein the second switch pair and the third switch pair are connected in parallel; a boost capacitor arranged in parallel with the second switch pair and the third switch pair; and A bypass connection connects the power switch of the first switch pair directly to the battery pack via the output node.

14. The circuit of claim 13, further comprising: transformer; and an inductor-capacitor LC circuit, wherein the first power module is connected to the second power module via the transformer and the LC circuit.

15. The circuit of claim 14 , wherein the four power switches of the first power module and four of the power switches of the second power module form a dual active bridge (DAB) having eight power switches, the circuit further comprising: a first control processor operable to control corresponding conductive states of the eight power switches and a state of the isolation circuit using respective control processes, wherein the respective control processes include single phase shift control and variable frequency control; and A second control processor is operable to vary a duty cycle of a power switch in the third switch pair. 16 . The circuit of claim 13 , wherein voltage ratings of the power switches in the second and third switch pairs exceed voltage ratings of the power switches in the first switch pair.

17. The circuit of claim 16, wherein the power switches in the second and third switch pairs are silicon carbide (SiC) switches.

18. A power module comprising: Output node; three switch pairs, the three switch pairs comprising a first switch pair, a second switch pair, and a third switch pair, wherein the second switch pair and the third switch pair are connected in parallel, and wherein a rated voltage of the power switches in the second switch pair and the third switch pair exceeds a rated voltage of the power switch in the first switch pair; a boost capacitor arranged in parallel with the second switch pair and the third switch pair; a bypass connection connecting the power switch of the first switch pair directly to a battery pack via an output node; and A control processor is configured to control states of the power switches in the three switch pairs.

19. The power module according to claim 18, wherein: The control processor is operable to selectively maintain the power switch of the third switch pair in an on conductive state.

20. The power module of claim 18, wherein the power switches in the first and second switch pairs comprise silicon carbide (SiC) power switches.

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