Off-line parameter identification method of position-sensorless permanent magnet synchronous motor

Abstract

The invention relates to an off-line parameter identification method of a position-sensorless permanent magnet synchronous motor, which belongs to the technical field of permanent magnet synchronous motor control and parameter identification, and comprises the following steps: S1, constructing a voltage mathematical model of an embedded permanent magnet synchronous motor IPMSM under d and q coordinate systems; s2, establishing an inverter nonlinear compensation strategy considering parasitic capacitance; s3, considering an inverter non-linear compensation strategy considering parasitic capacitance into resistance identification, and identifying the resistance; s4, optimizing a high-frequency injection scheme, and identifying dq-axis inductance by adopting a simplified inductance identification method based on signal filtering and peak value extraction; and S5, considering an inverter non-linear compensation strategy considering stray capacitance into flux linkage identification, and proposing a dq-axis flux linkage decomposition method without a position to identify the flux linkage.

Description

Technical Field

[0001] The present invention belongs to the technical field of permanent magnet synchronous motor control and parameter identification, and relates to an off-line parameter identification method for a position sensorless permanent magnet synchronous motor. Background Art

[0002] Permanent magnet synchronous motors (PMSMs) are widely used in industry, transportation, aerospace, and renewable energy power generation due to their high efficiency, high power density, and high reliability. Achieving high-performance control (such as field-oriented control) requires precise measurement of key parameters such as stator resistance, rotor flux, and dq-axis inductance. However, these parameters are difficult to measure directly and vary with temperature, manufacturing tolerances, and magnetic saturation. Therefore, high-precision offline parameter identification before operation is crucial. However, current offline identification algorithms are not universally applicable. For stator resistance identification, both DC injection and gradient current differential methods, while theoretically able to eliminate voltage errors caused by inverter nonlinearity, ignore the nonlinear effects of parasitic capacitance on the inverter output voltage at varying currents, resulting in reduced identification accuracy in the low-current region. Pulse injection and harmonic injection are commonly used to identify dq-axis inductance. The pulse injection method requires combining the injected voltage and induced current signals to first identify the resistance, and then calculate the inductance through differentiation. This can cause resistance errors to accumulate in the inductance identification results. Furthermore, when identifying the q-axis inductance, the motor may not be able to remain stationary, resulting in identification failure. The high-frequency sinusoidal voltage injection method calculates the inductance by using the relationship between the injected voltage amplitude and frequency and the induced current. When identifying the q-axis inductance, a current bias is typically injected into the d-axis to fix the rotor position. However, this method requires discrete Fourier transform extraction of voltage and current information, which is computationally complex. When identifying the flux, the lack of position information means that the identified flux is not distributed solely along the q-axis, but rather has flux components on both the d and q axes.

[0003] Therefore, existing inverter nonlinearity compensation schemes primarily target high-current operating conditions and are unable to effectively compensate for the voltage loss caused by the parasitic capacitance of switching devices in the low-current range. This voltage loss introduces significant errors in stator resistance identification. Furthermore, existing inductance identification methods often rely on complex discrete Fourier transform signal preprocessing, which is cumbersome. Furthermore, in sensorless scenarios, the accuracy of rotor permanent magnet flux identification is difficult to guarantee. Summary of the Invention

[0004] In view of this, the purpose of the present invention is to provide an offline parameter identification method for a position sensorless permanent magnet synchronous motor, which adopts a DC volt-ampere method considering nonlinear compensation of parasitic capacitance to identify the stator resistance of the motor, a sinusoidal high-frequency voltage injection method to identify the dq-axis inductance of the motor stator, and a virtual dq-axis flux decomposition method to realize permanent magnet flux identification without the need for a position signal; through modeling and analysis of the inverter, an error voltage compensation method under arbitrary current conditions considering parasitic capacitance is proposed; offline identification under easy working conditions is realized, and the accuracy of offline identification is improved to provide accurate parameters and initial values ​​for subsequent online parameter identification and control.

[0005] In order to achieve the above object, the present invention provides the following technical solutions:

[0006] An offline parameter identification method for a position sensorless permanent magnet synchronous motor comprises the following steps:

[0007] S1: Construct the voltage mathematical model of the embedded permanent magnet synchronous motor IPMSM in the d and q coordinate systems;

[0008] S2: Establish an inverter nonlinear compensation strategy considering parasitic capacitance;

[0009] S3: Taking the inverter non-linear compensation strategy considering parasitic capacitance into consideration in the resistance identification, the resistance is identified;

[0010] S4: Optimize the high-frequency injection scheme and use a simplified inductance identification method based on signal filtering and peak extraction to identify the dq-axis inductance;

[0011] S5: The inverter nonlinear compensation strategy considering parasitic capacitance is taken into account in the flux identification, and a dq-axis flux decomposition method without position is proposed to identify the flux.

[0012] Furthermore, the voltage mathematical model of IPMSM in the d, q coordinate system is:

[0013]

[0014] Among them, U d 、U q is the d and q axis voltage of the motor, R s is the stator resistance of the motor, i d 、i q is the motor d and q axis current, ψ q , ψ q is the motor d and q axis magnetic flux, ω e is the electrical angular velocity of the motor rotor, L d , L q is the motor d and q axis inductance

[0015] Furthermore, step S2 specifically includes the following steps:

[0016] S21: Considering the influence of parasitic capacitance, calculate the voltage loss caused by the nonlinearity of the inverter;

[0017] S22: Compensate the voltage by taking into account the dead zone effect.

[0018] Furthermore, step S21 specifically includes:

[0019] If only the control dead zone and the rise delay and turn-off delay of the device are considered, the ideal a phase to reference 0 voltage The actual phase a to reference 0 voltage U a0 Expressed as:

[0020]

[0021] Among them, T s is a control cycle, t D is the dead time, t rise-delay is the device rise delay time, t fall-delay is the device fall delay time, f(x)=sign(x) is the sign function, V dc is the DC bus voltage;

[0022] If the conduction saturation voltage drop of the power device and the voltage drop of the device's freewheeling diode are considered at the same time, the a-phase upper tube S a The duty cycle represents the tube voltage drop, and the neutral point 0 is defined as V dc / 2, the conduction conditions of the switch tube are divided into the following four conditions:

[0023] If i as ≥0, S a =1,U a0 =V dc / 2-V ce ;

[0024] If i as ≥0, S a =0, U a0 =-V dc / 2-V d ;

[0025] If i as <0, S a =0, U a0 =V dc / 2+V d ;

[0026] If i as <0, S a =0, U a0 =-V dc / 2+Vce ;

[0027] For all of the above cases, U a0 =(V dc -V ce +V d )(S a -0.5)-0.5(V ce +V d )sgn(i a )

[0028] Where V ce is the forward voltage drop of the power device, V d is the freewheeling diode voltage drop, S a is the duty cycle of the upper tube of phase a, i a is the a-phase current;

[0029] The voltage change to the neutral point of the motor finally gives the three-phase voltage compensation formula, which is the voltage loss caused by the nonlinearity of the inverter:

[0030]

[0031] M=t D +t rise-relay -t fall-delay

[0032] Where U' an is the voltage lost in phase a, i b is the b-phase current, i c is the c-phase current, U′ bn is the voltage lost in phase b, U′ cn is the voltage lost in phase c, V c is the intermediate variable, M is the intermediate variable; R ce is the saturation equivalent resistance of the power device, R d is the freewheeling diode equivalent resistance.

[0033] Further, step S22 specifically includes:

[0034] After considering the influence of parasitic capacitance, it is necessary to consider the tube voltage drop caused by parasitic capacitance during the dead zone and compensate for the dead zone effect. For phase a, the upper bridge arm of phase a is turned on and off once in one control cycle. Analyzing it based on the duty cycle, when the current of phase a is greater than 0, when the upper bridge arm of tube a is turned off and the lower bridge arm of tube a is turned on, the voltage will ramp down at a rate within the dead zone. At the same time, when the current of phase a is less than 0, when the upper bridge arm of tube a is turned off and the lower bridge arm of tube a is turned on, the voltage will ramp down at a rate within the dead zone.

[0035] When the current of phase a is greater than 0, the upper tube of phase a is turned off and the lower tube is turned on. The voltage decreases during the dead zone according to the current. Let |i lim | is the current at which the voltage just drops to 0 during the entire dead time:

[0036]

[0037] (1) When 0<i a <|i lim |When the voltage Sa is increased by the parasitic capacitance + for:

[0038]

[0039] Where C is the size of the parasitic capacitance;

[0040] (2) When |i lim |<i a When the voltage Sa is increased by the parasitic capacitance + for:

[0041]

[0042] When the a-phase current is less than 0, the following compensation formula is obtained:

[0043] (1)-|i lim |<ia<0, the voltage Sa increased by the parasitic capacitance + for:

[0044]

[0045] (2)ia<-|i lim |<0, the voltage Sa increased by the parasitic capacitance + for:

[0046]

[0047] Similarly, we get S b+ With S c+ ;

[0048] The compensation voltage converted to the neutral point of the motor is:

[0049]

[0050] Furthermore, the resistance identification formula in step S3 is:

[0051]

[0052] in Given the d-axis voltage for the control link, U ascTo consider the parasitic voltage a phase needs to compensate the voltage, i d is a direct current.

[0053] Furthermore, in step S4, when identifying the stator dq-axis inductance, ensure that the rotor is stationary, and then inject a high-frequency voltage signal with a fundamental angular frequency of ω into the d-axis and q-axis respectively. By extracting the fundamental components from the voltage and current signals, the d-axis and q-axis inductances are solved;

[0054] The expression of the d-axis injection signal is as follows:

[0055] U fd =U dm_dc sin(ωt)U dm_dc is the d-axis high-frequency voltage amplitude, and the three-phase currents abc are measured. After d- and q-axis transformation, the d-axis current signal under the actual high-frequency input is obtained. After filtering out the signal with more than twice the injection frequency, the maximum d-axis current is obtained. The d-axis inductance value is expressed as:

[0056]

[0057] Among them I dm_dc is the amplitude of the d-axis high-frequency current

[0058] When identifying the q-axis inductance, a DC voltage bias is injected into the d-axis to ensure that the rotor position remains unchanged. The expression of the q-axis injection signal is as follows:

[0059]

[0060] Among them, U fq is the q-axis high-frequency voltage fundamental effective value, U d_dc is the d-axis DC voltage bias, U qm_dc is the q-axis high-frequency voltage amplitude, U q_ac is the effective value of the high-frequency voltage on the q-axis;

[0061] The expression of q-axis inductance Lq is as follows:

[0062]

[0063] Among them, I qm_ac is the q-axis high-frequency current amplitude, U qm_ac is the q-axis high-frequency voltage amplitude, I q_ac is the effective value of the q-axis high-frequency current, U q_ac is the effective value of the q-axis high-frequency voltage.

[0064] Furthermore, in step S5, without position sensor, the actual magnetic flux ψ dreal Decomposed into ψ on the d and q axes of the model dref , ψ qref; Combining the stator resistance Rs and the d-axis and q-axis inductance parameters Ld and Lq obtained by identification, the flux linkage expressions on the d-axis and q-axis of the model are obtained:

[0065]

[0066] Where R s The resistance to be identified, L d , L q is the identified resistance, ω e The electrical angular velocity given by the open-loop virtual axis of the circuit.

[0067] The beneficial effects of the present invention are:

[0068] (1) An inverter nonlinearity compensation strategy considering parasitic capacitance is established, which achieves accurate compensation for voltage loss caused by inverter nonlinearity at low current.

[0069] (2) The inverter non-linear compensation scheme considering parasitic capacitance is taken into account in the resistance identification, and the resistance identification scheme is optimized, making the resistance identification method applicable to the identification of currents under various working conditions and improving the identification accuracy;

[0070] (3) Optimize the high-frequency injection scheme to achieve accurate identification of dq-axis inductance

[0071] (4) The inverter nonlinear compensation scheme considering parasitic capacitance is taken into account in the flux identification, and a scheme for dq axis flux decomposition without position is proposed to improve the identification accuracy of flux without position signal.

[0072] Other advantages, objects, and features of the present invention will be described in part in the following description and, in part, will be apparent to those skilled in the art upon examination of the following description or may be learned from practice of the present invention. The objects and other advantages of the present invention may be realized and obtained through the following description. BRIEF DESCRIPTION OF THE DRAWINGS

[0073] In order to make the purpose, technical solutions and advantages of the present invention more clear, the present invention will be described in detail below with reference to the accompanying drawings, in which:

[0074] Figure 1 (a) is the ideal output voltage, and (b) is the effect of the dead zone on the output voltage.

[0075] Figure 2 The conduction status of the switch tube;

[0076] Figure 3 This is a model that takes parasitic capacitance into account;

[0077] Figure 4Switching state analysis considering parasitic capacitance;

[0078] Figure 5 Equivalent circuit diagram for identifying motor resistance;

[0079] Figure 6 is the d-axis AC steady-state equivalent circuit;

[0080] Figure 7 It is the decomposition of magnetic chain;

[0081] Figure 8 is the effect of parasitic capacitance on output voltage as current a increases;

[0082] Figure 9 To identify the d-axis voltage, current, and compensation voltage diagram of the resistor;

[0083] Figure 10 is the injected high-frequency d-axis voltage and d-axis current;

[0084] Figure 11 is the d-axis inductance identification value;

[0085] Figure 12 is the injected q-axis high-frequency voltage and dq-axis current;

[0086] Figure 13 is the q-axis inductance identification result;

[0087] Figure 14 This is the result of magnetic linkage identification. DETAILED DESCRIPTION

[0088] The following describes the embodiments of the present invention by means of specific examples, and those skilled in the art can easily understand other advantages and effects of the present invention from the contents disclosed in this specification. The present invention can also be implemented or applied through other different specific embodiments, and the details in this specification can also be modified or changed in various ways based on different viewpoints and applications without departing from the spirit of the present invention. It should be noted that the illustrations provided in the following embodiments are only schematic illustrations of the basic concept of the present invention, and the following embodiments and features in the embodiments can be combined with each other without conflict.

[0089] It should be noted that the illustrations provided in the following embodiments are merely schematic illustrations of the basic concept of the present invention. Therefore, the illustrations only show components related to the present invention and are not drawn according to the number, shape, and size of components in actual implementation. In actual implementation, the type, quantity, and proportion of each component may be changed arbitrarily, and the component layout may also be more complex.

[0090] In the following description, numerous details are discussed to provide a more thorough explanation of the embodiments of the present invention. However, it will be apparent to those skilled in the art that the embodiments of the present invention may be practiced without these specific details. In other embodiments, well-known structures and devices are shown in block diagram form rather than in detail to avoid obscuring the embodiments of the present invention.

[0091] Example 1:

[0092] The present invention provides an offline parameter identification method for an IPMSM without position signal considering the nonlinearity of an inverter, the method comprising the following steps:

[0093] S1: The original three-phase dynamic model of an internal permanent magnet synchronous motor (IPMSM) is relatively complex. To simplify analysis, the model is usually transformed into a two-phase rotating coordinate system using coordinate transformation. The mathematical model of the IPMSM voltage in the d and q coordinate systems is:

[0094]

[0095] Among them, U d 、U q is the d and q axis voltage of the motor, R s is the stator resistance of the motor, i d 、i q is the motor d and q axis current, ψ d , ψ q is the motor d and q axis magnetic flux, ω e is the electrical angular velocity of the motor rotor, L d , L q is the motor d and q axis inductance

[0096] S2: Inverter nonlinear compensation scheme.

[0097] S2.1: If Figure 1 As shown in (a) and (b), if only the control dead zone and the rise delay and turn-off delay of the device are considered, the ideal phase a to reference 0 voltage The actual phase a to reference 0 voltage U a0 It can be expressed as:

[0098]

[0099] Among them, T s , t D , t rise-delay , t fall-delay is a control cycle, dead time, device rise delay time, device fall delay time, f(x)=sign(x) is the sign function, V dc is the DC bus voltage.

[0100] If we want to consider both the saturation voltage drop of the power device and the voltage drop of the device's freewheeling diode, we can use the duty cycle of the a-phase upper tube Sa to represent the tube voltage drop. Define the neutral point 0 as V dc / 2, and then divide the switch conduction conditions into Figure 2 Four situations:

[0101] If i as ≥0, S a =1, U a0 =V dc / 2-V ce ;

[0102] If i as ≥0, S a =0, U a0 =-V dc / 2-V d ;

[0103] If i as <0, S a =0, U a0 =V dc / 2+V d ;

[0104] If i as <0, S a =0, U a0 =-V dc / 2+V ce ;

[0105] For all of the above cases, U a0 =(V dc -V ce +V d )(S a -0.5)-0.5(V ce +V d )sgn(i a )

[0106] Where V ce is the forward voltage drop of the power device, V d is the freewheeling diode voltage drop, S a is the duty cycle of the upper tube of phase a, i a is the a-phase current;

[0107] The voltage change to the neutral point of the motor can finally be used to obtain the three-phase voltage compensation formula, which is the voltage loss caused by the nonlinearity of the inverter:

[0108]

[0109] M=t D +t rise-relay-t fall-delay (3)

[0110] Where U' an is the voltage lost in phase a, i b is the b-phase current, i c is the c-phase current, U′ bn is the voltage lost in phase b, U′ cn is the voltage lost in phase c, V c is the intermediate variable, M is the intermediate variable; R ce is the saturation equivalent resistance of the power device, V d is the freewheeling diode voltage drop, R d is the freewheeling diode equivalent resistance.

[0111] S2.2: After considering the influence of parasitic capacitance, it is necessary to consider the tube voltage drop caused by parasitic capacitance during the dead zone. The model of the tube considering parasitic capacitance is as follows: Figure 3 shown.

[0112] Considering the compensation of dead zone effect, take phase a as an example. In one control cycle, the upper bridge arm of phase a will be turned on and off once. Figure 4 As shown in the figure, based on the duty cycle, when the phase a current is greater than 0, when the upper arm of tube a is turned off and the lower arm of tube a is turned on, the voltage will ramp down at a rate within the dead zone. At the same time, when the phase a current is less than 0, when the upper arm of tube a is turned off and the lower arm of tube a is turned on, the voltage will ramp down at a rate within the dead zone.

[0113] Taking the moment when the current of phase a is greater than 0 as an example, when the upper tube of phase a is turned off and the lower tube is turned on, the voltage will drop during the dead zone according to the magnitude of the current. When the current is less than the current at which the voltage just drops to 0 during the entire dead zone, the voltage will drop to 0. lim |, that is, under this current, the voltage cannot drop to 0 during the entire dead time period. The voltage drop state is as follows: Figure 4 If the current value satisfies the voltage drop to 0 during the dead zone, the voltage drop state is as shown in state 1. Figure 4 As shown in state 2.

[0114]

[0115] (1) Therefore when 0<ia<|i lim |When the voltage Sa is increased by the parasitic capacitance + for:

[0116]

[0117] Where C is the size of the parasitic capacitance.

[0118] (2) Therefore, when |i lim|<ia, the voltage Sa increased by the parasitic capacitance + for:

[0119]

[0120] Similarly, when the a-phase current is less than 0, the following compensation formula is obtained:

[0121] (1)-|i lim |<ia<0, the voltage Sa increased by the parasitic capacitance + for:

[0122]

[0123] (2)ia<-|i lim |<0, the voltage Sa increased by the parasitic capacitance + for:

[0124]

[0125] Similarly, we can get S b+ With S c+ Therefore, the compensation voltage converted to the neutral point of the motor is:

[0126]

[0127] The following methods are all based on the compensation in this step. The voltages in the following schemes are all considered to have been compensated in this step and accurate voltages have been obtained.

[0128] S3: Parameter Identification Scheme

[0129] S3.1: Identify resistance

[0130] When identifying the stator resistance, a DC current I is usually injected into the d-axis. d , there is no alternating current on the stator side of the motor, and no rotating magnetic field is generated, so the motor rotor does not rotate. The three-phase stator current generated at this time will quickly reach a stable value. After the motor is locked to the d-axis, the motor will be in a stationary state, that is, the electrical angular velocity ωe is 0. The resistance parameters of the motor are identified, and the equivalent circuit is as follows Figure 5 As shown, the resistance identification formula can be obtained from formula (1).

[0131]

[0132] in Given the d-axis voltage for the control link, U asc To consider the parasitic voltage a phase needs to compensate the voltage, i d is the d-axis current.

[0133] S3.2: Identify dq-axis inductance

[0134] When identifying the stator dq-axis inductance, the rotor should be kept stationary. Then, a high-frequency voltage signal with a fundamental angular frequency of ω is injected into the d-axis and q-axis, respectively. When the injected fundamental angular frequency is high, the torque ripple can be almost offset, and the motor is in a stationary state. By extracting the fundamental components from the voltage and current signals, the d- and q-axis inductances are solved. Taking the identification of the d-axis inductance as an example, the d-axis AC steady-state equivalent circuit is as follows: Figure 6 shown.

[0135] When identifying the d-axis inductance, the motor's electromagnetic torque is primarily determined by the q-axis current. The high-frequency voltage injected into the d-axis excites the d-axis high-frequency current, but since the q-axis voltage is approximately zero in steady state, the average torque is zero. Therefore, the d-axis injection signal is expressed as follows:

[0136] U fd =U dm_dc sin(ωt) (11)U dm_dc is the d-axis high-frequency voltage amplitude, and the three-phase currents abc are measured. After d- and q-axis transformation, the d-axis current signal under the actual high-frequency input is obtained. After filtering out the signal with more than twice the injection frequency, the maximum d-axis current is obtained. The d-axis inductance value can be expressed as:

[0137]

[0138] Among them I dm_dc is the amplitude of the d-axis high-frequency current.

[0139] Similarly, the q-axis inductance L q The injection and calculation methods for the q-axis are basically the same as those for the d-axis. When identifying the q-axis inductance, a sufficiently large DC voltage bias is also injected into the d-axis to ensure that the rotor position remains unchanged. The expression for the q-axis injection signal is as follows:

[0140]

[0141] Among them, U fq is the q-axis high-frequency voltage fundamental effective value, U d_dc is the d-axis DC voltage bias, U qm_ac is the q-axis high-frequency voltage amplitude, U q_dc is the effective value of the q-axis high-frequency voltage.

[0142] The expression of q-axis inductance Lq is as follows:

[0143]

[0144] Among them, I qm_ac is the q-axis high-frequency current amplitude, U qm_ac is the q-axis high-frequency voltage amplitude, I q_acis the effective value of the q-axis high-frequency current, U q_ac is the effective value of the q-axis high-frequency voltage.

[0145] S3.3: Identify magnetic flux

[0146] Without position sensor, such as Figure 7 As shown in the figure, since the motor speed cannot be obtained in real time, there is a certain angle deviation between the actual dq axis and the dq axis of the model. dreal Decomposed into ψ on the d and q axes of the model dref , ψ qref .

[0147] Combining the stator resistance Rs and the d-axis and q-axis inductance parameters Ld and Lq obtained by identification, the flux linkage expressions on the d-axis and q-axis of the model are obtained.

[0148]

[0149] Where R s The resistance to be identified, L d , L q is the identified resistance, ω e The electrical angular velocity given by the open-loop virtual axis of the circuit.

[0150] To further verify the actual identification effect of the proposed algorithm, the resistance, inductance, and flux linkage of the experimental motor were identified offline. The experimental motor parameters are shown in Table 1.

[0151] Table 1

[0152]

[0153] In the process of identifying resistance, the compensation term of parasitic capacitance is considered in combination with the traditional identification, such as Figure 8 As shown in Figure 1, the effect of parasitic capacitance on the output voltage increases with the current in item a. It can be seen that the effect of parasitic capacitance on the voltage is more obvious when the current is small. As the current gradually increases, the effect of parasitic capacitance on the voltage gradually stabilizes and decreases. Therefore, considering the influence of parasitic capacitance is necessary for offline parameter identification.

[0154] First, the resistance is identified. A 1A current is injected into the motor's D-axis for positioning. Then a 6A current is injected into the D-axis for resistance parameter identification. Figure 9 As shown in the figure, after 4 seconds, the motor enters a steady state, and the measured d-axis output voltage is 5.34 V, the d-axis voltage is 2.919 V, and the d-axis current is 6 A. The resistance obtained is 0.4035 Ω, with an error of 0.875%.

[0155] To identify the d-axis inductance, a high-frequency d-axis voltage with a frequency of 312.5 Hz and an amplitude of 30 V is injected into the d-axis, as shown in Figure 10By measuring the high-frequency fundamental amplitude of the d-axis high-frequency current, the d-axis inductance parameters are obtained, as shown in Figure 11 As shown, the identification result is 1.425mH, and the identification error is 1.72%.

[0156] Similarly, if Figure 12 As shown, a high-frequency q-axis voltage with a frequency of 312.5Hz and an amplitude of 15V is injected into the q-axis; a DC voltage with an amplitude of 2V is injected into the d-axis, the d-axis current is DC, and the q-axis is a high-frequency current with a frequency of 312.5. Figure 13 As shown in the figure, the q-axis inductance identification parameters are obtained, and the identification result is 1.74mH with an error of 3.3%.

[0157] When identifying the flux linkage, the motor needs to be rotated, and through the current closed loop, the q-axis current is set to 1A and the d-axis current is set to 0 to increase the motor speed. Figure 14 As shown in the flux identification result diagram, the speed of the given current closed-loop motor is 300r / min.

[0158] Compared with the actual parameters: the identification error of resistance is 0.875%, the identification error of d-axis inductance is 1.72%, the identification error of q-axis inductance is 3.3%, and the identification error of magnetic flux is 14%.

[0159] Example 2:

[0160] An electronic device comprising a memory and a processor;

[0161] The memory is used to store computer programs;

[0162] The processor is configured to implement the method described in Example 1 when executing the computer program.

[0163] Example 3:

[0164] A computer-readable storage medium stores a computer program, and when the computer program is executed by a processor, the method described in Example 1 is implemented.

[0165] Example 4:

[0166] A computer program product includes a computer program, which implements the method described in embodiment 1 when executed by a processor.

[0167] In the above embodiments, references to "this embodiment" in the specification indicate that a particular feature, structure, or characteristic described in conjunction with the embodiment is included in at least some embodiments, but not necessarily all embodiments. Multiple occurrences of "this embodiment" do not necessarily refer to the same embodiment.

[0168] In the above embodiments, although the invention has been described in conjunction with specific embodiments thereof, many alternatives, modifications, and variations of these embodiments will be apparent to those skilled in the art based on the foregoing description. For example, other memory structures (e.g., dynamic RAM (DRAM)) may be used with the embodiments discussed. The embodiments of the present invention are intended to encompass all such alternatives, modifications, and variations that fall within the broad scope of the appended claims.

[0169] Regarding the computer-readable storage medium in this embodiment, those skilled in the art will appreciate that all or part of the steps in the aforementioned method embodiments can be implemented using hardware associated with the computer program. The aforementioned computer program can be stored in a computer-readable storage medium. When executed, the program performs the steps in the aforementioned method embodiments. The aforementioned storage medium includes various media capable of storing program code, such as ROM, RAM, magnetic disks, or optical disks.

[0170] The electronic terminal provided in this embodiment includes a processor, a memory, a transceiver and a communication interface. The memory and the communication interface are connected to the processor and the transceiver and complete communication with each other. The memory is used to store computer programs, the communication interface is used for communication, and the processor and the transceiver are used to run computer programs so that the electronic terminal executes the various steps of the above method.

[0171] In this embodiment, the memory may include a random access memory (RAM), and may also include a non-volatile memory (non-volatile memory), such as at least one disk storage.

[0172] The above-mentioned processor can be a general-purpose processor, including a central processing unit (CPU), a network processor (NP), etc.; it can also be a digital signal processor (DSP), an application-specific integrated circuit (ASIC), a field-programmable gate array (FPGA) or other programmable logic devices, discrete gate or transistor logic devices, and discrete hardware components.

[0173] The present invention can be used in a wide variety of general-purpose or special-purpose computing system environments or configurations, such as personal computers, server computers, handheld or portable devices, tablet devices, multiprocessor systems, microprocessor-based systems, set-top boxes, programmable consumer electronics, network PCs, minicomputers, mainframe computers, and distributed computing environments that include any of the above.

[0174] The present invention may be described in the general context of computer-executable instructions, such as program modules, executed by a computer. Generally, program modules include routines, programs, objects, components, data structures, and the like that perform specific tasks or implement specific abstract data types. The present invention may also be practiced in distributed computing environments where tasks are performed by remote processing devices connected via a communications network. In a distributed computing environment, program modules may be located in both local and remote computer storage media, including storage devices.

[0175] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention and are not limiting. Although the present invention has been described in detail with reference to the preferred embodiments, those skilled in the art should understand that the technical solutions of the present invention can be modified or replaced by equivalents without departing from the purpose and scope of the technical solutions, which should all be included in the scope of the claims of the present invention.

Claims

1. An off-line parameter identification method for a position sensorless permanent magnet synchronous motor, characterized by: The following steps are involved: S1: Construct the voltage mathematical model of the embedded permanent magnet synchronous motor IPMSM in the d and q coordinate systems; S2: Establish an inverter nonlinear compensation strategy considering parasitic capacitance; S3: Taking the inverter non-linear compensation strategy considering parasitic capacitance into consideration in the resistance identification, the resistance is identified; S4: Optimize the high-frequency injection scheme and use a simplified inductance identification method based on signal filtering and peak extraction to identify the dq-axis inductance; S5: The inverter nonlinear compensation strategy considering parasitic capacitance is taken into account in the flux identification, and a dq-axis flux decomposition method without position is proposed to identify the flux.

2. The offline parameter identification method of a position sensorless permanent magnet synchronous motor according to claim 1, characterized in that: The voltage mathematical model of IPMSM in the d, q coordinate system is: Among them, U d 、U q is the d and q axis voltage of the motor, R s is the stator resistance of the motor, i d 、i q is the motor d and q axis current, ψ d , ψ q is the motor d and q axis magnetic flux, ω e is the electrical angular velocity of the motor rotor, L d , L q is the motor d-axis and q-axis inductance.

3. The offline parameter identification method of a position sensorless permanent magnet synchronous motor according to claim 1, characterized in that: Step S2 specifically includes the following steps: S21: Considering the influence of parasitic capacitance, calculate the voltage loss caused by the nonlinearity of the inverter; S22: Compensate the voltage by taking into account the dead zone effect.

4. The offline parameter identification method of a position sensorless permanent magnet synchronous motor according to claim 3, characterized in that: Step S21 specifically includes: If only the control dead zone and the rise delay and turn-off delay of the device are considered, the ideal a phase to reference 0 voltage The actual phase a to reference 0 voltage U a0 Expressed as: Among them, T s is a control cycle, t D is the dead time, t rise-delay is the device rise delay time, t fall-delay is the device fall delay time, f(x)=sign(x) is the sign function, V dc is the DC bus voltage; If the conduction saturation voltage drop of the power device and the voltage drop of the device's freewheeling diode are considered at the same time, the a-phase upper tube S a The duty cycle represents the tube voltage drop, and the neutral point 0 is defined as V dc / 2, the conduction conditions of the switch tube are divided into the following four conditions: If i as ≥ 0, S a = 1, U a0 = V dc / 2 - V ce ; If i as ≥ 0, S a = 0, U a0 = -V dc / 2 - V d ; If i as < 0, S a = 0, U a0 = V dc / 2 + V d ; If i as < 0, S a = 0, U a0 = -V dc / 2 + V ce ; For all of the above cases, U a0 =(V dc -V ce +V d )(S a -0.5)-0.5(V ce +V d )sgn(i a ) Where V ce is the forward voltage drop of the power device, V d is the freewheeling diode voltage drop, S a is the duty cycle of the upper tube of phase a, i a is the a-phase current; The voltage change to the neutral point of the motor finally gives the three-phase voltage compensation formula, which is the voltage loss caused by the nonlinearity of the inverter: M=t D +t rise-relay -t fall-delay Where U' an is the voltage lost in phase a, i b is the b-phase current, i c is the c-phase current, U' bn is the voltage lost in phase b, U' cn is the voltage lost in phase c, V c is the intermediate variable, M is the intermediate variable; R ce is the saturation equivalent resistance of the power device, R d is the freewheeling diode equivalent resistance.

5. The offline parameter identification method of a position sensorless permanent magnet synchronous motor according to claim 3, characterized in that: Step S22 specifically includes: After considering the influence of parasitic capacitance, it is necessary to consider the tube voltage drop caused by parasitic capacitance during the dead zone and the compensation of the dead zone effect. For phase a, the upper bridge arm of phase a is turned on and off once in one control cycle. Analyzing it based on the duty cycle, when the current of phase a is greater than 0, when the upper bridge arm of tube a is turned off and the lower bridge arm of tube a is turned on, the voltage will decrease at a rate ramp within the dead zone. At the same time, when the current of phase a is less than 0, when the upper bridge arm of tube a is turned off and the lower bridge arm of tube a is turned on, the voltage will decrease at a rate ramp within the dead zone. When the current of phase a is greater than 0, when the upper tube of phase a is turned off and the lower tube is turned on, the voltage decreases during the dead zone according to the current. Let |i lim | is the current at which the voltage just drops to 0 during the entire dead time: (1) When 0<i a <|i lim |When the voltage Sa is increased by the parasitic capacitance + for: Where C is the size of the parasitic capacitance; (2) When |i lim |<i a When the voltage Sa is increased by the parasitic capacitance + for: When the a-phase current is less than 0, the following compensation formula is obtained: (1)-|i lim |<ia<0, the voltage Sa increased by the parasitic capacitance + for: (2)ia<-|i lim |<0, the voltage Sa increased by the parasitic capacitance + for: Similarly, we get S b+ With S c+ ; The compensation voltage converted to the neutral point of the motor is:

6. The offline parameter identification method of a position sensorless permanent magnet synchronous motor according to claim 1, characterized in that: The resistance identification formula in step S3 is: in Given the d-axis voltage for the control link, U asc To consider the parasitic voltage a phase needs to compensate the voltage, i d is the d-axis current.

7. The offline parameter identification method of a position sensorless permanent magnet synchronous motor according to claim 1, characterized in that: In step S4, when identifying the stator dq-axis inductance, ensure that the rotor is stationary. Then, a high-frequency voltage signal with a fundamental angular frequency of ω is injected into the d-axis and q-axis, respectively. By extracting the fundamental components from the voltage and current signals, the d-axis and q-axis inductances are solved. The expression of the d-axis injection signal is as follows: U fd =U dm_dc sin(ωt) U dm_dc is the d-axis high-frequency voltage amplitude, and the three-phase currents abc are measured. After d- and q-axis transformation, the d-axis current signal under the actual high-frequency input is obtained. After filtering out the signal with more than twice the injection frequency, the maximum d-axis current is obtained. The d-axis inductance value is expressed as: Among them I dm_dc is the amplitude of the d-axis high-frequency current; When identifying the q-axis inductance, a DC voltage bias is injected into the d-axis to ensure that the rotor position remains unchanged. The expression of the q-axis injection signal is as follows: Among them, U fq is the q-axis high-frequency voltage fundamental effective value, U d_dc is the d-axis DC voltage bias, U qm_ac is the q-axis high-frequency voltage amplitude, U q_ac is the effective value of the high-frequency voltage on the q-axis; The q-axis inductance Lq is expressed as follows: Among them, I qm_ac is the q-axis high-frequency current amplitude, U qm_ac is the q-axis high-frequency voltage amplitude, I q_ac is the effective value of the q-axis high-frequency current, U q_ac is the effective value of the q-axis high-frequency voltage.

8. The offline parameter identification method of a position sensorless permanent magnet synchronous motor according to claim 1, characterized in that: In step S5, without position sensor, the actual magnetic flux ψ dreal Decomposed into ψ on the d and q axes of the model dref , ψ qref ; Combining the stator resistance Rs and the d-axis and q-axis inductance parameters Ld and Lq obtained by identification, the flux linkage expressions on the d-axis and q-axis of the model are obtained: Where R s The resistance to be identified, L d , L q is the identified resistance, ω e The electrical angular velocity given by the open-loop virtual axis of the circuit.

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