Ripple real-time offset circuit of self-excitation fluxgate sensor

Abstract

The invention relates to a ripple real-time offset circuit of a self-oscillation fluxgate sensor. The ripple real-time offset circuit comprises a measuring winding Lm, a measuring resistor Rm, a current sampling device, a ripple signal extraction module and a real-time offset module, the current sampling device is used for sampling a current signal flowing through the measuring winding in real time and outputting the signal to the ripple signal extraction module; the ripple signal extraction module samples a mixed signal flowing through the measurement winding in real time through the current sampling device, separates a ripple component from the mixed signal, and outputs a ripple reference signal representing the ripple component to the real-time offset module; and the real-time counteracting module is configured to generate counteracting current which is equal to the ripple current in amplitude and opposite to the ripple current in phase on the basis of the ripple reference signal, inject the counteracting current onto the measuring resistor Rm, and counteract the ripple current on the measuring resistor Rm in real time. The ripple current can be counteracted in real time, and the measurement error is remarkably reduced.

Description

Technical Field

[0001] This invention relates to the field of self-excited fluxgate sensor technology, and particularly to a real-time ripple cancellation circuit for a self-excited fluxgate sensor. Background Technology

[0002] Fluxgate technology is a technique that measures magnetic fields by utilizing the nonlinear relationship between the magnetic flux density and the magnetic field strength of a high-permeability magnetic core under saturation excitation of an alternating magnetic field. Compared with traditional fluxgate technology, self-excited oscillating fluxgate technology has the advantage of simpler circuitry and can theoretically achieve low-cost, precise measurements.

[0003] There are two basic types of self-excited fluxgates: open-loop and closed-loop configurations. In the open-loop configuration, the excitation winding also serves as the measurement winding; the same coil generates an alternating magnetic field and detects signals. In the closed-loop configuration, an independent feedback winding acts as the measurement winding. The feedback coil detects changes in the magnetic field and adjusts the excitation parameters, forming a closed-loop control to improve measurement accuracy.

[0004] For open-loop self-excited fluxgate magnetometers, the large alternating saturation current in the excitation coil generates significant excitation interference, which varies with the magnitude of the measured magnetic field. Theoretically, when the measured magnetic field is zero, the excitation interference is symmetrical; however, when the measured magnetic field is not zero, the excitation interference is no longer symmetrical. Therefore, excitation interference has a significant impact on the measurement accuracy of open-loop self-excited fluxgate magnetometers.

[0005] A major reason why the measurement accuracy of existing closed-loop self-oscillating fluxgate technology is difficult to improve is the ripple caused by the excitation current. To reduce this effect, a common method is to add an excitation core and an excitation winding to cancel out the excitation interference ripple [N. Wang et al., "Self-Oscillating Fluxgate-Based Quasi-Digital Sensor for DC High-Current Measurement," in IEEE Transactions on Instrumentation and Measurement, vol. 64, no. 12, pp. 3555-3563, Dec. 2015]. However, adding a core and coil increases the complexity and cost of the sensor, and it is difficult to make the two cores and coils completely identical. Inconsistencies in the core and coil reduce the ability to cancel out the excitation interference ripple.

[0006] The existing technology [Fluorescence Gate Current Sensor and Electronic Equipment, Application Publication No. CN118795200A] uses a "compensation circuit to generate a compensation signal based on the filtered signal when the measured current is zero, and the compensation signal is used to compensate the excitation signal when the measured current is not zero." This compensation circuit is used to "eliminate the fundamental frequency ripple signal within the excitation signal." However, the fundamental frequency ripple of the current changes with the operating conditions. For example, when the measured current is not zero, especially when the measured current is much greater than zero, the fundamental frequency ripple will be different; similarly, when the measured current changes rapidly, the fundamental frequency ripple signal will also be different. Furthermore, current sensors are easily affected by high-frequency interference in practical applications. Therefore, the technology in [Fluorescence Gate Current Sensor and Electronic Equipment, Application Publication No. CN118795200A] cannot adapt to changes in the fundamental frequency ripple signal in practical applications because the compensation signal is fixed, thus failing to guarantee the compensation effect.

[0007] The existing technology [Method for Compensating Ripple Current of Fluxgate Current Sensor Sampling Resistor, Application Publication No. CN113189384A] "collects noise on the sampling port, obtains a cancellation signal after signal processing, and then inputs it to the sampling port for cancellation." In this patent, the sampling port is used for noise acquisition and also for inputting the cancellation signal. Therefore, noise acquisition and compensation cannot be performed simultaneously. Furthermore, "digital processing is performed on the amplified ripple noise signal of several acquisition cycles to convert it into a noise cancellation signal." Since several cycles of signal processing are required, the compensation has a lag. Therefore, the technology in [Method for Compensating Ripple Current of Fluxgate Current Sensor Sampling Resistor, Application Publication No. CN113189384A] cannot adapt to the real-time changes of the ripple signal in actual applications due to the lag, and therefore cannot guarantee the compensation effect. The "beneficial effects" of this patent are: (1) It can specifically cancel the ripple noise during each normal operation, preventing the difference in the ripple signal at each power-on from affecting the cancellation effect. This patent only considers the difference in ripple signal each time it is powered on. The ripple signal used for compensation remains unchanged. It does not consider other factors, such as the magnitude of the current, on the difference in ripple signal. Therefore, it cannot guarantee the compensation effect. Summary of the Invention

[0008] To address the shortcomings of existing technologies, the technical problem this invention aims to solve is to provide a real-time ripple cancellation circuit for a self-excited fluxgate sensor, which solves the ripple interference problem caused by the excitation current in traditional self-excited fluxgate sensors.

[0009] The technical solution adopted by the present invention to solve the aforementioned technical problem is as follows:

[0010] A real-time ripple cancellation circuit for a self-excited oscillating fluxgate sensor includes a measuring winding Lm and a measuring resistor Rm. The measuring resistor Rm is connected in series in the branch where the measuring winding Lm is located, and one end of it is grounded. The voltage signal across the measuring resistor Rm serves as the output signal of the self-excited oscillating fluxgate sensor. The circuit also includes a current sampling device, a ripple signal extraction module, and a real-time cancellation module.

[0011] The current sampling device is used to sample the current signal flowing through the measurement winding in real time and output the signal to the ripple signal extraction module.

[0012] The ripple signal extraction module samples the mixed signal (including ripple signal) flowing through the measurement winding in real time through the current sampling device, separates the ripple component from the mixed signal, and outputs a ripple reference signal representing the ripple component to the real-time cancellation module.

[0013] The real-time cancellation module is configured to generate a cancellation current with the same amplitude and opposite phase as the ripple current based on the ripple reference signal, and inject the cancellation current into the measuring resistor Rm, thereby canceling the ripple current in the measuring resistor Rm in real time.

[0014] In one embodiment, the current sampling device is selected from the sampling resistor Rs; the ripple signal extraction module includes: a differential amplifier U1 and a high-pass filter HPF;

[0015] One end of the sampling resistor Rs is connected to one end of the measuring winding Lm at node N1, and the other end of the sampling resistor Rs is connected to the ungrounded terminal of the measuring resistor Rm at node N2.

[0016] The inverting and non-inverting input terminals of the differential amplifier U1 are respectively connected to the two ends of the sampling resistor Rs to amplify the voltage difference across the resistor and output a voltage signal proportional to the current flowing through the sampling resistor Rs.

[0017] The input terminal of the high-pass filter HPF is connected to the output terminal of the differential amplifier U1. Its cutoff frequency fc is set to be lower than the ripple frequency fr of the self-excited fluxgate but higher than the target current signal frequency, in order to filter out DC and low-frequency components while retaining high-frequency ripple components; the output terminal of the HPF provides a ripple reference signal.

[0018] The voltage gain of the ripple signal extraction module is set to G1, and the transconductance gain G2 of the real-time cancellation module is defined as the ratio of the cancellation current i2(t) output by the module to the ripple reference signal voltage input to the module; the two satisfy the condition: Rs×G1×G2=-1.

[0019] In one embodiment, the differential amplifier U1 is composed of an operational amplifier A1, resistors R1, R2, R3, and R4; one end of resistor R1 serves as the inverting input of the differential amplifier U1 and is connected to node N2 (i.e., the ungrounded common connection node of the sampling resistor Rs and the measuring resistor Rm); the other end of resistor R1 is connected to the inverting input of operational amplifier A1; one end of resistor R2 serves as the non-inverting input of the differential amplifier U1 and is connected to node N1 (i.e., the common connection node of the sampling resistor Rs and the measuring winding Lm); the other end of resistor R2 is connected to the non-inverting input of operational amplifier A1; resistor R3 is connected between the inverting input and the output of operational amplifier A1; resistor R4 is connected between the non-inverting input and ground of operational amplifier A1; the output of operational amplifier A1 serves as the output of the differential amplifier U1 and is connected to the input of the high-pass filter HPF.

[0020] In one embodiment, the differential amplifier U1 includes: operational amplifiers A3, A4, and A5, and resistors R7 to R13;

[0021] The non-inverting input (+) of the operational amplifier A3 serves as the inverting input of the differential amplifier U1 and is connected to the common node N1 of the sampling resistor Rs and the measurement winding; the inverting input (-) of the operational amplifier A3 is connected to its own output through resistor R8; and the output is connected to the inverting input of the operational amplifier A5 through resistor R10.

[0022] The non-inverting input (+) of the operational amplifier A4 serves as the non-inverting input of the differential amplifier U1 and is connected to the common node N2 of the sampling resistor Rs and the measuring resistor Rm; the inverting input (-) of the operational amplifier A4 is connected to its own output through resistor R9; and the output is connected to the non-inverting input (+) of the operational amplifier A5 through resistor R11.

[0023] The inverting input terminal (-) of operational amplifier A3 is connected to the inverting input terminal (-) of operational amplifier A4 via R7.

[0024] The inverting input (-) of operational amplifier A5 receives the output signal of A3 through resistor R10 and is connected to its own output through feedback resistor R12; the non-inverting input (+) of operational amplifier A5 receives the output signal of A4 through resistor R11 and is grounded through resistor R13; the output terminal serves as the output terminal of differential amplifier U1 and is connected to the input terminal of high-pass filter HPF.

[0025] The operational amplifiers A3 and A4 serve as input buffer stages, providing high input impedance and suppressing common-mode interference.

[0026] The operational amplifier A5 forms a differential amplifier stage to amplify the input differential signal, with a voltage gain of R12 / R10.

[0027] This circuit achieves a high common-mode rejection ratio (CMRR) through matching resistors (R8=R9, R10=R11, R12=R13), effectively extracting differential small signals. Resistor R7 is connected between the inverting input terminals of A3 and A4 to balance the input impedance.

[0028] The voltage gain of the differential amplifier U1 is G = -(1 + 2 × R8 / R7) × (R12 / R10), and ideally G is approximately G1.

[0029] In one embodiment, the current sampling device is selected as a sampling inductor Ls; the ripple signal extraction module includes a differential integration circuit U4 and a high-pass filter HPF.

[0030] The sampling inductor Ls is connected in series in the branch where the measuring winding is located. The sampling inductor Ls converts the real-time acquired mixed current signal i1(t) flowing through the measuring winding into a voltage signal v. s (t), to achieve low-loss current sampling:

[0031]

[0032] In the formula, i 1-AC (t) is the AC component of i1(t);

[0033] One end of the sampling inductor Ls is connected to one end of the measuring winding Lm at node N1, and the other end of the sampling inductor Ls is connected to the ungrounded terminal of the measuring resistor Rm at node N2.

[0034] The non-inverting input of the differential integrator circuit U4 is connected to the common node N1 of Ls and the measuring winding, and the inverting input is connected to the common node N2 of Ls and Rm.

[0035] In one embodiment, the differential integration circuit U4 includes: operational amplifier A8, resistor R18, resistor R19, capacitor C1, and capacitor C2.

[0036] One end of resistor R18 is connected to the inverting input of differential integrator circuit U4 and the common node N2 of Ls and Rm. The other end of resistor R18 is connected to the inverting input of operational amplifier A8. One end of resistor R19 is connected to the non-inverting input of differential integrator circuit U4 and the common node N1 of Ls and the measurement winding. The other end of resistor R19 is connected to the non-inverting input of operational amplifier A8. One end of capacitor C1 is connected to the inverting input of operational amplifier A8, and the other end of capacitor C1 is connected to the output of operational amplifier A8. One end of capacitor C2 is connected to the non-inverting input of operational amplifier A8, and the other end of capacitor C2 is grounded. The output of operational amplifier A8 is the output of differential integrator circuit U4 and is connected to the input of high-pass filter HPF.

[0037] In this embodiment, R18 = R19 = R, C1 = C2 = C.

[0038] The input v is processed by the differential integrator circuit U4. s The signal v1(t) is processed to obtain the output signal v1(t):

[0039]

[0040] In the formula, V o1 The DC component, The AC component includes ripple signals.

[0041] The input terminal of the high-pass filter HPF is connected to the output terminal of the differential amplifier U4. The cutoff frequency fc is set to be lower than the ripple frequency fr of the self-excited fluxgate but higher than the target current signal frequency, so as to filter out the DC bias and low-frequency target signal and retain the high-frequency ripple component. The output terminal of HPF provides the inverted ripple reference signal for driving the real-time cancellation module.

[0042] The transconductance gain of the real-time cancellation module is set to G2, which is defined as the ratio of the cancellation current i2(t) output by the module to the ripple reference signal voltage input to the module; the condition Ls / (R×C)×G2=-1 is satisfied, which enables real-time cancellation.

[0043] In one embodiment, the real-time cancellation module includes: a proportional amplifier circuit U2 and a resistor Rf;

[0044] The proportional amplifier circuit U2 includes an operational amplifier A2, resistor R5, and resistor R6;

[0045] The output of the ripple signal extraction module is connected to the inverting input of operational amplifier A2 via resistor R6; the inverting input of operational amplifier A2 is also connected to its output via resistor R5; the non-inverting input of operational amplifier A2 is grounded; the output of operational amplifier A2 is connected to N2 via resistor Rf, and ripple current is injected in real time through resistor Rf.

[0046] The transconductance gain of the real-time cancellation module is: G2=-R5 / [R6×(Rm+Rf)].

[0047] In one embodiment, the real-time cancellation module further includes a DC blocking capacitor Co and a bleed resistor Rb;

[0048] The DC blocking capacitor Co is used to eliminate the DC bias error at the output of the real-time cancellation module (such as DC offset caused by operational amplifier offset voltage and bias current), preventing it from being superimposed on the measurement branch and affecting the sensor accuracy; C o A series resistor Rf and a measuring resistor Rm are connected in series to form an AC coupling path; one end of Co is connected to resistor Rf, and the other end is connected to the injection node N2. The capacitance value must be selected to satisfy [1 / (2π×fr×Co)] << Rm to ensure that the impedance of Co is negligible at the ripple frequency. The ripple frequency fr is determined by the following formula:

[0049] fr=Ve / (4×Bsat×Ae×Ne)

[0050] In the formula, Ve is the voltage (V) of the self-excited fluxgate excitation power supply, Bsat is the saturation magnetic induction intensity (T) of the excitation core, and Ae is the effective cross-sectional area (m²) of the excitation core. 2 ), where Ne is the number of turns in the excitation winding.

[0051] The discharge resistor Rb is connected in parallel across Co to discharge the charge accumulated in the capacitor; Co is selected as a ceramic capacitor or film capacitor with low series equivalent resistance (ESR).

[0052] In one embodiment, the real-time cancellation module further includes a power amplifier circuit;

[0053] The power amplifier circuit serves as a current enhancement stage to improve the load capacity of the real-time cancellation module, ensuring that sufficient cancellation current can still be output under large ripple current scenarios to avoid signal clipping or distortion.

[0054] The input terminal of the power amplifier circuit is connected to the output terminal of the proportional amplifier circuit U2, and the output terminal is connected to node N2 via resistor Rf, which is used to inject high-power cancellation current into the measurement branch; at this time, the transconductance gain of the real-time cancellation module satisfies: G2=-G3×R5 / [R6×(Rm+Rf)], where: G3 is the gain of the power amplifier circuit.

[0055] In one embodiment, the real-time cancellation module further includes a power amplifier circuit, a DC blocking capacitor Co, and a bleed resistor Rb; the power amplifier circuit is disposed between the resistor Rf and the output of the proportional amplifier circuit U2, and the DC blocking capacitor Co and the bleed resistor Rb are connected in parallel and then in series between the resistor Rf and node N2.

[0056] In one embodiment, the real-time cancellation module includes: a voltage / current conversion circuit U3.

[0057] The voltage / current conversion circuit U3 includes: two operational amplifiers A6 and A7, four resistors R14, R15, R16, R17 and resistor Ro;

[0058] The operational amplifiers A6 and A7 constitute a two-stage amplification and feedback control; resistors R14, R15, R16, and R17 are matching resistors. o For transimpedance feedback resistor; input terminal u i Receive the ripple reference signal, connect the output terminal to the measurement winding injection node N2, and inject a reverse canceling current i2(t), i2(t) = i o (t);

[0059] The inverting input terminal (-) of the operational amplifier A6 is grounded through resistor R14 and connected to the output terminal u of the operational amplifier A6 through resistor R15. o1 (t); The non-inverting input terminal (+) is connected to the input u through resistor R16. i And connect it to the output terminal u of operational amplifier A7 through resistor R17. o2 (t);

[0060] The non-inverting input (+) of operational amplifier A7 is directly connected to the output node of the voltage / current conversion circuit; the inverting input (-) of operational amplifier A7 is connected to its output terminal u. o2 (t), forming unit negative feedback;

[0061] One end of resistor Ro is connected to the output terminal u of operational amplifier A6. o1 (t), the other end is connected to node N2, and the output current i o (t)=(u o1 (t)-u o2 (t)) / Ro, where the transconductance gain of the real-time cancellation module is G2=1 / Ro.

[0062] The constraints of the voltage / current conversion circuit are:

[0063] The four resistors R14, R15, R16, and R17 are all the same value, i.e., R14 = R15 = R16 = R17.

[0064] Compared with the prior art, the beneficial effects of the present invention are as follows:

[0065] As mentioned earlier, the reference signal for the compensation current in the existing technology [Fluorescence Gate Current Sensor and Electronic Equipment, CN118795200A; Fluorescence Gate Current Sensor Sampling Resistor Ripple Current Compensation Method, CN113189384A] is based on a fixed compensation waveform, which cannot well match the actual ripple waveform under different operating conditions, resulting in unsatisfactory compensation effect. The technology invented in [Fluorescence Gate Current Sensor Sampling Resistor Ripple Current Compensation Method, CN113189384A] can reduce ripple to about 1 / 3 of the original. Because the present invention can perform real-time cancellation, it can almost completely eliminate ripple.

[0066] Therefore, the present invention can cancel ripple current in real time and significantly reduce measurement error. Attached Figure Description

[0067] Figure 1 This is a block diagram of a real-time ripple cancellation circuit for a self-excited fluxgate sensor according to the present invention.

[0068] Figure 2 This is a schematic diagram of the ripple real-time cancellation circuit of the self-excited fluxgate sensor according to the first embodiment of the present invention.

[0069] Figure 3 This is a schematic diagram of the differential amplifier U1 structure according to an embodiment of the present invention;

[0070] Figure 4 This is a schematic diagram of the proportional amplifier circuit U2 according to an embodiment of the present invention;

[0071] Figure 5 This is a schematic diagram of the ripple real-time cancellation circuit of the self-excited fluxgate sensor according to the second embodiment of the present invention.

[0072] Figure 6 This is a schematic diagram of the differential amplifier U1 structure according to an embodiment of the present invention;

[0073] Figure 7 This is a schematic diagram of the structure of a real-time cancellation module (voltage / current conversion circuit) according to an embodiment of the present invention;

[0074] Figure 8 This is a schematic diagram of the ripple real-time cancellation circuit of the self-excited fluxgate sensor according to the third embodiment of the present invention.

[0075] Figure 9 This is a schematic diagram of the ripple real-time cancellation circuit of the self-excited fluxgate sensor according to the fourth embodiment of the present invention.

[0076] Figure 10This is a schematic diagram of the ripple real-time cancellation circuit of the self-excited fluxgate sensor according to the fifth embodiment of the present invention;

[0077] Figure 11 This is a schematic diagram of the differential integrator circuit U4 according to an embodiment of the present invention;

[0078] Figure 12 This is a schematic diagram of the ripple real-time cancellation circuit of the self-excited fluxgate sensor according to the sixth embodiment of the present invention.

[0079] Figure 13 The spectrum of the current ripple before cancellation was experimentally measured when the measured current was 0A.

[0080] Figure 14 The spectrum of the canceling current was experimentally measured when the measured current was 0A;

[0081] Figure 15 The spectrum of the canceled current ripple was experimentally measured when the measured current was 0A.

[0082] Figure 16 The spectrum of the current ripple before cancellation was experimentally measured when the measured current was 600A.

[0083] Figure 17 The spectrum of the canceling current was experimentally measured when the measured current was 600A;

[0084] Figure 18 The spectrum of the canceled current ripple was experimentally measured when the measured current was 600A. Detailed Implementation

[0085] To more clearly explain the present invention, the present invention will be further described in detail below with reference to the accompanying drawings and embodiments.

[0086] It should be noted that the specific embodiments described herein are merely illustrative of the invention and are not intended to limit the invention.

[0087] First Embodiment

[0088] Please see Figure 1 , Figure 2 This invention provides a real-time ripple cancellation circuit for a self-excited fluxgate sensor, including a measuring winding Lm, a measuring resistor Rm, a current sampling device, a ripple signal extraction module, and a real-time cancellation module; the ripple signal extraction module does not introduce any devices that cause phase changes.

[0089] The measuring resistor Rm is connected in series in the branch where the measuring winding Lm is located, and one end of it is grounded. The voltage signal across the measuring resistor Rm serves as the output signal of the self-excited oscillating fluxgate sensor.

[0090] In this case, the current sampling device is a sampling resistor Rs; one end of the sampling resistor Rs is connected to one end of the measuring winding Lm at node N1, and the other end of the sampling resistor Rs is connected to the ungrounded terminal of the measuring resistor Rm at node N2; it is used to sample the current signal flowing through the measuring winding in real time and output the signal to the ripple signal extraction module.

[0091] In this case, the ripple signal extraction module includes: a differential amplifier U1 and a high-pass filter HPF;

[0092] The inverting and non-inverting input terminals of the differential amplifier U1 are respectively connected to the two ends of the sampling resistor Rs to amplify the voltage difference across the resistor and output a voltage signal proportional to the current flowing through the sampling resistor Rs.

[0093] In this case, such as Figure 3 As shown, the differential amplifier U1 consists of operational amplifier A1, resistors R1, R2, R3, and R4. One end of resistor R1 is the inverting input of the differential amplifier U1, connected to node N2 (i.e., the ungrounded common connection node of sampling resistor Rs and measurement resistor Rm); the other end of resistor R1 is connected to the inverting input of operational amplifier A1. One end of resistor R2 serves as the non-inverting input of the differential amplifier U1, connected to node N1 (i.e., the common connection node of sampling resistor Rs and measurement winding Lm); the other end of resistor R2 is connected to the non-inverting input of operational amplifier A1. Resistor R3 is connected between the inverting input and output of operational amplifier A1. Resistor R4 is connected between the non-inverting input and ground of operational amplifier A1. The output of operational amplifier A1 serves as the output of the differential amplifier U1, connected to the input of the high-pass filter HPF.

[0094] The input terminal of the high-pass filter HPF is connected to the output terminal of the differential amplifier U1. The cutoff frequency fc of the high-pass filter HPF is set to be lower than the ripple frequency fr of the self-excited fluxgate but higher than the target current signal frequency, in order to filter out DC and low-frequency components while retaining high-frequency ripple components. The output terminal of the HPF provides a ripple reference signal for driving the real-time cancellation module.

[0095] The ripple frequency fr of the self-excited fluxgate is obtained by the following formula:

[0096] fr=Ve / (4×Bsat×Ae×Ne)

[0097] In the formula, Ve is the voltage (V) of the self-excited fluxgate excitation power supply, Bsat is the saturation magnetic induction intensity (T) of the excitation core, and Ae is the effective cross-sectional area (m²) of the excitation core. 2Ne represents the number of turns in the excitation winding. A high-pass filter is selected based on this frequency to ensure that the ripple is sampled without distortion and can be effectively and almost completely canceled in the later stages.

[0098] In this case, the real-time cancellation module consists of a proportional amplifier circuit U2 and a resistor Rf;

[0099] In this case, such as Figure 4 As shown, the proportional amplifier circuit U2 includes an operational amplifier A2, resistor R5, and resistor R6;

[0100] The output of the ripple signal extraction module is connected to the inverting input of operational amplifier A2 via resistor R6; the inverting input of operational amplifier A2 is also connected to its output via resistor R5; the non-inverting input of operational amplifier A2 is grounded; the output of operational amplifier A2 is connected to N2 via resistor Rf, and ripple current is injected in real time through resistor Rf.

[0101] In this case, the transconductance gain of the real-time cancellation module is: G2 = -R5 / [R6×(Rm+Rf)].

[0102] The voltage gain of the ripple signal extraction module is set to G1, and the transconductance gain G2 of the real-time cancellation module is defined as the ratio of the cancellation current i2(t) output by the module to the ripple reference signal voltage input to the module, satisfying the condition: Rs×G1×G2=-1.

[0103] Second Embodiment

[0104] Please see Figure 5 This invention provides a real-time ripple cancellation circuit for a self-excited fluxgate sensor.

[0105] The difference from the first embodiment is that the ripple signal extraction module and the real-time cancellation module are implemented differently in this case.

[0106] In this case, such as Figure 6 As shown, the differential amplifier U1 of the ripple signal extraction module is composed of operational amplifier A3, operational amplifier A4, operational amplifier A5, resistor R7, resistor R8, resistor R9, resistor R10, resistor R11, resistor R12, and resistor R13.

[0107] The non-inverting input (+) of the operational amplifier A3 serves as the inverting input of the differential amplifier U1 and is connected to the common node N1 of the sampling resistor Rs and the measurement winding; the inverting input (-) of the operational amplifier A3 is connected to its own output through resistor R8; and the output is connected to the inverting input of the operational amplifier A5 through resistor R10.

[0108] The non-inverting input (+) of the operational amplifier A4 serves as the non-inverting input of the differential amplifier U1 and is connected to the common node N2 of the sampling resistor Rs and the measuring resistor Rm; the inverting input (-) of the operational amplifier A4 is connected to its own output through resistor R9; and the output is connected to the non-inverting input (+) of the operational amplifier A5 through resistor R11.

[0109] The inverting input terminal (-) of operational amplifier A3 is connected to the inverting input terminal (-) of operational amplifier A4 via R7.

[0110] The inverting input (-) of operational amplifier A5 receives the output signal of A3 through resistor R10 and is connected to its own output through feedback resistor R12; the non-inverting input (+) of operational amplifier A5 receives the output signal of A4 through resistor R11 and is grounded through resistor R13; the output terminal serves as the output terminal of differential amplifier U1 and is connected to the input terminal of high-pass filter HPF.

[0111] The operational amplifiers A3 and A4 serve as input buffer stages, providing high input impedance and suppressing common-mode interference.

[0112] The operational amplifier A5 forms a differential amplifier stage to amplify the input differential signal, with a voltage gain of R12 / R10.

[0113] This circuit achieves a high common-mode rejection ratio (CMRR) through matching resistors (R8=R9, R10=R11, R12=R13), effectively extracting differential small signals. Resistor R7 is connected between the inverting input terminals of A3 and A4 to balance the input impedance.

[0114] In this case, the voltage gain G of the differential amplifier U1 is -(1+2×R8 / R7)×(R12 / R10).

[0115] In this case, such as Figure 7 As shown, the real-time cancellation module is composed of a voltage / current conversion circuit U3;

[0116] The voltage / current conversion circuit U3 is composed of operational amplifier A6, operational amplifier A7, resistor R14, resistor R15, resistor R16, resistor R17, and resistor Ro.

[0117] The operational amplifiers A6 and A7 constitute a two-stage amplification and feedback control; resistors R14, R15, R16, and R17 are matching resistors. o For transimpedance feedback resistor; input terminal u i Receive the ripple reference signal, connect the output terminal to the measurement winding injection node N2, and inject a reverse canceling current i2(t), i2(t) = i o (t);

[0118] The inverting input terminal (-) of the operational amplifier A6 is grounded through resistor R14 and connected to the output terminal u of the operational amplifier A6 through resistor R15. o1 (t); The non-inverting input terminal (+) is connected to the input u through resistor R16. i (t), and connected to the output terminal u of operational amplifier A7 via resistor R17. o2 (t);

[0119] The non-inverting input (+) of operational amplifier A7 is directly connected to the output node of the voltage / current conversion circuit; the inverting input (-) of operational amplifier A7 is connected to its output terminal u. o2 (t), forming unit negative feedback;

[0120] One end of resistor Ro is connected to the output terminal u of operational amplifier A6. o1 (t), the other end is connected to node N2, and the output current i o (t)=(u o1 (t)-u o2 (t)) / Ro, where the transconductance gain of the real-time cancellation module is G2=1 / Ro.

[0121] The constraints of the voltage / current conversion circuit are:

[0122] The four resistors R14, R15, R16, and R17 are all the same value, i.e., R14 = R15 = R16 = R17.

[0123] The voltage gain of the ripple signal extraction module is set to G1, and the transconductance gain G2 of the real-time cancellation module is defined as the ratio of the cancellation current i2(t) output by the module to the ripple reference signal voltage input to the module, satisfying the condition: Rs×G1×G2=-1.

[0124] Compared with the first embodiment, the ripple signal extraction module used in the second embodiment extracts ripple signals with higher accuracy.

[0125] Compared with the first embodiment, the real-time cancellation module used in the second embodiment has stronger applicability and is easier to design.

[0126] Third Embodiment

[0127] Please see Figure 8 This invention provides a real-time ripple cancellation circuit for a self-excited fluxgate sensor.

[0128] The difference from the first embodiment is that, in this case, the real-time cancellation module further includes a DC blocking capacitor Co and a bleed resistor Rb. The DC blocking capacitor Co is used to eliminate the DC bias error at the output of the real-time cancellation module (such as DC offset caused by operational amplifier offset voltage and bias current), preventing it from being superimposed on the measurement branch and affecting the sensor accuracy; C o A series resistor Rf and a measuring resistor Rm are connected in series to form an AC coupling path; one end of Co is connected to resistor Rf, and the other end is connected to the injection node N2. The capacitance value must be selected to satisfy [1 / (2π×fr×Co)] << Rm to ensure that the impedance of Co is negligible at the ripple frequency. The ripple frequency fr is determined by the following formula:

[0129] fr=Ve / (4×Bsat×Ae×Ne)

[0130] In the formula, Ve is the voltage (V) of the self-excited fluxgate excitation power supply, Bsat is the saturation magnetic induction intensity (T) of the excitation core, and Ae is the effective cross-sectional area (m²) of the excitation core. 2 ), where Ne is the number of turns in the excitation winding.

[0131] The discharge resistor Rb is connected in parallel across Co to discharge the charge accumulated in the capacitor; Co is selected as a ceramic capacitor or film capacitor with low series equivalent resistance (ESR).

[0132] The voltage gain of the ripple signal extraction module is set to G1; the transconductance gain G2 of the real-time cancellation module is defined as the ratio of the cancellation current i2(t) output by the module to the ripple reference signal voltage input to the module; the condition is satisfied: Rs×G1×G2=-1.

[0133] Fourth embodiment

[0134] Please see Figure 9 This invention provides a real-time ripple cancellation circuit for a self-excited fluxgate sensor.

[0135] The difference from the first embodiment is that, in this case, the real-time cancellation module further includes a power amplifier circuit;

[0136] The power amplifier circuit serves as a current enhancement stage to improve the load capacity of the real-time cancellation module, ensuring that sufficient cancellation current can still be output under large ripple current scenarios to avoid signal clipping or distortion.

[0137] The input terminal of the power amplifier circuit PA is connected to the output terminal of the proportional amplifier circuit U2, and the output terminal is connected to node N2 via resistor Rf, which is used to inject high-power cancellation current into the measurement branch; the proportional coefficient of the real-time cancellation module satisfies: G2=-G3×R5 / [R6×(Rm+Rf)], where: G3 is the gain of the power amplifier circuit.

[0138] The voltage gain of the ripple signal extraction module is set to G1; the transconductance gain G2 of the real-time cancellation module is defined as the ratio of the cancellation current i2(t) output by the module to the ripple reference signal voltage input to the module; the condition is satisfied: Rs×G1×G2=-1.

[0139] Fifth embodiment

[0140] Please see Figure 10 This invention provides a real-time ripple cancellation circuit for a self-excited fluxgate sensor.

[0141] The difference from the first embodiment is that the current sampling device and the ripple signal extraction module are different in this case.

[0142] In this case, the current sampling device is a sampling inductor Ls; the ripple signal extraction module includes a differential integration circuit U4 and a high-pass filter HPF.

[0143] The sampling inductor Ls is connected in series in the branch where the measuring winding is located. The sampling inductor Ls converts the real-time acquired mixed current signal i1(t) flowing through the measuring winding into a voltage signal v. s (t), to achieve low-loss current sampling:

[0144]

[0145] In the formula, i 1-AC (t) is the AC component of i1(t);

[0146] One end of the sampling inductor Ls is connected to one end of the measuring winding Lm at node N1, and the other end of the sampling inductor Ls is connected to the ungrounded terminal of the measuring resistor Rm at node N2.

[0147] The non-inverting input of the differential integrator circuit U4 is connected to the common node N1 of Ls and the measuring winding, and the inverting input is connected to the common node N2 of Ls and Rm.

[0148] In this case, such as Figure 11 As shown, the differential integration circuit U4 is composed of operational amplifier A8, resistor R18, resistor R19, capacitor C1, and capacitor C2.

[0149] One end of resistor R18 is connected to the inverting input of differential integrator circuit U4 and the common node N2 of Ls and Rm. The other end of resistor R18 is connected to the inverting input of operational amplifier A8. One end of resistor R19 is connected to the non-inverting input of differential integrator circuit U4 and the common node N1 of Ls and the measurement winding. The other end of resistor R19 is connected to the non-inverting input of operational amplifier A8. One end of capacitor C1 is connected to the inverting input of operational amplifier A8, and the other end of capacitor C1 is connected to the output of operational amplifier A8. One end of capacitor C2 is connected to the non-inverting input of operational amplifier A8, and the other end of capacitor C2 is grounded. The output of operational amplifier A8 is the output of differential integrator circuit U4 and is connected to the input of high-pass filter HPF.

[0150] In this embodiment, R18 = R19 = R, C1 = C2 = C.

[0151] The input v is processed by the differential integrator circuit. s (t) is processed to obtain the output signal.

[0152]

[0153] In the formula V o1 The DC component is the component, while the AC component is the component. The AC component includes ripple signals;

[0154] The input terminal of the high-pass filter HPF is connected to the output terminal of the differential amplifier U4. The cutoff frequency fc of the high-pass filter HPF is set to be lower than the ripple frequency fr of the self-excited fluxgate but higher than the target current signal frequency, so as to filter out the DC bias and low-frequency target signal and retain the high-frequency ripple component. The output terminal of HPF provides the inverted ripple reference signal for driving the real-time cancellation module.

[0155] The transconductance gain of the real-time cancellation module is set to G2, which is defined as the ratio of the cancellation current i2(t) output by the module to the ripple reference signal voltage input to the module; the condition must be met: Ls / (R×C)×G2=-1.

[0156] Compared with the first embodiment, the ripple signal extraction module used in the fifth embodiment uses a sampling inductor, which can reduce power consumption, but the design is slightly more complicated than that of a sampling resistor.

[0157] Sixth Embodiment

[0158] Please see Figure 12 This invention provides a real-time ripple cancellation circuit for a self-excited fluxgate sensor.

[0159] The difference from the first embodiment is that, in this case, the real-time cancellation module further includes a power amplifier circuit, a DC blocking capacitor Co, and a bleed resistor Rb.

[0160] The input terminal of the power amplifier circuit is connected to the output terminal of the proportional amplifier circuit U2. The output terminal of the power amplifier circuit is connected to one end of the resistor Rf. The other end of the resistor Rf is connected to one end of the DC blocking capacitor Co. The other end of the DC blocking capacitor Co is connected to the injection node N2.

[0161] The voltage gain of the ripple signal extraction module is set to G1, and the transconductance gain G2 of the real-time cancellation module is defined as the ratio of the cancellation current i2(t) output by the module to the ripple reference signal voltage input to the module; the condition must be met: Rs×G1×G2=-1.

[0162] The applicant fabricated a prototype based on the circuit of the first embodiment and conducted tests. The experimental results are as follows: Figures 13 to 18 As shown.

[0163] The ripple current is measured using a current-to-voltage converter with a transimpedance gain of 1000, and its corresponding voltage spectrum is as follows: Figure 13 , Figure 15 , Figure 16 and Figure 18 As shown; the cancellation current is directly measured through resistor Rf (transimpedance gain is 10000 times), and its corresponding voltage spectrum is as follows. Figure 14 and Figure 17 As shown.

[0164] By comparison Figure 13 and Figure 16 It can be seen that the spectrum of the ripple current changes significantly when the measured current is different, specifically manifested in the differences in the peak values ​​and frequencies of the first and third harmonics in the figure. Therefore, the canceling current cannot be fixed; it must track the changes in the ripple current in real time, that is, maintain it to be equal in magnitude and opposite in phase to the measured ripple current (e.g., ...). Figure 14 and Figure 17 (As shown). Figure 15 and Figure 18 The results show that the real-time cancellation circuit of this application achieves good ripple cancellation effect under different measured current conditions.

[0165] Similarly, the present invention can also perform real-time cancellation when the circuit environment changes.

[0166] Any aspects not covered in this invention are applicable to existing technologies.

Claims

1. A real-time ripple cancellation circuit for a self-excited oscillating fluxgate sensor, comprising a measuring winding Lm and a measuring resistor Rm, wherein the measuring resistor Rm is connected in series in the branch containing the measuring winding Lm, and one end of the resistor Rm is grounded, and the voltage signal across the measuring resistor Rm serves as the output signal of the self-excited oscillating fluxgate sensor; characterized in that, Also includes: Current sampling device, ripple signal extraction module, and real-time cancellation module; The current sampling device is used to sample the current signal flowing through the measurement winding in real time and output the signal to the ripple signal extraction module. The ripple signal extraction module samples the mixed signal flowing through the measurement winding in real time through the current sampling device, separates the ripple component from the mixed signal, and outputs a ripple reference signal representing the ripple component to the real-time cancellation module. The real-time cancellation module is configured to generate a cancellation current with the same amplitude and opposite phase as the ripple current based on the ripple reference signal, and inject the cancellation current into the measuring resistor Rm, thereby canceling the ripple current in the measuring resistor Rm in real time.

2. The ripple real-time cancellation circuit according to claim 1, characterized in that, The current sampling device is selected from the sampling resistor Rs; The ripple signal extraction module includes: a differential amplifier U1 and a high-pass filter HPF; One end of the sampling resistor Rs is connected to one end of the measuring winding Lm at node N1, and the other end of the sampling resistor Rs is connected to the ungrounded terminal of the measuring resistor Rm at node N2. The inverting and non-inverting input terminals of the differential amplifier U1 are respectively connected to the two ends of the sampling resistor Rs to amplify the voltage difference across the resistor and output a voltage signal proportional to the current flowing through the sampling resistor Rs. The input terminal of the high-pass filter HPF is connected to the output terminal of the differential amplifier U1. Its cutoff frequency fc is set to be lower than the ripple frequency fr of the self-excited fluxgate but higher than the target current signal frequency, in order to filter out DC and low-frequency components while retaining high-frequency ripple components; the output terminal of the HPF provides a ripple reference signal. The voltage gain of the ripple signal extraction module is set to G1, and the transconductance gain G2 of the real-time cancellation module is defined as the ratio of the output cancellation current i2(t) to the ripple reference signal voltage input to the real-time cancellation module, satisfying the condition: Rs×G1×G2=-1.

3. The ripple real-time cancellation circuit according to claim 2, characterized in that, The differential amplifier U1 is composed of operational amplifier A1, resistors R1, R2, R3, and R4. One end of resistor R1 serves as the inverting input of the differential amplifier U1 and is connected to node N2; the other end of resistor R1 is connected to the inverting input of operational amplifier A1. One end of resistor R2 serves as the non-inverting input of the differential amplifier U1 and is connected to node N1; the other end of resistor R2 is connected to the non-inverting input of operational amplifier A1. Resistor R3 is connected between the inverting input and output of operational amplifier A1. Resistor R4 is connected between the non-inverting input of operational amplifier A1 and ground. The output of operational amplifier A1 serves as the output of the differential amplifier U1 and is connected to the input of the high-pass filter HPF.

4. The ripple real-time cancellation circuit according to claim 2, characterized in that, The differential amplifier U1 includes operational amplifiers A3, A4, A5 and resistors R7 to R13; The non-inverting input (+) of the operational amplifier A3 is connected to the common node N1 of Rs and the measurement winding, and the inverting input (-) is connected to its output through R8. The output is connected to the inverting input (-) of A5 through R10. The non-inverting input (+) of the operational amplifier A4 is connected to the common node N2 of Rs and Rm, the inverting input (-) is connected to its output through R9, and the output is connected to the non-inverting input (+) of A5 through R11. The inverting input terminal (-) of the operational amplifier A3 is connected to the inverting input terminal (-) of the operational amplifier A4 via R7; The inverting input terminal (-) of the operational amplifier A5 is connected to its output terminal via R12, and the non-inverting input terminal (+) is grounded via R13. The output terminal of the operational amplifier A5 serves as the output of the differential amplifier U1. The resistor matching conditions are: R8 = R9, R10 = R11, R12 = R13; The voltage gain G of the differential amplifier U1 is -(1+2×R8 / R7)×(R12 / R10).

5. The ripple real-time cancellation circuit according to claim 1, characterized in that, The current sampling device is selected from the sampling inductor Ls; the ripple signal extraction module includes a differential integration circuit U4 and a high-pass filter HPF. One end of the sampling inductor Ls is connected to one end of the measuring winding Lm at node N1, and the other end of the sampling inductor Ls is connected to the ungrounded terminal of the measuring resistor Rm at node N2. The non-inverting input of the differential integrator circuit U4 is connected to the common node N1 of Ls and the measuring winding, and the inverting input is connected to the common node N2 of Ls and Rm. The differential integration circuit U4 is composed of operational amplifier A8, resistor R18, resistor R19, capacitor C1, and capacitor C2. The inverting input terminal of operational amplifier A8 is connected to node N2 through resistor R18, the non-inverting input terminal is connected to node N1 through resistor R19, capacitor C1 is connected between the inverting input terminal and the output terminal, and the non-inverting input terminal is grounded through capacitor C2. Where R18 = R19 = R, C1 = C2 = C; The output v1(t) of the differential integrator circuit: In the formula, V o1 This is the DC component; For AC components, including ripple signals; i 1-AC (t) is the AC component of i1(t); t represents time; i1(t) is the mixed current signal flowing through the measuring winding; The transconductance gain of the real-time cancellation module is set to G2, which is defined as the ratio of the cancellation current i2(t) output by the module to the ripple reference signal voltage input to the module. It must satisfy the condition: Ls / (R×C)×G2=-1.

6. The ripple real-time cancellation circuit for the self-excited fluxgate sensor according to any one of claims 1-5, characterized in that, The real-time cancellation module consists of a proportional amplifier circuit U2 and a resistor Rf. The proportional amplifier circuit U2 includes an operational amplifier A2, a resistor R5, and a resistor R6. The output of the ripple signal extraction module is connected to the inverting input of operational amplifier A2 via resistor R6; the inverting input of operational amplifier A2 is also connected to its output via resistor R5; the non-inverting input of operational amplifier A2 is grounded; the output of operational amplifier A2 is connected to N2 via resistor Rf, and ripple current is injected in real time through resistor Rf. The transconductance gain of the real-time cancellation module is: G2=-R5 / [R6×(Rm+Rf)].

7. The ripple real-time cancellation circuit according to claim 6, characterized in that, The real-time cancellation module also includes a DC blocking capacitor Co and a bleed resistor Rb; The DC blocking capacitor Co is connected in series between the resistor Rf and node N2, and its capacitance value satisfies [1 / (2π×fr×Co)]<<Rm, where the ripple frequency fr is determined by the following formula: fr=Ve / (4×Bsat×Ae×Ne) In the formula, Ve is the voltage (V) of the self-excited fluxgate excitation power supply, Bsat is the saturation magnetic induction intensity (T) of the excitation core, and Ae is the effective cross-sectional area (m²) of the excitation core. 2 Ne is the number of turns in the excitation winding; The discharge resistor Rb is connected in parallel across Co to discharge the charge accumulated in the capacitor; Co is selected as a ceramic capacitor or film capacitor with low series equivalent resistance (ESR).

8. The ripple real-time cancellation circuit according to claim 6, characterized in that, The real-time cancellation module also includes a power amplifier circuit, whose input is connected to the output of the proportional amplifier circuit U2, and whose output is connected to node N2 via resistor Rf, for injecting high-power cancellation current into the measurement branch; at this time, the transconductance gain of the real-time cancellation module satisfies: G2=-G3×R5 / [R6×(Rm+Rf)], where: G3 is the gain of the power amplifier circuit.

9. The ripple real-time cancellation circuit according to claim 6, characterized in that, The real-time cancellation module includes a power amplifier circuit, a DC blocking capacitor Co, and a bleed resistor Rb. The power amplifier circuit is located between the resistor Rf and the output of the proportional amplifier circuit U2. The DC blocking capacitor Co and the bleed resistor Rb are connected in parallel and then in series between the resistor Rf and node N2.

10. The ripple real-time cancellation circuit for the self-excited fluxgate sensor according to any one of claims 1-5, characterized in that, The real-time cancellation module is a voltage / current conversion circuit U3, including operational amplifiers A6 and A7, resistors R14 to R17, and resistor Ro; R14 = R15 = R16 = R17. The inverting input terminal (-) of the operational amplifier A6 is grounded through resistor R14 and connected to the output terminal u of the operational amplifier A6 through resistor R15. o1 (t); The non-inverting input terminal (+) is connected to the input u through resistor R16. i (t), and connected to the output terminal u of operational amplifier A7 via resistor R17. o2 (t); The non-inverting input (+) of operational amplifier A7 is directly connected to the output node of the voltage / current conversion circuit; the inverting input (-) of operational amplifier A7 is connected to its output terminal u. o2 (t), forming unit negative feedback; One end of resistor Ro is connected to the output terminal u of operational amplifier A6. o1 (t), the other end is connected to node N2, and the output current i o (t)=(u o1 (t)-u o2 (t)) / Ro, where the transconductance gain of the real-time cancellation module is G2=1 / Ro.

Images