A switching control method based on SPWM soft start and pulse type load LLC resonant converter

Abstract

The application discloses a switching control method based on SPWM soft starting and a pulse type load LLC resonant converter. k The converter mainly comprises an LLC resonant converter, an H-bridge inverter circuit and a load thereof, and the control method is as follows: k For any positive integer, the load state is accurately judged through an H-bridge switching logic unit, and burst intermittent control and linear frequency control are respectively adopted in the no-load state and the full-load state in the steady state; when the load is switched, two-pulse optimal control is carried out by using a nonlinear SOTC optimal trajectory controller, so that the state is transitioned to a new mode along the shortest trajectory, and a dynamic response optimizer is introduced to compensate the frequency after switching according to the state radius error, so as to suppress voltage secondary fluctuation, solve the problems of slow dynamic response and large output voltage fluctuation of a traditional linear control facing the pulse type load LLC resonant converter, and improve the output voltage stability and anti-interference ability of the system.

Description

Technical Field

[0001] This invention relates to the field of power electronic converter technology, and in particular to a switching control method based on SPWM soft start and pulse-type load LLC resonant converter. Background Technology

[0002] With the continuous development of power electronics technology, LLC resonant converters, due to their ability to achieve zero-voltage turn-on of the primary-side switch and zero-current turn-off of the secondary-side rectifier, are widely used in applications requiring high efficiency and high power density, such as data centers, new energy power generation, and industrial power supplies. This solves the problems of high switching losses and severe electromagnetic interference inherent in traditional hard-switching converters. However, traditional LLC resonant converters still have significant shortcomings during startup and under extreme conditions of sudden load changes from no-load to full-load. Limited by the finite bandwidth of linear control methods, the system's dynamic response is slow. When dealing with pulse-type loads, such as the front-end DC-DC isolation unit in a cascaded H-bridge photovoltaic inverter, the output voltage exhibits significant overshoot or drop, severely affecting the stable operation of downstream equipment and the overall reliability of the system.

[0003] This invention discloses a switching control method for an LLC resonant converter based on SPWM soft start and pulse-type load. Addressing the issues of slow response and large fluctuations in traditional control methods under dynamic conditions, this method applies SPWM modulation to the soft start stage of the LLC resonant converter. The converter load state is determined based on the H-bridge switching modes, and Burst control, linear control, and nonlinear SOTC optimal trajectory control are dynamically switched. A dynamic response optimizer is introduced for frequency compensation to suppress secondary voltage fluctuations. This invention achieves smooth start-up of the LLC resonant converter output voltage and rapid stabilization of the output voltage under no-load and full-load switching conditions, significantly improving the steady-state accuracy and operational reliability of the system. Summary of the Invention

[0004] This invention discloses a switching control method for an LLC resonant converter based on SPWM soft start and pulsed load, and proposes a corresponding multi-mode switching control mechanism. It employs an SPWM modulation strategy to achieve voltage soft start, and utilizes H-bridge load state identification for Burst control, linear control, and nonlinear SOTC optimal trajectory control. The converter mainly includes an LLC resonant converter, a subsequent H-bridge inverter circuit, and the load. This invention combines SPWM modulation with H-bridge load state identification to achieve stable control of the LLC resonant converter during soft start and pulsed load switching. Unlike traditional linear control, this invention, through SPWM soft start and multi-mode switching, offers advantages such as fast dynamic response, small output voltage fluctuation, low switching stress, and high reliability. It aims to solve the problems of slow dynamic response, large voltage overshoot, and poor system stability in traditional LLC resonant converter control methods under startup and pulsed loads, expanding the application of LLC resonant converters in high-end industrial power supplies and new energy power generation.

[0005] An LLC resonant converter based on SPWM soft start and pulse-type load includes: a main circuit unit and a main control unit;

[0006] The main circuit unit includes: a DC power supply. V in Resonant capacitor C r Resonant inductor L r ,transformer T Including magnetizing inductor L m Input capacitor C in Output capacitor C o First switching transistor S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S 4. Fifth switching transistor S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S 8. First diode D 1. Second diode D 2. Third diode D 3. Fourth diode D 4. Load R L ;

[0007] The main control unit includes: an SPWM soft-start controller, a drive circuit, an H-bridge mode recognizer, a Burst controller, a PI controller, an SOTC controller, an adder SUM, a state observer, a dynamic response optimizer, and a voltage-controlled oscillator.

[0008] Preferably, the first switching transistor S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S 4 is a high-frequency MOSFET; preferably, the fifth switching transistor... S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S 8 represents low-frequency IGBTs;

[0009] Preferably, the first switching transistor S 1. Third switching transistor S Three form a group, the second switching transistor S 2. Fourth switching transistor S 4 form another group, and the two groups of switching transistors are turned on alternately;

[0010] Preferably, the fifth switching transistor S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S 8 form an H-bridge inverter circuit. The H-bridge uses low-frequency square wave modulation to provide a pulsed load.

[0011] Preferably, the resonant capacitor C r Resonant inductor L r Magnetizing inductor L m It forms a resonant cavity.

[0012] A specific control method based on SPWM soft start and pulse-type load LLC resonant converter is as follows:

[0013] S1: The SPWM soft-start controller generates according to the preset switching point. k The SPWM modulation signal is passed through the driver circuit to the first switch transistor. S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S The gate of 4 is driven to drive the output capacitor. C o Pre-charge to achieve output voltage V oThe process proceeds smoothly, completing the soft start;

[0014] S2: The fifth switching transistor S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S 8. Inverter conduction begins; the H-bridge mode recognizer detects the fifth switch. S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S The gate drive signal of 8 is used for mode recognition to determine the load. R L It is equivalent to being in no-load, full-load, or no-full-load switching state on the LLC resonant converter side, and outputs a mode indication signal model. The mode indication signal mode is used to indicate three working modes: no-load (model=0), full-load (model=1), and no-full-load switching (model=2).

[0015] S3: The H-bridge mode recognizer sends an enable command to the Burst controller, PI controller or SOTC controller according to the mode indication signal model;

[0016] If model=0 (no load), the Burst controller is enabled, and it switches between burst-on and burst-off states according to the voltage hysteresis control principle. In the burst-on state, the first switching transistor is driven. S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S 4. To maintain the output voltage, the output voltage V o Rise; in the burst-off state, turn off the first switch. S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S 4. Cut off the energy transfer from the primary to the secondary side of the transformer, and the output voltage... V o decline;

[0017] If model=1 (full load), the PI controller is enabled, and the output voltage... V o With reference voltage V ref The adder SUM compares the output error voltage. V e As the input signal to the PI controller, the error voltageV e The PI controller outputs an adjustment signal, which is then passed through the voltage-controlled oscillator to the first switching transistor. S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S The drive signal of 4 is used by the LLC resonant converter to perform frequency conversion regulation by pulse frequency modulation.

[0018] If model=2 (empty-full switching), the SOTC controller is enabled, and two optimal switching pulse widths are calculated based on the simplified optimal trajectory control theory, driving the first switching transistor respectively. S 1. Third switching transistor S 3 and the second switching transistor S 2. Fourth switching transistor S 4. Enables rapid switching between empty and full modes;

[0019] S4: After the two switching pulses of the SOTC controller have been executed, the dynamic response optimizer is enabled, and the state observer calculates the input resonant inductor current. i Lr Output normalized resonant capacitor voltage v CrN and resonant inductor current i LrN Constituting the radius of the state ρ s The dynamic response optimizer is based on the state radius. ρ s With the target radius ρ ref deviation Δ e Generate frequency compensation command u T This suppresses potential secondary voltage oscillations that may occur after SOTC switching. Furthermore, if the radius deviation Δ... e If the value is within the preset range, the dynamic response optimizer will exit and the PI controller will be entered directly.

[0020] As can be seen from the above technical methods, the embodiments of the present invention have the following beneficial effects:

[0021] This invention discloses a switching control method based on SPWM soft start and pulse-type load LLC resonant converter, and proposes a corresponding multi-mode switching control mechanism. k The multi-stage SPWM modulation strategy suppresses the startup inrush voltage. kThe input is any positive integer greater than 1. The load state is accurately determined by detecting the state of the H-bridge switches. This allows for switching between Burst control, linear control, and nonlinear SOTC optimal trajectory control. A dynamic response optimizer is introduced after SOTC control, generating a frequency compensation signal based on the system state trajectory radius deviation. Compared to traditional methods, this invention aims to improve the dynamic response performance of LLC resonant converters for pulsed loads, suppress output voltage fluctuations, reduce switching stress, and improve system reliability. It is suitable for applications such as photovoltaic inverters and industrial power supplies where pulsed loads are caused by the periodic switching of the H-bridge, thus expanding the application prospects of LLC resonant converters in the fields of new energy and high-efficiency power conversion. Attached Figure Description

[0022] To more clearly illustrate the technical solutions in the embodiments of the present invention, the accompanying drawings will be briefly introduced below. The technical solutions in the embodiments of the present invention will be clearly and completely described in conjunction with the accompanying drawings. Obviously, the described embodiments are some embodiments of the present invention. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0023] Figure 1 This is a schematic diagram of a SPWM soft-start and pulse-type load LLC resonant converter in an embodiment of the present invention;

[0024] Figure 2 This is a control flowchart of an LLC resonant converter based on SPWM soft start and pulse-type load in an embodiment of the present invention;

[0025] Figure 3a This is a schematic diagram illustrating the working principle of an SPWM soft-start converter based on a pulse-type load LLC resonant converter in an embodiment of the present invention.

[0026] Figure 3b This is a simulation waveform diagram of SPWM soft-start based on an LLC resonant converter with pulse-type load in an embodiment of the present invention.

[0027] Figure 4 This is a schematic diagram of the working waveform of an idle intermittent Burst controller based on SPWM soft start and pulse-type load LLC resonant converter in an embodiment of the present invention.

[0028] Figure 5 This is a schematic diagram illustrating the optimal state trajectory analysis of no-load switching for an LLC resonant converter based on SPWM soft start and pulsed load in an embodiment of the present invention.

[0029] Figure 6This is a structural diagram of a common DC bus direct-connected CHB photovoltaic inverter system based on an SPWM soft-start and pulse-type load LLC resonant converter application scenario in an embodiment of the present invention. Detailed Implementation

[0030] To clearly illustrate the objectives, technical solutions, and features of the embodiments of the present invention, the technical solutions of the embodiments of the present invention will be further described below in conjunction with the prior art and the accompanying drawings. Obviously, the described embodiments are only a part of the embodiments of the present invention, and not all of them. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort should fall within the scope of protection of the present invention.

[0031] Please refer to Figure 1 , Figure 1 This is a schematic diagram of the structure of an LLC resonant converter based on SPWM soft start and pulse-type load according to the present invention, including: a main circuit unit and a main control unit;

[0032] The main circuit unit includes: a DC power supply. V in Resonant capacitor C r Resonant inductor L r ,transformer T Including magnetizing inductor L m Input capacitor C in Output capacitor C o First switching transistor S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S 4. Fifth switching transistor S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S 8. First diode D 1. Second diode D 2. Third diode D 3. Fourth diode D 4. Load R L The specific connection method is as follows:

[0033] The DC power supply V in The positive terminal and the input capacitor C in The first terminal, the first switching transistorS The drain of 1, the third switching transistor S 3. Drain connection; the DC power supply V in The negative terminal and the input capacitor C in The second terminal, the second switching transistor S The source of 2, the fourth switch transistor S 4 source connections;

[0034] The resonant capacitor C r The first end and the first switch transistor S The source of 1, the second switch transistor S 2. Drain connection; the resonant capacitor C r The second end is connected to the resonant inductor L r The first end is connected;

[0035] The transformer T The first end of the primary winding is connected to the resonant inductor L r The second end is connected; the transformer T The second end of the primary winding and the third switching transistor S The source of 3, the fourth switch transistor S 4 drain connection;

[0036] The excitation inductor L m Parallel connection to transformer T Primary winding;

[0037] The transformer T The first end of the secondary winding is connected to the first diode. D anode of 1, second diode D 2. Cathode connection; the transformer T The second end of the secondary winding is connected to the third diode. D The anode of 3, the fourth diode D 4. Cathode connection;

[0038] The first diode D The cathode of 1 and the third diode D 3's cathode, the output capacitor C o The first terminal, the fifth switching transistor S The collector of 5, the seventh switch transistor S 7. Collector connection;

[0039] The second diodeD The anode of 2 and the fourth diode D 4's anode, the output capacitor C o The second end, the sixth switch tube S The emitter of 6, the eighth switch transistor S 8 emitter connection;

[0040] The load R L The first terminal and the fifth switch tube S The emitter of 5, the sixth switch transistor S 6. Collector connection; the load R L The second terminal is connected to the seventh switch. S The emitter of 7, the eighth switch transistor S 8's collector connection.

[0041] The main control unit includes: an SPWM soft-start controller, a drive circuit, an H-bridge mode recognizer, a Burst controller, a PI controller, an SOTC controller, an adder SUM, a state observer, a dynamic response optimizer, and a voltage-controlled oscillator. The specific connection method is as follows:

[0042] The first input terminal of the drive circuit is connected to the output terminal of the SPWM soft-start controller, the second input terminal of the drive circuit is connected to the output terminal of the Burst controller, the third input terminal of the drive circuit is connected to the output terminal of the voltage-controlled oscillator, and the fourth input terminal of the drive circuit is connected to the first output terminal of the SOTC controller; the first output terminal of the drive circuit is connected to the first switching transistor. S The gate of 1 is connected, and the second output terminal of the driving circuit is connected to the second switching transistor. S The gate of 2 is connected, and the third output terminal of the driving circuit is connected to the third switching transistor. S The gate of 3 is connected, and the fourth output terminal of the driving circuit is connected to the fourth switching transistor. S 4. Gate connection; the first input terminal of the H-bridge mode recognizer is connected to the fifth switch. S The gate connection of 5, the second input terminal of the H-bridge mode recognizer is connected to the sixth switch. S The gate connection of 6, the third input terminal of the H-bridge mode recognizer is connected to the seventh switch. S The gate connection of 7, the fourth input terminal of the H-bridge mode recognizer is connected to the eighth switch. SThe gate of the 8 is connected; the first output of the H-bridge mode recognizer is connected to the input of the Burst controller, the second output of the H-bridge mode recognizer is connected to the negative input of the adder SUM, and the third output of the H-bridge mode recognizer is connected to the input of the SOTC controller; the input of the PI controller is connected to the output of the adder SUM; the output of the PI controller is connected to the first input of the voltage-controlled oscillator; the first input of the dynamic response optimizer is connected to the output of the state observer, the second input of the dynamic response optimizer is connected to the second output of the SOTC controller, and the output of the dynamic response optimizer is connected to the second input of the voltage-controlled oscillator.

[0043] Preferably, the first switching transistor S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S 4 is a high-frequency MOSFET; preferably, the fifth switching transistor... S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S 8 represents low-frequency IGBTs;

[0044] Preferably, the first switching transistor S 1. Third switching transistor S Three form a group, the second switching transistor S 2. Fourth switching transistor S 4 form another group, and the two groups of switching transistors are turned on alternately;

[0045] Preferably, the fifth switching transistor S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S 8 form an H-bridge inverter circuit. The H-bridge uses low-frequency square wave modulation to provide a pulsed load.

[0046] Preferably, the resonant capacitor C r Resonant inductor L r Magnetizing inductor L m It forms a resonant cavity.

[0047] Please refer to Figure 2 , Figure 2 This is a control flowchart of an LLC resonant converter based on SPWM soft start and pulse-type load according to the present invention. The specific control method is as follows:

[0048] S1: As Figure 3a The diagram shows the SPWM soft-start controller generating power according to a preset switching point. k The working principle diagram of the SPWM modulation signal is shown. The first switch is driven by the driving circuit. S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S The gate of 4 is driven, in the first k One switching point resonant cavity V AB ( t The voltage of ) satisfies the following equation:

[0049]

[0050] In the formula m k For the first k SPWM modulation level, w m The carrier angular frequency is given, where the SPWM carrier frequency is equal to the steady-state switching frequency of the LLC resonant converter. Multi-stage SPWM switching affects the output capacitor. C o Pre-charge to achieve output voltage V o After a smooth rise and completion of the soft-start process, the LLC resonant converter enters pulse frequency modulation with a 50% duty cycle. For example... Figure 3b The simulation waveform diagram shown indicates that the present invention k The SPMW modulation strategy can significantly reduce the startup time and output voltage surge of the LLC resonant converter. Furthermore, by rationally setting the modulation magnitude and switching time, k It can make the output voltage V o To achieve a theoretically smooth ascent, k It is any positive integer greater than 1.

[0051] S2: The fifth switching transistor S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S 8. Inverter conduction begins; the H-bridge mode recognizer detects the fifth switch. S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S The gate drive signal of 8 is used for mode recognition to determine the load. R LThe equivalent state on the LLC resonant converter side is no-load, full-load, or no-full-load switching state, and outputs a mode indication signal model; the mode indication signal mode is used to indicate three working modes: no-load (model=0), full-load (model=1), and no-full-load switching (model=2).

[0052] If the fifth switching transistor S 5. Seventh switching transistor S 7 or the sixth switch tube S 6. Eighth switching transistor S If switch 8 is on, then model = 0; if the fifth switch is on... S 5. Eighth switching transistor S 8 or the sixth switch tube S 6. Seventh switching transistor S If signal 7 is turned on, then model=1; if signal model changes from 0 to 1, then model=2.

[0053] S3: The H-bridge mode recognizer sends an enable command to the Burst controller, PI controller or SOTC controller according to the mode indication signal model;

[0054] If model=0 (no load), the Burst controller is enabled, and the LLC resonant converter switches between burst-on and burst-off states according to voltage hysteresis control. Figure 4 As shown. In burst-on state, the output voltage... V o Below the lower limit V omin Drive the first switch transistor S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S 4. To maintain the output voltage, the output voltage V o Rise; in burst-off state, output voltage V o Higher than the upper limit V omax Turn off the first switch transistor S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S 4. Cut off the energy transfer from the primary to the secondary side of the transformer, and the output voltage... V o The descent, intermittent Burst control mode is shown below:

[0055]

[0056] In the formula V omin This represents the lower limit of the no-load output voltage. V omax This is the upper limit of the no-load output voltage.

[0057] If model=1 (full load), the PI controller is enabled, and the output voltage... V o With reference voltage V ref The adder SUM compares the output error voltage. V e As the input signal to the PI controller, the error voltage V e The PI controller outputs an adjustment signal, which is then passed through the voltage-controlled oscillator to the first switching transistor. S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S The drive signal of 4 is used for closed-loop frequency conversion regulation of the LLC resonant converter by pulse frequency modulation.

[0058] If model=2 (empty-full switching), the SOTC controller is enabled. Based on simplified optimal trajectory control theory, the SOTC method adjusts the resonant capacitor voltage. v Cr and resonant inductor current i Lr After normalization, the state plane trajectory diagrams of these two are established, as follows: Figure 5 The working process of the LLC resonant converter is analyzed as shown.

[0059] The equation of the circular trajectory in the state plane under full load is:

[0060]

[0061] In the formula v CrN Resonant capacitor voltage v Cr The per-unit value, i.e. v CrN= v Cr / V in , i LrN For the resonant inductor current i Lr The per-unit value, i.e. i LrN=i Lr / ( V in / Z r ), The characteristic impedance is a binary element. The radius of the fully loaded circular trajectory. ρ Equal to the resonant inductor current i Lr The per-unit value of the peak value, i.e. .

[0062] I rms The resonant inductor current under full load i Lr Valid values:

[0063]

[0064] In the formula T The period of resonance, n This refers to the number of transformer turns. i o For load R L Electric current.

[0065] The equation of the elliptical trajectory in the state plane under no-load conditions is:

[0066]

[0067] In the formula It is a ternary characteristic impedance.

[0068] The two optimal switching pulse widths were calculated using the optimal trajectory method. t 1- t 0 and t 2- t 1. As follows:

[0069]

[0070] In the formula v CrNC The voltage across the resonant capacitor at point C. v Cr per unit value, i LrNB The resonant inductor current at point B i Lr per unit value, w r This is the full-load angular frequency. Based on two optimal switching pulse widths... t 1- t 0 and t 2- t 1 drives the first switching transistor respectively S 1. Second switching transistor S2 and the third switching transistor S 3. Fourth switching transistor S 4. Enables rapid switching between empty and full modes;

[0071] S4: After the two switching pulses of the SOTC controller are executed, the dynamic response optimizer is enabled, and the system enters the transient recovery period. The state observer calculates the input resonant inductor current. i Lr Output normalized resonant capacitor voltage v CrN and resonant inductor current i LrN Constituting the radius of the state ρ s The dynamic response optimizer is based on the state radius. ρ s With the target radius ρ ref deviation Δ e Generate frequency compensation command u T Deviation Δ e for:

[0072]

[0073] In the formula ρ ref Let the target steady-state operating radius be the state radius. ρ s Depend on v CrN and i LrN Composition, that is , v CrN Resonant capacitor voltage v Cr per-unit value, i LrN Resonant current i Lr The per-unit value. The dynamic response optimizer is based on the deviation Δ e Generate frequency compensation command u T for:

[0074]

[0075] In the formula f ref The target steady-state resonant frequency, s G ∈{+1, 1} represents the monotonic direction symbol for the work area. k p>0 represents the proportional gain. Furthermore, if Δ e Meets the preset radius deviation tolerance ɛ Then the transient recovery period ends immediately, and the dynamic response optimizer exits, i.e.:

[0076]

[0077] In the formula ɛ Preset radius deviation tolerance.

[0078] Please refer to Figure 6 The control method of this invention is applied to a common DC bus direct-connected cascaded H-bridge (CHB) photovoltaic inverter system. This system consists of multiple power modules connected in parallel to a common DC bus. Each module's preceding stage is an isolated LLC resonant DC-DC converter, providing DC voltage to the subsequent H-bridge inverter unit. During system startup, k The multi-stage SPWM modulation strategy enables the LLC resonant converter to achieve fast soft-start independent of load conditions. k It is any positive integer greater than 1. During grid-connected operation, the cascaded H-bridge operates in power frequency square wave modulation mode. Its periodic switching causes the load of the LLC resonant converter to jump between no-load and full-load, forming a typical pulse load condition. This invention determines the switching state of the H-bridge and dynamically switches between various control modes of the LLC resonant converter, effectively suppressing output voltage fluctuations and switching stress under such extreme conditions, and ensuring stable and efficient system operation.

[0079] In summary, this invention discloses a switching control method based on SPWM soft start and pulse-type load LLC resonant converter. The start-up phase employs... k The system uses SPWM for pre-charging; during operation, it identifies the operating condition based on the H-bridge switching criterion and dynamically switches to Burst, linear PI, and SOTC two-pulse control. A dynamic response optimizer is introduced after SOTC to achieve high-frequency compensation, realizing the rapid dynamic response of the LLC resonant converter under startup and pulse load conditions. This solves the problems of slow dynamic response and large output voltage fluctuations in traditional linear control methods, significantly improving the stability and reliability of the system, reducing switching stress and electromagnetic interference, and expanding the application prospects of LLC resonant converters in high-end industrial power supplies, photovoltaic power generation, and other fields.

[0080] It should be noted that in this specification, relational terms such as first and second are used only to distinguish one entity or operation from another entity or operation, and do not necessarily require or imply any such actual relationship or order between these entities or operations.

[0081] While the specific embodiments of the present invention have been described above in conjunction with the accompanying drawings, this is not intended to limit the scope of protection of the present invention. The description of the embodiments disclosed in the present invention enables those skilled in the art to use or implement the present invention, and they can still modify the technical solutions described in the foregoing embodiments or make equivalent substitutions for some of the technical features. Such substitutions or modifications do not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of this application.

Claims

1. A resonant LLC converter based on SPWM soft start and pulse-type load, characterized in that, include: Main circuit unit and main control unit; The main circuit unit includes: a DC power supply. V in Resonant capacitor C r Resonant inductor L r ,transformer T Including magnetizing inductor L m Input capacitor C in Output capacitor C o First switching transistor S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S 4. Fifth switching transistor S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S 8. First diode D 1. Second diode D 2. Third diode D 3. Fourth diode D 4. Load R L ; The DC power supply V in The positive terminal and the input capacitor C in The first terminal, the first switching transistor S The drain of 1, the third switching transistor S 3. Drain connection; the DC power supply V in The negative terminal and the input capacitor C in The second terminal, the second switching transistor S The source of 2, the fourth switch transistor S 4 source connections; The resonant capacitor C r The first terminal and the first switch transistor S The source of 1, the second switch transistor S 2. Drain connection; the resonant capacitor C r The second end is connected to the resonant inductor L r The first end is connected; The transformer T The first end of the primary winding is connected to the resonant inductor L r The second end is connected; the transformer T The second end of the primary winding and the third switching transistor S The source of 3, the fourth switch transistor S 4 drain connection; The excitation inductor L m Parallel connection to transformer T Primary winding; The transformer T The first end of the secondary winding is connected to the first diode. D anode of 1, second diode D 2. Cathode connection; the transformer T The second end of the secondary winding is connected to the third diode. D The anode of 3, the fourth diode D 4. Cathode connection; The first diode D The cathode of 1 and the third diode D 3. Cathode, the output capacitor C o The first terminal, the fifth switching transistor S The collector of 5, the seventh switch transistor S 7. Collector connection; The second diode D The anode of 2 and the fourth diode D 4's anode, the output capacitor C o The second end, the sixth switch tube S The emitter of 6, the eighth switch transistor S 8 emitter connection; The load R L The first terminal and the fifth switch tube S The emitter of 5, the sixth switch transistor S 6. Collector connection; the load R L The second terminal is connected to the seventh switch. S The emitter of 7, the eighth switch transistor S 8 collector connection; The main control unit includes: an SPWM soft-start controller, a drive circuit, an H-bridge mode recognizer, a Burst controller, a PI controller, an SOTC controller, an adder SUM, a state observer, a dynamic response optimizer, and a voltage-controlled oscillator. The first input terminal of the drive circuit is connected to the output terminal of the SPWM soft-start controller, the second input terminal of the drive circuit is connected to the output terminal of the Burst controller, the third input terminal of the drive circuit is connected to the output terminal of the voltage-controlled oscillator, and the fourth input terminal of the drive circuit is connected to the first output terminal of the SOTC controller; the first output terminal of the drive circuit is connected to the first switching transistor. S The gate of 1 is connected, and the second output terminal of the driving circuit is connected to the second switching transistor. S The gate of 2 is connected, and the third output terminal of the driving circuit is connected to the third switching transistor. S The gate of 3 is connected, and the fourth output terminal of the driving circuit is connected to the fourth switching transistor. S 4. Gate connection; the first input terminal of the H-bridge mode recognizer is connected to the fifth switch. S The gate connection of 5, the second input terminal of the H-bridge mode recognizer is connected to the sixth switch. S The gate connection of 6, the third input terminal of the H-bridge mode recognizer is connected to the seventh switch. S The gate connection of 7, the fourth input terminal of the H-bridge mode recognizer is connected to the eighth switch. S The gate of the 8 is connected; the first output of the H-bridge mode recognizer is connected to the input of the Burst controller, the second output of the H-bridge mode recognizer is connected to the negative input of the adder SUM, and the third output of the H-bridge mode recognizer is connected to the input of the SOTC controller; the input of the PI controller is connected to the output of the adder SUM; the output of the PI controller is connected to the first input of the voltage-controlled oscillator; the first input of the dynamic response optimizer is connected to the output of the state observer, the second input of the dynamic response optimizer is connected to the second output of the SOTC controller, and the output of the dynamic response optimizer is connected to the second input of the voltage-controlled oscillator; The H-bridge mode identifier outputs a mode indication signal model to accurately determine the load state of the LLC resonant converter. The mode indication signal mode is used to indicate three operating modes, where mode=0 indicates no load, mode=1 indicates full load, and mode=2 indicates no-full-load switching.

2. The LLC resonant converter based on SPWM soft start and pulse-type load according to claim 1, characterized in that, The first switching transistor S 1. Third switching transistor S 3 as a group, the second switching transistor S 2. Fourth switching transistor S 4 is another group, with the two groups of switches conducting alternately, the fifth switch... S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S The 8 components form an H-bridge inverter circuit. The H-bridge uses low-frequency square wave modulation to provide a pulsed load for the LLC resonant converter.

3. A switching control method based on SPWM soft start and pulse-type load LLC resonant converter, characterized in that, The specific control method for the LLC resonant converter according to claim 1 is as follows: S1: The SPWM soft-start controller generates according to the preset switching point. k The SPWM modulation signal is passed through the driver circuit to the first switch transistor. S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S The gate of 4 is driven to drive the output capacitor. C o Pre-charge to achieve output voltage V o The process proceeds smoothly, completing the soft start; S2: The fifth switching transistor S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S 8. Inverter conduction begins; the H-bridge mode recognizer detects the fifth switch. S 5. Sixth switching transistor S 6. Seventh switching transistor S 7. Eighth switching transistor S The gate drive signal of 8 is used for mode recognition to determine the load. R L The equivalent state on the LLC resonant converter side is no-load, full-load, or no-full-load switching state, and outputs a mode indicator signal model. The mode indicator signal mode is used to indicate three operating modes, where mode=0 indicates no-load, mode=1 indicates full-load, and mode=2 indicates no-full-load switching. S3: The H-bridge mode recognizer sends an enable command to the Burst controller, PI controller or SOTC controller according to the mode indication signal model; If model=0, the Burst controller is enabled, and it switches between burst-on and burst-off states according to the voltage hysteresis control principle; in the burst-on state, the first switching transistor is driven. S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S 4. To maintain the output voltage, the output voltage V o Rise; in the burst-off state, turn off the first switch. S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S 4. Cut off the energy transfer from the primary to the secondary side of the transformer, and the output voltage... V o decline; If model=1, the PI controller is enabled, and the output voltage... V o With reference voltage V ref The adder SUM compares the output error voltage. V e As the input signal to the PI controller, the error voltage V e The PI controller outputs an adjustment signal, which is then passed through the voltage-controlled oscillator to the first switching transistor. S 1. Second switching transistor S 2. Third switching transistor S 3. Fourth switching transistor S The drive signal of 4 is used by the LLC resonant converter to perform frequency conversion regulation by pulse frequency modulation. If model=2, the SOTC controller is enabled, and two optimal switching pulse widths are calculated based on simplified optimal trajectory control theory, which then drive the first switching transistor. S 1. Third switching transistor S 3 and the second switching transistor S 2. Fourth switching transistor S 4. Enables rapid switching between empty and full modes; S4: After the two switching pulses of the SOTC controller have been executed, the dynamic response optimizer is enabled, and the state observer calculates the input resonant inductor current. i Lr Output normalized resonant capacitor voltage v CrN and resonant inductor current i LrN Constituting the radius of the state ρ s The dynamic response optimizer is based on the state radius. ρ s With the target radius ρ ref deviation Δ e Generate frequency compensation command u T To suppress secondary voltage oscillations that may occur after SOTC switching; further, if the radius deviation Δ e If the value is within the preset range, the dynamic response optimizer will exit and the PI controller will be entered directly.

4. The soft-start method for an LLC resonant converter based on SPWM soft-start and pulse-type load according to claim 3, characterized in that, The SPWM soft-start process is an open-loop control, and its startup logic is independent of the load state. It does not rely on feedback from the load current or output voltage for adjustment, and only executes at a preset switching point after the system is powered on. k The SPWM drive signal output is further optimized by setting the modulation index and the number of switching time points. k Make the output voltage V o To achieve a theoretically smooth ascent, k It is any positive integer greater than 1.

5. The soft-start method for an LLC resonant converter based on SPWM soft-start and pulse-type load according to claim 4, characterized in that, S1's k One switching point resonant cavity V AB ( t The voltage of ) satisfies the following equation: In the formula m k For the first k SPWM modulation level w m The carrier angular frequency is equal to the steady-state switching angular frequency of the converter, where the carrier frequency of the SPWM is equal to the steady-state switching frequency of the LLC resonant converter.

6. The switching control method based on SPWM soft start and pulse-type load LLC resonant converter according to claim 3, characterized in that, The load state of the LLC resonant converter is accurately determined by the output mode indication signal "model" of the H-bridge mode identifier, and the fifth switch... S 5. Seventh switching transistor S 7 or the sixth switch tube S 6. Eighth switching transistor S 8 is on, indicating no-load mode = 0; the fifth switch transistor S 5. Eighth switching transistor S 8 or the sixth switch tube S 6. Seventh switching transistor S When 7 is on, it indicates full load (model=1); when the mode indicator signal changes from 0 to 1, it indicates empty-full switching (model=2).

7. The switching control method based on SPWM soft start and pulse-type load LLC resonant converter according to claim 3, characterized in that, S3 has two optimal switching pulse widths. t 1- t 0 and t 2- t 1 is: In the formula v CrNC The voltage of the resonant capacitor at point C v Cr per-unit value, i LrNB The resonant inductor current at point B i Lr per-unit value, w r The full-load angular frequency, ρ For the resonant inductor current i Lr The per-unit value of the peak value, i.e. , Z r It is a binary characteristic impedance, i.e. , I rms The resonant inductor current under full load i Lr Valid values, i.e.: In the formula T The period of resonance, n This refers to the number of transformer turns. i o For load R L Electric current.

8. The switching control method based on SPWM soft start and pulse-type load LLC resonant converter according to claim 3, characterized in that, The state radius of S4 ρ s With the target radius ρ ref deviation Δ e for: In the formula ρ ref Let the target steady-state radius be the state radius. ρ s Depend on v CrN and i LrN Composition, that is , v CrN Resonant capacitor voltage v Cr per-unit value, i LrN For the resonant inductor current i Lr The per-unit value; The dynamic response optimizer is based on the deviation Δ e Generate frequency compensation command u T for: In the formula f ref The target steady-state resonant frequency, s G ∈{+1, 1} represents the monotonic direction symbol for the work area. k p >0 represents the proportional gain; if Δ e Meets the preset radius deviation tolerance ɛ Then exit the dynamic response optimizer, that is: In the formula ɛ This is the preset radius deviation tolerance.

Images