Reverse control method of LLC resonant converter, converter and electronic device

Abstract

The application relates to a conversion technology between direct currents and direct currents, and discloses a reverse control method of an LLC resonant converter, the converter and electronic equipment. The method comprises the following steps: S1, setting the conduction time Ton of a main switch tube in a secondary side switching network as a constant conduction time; S2, collecting an input voltage V2 of the secondary side switching network and an output voltage V1 of a primary side switching network, and calculating the gain G of the input voltage V2 and the output voltage V1; and S3, adjusting the off time Toff of the main switch tube in the secondary side switching network according to the gain G, so that the off time Toff monotonously changes with the change of the gain G. The application does not need to additionally increase a regulation resonant element, can retain the original LLC topology low cost and small size advantages, improves the problem that a traditional reverse mode is easy to enter a discontinuous mode, reduces the complexity of loop control, and is suitable for multi-working condition requirements.

Description

Technical Field

[0001] This application relates to the field of DC-DC conversion technology, and in particular to an inverse control method for an LLC resonant converter, the converter, and electronic equipment. Background Technology

[0002] With the explosive growth in demand from the power electronics industry in recent years, especially in the new energy field, such as photovoltaics, charging piles, energy storage, electric vehicle (EV) on-board chargers (OBC), and on-board DC-DC converters, the demand for high-performance, high-efficiency, high-power-density, and low-cost bidirectional DC-DC converters has become increasingly urgent.

[0003] Taking a bidirectional OBC application as an example, a bidirectional OBC simultaneously meets the charging and discharging requirements of the power battery. In charging mode, the bidirectional OBC rectifies the AC mains input voltage through a pre-stage PFC circuit to obtain a primary-side intermediate DC voltage. The subsequent bidirectional DC-DC converter transforms this primary-side intermediate DC voltage into the voltage and current required by the EV power battery, achieving electrical isolation between the two. In discharging mode, the bidirectional DC-DC converter transforms the power battery voltage into the required intermediate DC voltage. This intermediate DC voltage is then inverted and converted by the PFC circuit to output AC power.

[0004] With the development of power electronics technology, there is a growing demand for higher power density and higher efficiency in DC-DC converters. Common topologies for isolated DC-DC converters include LLC circuits, phase-shifted full-bridge circuits, hard-switching full-bridge circuits, and dual active bridge (DAB) circuits, among others. LLC circuits, in particular, are widely used in high-efficiency, high-frequency DC-DC converters. They are well-suited for applications requiring high efficiency and high power density.

[0005] When an LLC circuit is directly applied to a bidirectional DC-DC converter, its forward characteristics are those of a conventional LLC circuit. However, in reverse operation, it functions as a series resonant converter (SRC) with the resonant cavity on the output side. Its characteristics differ from those of an LLC; the maximum output gain is 1, while the output gain of an LLC circuit can exceed 1 (i.e., at a unity turns ratio, the output voltage can be higher than the input voltage). This characteristic, through proper design, can theoretically be applied to bidirectional DC-DC converters with a wide voltage range. However, in actual reverse operation, due to limitations imposed by the LLC parameters (usually meeting the requirements for forward operation first), the output voltage gain characteristics of an LLC in SRC mode cannot meet the needs of practical applications. Within the same switching frequency range as the LLC mode, the voltage gain characteristics of the SRC mode typically exhibit the following problems:

[0006] 1) The voltage gain characteristic range is relatively narrow, and the gain does not meet the requirements of battery voltage applications;

[0007] 2) The voltage gain characteristic range is relatively narrow, and the switching frequency changes very widely during gain adjustment. Due to the limitation of the device boundary operating frequency, it is easy to reach the boundary frequency, which causes the converter to enter the burst mode. The output current is discontinuous, the output voltage and current characteristics are poor, and it also has an adverse effect on the life of the output capacitor.

[0008] 3) The voltage gain characteristics are nonlinear under different load conditions. The gain changes nonlinearly and monotonically near the boundary frequency, which further degrades its output characteristics. Moreover, the control loop is not easy to stabilize, making the design very difficult.

[0009] Because LLC circuits have the above characteristics when operating in reverse state (SRC mode), their output voltage characteristics are poor in practical applications. They are prone to entering discontinuous mode under high voltage input or non-heavy load conditions. The switching transistors and transformers operate intermittently at the highest frequency, resulting in low overall efficiency, severe heat generation of components, and the generation of input inrush current, which greatly limits their practical applications.

[0010] To meet the reverse application requirements of DC-DC converters with actual battery input, a common approach is to modify the LLC topology by adding components to improve voltage gain in reverse mode. Topologies such as CLLC and CLLLC improve this by adding a resonant capacitor (C) or resonant capacitor and inductor (C, L) on the output side of the LLC, resulting in a better voltage gain curve and higher efficiency compared to LLC in reverse mode. However, like CLLC, CLLLC increases the number of components, thus increasing cost and size. Furthermore, adding resonant components complicates the gain curve, increasing the design difficulty of the control loop.

[0011] Therefore, there is an urgent need for a technical solution to optimize the reverse working performance of LLC. Summary of the Invention

[0012] To address the aforementioned technical problems, this application provides an inverse control method for an LLC resonant converter, the converter itself, and an electronic device.

[0013] Firstly, this application provides a reverse control method for an LLC resonant converter, which employs the following technical solution:

[0014] A reverse control method for an LLC resonant converter, the LLC resonant converter including a primary-side switching network, a secondary-side switching network, and a resonant network disposed between the primary-side and secondary-side switching networks, the method includes the following steps: S1, setting the on-time Ton of the main switch in the secondary-side switching network to a constant on-time; S2, acquiring the input voltage V2 of the secondary-side switching network and the output voltage V1 of the primary-side switching network, and calculating the gain G of the input voltage V2 and the output voltage V1; S3, adjusting the off-time Toff of the main switch in the secondary-side switching network according to the gain G, so that the off-time Toff changes monotonically with the change of the gain G.

[0015] By adopting the above technical solution, the on-time of the main switch is fixed as a constant on-time, which can achieve zero-current turn-on of the main switch, reduce switching losses, and meet the power supply requirements of the load. By dynamically adjusting the off-time of the main switch to adapt to the changes in the secondary input voltage, the switching cycle is monotonically related to the voltage gain, preventing switching frequency fluctuations and improving the problem of traditional reverse mode easily entering discontinuous waveform mode. No additional resonant components are needed, which improves output stability and conversion efficiency while maintaining the low cost and small size advantages of the original LLC topology.

[0016] Optionally, in step S3, making the turn-off time Toff change monotonically with the gain G includes: when the gain G increases, decreasing the turn-off time Toff to make the output voltage V1 return to the preset target; when the gain G decreases, increasing the turn-off time Toff to make the output voltage V1 return to the preset target; wherein, the turn-off time Toff increases monotonically with the secondary side input voltage V2 to maintain the primary side output voltage V1.

[0017] By adopting the above technical solution, the switching cycle is adjusted according to the changes in the gain G of the input voltage V2 and the output voltage V1, so as to achieve stable control of the output voltage V1; the turn-off time is monotonically correlated with the secondary input voltage V2, ensuring that the switching frequency is in a relatively low frequency range when high voltage is input, thus improving the problem of excessive switching loss when high voltage is input in traditional solutions and further improving the efficiency of reverse working mode.

[0018] Optionally, the LLC resonant converter also includes a resonant network disposed between the primary-side switching network and the secondary-side switching network. The resonant network includes a resonant inductor Lr and a resonant capacitor Cr, with a resonant period of... Set the on-time Ton of the main switch in the secondary switch network to be equal to 1 / 2Tr.

[0019] By adopting the above technical solution, the conduction time of the fixed switch is half of the resonant period of the resonant network. Based on the parameters of the resonant inductor and the resonant capacitor, the conduction time can be obtained to be equal to 1 / 2Tr, which can further achieve zero current turn-off and further reduce switching losses.

[0020] Optionally, the turn-off time Toff includes a first dead time t1, a symmetrical turn-on time t2, and a second dead time t3, which are set continuously. The symmetrical turn-on time t2 is the conduction duration of the complementary switch in the secondary-side switching network. The symmetrical turn-on time t2 is equal to the conduction time Ton of the main switch. The complementary switch and the main switch conduct alternately. The turn-off time Toff is adjusted by adjusting the second dead time t3.

[0021] By adopting the above technical solution, the turn-off time is decomposed into multiple time periods, and the symmetrical conduction time is matched with the conduction time of the main switch to ensure efficient energy transmission of the resonant circuit. The turn-off time can be dynamically adjusted by adjusting the second dead time, which is flexible. At the same time, the first dead time can prevent the main switch and the complementary switch from being directly connected, thus ensuring circuit safety.

[0022] Optionally, when the load and output voltage V1 of the primary-side switching network are fixed, the peak value of the current Id of the main switch and the complementary switch in the secondary-side switching network increases monotonically with the increase of the secondary-side input voltage.

[0023] By adopting the above technical solution, the peak current is monotonically correlated with the secondary input voltage, ensuring that the circuit can work stably under different input voltages, further optimizing the output characteristics and improving the stability of the control loop.

[0024] Optionally, both the main switch and the complementary switch can be MOSFET, IGBT, SiC MOSFET, or GaNHEMT.

[0025] By adopting the above technical solutions, multiple power switching transistors can be selected. MOSFETs have fast switching speed and low on-resistance, IGBTs have large current carrying capacity, and SiC MOSFETs have high temperature resistance and low loss. They can be flexibly selected according to different power levels and operating conditions, enhancing the adaptability and practicality of the solution.

[0026] Optionally, the LLC resonant converter includes a full-bridge LLC resonant converter, a half-bridge LLC resonant converter, or a multiphase interleaved LLC resonant converter.

[0027] By adopting the above technical solutions, the control method can be adapted to various LLC topologies. Whether it is a basic topology of full bridge or half bridge, or an extended topology with multiple interleaved phases such as three-phase and four-phase, it can improve the performance of the reverse working mode, broaden the application range of the control method, and enhance the versatility of the technical solution.

[0028] Optionally, it also includes a soft-start step S0, controlling the secondary-side switching network to start at a preset minimum switching frequency, and gradually increasing the switching frequency according to a preset frequency change rate until the switching frequency reaches the initial operating frequency required to enter steady-state control, thereby suppressing the surge current during the startup phase through a smooth frequency transition.

[0029] By adopting the above technical solution, a soft-start stage is added to prevent surge current impact caused by frequency changes during the startup phase, protect devices such as switching transistors and transformers, extend the service life of the circuit, and ensure a smooth startup process, thereby further improving system reliability.

[0030] Secondly, the LLC resonant converter provided in this application adopts the following technical solution:

[0031] An LLC resonant converter includes a primary-side switching network, a secondary-side switching network, a resonant network, a transformer, and a control unit. The transformer includes a primary winding and a secondary winding. The primary winding is connected to the primary-side switching network, which is used to connect to a first power supply side or a first load side. The secondary winding is connected to the secondary-side switching network, which is used to connect to a second power supply side or a second load side. The resonant network is connected in series between the primary-side switching network and the primary winding of the transformer, or in series between the secondary-side switching network and the secondary winding of the transformer. The resonant network includes a resonant inductor Lr and a resonant capacitor Cr, which are used to form an LLC resonant circuit. The signal input terminal of the control unit is connected to the voltage output terminal of the primary-side switching network and the voltage input terminal of the secondary-side switching network, for acquiring the primary-side output voltage and the secondary-side input voltage. The signal output terminal of the control unit is connected to the control terminal of the secondary-side switching network, for outputting a turn-on / turn-off control signal. The control unit is configured to execute the reverse control method as described in any of the first aspects above.

[0032] By adopting the above technical solution, the converter relies on the control unit to execute the reverse control method, without modifying the original LLC topology, and can achieve wide gain adjustment, stable output and high-efficiency conversion in reverse working mode; the primary and secondary switching networks are adapted to bidirectional power transmission requirements, the resonant network ensures stable resonant characteristics, the overall structure is simple and cost controllable, and it is suitable for various new energy application scenarios such as photovoltaics, energy storage, and electric vehicles.

[0033] Thirdly, the electronic device provided in this application adopts the following technical solution:

[0034] An electronic device comprising an LLC resonant converter as described in the second aspect above.

[0035] By adopting the above technical solutions, electronic devices integrate LLC resonant converters, which have efficient bidirectional power conversion capabilities. In reverse working mode, the output is stable and the loss is low, which can meet the application requirements of bidirectional switching of charging and discharging in the new energy field and improve the overall performance and reliability of electronic devices.

[0036] In summary, this application includes at least one of the following beneficial technical effects:

[0037] 1. By adopting the above technical solution, the on-time of the main switch is fixed as a constant on-time, which can achieve zero-current turn-on of the main switch and reduce switching losses; by dynamically adjusting the off-time to adapt to the changes in the secondary input voltage, the switching cycle is monotonically correlated, preventing switching frequency fluctuations and improving the problem of traditional reverse mode easily entering discontinuous waveform mode; no additional resonant components are needed, while maintaining the advantages of low cost and small size of the original LLC topology, the output stability and conversion efficiency are improved;

[0038] 2. By adopting the above technical solution, the switching cycle is adjusted according to the change of the gain G of the input voltage V2 and the output voltage V1, so as to achieve stable control of the output voltage V1; the turn-off time is monotonically correlated with the secondary input voltage V2, ensuring that the switching frequency is in the low frequency range when the high voltage input is used, thereby improving the problem of excessive switching loss when the high voltage input is used in the traditional solution and further improving the efficiency of the reverse working mode.

[0039] 3. By adopting the above technical solution, the turn-off time is decomposed into multiple time periods, and the symmetrical turn-on time is matched with the turn-on time of the main switch to ensure efficient energy transmission of the resonant circuit; the turn-off time can be dynamically adjusted by adjusting the second dead time, and the adjustment method is flexible. At the same time, the first dead time can prevent the main switch and the complementary switch from being directly connected, ensuring circuit safety. Attached Figure Description

[0040] Figure 1 This is a structural diagram of the LLC resonant converter provided in the embodiments of this application;

[0041] Figure 2 This is a flowchart of the inverse control method for an LLC resonant converter provided in the embodiments of this application;

[0042] Figure 3 This is the timing logic diagram of the LLC resonant converter in reverse operating mode provided in the embodiments of this application;

[0043] Figure 4 This is a structural diagram of another LLC resonant converter provided in the embodiments of this application.

[0044] Explanation of reference numerals in the attached figures:

[0045] 1. Primary-side switching network; 2. Resonant network; 3. Transformer; 4. Secondary-side switching network. Detailed Implementation

[0046] The following is in conjunction with the appendix Figure 1-4 This application will be described in further detail.

[0047] Reference Figure 1 This application discloses an LLC resonant converter. The LLC resonant converter includes a primary-side switching network 1, a secondary-side switching network 4, a resonant network 2, a transformer 3, and a control unit. The transformer 3 (T1) includes a primary winding and a secondary winding. The primary winding is connected to the primary-side switching network 1, which is used to connect to a first power supply side (V1) or a first load side (not shown in the figure). The secondary winding is connected to the secondary-side switching network 4, which is used to connect to a second load side (not shown in the figure) or a second power supply side (V2). The resonant network 2 is connected in series between the primary-side switching network 1 and the transformer. The primary windings of transformer 3 are connected in series, either between the primary windings of the primary winding and the secondary windings of the secondary winding, or between the secondary windings of the secondary winding and the secondary winding of transformer 3. The resonant network 2 includes a resonant inductor Lr and a resonant capacitor Cr, used to form an LLC resonant circuit. The primary winding switching network 1 includes switching transistors Q1 to Q4, operating in forward mode, where energy is transferred from the primary side to the secondary side (V1 to V2). Q1 and Q4 are synchronously turned on / off, and Q2 and Q3 are synchronously turned on / off. The two sets of switching transistors are alternately turned on, inverting the primary voltage V1 into an AC voltage, which is then transmitted through the resonant network 2 (Lr, Cr) to the transformer. Energy is transferred from the secondary side to the primary side (V2 to V1) in reverse operation mode. Q1 to Q4 remain off, acting only as rectifiers on the primary side. Passive rectification is achieved using a body diode connected in parallel with the switches, receiving the energy (voltage V2) transferred from the secondary side and converting it into the primary side output voltage V1. The secondary-side switching network 4 includes switches Q5 to Q8, where Q5 and Q8 are the main switches, and Q6 and Q7 are complementary switches. In forward operation mode, Q5 to Q8 remain off, and energy is transferred only through the body diode connected in parallel. Passive rectification is achieved, converting the AC voltage on the secondary side of transformer 3 into the DC voltage V2 on the secondary side. In reverse operation mode, the main switches Q5 and Q8 are synchronously turned on / off, while the complementary switches Q6 and Q7 are turned on alternately with the main switches (Q6 / Q7 is off when Q5 / Q8 is on, and Q6 / Q7 is on when Q5 / Q8 is off). By fixing the on-time Ton of Q5 / Q8 and dynamically adjusting its off-time Toff, controllable energy transfer from the secondary side V2 to the primary side V1 is achieved, while the alternating on-time of Q6 / Q7 completes symmetrical conduction.

[0048] The control unit (not shown in the figure) is connected to the primary-side switching network 1 and the secondary-side switching network 4. Specifically, it may include a sampling circuit and a driving circuit. The sampling circuit is connected to the voltage output terminal (V1) of the primary-side switching network 1 and the voltage input terminal (V2) of the secondary-side switching network 4. In reverse working mode, it is used to collect the output voltage (V1) of the primary side and the input voltage (V2) of the secondary side. The driving circuit is connected to the control terminal (control terminal of switching transistors Q5 to Q8) of the secondary-side switching network 4 and is used to output the turn-on / turn-off control signal.

[0049] The implementation principle of an LLC resonant converter according to an embodiment of this application is as follows: In the forward operating mode, the switches Q1-Q4 in the primary-side switching network 1 are alternately turned on, inverting the output voltage V1 on the primary side into an AC voltage, which is then transmitted to the secondary side via the resonant network 2 and transformer 3. The switches Q5-Q8 in the secondary-side switching network 4 output a DC voltage V2 through passive rectification by the body diode. In the reverse operating mode, Q1-Q4 in the primary-side switching network 1 are turned off and passively rectified by the body diode. The main switches Q5 / Q8 and the complementary switches Q6 / Q7 in the secondary-side switching network 4 are alternately turned on, controlling the output voltage V2. The control unit sets the on-time Ton of the main switch in the secondary-side switching network 4 to a constant on-time. Then, it collects the input voltage V2 and the primary-side output voltage V1 of the secondary-side switching network 4 through the sampling circuit, calculates the gain G of the input voltage V2 and the output voltage V1, and adjusts the off-time Toff of the main switch in the secondary-side switching network 4 according to the change of gain G to change the switching period T, so as to realize the controllable transfer of energy from the secondary side to the primary side (V2 to V1). At the same time, the alternating conduction of Q6 / Q7 completes symmetrical conduction, reduces freewheeling loss, and finally completes the stable energy conversion in reverse mode.

[0050] It is understood that the LLC resonant converter of this application relies on the control unit to execute the reverse control method, and can achieve wide gain adjustment, stable output and high efficiency conversion in reverse working mode without changing the original LLC topology. The primary-side switch network 1 and the secondary-side switch network 4 are adapted to bidirectional power transmission requirements, and the resonant network 2 ensures stable resonant characteristics. The overall structure is simple and the cost is controllable, making it suitable for various new energy application scenarios such as photovoltaics, energy storage and electric vehicles.

[0051] Reference Figure 2 The reverse control method for the LLC resonant converter provided in this application includes the following steps: S0, soft start step; S1, setting the on-time Ton of the main switch in the secondary-side switching network 4 to a constant on-time; S2, acquiring the input voltage V2 of the secondary-side switching network 4 and the output voltage V1 of the primary-side switching network 1, and calculating the gain G of the input voltage V2 and the output voltage V1; S3, adjusting the off-time Toff of the main switch in the secondary-side switching network 4 according to the gain G, so that the off-time Toff changes monotonically with the change of the gain G.

[0052] In one embodiment, step S0 specifically involves controlling the secondary-side switching network 4 to start at a preset minimum switching frequency, and gradually increasing the switching frequency according to a preset frequency change rate until the switching frequency reaches the initial operating frequency required to enter steady-state control. The surge current during the startup phase is suppressed through a smooth frequency transition. The preset minimum switching frequency is the lower limit frequency at which the converter is allowed to operate, and the minimum switching frequency f is lower than the resonant frequency fr of the resonant network 2. In the reverse operating mode, the gain G = V1 / V2 (V1 is the primary-side output voltage, and V2 is the secondary-side input voltage), and the gain G is positively correlated with the switching frequency f. The lower the switching frequency, the smaller the gain G, and the lower the corresponding primary-side output voltage V1. Therefore, starting at the minimum switching frequency can keep the initial value of V1 at a low level and avoid V1 overshoot at startup. The preset frequency change rate characterizes the rate at which the frequency rises, i.e., the magnitude of frequency increase per unit time. Its value can be set according to the power rating and resonant parameters of the converter. Converters with higher power usually correspond to a smaller frequency change rate to ensure a continuous and smooth frequency rise, allowing the gain G to increase synchronously and smoothly, and V1 to rise smoothly to the preset target value, preventing V1 fluctuations and current surges caused by frequency abrupt changes. The initial operating frequency required to enter steady-state control is usually set near the resonant frequency fr of resonant network 2, and can be pre-calibrated based on the converter's rated power, input voltage range, and other operating conditions. For example, in a bidirectional LLC resonant converter with a rated power of 10kW, the resonant frequency fr of the resonant network 2 is 50kHz. If the preset minimum switching frequency is 30kHz and the frequency change rate is 2kHz / ms, then after the converter starts in reverse, the switching frequency smoothly increases from 30kHz at a rate of 2kHz per millisecond. After 10ms, the frequency increases to 50kHz. At this time, the gain G synchronously rises to the steady-state value, V1 reaches the preset target voltage, the conduction time of the main switch in the secondary switching network 4 matches the preset constant conduction time Ton, the soft start step ends, and the converter switches to the steady-state control process (fixed Ton, adjusted Toff).

[0053] Through the above-mentioned frequency smooth transition control method, the resonant current can slowly increase to a steady-state value, preventing the surge current impact caused by the sudden energy change of the resonant network 2 at the moment of startup, thereby protecting the switching transistor, transformer 3 and other devices, extending the service life of the converter, and improving the reliability of the system during the startup phase.

[0054] Reference Figure 3 In one embodiment, in step S1, the on-time Ton of the main switch in the secondary-side switch network 4 is set to a constant on-time.

[0055] Specifically, the on-time Ton is a fixed value preset and stored in the control unit. The setting of Ton determines the duration of the "smallest unit" or "single pulse energy packet" of energy transmission by the converter. During this duration, the secondary-side switch is turned on, loading the secondary-side input voltage V2 onto the resonant network, causing the resonant current to change sinusoidally and transfer energy to the primary side, completing a quantitative energy transfer. At this time, the control unit no longer controls the power by changing the on-time, but uses Ton as a fixed time reference to ensure that the current can change and be transmitted according to the expected trajectory in each switching action, adapting to different input voltages to meet the power supply requirements of the load, while avoiding resonant frequency shifts caused by Ton fluctuations, thus improving the operating stability of the converter.

[0056] In one embodiment, the value of the fixed on-time Ton can be determined comprehensively based on the hardware parameters of the bidirectional DC-DC converter and the expected performance indicators. For example, it can be determined by considering factors such as the maximum operating frequency, resonant cavity parameter matching, and peak current limit. The maximum operating frequency limit is: in reverse operating mode, the switching period T = 2Ton + Toff (assuming symmetrical operation of the half-bridge / full-bridge, or a single operation cycle). When Toff approaches 0, the converter reaches the maximum allowable switching frequency fmax, which is approximately equal to 1 / (2Ton). Therefore, the setting of Ton first determines the upper limit of the system's maximum operating frequency, which should be lower than the physical frequency limit of the switching devices (such as MOSFETs or IGBTs) and the response limit of the drive circuit. The resonant cavity parameter matching is as follows: the value of Ton is usually referenced to the resonant period Tr of the LC resonant cavity. The value of Ton is usually set near the resonant half-cycle. If Ton is too small, the energy transferred in a single pulse will be insufficient, failing to meet the full-load power requirement; if Ton is too large, it may lead to excessive resonant current or saturation of magnetic components. The peak current limit is: a fixed Ton determines the peak value of the resonant inductor current rise in each switching cycle. The setting of Ton must ensure that the peak current flowing through the switching transistor does not exceed the device's safe operating area (SOA) under various input voltage conditions. In practice, a suitable fixed time can be selected as Ton through simulation or experimentation, provided that the maximum output power is guaranteed and the switching transistor does not overheat. Once selected, Ton remains unchanged in the subsequent voltage regulation control process (step S3), and only the off-time Toff is adjusted to adapt to changes in voltage gain.

[0057] Reference Figure 3 In one embodiment, in step S2, the input voltage V2 of the secondary-side switching network 4 and the output voltage V1 of the primary-side switching network 1 are collected, and the gain G of the input voltage V2 and the output voltage V1 is calculated.

[0058] Specifically, since this embodiment describes a buck operating mode where energy flows from the secondary side to the primary side (i.e., the secondary side acts as a high-voltage input source, and the primary side acts as a low-voltage output source), the secondary side voltage V2 is defined as the input voltage (e.g., the high-voltage battery voltage), and the primary side voltage V1 is defined as the output voltage (e.g., the low-voltage bus voltage). The control unit samples these two voltage values ​​in real time and calculates the required voltage gain under the current operating condition according to the formula G=V1 / V2 (i.e., the ratio of the output voltage to the input voltage). This gain G reflects the current voltage difference between the input and output and is the basis for the system's feedforward control. Based on this, the control unit can further combine the error feedback signal between the output voltage V1 and the preset target voltage (Vt) to dynamically adjust the subsequent turn-off time Toff to achieve accurate closed-loop control.

[0059] Reference Figure 3 In one embodiment, step S3, making the turn-off time Toff monotonically change with the gain G, includes:

[0060] When the gain G increases, the off-time Toff decreases, causing the output voltage V1 to return to the preset target.

[0061] When the gain G decreases, the off-time Toff is increased so that the output voltage V1 returns to the preset target.

[0062] The turn-off time Toff increases monotonically as the secondary side input voltage V2 increases, in order to maintain the primary side output voltage V1.

[0063] Specifically, since Ton is fixed, the energy transferred within a single switching cycle is mainly determined by the input voltage V2. When the secondary-side input voltage V2 increases (i.e., under high-voltage conditions), the energy injected into the resonant cavity at the input terminal will increase within the same Ton time. To prevent the output voltage V1 from overshooting and maintain stability, the system must reduce the frequency of energy transfer. Therefore, the control unit monotonically increases the turn-off time Toff (i.e., extends the non-conduction waiting time of the switching transistor), reducing the number of energy pulses transferred per unit time, thereby offsetting the excess energy caused by the increase in input voltage V2.

[0064] Conversely, when the load on the primary side becomes heavier (i.e., the output current demand increases) or the input voltage on the secondary side decreases, the control unit reduces the off-time Toff, thereby increasing the switching frequency to deliver more energy pulses per unit time to maintain the balance of the output voltage V1. This control method utilizes the wide-range adjustment capability of the off-time Toff (theoretically adjustable from 0 to infinity) to solve the problem that traditional SRC / LLC converters cannot effectively reduce gain (i.e., insufficient gain adjustment range) under high-voltage input on the secondary side or light-load conditions on the primary side, and ensures that the inductor current has enough time to naturally return to zero before each turn-on, achieving soft switching.

[0065] Based on the above control logic, this application can achieve a monotonically related (linear or quasi-linear) relationship between the switching period T and the input voltage V2, and avoid the traditional "intermittent waveform" problem. The specific principle analysis is as follows:

[0066] Reference Figure 3 The figure illustrates the timing logic and voltage-current relationship of the control method of this application in reverse operating mode. In the figure, Vgs is the gate control voltage of the switch, Id is the switch current, and T is the switching period.

[0067] During voltage regulation, assuming the primary side output voltage V1 needs to remain stable (i.e., the target voltage Vt remains unchanged), when the secondary side input voltage V2 changes, the control unit will adjust the response through step S3:

[0068] As the secondary side input voltage V2 gradually increases from low to high, the control unit automatically increases the off-time Toff in order to maintain V1 stability. Since Ton remains constant (e.g., fixed at Tr / 2 or other constants), the switching period T (T=Ton+Toff) will monotonically increase with the increase of input voltage V2. In other words, macroscopically, the switching frequency f smoothly decreases as the input voltage V2 increases.

[0069] It should be noted that traditional LLC reverse operation mode typically requires a higher operating frequency to adjust the gain under high voltage input, leading to increased switching losses. In contrast, this application adjusts the gain by increasing Toff, so that the switching frequency is actually in the low-frequency range under high voltage input (because Toff is longer). Combined with zero-turn-on characteristics, this improves the switching loss problem under high voltage input. Furthermore, when traditional bidirectional LLC converters operate in reverse, limited by their gain characteristics, they often cannot further change the gain under light load or high voltage input, forcing them into "Burst Mode" (or hiccup mode), i.e., maintaining the voltage through intermittent "burst-stop-burst," which causes discontinuous fluctuations in input and output current. In contrast, this application, by adjusting Toff, can theoretically extend the turn-off time indefinitely. Even under extremely light load or extremely high input voltage, the control unit only needs to make Toff very long, putting the circuit into an "ultra-low frequency continuous operation state," without stopping the burst generation. This method maintains the discontinuous resonant current (DCM, which is beneficial for soft switching) on ​​a microscopic level, but avoids the intermittent shutdown of the control unit on a macroscopic level, thus achieving smooth monotonic regulation and obtaining voltage regulation effect.

[0070] Understandably, the main switches Q5 / Q8 and complementary switches Q6 / Q7 in the secondary-side switching network 4 use an alternating conduction control sequence. To prevent bridge arm shoot-through faults, the conduction sequences of the two sets of switches are symmetrically constrained. The minimum switching cycle of a complete main switch conduction-complementary switch conduction is 2Ton (main switch conduction Ton, complementary switch conduction Ton, ignoring dead time). According to the frequency-cycle conversion formula f=1 / T, when the switching cycle takes the minimum value of 2Ton, the switching frequency reaches its maximum value, i.e., the maximum operating frequency fmax=1 / (2Ton). In this mode, the control unit locks the growth duration of the resonant network 2 current by fixing the Ton, so that the resonant network 2 current changes sinusoidally during the main switch conduction phase. Combined with the turn-off time Toff, the switch operates in "discontinuous resonant current mode" (DCM). That is, in each switching cycle, the resonant current returns to zero after the end of the Ton phase or at the beginning of the Toff phase, and remains in a zero-current state for the remaining Toff time until the next cycle arrives. When the main switch receives the turn-on signal again, its initial current value has stabilized at zero, thus achieving zero-current turn-on (ZCS), eliminating the current surge at the moment of turn-on and reducing losses.

[0071] Reference Figure 3 Furthermore, in one embodiment, the conduction time Ton is set to half of the inherent resonant period Tr of the resonant network 2 (i.e., Ton = Tr / 2). The control unit, using pre-stored parameters of the resonant inductance Lr and resonant capacitance Cr, calculates the conduction time according to the formula... Tr can be calculated; or the voltage / current signal of the resonant network 2 can be collected by the hardware detection circuit and Tr can be calculated in real time. When Ton is fixed at half of the resonant period, the switching transistors on the secondary side (main switching transistors Q5, Q8 and corresponding complementary switching transistors Q6, Q7) can not only achieve zero-current turn-on, but also zero-current turn-off.

[0072] Understandably, at the instant the conduction phase ends, the resonant current completes half a sine wave change from "0 → peak → 0" and falls back to zero. At this moment, the control unit sends a turn-off signal, and the main switch turns off when the current is zero, resulting in neither current surge nor turn-off loss. Under this dual zero-current switching (ZCS) condition, the converter's switching losses are further reduced, and efficiency is further improved. Furthermore, this soft-switching characteristic allows for a wider range of switch device selection; both MOSFETs and IGBTs can be used and maintain high-efficiency operation.

[0073] Understandably, fixing the on-time of the main switch to half of the resonant period enables zero-current turn-on and turn-off of the main switch, reducing switching losses. By dynamically adjusting the turn-off time to adapt to changes in the secondary input voltage and primary load, the switching period becomes monotonically correlated, preventing switching frequency fluctuations and improving the problem of traditional reverse mode easily entering discontinuous waveform mode. No additional resonant components are needed, which improves output stability and conversion efficiency while maintaining the low cost and small size advantages of the original LLC topology.

[0074] Reference Figure 3 In one embodiment, the turn-off time Toff includes a first dead time t1 with continuous time, a symmetrical turn-on time t2, and a second dead time t3. The symmetrical turn-on time t2 is the turn-on duration of the complementary switch in the secondary-side switch network 4, and t2 is equal to Ton. The complementary switch and the main switch are turned on alternately. The turn-off time Toff is adjusted by adjusting the second dead time t3.

[0075] Specifically, the first dead time t1 is the period between the turn-off time of the main switches Q5 / Q8 and the turn-on time of the complementary switches Q6 / Q7 (corresponding to...). Figure 3 The interval between the falling edge of Vgs of Q5 / Q8 and the rising edge of Vgs of Q6 / Q7 is used to prevent the main switch and the complementary switch from conducting simultaneously, reducing bridge arm shoot-through faults and ensuring circuit safety. The symmetrical conduction time t2 is the period immediately following t1 (corresponding to...). Figure 3The duration of the high level Vgs of Q6 / Q7 is the conduction time of the complementary switch. In this embodiment, t2 is set to 1 / 2Tr, which matches the conduction time Ton(1 / 2Tr) of the main switch. Its function is to allow the complementary switch to conduct within the reverse half-cycle of the resonant current, thus completing symmetrical conduction. The second dead time t3 is the period between the turn-off time of the complementary switch Q6 / Q7 and the next turn-on time of the main switch Q5 / Q8 (corresponding to...). Figure 3 The interval between the falling edge of Vgs of Q6 / Q7 and the next rising edge of Vgs of Q5 / Q8 is used as the "adjustable part" of the turn-off time Toff. By extending or shortening the duration of t3, the control unit can dynamically change the entire turn-off time Toff, thereby adjusting the switching period T and achieving flexible control of the voltage gain.

[0076] Understandably, the turn-off time is broken down into multiple time periods, and the symmetrical turn-on time is matched with the turn-on time of the main switch to ensure efficient energy transfer in the resonant circuit. The turn-off time is dynamically adjusted by adjusting the second dead time, which is flexible in adjustment. At the same time, the first dead time can prevent the main switch and the complementary switch from shooting through, ensuring circuit safety.

[0077] Reference Figure 3 In one embodiment, when the load and output voltage V1 of the primary-side switching network 1 are fixed, the peak value of the current Id of the main switch and the complementary switch in the secondary-side switching network 4 increases monotonically with the increase of the secondary-side input voltage V2.

[0078] The implementation principle of this application embodiment is as follows:

[0079] In reverse operation, the secondary input voltage V2 determines the excitation voltage amplitude of resonant network 2. This embodiment uses a fixed Ton = 1 / 2Tr, meaning that during the conduction phase (i.e., energy transfer phase) of each switching cycle, resonant network 2 always completes a full half-wave resonance with a fixed duration. Because the time window is locked, the peak value (i.e., peak height) of the resonant current depends only on the current input voltage (higher input voltage, greater thrust, higher peak height), and is independent of subsequent actions. Frequency adjustment is achieved by changing the off-time Toff. Toff is in the "waiting phase" after the resonant current returns to zero. Adjusting Toff only changes the waiting time before the next energy transfer begins (i.e., changes the frequency of energy transfer), without reverting back to interfere with the current waveform and peak value already formed in the Ton stage. Unlike traditional control strategies, this application achieves decoupling of the current peak value and the switching frequency. Traditional schemes often compress or stretch the conduction time when adjusting the frequency, destroying the resonant state and causing the current peak value to drift nonlinearly with the frequency (i.e., the frequency and peak value are "bound" to interference). This application ensures through a decoupling mechanism that the peak value of the current Id increases monotonically with the increase of the secondary input voltage V2, thus improving the interference caused by frequency adjustment.

[0080] Understandably, the peak current is monotonically correlated with the secondary input voltage, ensuring that the circuit can operate stably under different input voltages, preventing output fluctuations caused by sudden current changes, further optimizing output characteristics, and improving the stability of the control loop.

[0081] In one embodiment, the main switch and complementary switch in the secondary-side switch network 4 are both of the following types: MOSFET, IGBT, SiC MOSFET, or GaN HEMT. MOSFET is suitable for low-to-medium voltage, high-frequency, and low-power applications, such as on-board chargers and small energy storage inverters. IGBT is suitable for high-voltage, high-current, and high-power applications, such as industrial-grade energy storage converters and electric vehicle drive power supplies. SiC MOSFET and GaN HEMT are suitable for high-temperature, high-frequency, and high-voltage applications, such as ultra-fast charging piles for new energy vehicles and aerospace power systems.

[0082] It is understood that the control method of this application supports the selection of various power switching transistors. MOSFETs have fast switching speed and low on-resistance, IGBTs have large current carrying capacity, and SiC MOSFETs have high temperature resistance and low loss. They can be flexibly selected according to different power levels and operating conditions, thereby enhancing the adaptability and practicality of the solution.

[0083] In one embodiment, the LLC resonant converter includes a full-bridge LLC resonant converter, a half-bridge LLC resonant converter, or a multiphase interleaved LLC resonant converter.

[0084] The control method described in this application is universal and can be applied to various fields besides [specific applications]. Figure 1 In addition to the conventional full-bridge LLC converter shown, it can also be applied to the reverse working mode control of half-bridge LLC converters and multi-phase interleaved LLC resonant converters (such as three-phase, four-phase, and six-phase converters).

[0085] Specifically, such as Figure 4 As shown, a three-phase interleaved LLC circuit is used as an example. Q1~Q6 are the switching transistors of the primary-side switching network 1, Q7~Q12 are the switching transistors of the secondary-side switching network 4, Lr1~Lr3 are the resonant inductors of each phase resonant network 2, and Cr1~Cr3 are the resonant capacitors of each phase resonant network 2. When the circuit enters the reverse operating mode, Q7~Q12 participate in the control as the switching transistors of the secondary-side switching network 4. According to the actual power requirements, the control unit can be flexibly configured as a "three-arm reverse mode" or a "two-arm reverse mode", and adopt the "fixed Ton + adjusted Toff" control strategy in the above embodiment in different modes. For example, in the three-arm reverse operating mode, all three arms of the three-phase interleaved LLC resonant converter are engaged. The control unit controls the switching transistors Q7~Q12 of the secondary-side switching network 4, so that the conduction phases of the three half-bridge arms are staggered by 120 degrees. Under this phase constraint, the control unit executes the control logic described in the above embodiment for each bridge arm, that is, fixes the on-time Ton of the switch (e.g., Ton=Tr / 2, Tr / 4 is greater than or less than Tr), and adjusts the off-time Toff of the switch according to the voltage gain G, thereby achieving reverse voltage regulation control in the full power range.

[0086] For example, in the dual-arm reverse operating mode, this mode is suitable for situations where the reverse power requirement is less than the forward power requirement (e.g., only 2 / 3 of the power is needed). The control unit randomly selects two half-bridge arms from the three-phase arms to operate, and controls the conduction phases of the switching transistors of these two arms to be staggered by 180 degrees. At this time, the operating characteristics of the three-phase circuit are equivalent to the reverse circuit of a conventional full-bridge LLC circuit. The strategy of "fixing the Ton of the main switch and adjusting the Toff" in the above embodiment is also applied for control. In application scenarios using this derating operating mode, the switching transistor of the remaining unused bridge arm (e.g., the originally configured MOSFET or IGBT) can be replaced with a diode to further reduce device costs.

[0087] It should be noted that, in cases such as Figure 1 In the full-bridge LLC topology shown, the main switch refers to the conducting group consisting of two diagonally opposite switches (e.g., Q5 and Q8), and the complementary switch refers to another diagonally opposite set of switches (e.g., Q6 and Q7). In this case, the control unit simultaneously drives both diagonally opposite switches; in... Figure 4In the three-phase interleaved LLC topology shown, the circuit can be equivalent to three parallel half-bridges. In this case, each phase arm operates independently. For each phase, the upper transistor connected to the positive input (e.g., Q7) is the main switch, and the lower transistor connected to the negative input (e.g., Q8) is the complementary switch. Although the physical combinations differ, the control strategy of this application (fixed on-time Ton, adjustable off-time Toff) is applicable to the above topology and can improve its performance in reverse operating mode, thereby broadening the application range of the control method and enhancing the versatility of the technical solution.

[0088] It is understandable that the reverse control method of this application can be adapted to various LLC topologies. Whether it is a basic topology of full bridge or half bridge, or an extended topology with multiple interleaved phases such as three-phase and four-phase, it can improve the performance of the reverse working mode, broaden the application scope of the control method, and enhance the versatility of the technical solution.

[0089] This application also discloses an electronic device that integrates an LLC resonant converter. The topology of the LLC resonant converter includes, but is not limited to, the full-bridge LLC resonant converter, half-bridge LLC resonant converter, or multiphase interleaved LLC resonant converter described in the above embodiments. The LLC resonant converter is equipped with a control unit, which is configured to execute the reverse control method of the LLC resonant converter in any of the above embodiments. That is, by setting the on-time of the main switch of the secondary-side switch network 4 to a constant on-time, collecting and calculating the gain G of the input voltage V2 and the output voltage V1, and dynamically adjusting the off-time Toff of the main switch, stable control in the reverse working mode is achieved. At the same time, a soft-start step can be selectively integrated to further improve the reliability of the startup phase. The types of electronic devices include, but are not limited to, devices that require bidirectional power transmission, such as on-board chargers for new energy vehicles, portable energy storage power supplies, photovoltaic inverters, bidirectional energy storage converters, and rail transit auxiliary power supply systems.

[0090] It is understood that the electronic device of this application integrates an LLC resonant converter, which has bidirectional power conversion capability. In reverse working mode, the output is stable and the loss is low, which can meet the application requirements of bidirectional switching of charging and discharging in the new energy field and improve the overall performance and reliability of the electronic device.

[0091] The above are all preferred embodiments of this application, and are not intended to limit the scope of protection of this application. Therefore, all equivalent changes made in accordance with the structure, shape and principle of this application should be covered within the scope of protection of this application.

Claims

1. A reverse control method for an LLC resonant converter, the LLC resonant converter comprising a primary-side switching network (1), a secondary-side switching network (4), a resonant network (2), and a transformer (3), the transformer (3) comprising a primary winding and a secondary winding, the primary winding being connected to the primary-side switching network (1), the secondary winding being connected to the secondary-side switching network (4), and the resonant network (2) being connected in series between the primary-side switching network (1) and the primary winding of the transformer (3), characterized in that, The method includes the following steps: S1. Set the on-time Ton of the main switch in the secondary-side switch network (4) to a constant on-time. S2. Collect the input voltage V2 of the secondary-side switching network (4) and the output voltage V1 of the primary-side switching network (1), and calculate the gain G of the input voltage V2 and the output voltage V1; S3. Adjust the turn-off time Toff of the main switch in the secondary-side switching network (4) according to the gain G, so that the turn-off time Toff changes monotonically with the change of the gain G. Specifically, when the gain G increases, the off-time Toff decreases, so that the output voltage V1 returns to the preset target. When the gain G decreases, the off-time Toff is increased so that the output voltage V1 returns to the preset target. The turn-off time Toff increases monotonically as the input voltage V2 on the secondary side increases, in order to maintain the output voltage V1 on the primary side.

2. The reverse control method according to claim 1, characterized in that, The LLC resonant converter further includes a resonant network (2) disposed between the primary-side switching network (1) and the secondary-side switching network (4). The resonant network (2) includes a resonant inductor Lr and a resonant capacitor Cr. The resonant period of the resonant network is... The conduction time Ton of the main switch in the secondary switch network (4) is set to be equal to 1 / 2Tr.

3. The reverse control method according to claim 1, characterized in that, The turn-off time Toff includes a first dead time t1, a symmetrical turn-on time t2, and a second dead time t3, which are set continuously. The symmetrical turn-on time t2 is the turn-on duration of the complementary switch in the secondary switch network (4). The symmetrical turn-on time t2 is equal to the turn-on time Ton of the main switch. The complementary switch and the main switch are turned on alternately. The turn-off time Toff is adjusted by adjusting the second dead time t3.

4. The reverse control method according to claim 3, characterized in that, When the load of the primary-side switching network (1) and the output voltage V1 are fixed, the peak value of the current Id of the main switch and the complementary switch in the secondary-side switching network (4) increases monotonically with the increase of the input voltage V2 on the secondary side.

5. The reverse control method according to claim 3, characterized in that, Both the main switch and the complementary switch are of the types including MOSFET, IGBT, SiC MOSFET, or GaN HEMT.

6. The reverse control method according to claim 1, characterized in that, The LLC resonant converter includes a full-bridge LLC resonant converter, a half-bridge LLC resonant converter, or a multiphase interleaved LLC resonant converter.

7. The reverse control method according to claim 1, characterized in that, It also includes a soft start step S0, controlling the secondary switch network (4) to start at a preset minimum switching frequency, and gradually increasing the switching frequency according to a preset frequency change rate until the switching frequency reaches the initial operating frequency required to enter steady-state control, thereby suppressing the surge current during the start-up phase through a smooth frequency transition.

8. An LLC resonant converter, characterized in that, It includes a primary-side switching network (1), a secondary-side switching network (4), a resonant network (2), a transformer (3), and a control unit; The transformer (3) includes a primary winding and a secondary winding. The primary winding is connected to the primary switching network (1), which is used to connect to the first power supply side or the first load side. The secondary winding is connected to the secondary switching network (4), which is used to connect to the second power supply side or the second load side. The resonant network (2) is connected in series between the primary-side switching network (1) and the primary-side winding of the transformer (3). The resonant network (2) includes a resonant inductor Lr and a resonant capacitor Cr, which are used to form an LLC resonant circuit. The signal input terminal of the control unit is connected to the voltage output terminal of the primary-side switch network (1) and the voltage input terminal of the secondary-side switch network (4) for acquiring the primary-side output voltage V1 and the secondary-side input voltage V2. The signal output terminal of the control unit is connected to the control terminal of the secondary-side switch network (4) for outputting a turn-on / turn-off control signal. The control unit is configured to execute the reverse control method as described in any one of claims 1-7.

9. An electronic device, characterized in that, Including the LLC resonant converter as described in claim 8.

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