A power amplification driving method and system for long wave communication antenna

Abstract

The present application relates to the technical field of power amplification, and particularly relates to a power amplification driving method and system for long-wave communication antenna. The method comprises the following steps: collecting voltage and current data of the output end of the power amplifier; constructing an instantaneous reactive potential observation field, and calculating an instantaneous reactive potential representing residual oscillation of the antenna loop; extracting current frequency domain features, combining the instantaneous reactive potential, and using a multi-dimensional feature fusion inversion algorithm to calculate a tuning inductance thermal drift amount; based on the principle of charge conservation, using the thermal drift amount to correct the initial inductance, and combining real-time working conditions to dynamically calculate an updated dead time meeting zero-voltage turn-on. The present application accurately locks the parameter time-varying characteristics of the antenna loop through the above operation, and accordingly realizes adaptive compensation of the dead time, ensuring stable soft switching operation of the long-wave transmitter under full power working conditions.

Description

Technical Field

[0001] This invention relates to the field of power amplification technology, and in particular to a power amplification driving method and system for long-wave communication antennas. Background Technology

[0002] Long-wave communication systems are the core infrastructure for high-power radio transmission. Their power amplifier stages typically employ Class E or Class DE switching topologies and rely on high-quality Q-factor LC tuned circuits to achieve zero-voltage switching (ZVS). Existing technologies are generally based on the physical assumptions of linear time-invariant systems, assuming that the parameters of the inductors in the tuned circuit are controlled only by mechanical commands and remain constant for short periods. While some technologies introduce temperature compensation, they only assume that the inductance value changes linearly and monotonically with temperature. Therefore, control strategies often employ phase-locked loop (PLL) technology based on zero-crossing detection, maintaining the resonant state by locking the phase difference between voltage and current.

[0003] However, at the microscopic physics level and under high-power continuous wave transmission conditions, magnetic components exhibit magnetothermal dynamic hysteresis. Under high-frequency, high-flux excitation, the permeability of the tuned magnetic core of a long-wave transmitting antenna decays nonlinearly. As heat buildup increases within the core, domain flipping is hindered, leading to a surge in the third harmonic component, and the inductance decay exhibits nonlinear avalanche characteristics. At this point, the voltage zero-crossing point no longer accurately represents the zero potential energy point of the circuit, and the current waveform is distorted due to the presence of higher harmonics.

[0004] Therefore, existing technologies suffer from the drawback of using ideal sinusoidal steady-state operators to describe nonlinear magnetocaloric transient processes. This spatiotemporal mismatch between control parameters and physical entities is the root cause of zero-voltage turn-on failure. When the control system continues to switch according to the originally detected phase, the actual inductance has decreased due to thermal drift, reducing the characteristic impedance of the resonant tank circuit. This results in the charge on the junction capacitance of the switching transistor not being completely removed before the dead time ends. The residual charge is forcibly short-circuited and discharged through the channel at the moment the switching transistor turns on, triggering a hard-switching surge. This leads to a sharp increase in the switching losses of power devices, and in severe cases, it can damage the switching transistor due to transient thermal breakdown. Furthermore, to avoid hard switching, existing engineering methods often employ a strategy of reserving an ultra-large dead time, but this directly leads to a decrease in effective transmission power and a deterioration in waveform distortion, making it impossible to simultaneously ensure reliability and transmission efficiency. Summary of the Invention

[0005] To address the problem of equipment damage and efficiency degradation caused by the mismatch between control parameters and physical parameters leading to the switching transistor leaving the zero-voltage turn-on state under magnetic core thermal drift conditions, this invention provides a power amplification driving method and system for long-wave communication antennas.

[0006] In a first aspect, the present invention provides a power amplification driving method for a long-wave communication antenna, employing the following technical solution:

[0007] A power amplification driving method for a long-wave communication antenna includes the following steps:

[0008] Obtain the real-time voltage signal sequence and real-time current signal sequence of the power amplifier;

[0009] The real-time current signal sequence is quadrature phase-shifted to obtain quadrature current components, and the instantaneous reactive potential energy characterizing the loop oscillation is calculated by combining the real-time voltage signal sequence.

[0010] The fundamental amplitude and third harmonic amplitude of the real-time current signal sequence are extracted, and the fundamental amplitude, the third harmonic amplitude and the instantaneous reactive potential energy are calculated using a magnetic-thermal coupling inversion model to obtain the thermal drift of the tuned inductor.

[0011] The DC bus voltage, peak current, and pre-stored initial nominal inductance value are obtained. Based on the principle of charge conservation, the initial nominal inductance value is corrected using the thermal drift of the tuned inductor to obtain a real-time estimated inductance value. The update dead time that satisfies the zero-voltage turn-on condition is calculated based on the real-time estimated inductance value, the DC bus voltage, and the peak current.

[0012] The updated dead time is used to control the operation of the switching transistor to achieve power amplification drive.

[0013] This invention abandons the traditional simple phase detection logic and introduces instantaneous reactive potential energy and the thermal drift of the tuned inductor as intermediate calculation indicators. By constructing an observation field, it directly measures the microscopic energy oscillations within the circuit caused by the magnetocaloric effect. Thus, under the premise of using conventional sensors, it achieves the inversion of the nonlinear thermal drift of the magnetic core and adjusts the dead time accordingly, eliminating the risk of hard switching from its physical source.

[0014] Preferably, the instantaneous reactive potential energy satisfies the following relationship:

[0015]

[0016] In the formula, Characterizing the first The instantaneous reactive potential energy of each modulation cycle; Characterizes the total number of sampling points within a single period; Characterizing the first The modulation cycle number The instantaneous voltage value at each sampling point; Characterization of the first The modulation cycle number The corresponding values ​​in the orthogonal current components are obtained after phase-shifting the instantaneous current values ​​at each sampling point; Characterizes the preset sampling time interval.

[0017] This invention uses the discrete orthogonal integral method to map reactive power, which is originally only visible in the frequency domain, into a time-domain calculable cumulative energy value. This index has high sensitivity to small phase deviations and can effectively filter out random noise interference, providing a high signal-to-noise ratio input characteristic for subsequent thermal drift inversion.

[0018] Preferably, the orthogonal phase shifting of the real-time current signal sequence includes:

[0019] Construct a discrete Hilbert filter of a predetermined order;

[0020] The real-time current signal sequence is input into the discrete Hilbert filter for convolution operation, which causes the real-time current signal sequence to have a 90-degree phase lag, thus obtaining the orthogonal current components.

[0021] Preferably, the thermal drift of the tuned inductor satisfies the following relationship:

[0022]

[0023] In the formula, Characterizing the first The amount of thermal drift of the tuned inductor per modulation cycle; Characterizes the initial nominal inductance value; Characterizing the first The amplitude of the third harmonic in each modulation cycle; Characterizing the first The fundamental amplitude value for each modulation period; Characterizing the first The instantaneous reactive potential energy of each modulation cycle; The baseline energy storage constant characterizing the system; and Both represent the preset weighted coupling coefficients; Characterizes the preset zero-prevention parameter.

[0024] This invention establishes a multi-dimensional feature fusion model, in which the harmonic ratio characterizes the degree of hysteresis distortion of the magnetic core, while the energy ratio characterizes the degree of detuning of the overall impedance. Through weighted fusion, this scheme maps the thermal drift, which cannot be directly measured, into a calculable electrical characteristic quantity, solving the problem of parameter identification without temperature sensors.

[0025] Preferably, the method for determining the weighted coupling coefficient includes:

[0026] Temperature rise impedance characteristics of the tuned coil were tested to obtain inductance rate of change, harmonic distortion and residual energy data at different temperatures;

[0027] The weighted coupling coefficient is obtained by performing multiple linear regression analysis on the inductance rate of change data, the harmonic distortion data, and the residual energy data using the least squares method.

[0028] This invention uses offline data-driven calibration to solidify the physical properties of specific magnetic materials into the algorithm coefficients, enabling high estimation accuracy to be obtained with only simple algebraic operations during online calculations, thus reducing the computational burden on the real-time controller.

[0029] Preferably, the reference energy storage constant satisfies the following relationship:

[0030]

[0031] In the formula, Characterizes the effective value of the rated maximum current of the system; This characterizes the initial nominal inductance value.

[0032] Preferably, the update dead time satisfies the following relationship:

[0033]

[0034] In the formula, Characterizing the first The update dead time for each modulation cycle; Characterizes the preset equivalent output junction capacitance of the switching transistor; Characterizing the first The DC bus voltage for each modulation cycle; Characterizing the first The peak current for each modulation cycle; Characterizes the initial nominal inductance value; Characterizing the first The amount of thermal drift of the tuned inductor per modulation cycle; Characterizes the preset minimum safety timing margin of the system; Characterizes the preset zero-prevention parameter.

[0035] This invention introduces an adaptive correction factor based on the evolution of physical parameters. When the inductance decreases due to thermal decay, the correction factor automatically increases, extending the dead time and thus giving the junction capacitance sufficient time to complete charge extraction. This closed-loop control based on physical mechanisms ensures that the dead time adjustment is neither too conservative nor too aggressive.

[0036] Preferably, after calculating the update dead time that satisfies the zero-voltage turn-on condition based on the real-time estimated inductance value, the DC bus voltage, and the peak current, the method further includes:

[0037] Determine whether the update dead time exceeds the preset dead time limit;

[0038] When the update dead time exceeds the dead time limit, the update dead time is clamped to the boundary value of the dead time limit and a detuning alarm signal is issued.

[0039] Preferably, controlling the switching transistor's operation using the updated dead time includes:

[0040] The updated dead time is loaded into the pulse width modulation register of the digital signal processor;

[0041] At the start of the next modulation cycle, complementary gate drive signals are generated using the updated pulse width modulation register parameters.

[0042] Secondly, the present invention provides a power amplification driving system for long-wave communication antennas, employing the following technical solution:

[0043] A power amplifier driving system for a long-wave communication antenna includes a processor and a memory, wherein the memory stores computer program instructions, and when the computer program instructions are executed by the processor, the aforementioned power amplifier driving method for a long-wave communication antenna is implemented.

[0044] By adopting the above technical solution, a power amplification driving method for long-wave communication antennas is generated into a computer program and stored in a memory for loading and execution by a processor. This allows for the creation of terminal devices based on the memory and processor, facilitating their use.

[0045] The present invention has the following technical effects:

[0046] This invention, by introducing the instantaneous reactive potential energy as an energy domain index, successfully captures the weak detuning signal in the early stages of magnetic core thermal drift, exhibiting higher detection sensitivity than traditional phase detection methods. Combined with a magnetothermal coupling inversion model, it can acquire the physical health status of the antenna tuning circuit in real time, transforming the invisible thermal drift into a visible control variable.

[0047] Furthermore, this invention can adaptively adjust the dead-time window based on the real-time attenuation of the inductor, ensuring that the junction capacitance charge of the switching transistor can be completely extracted at different temperatures and power levels. This eliminates the risk of device failure caused by hard switching, extends the service life of power devices, and also enables the transmitter to always operate at a higher efficiency point, reducing the operating energy consumption and heat dissipation costs of long-wave communication systems. Attached Figure Description

[0048] Figure 1 This is a flowchart of a power amplification driving method for a long-wave communication antenna provided in an embodiment of the present invention;

[0049] Figure 2 The thermal drift inversion and tracking curves of the tuned inductor provided in the embodiments of the present invention;

[0050] Figure 3 A comparison diagram of the dead time dynamic correction waveform provided in the embodiments of the present invention. Detailed Implementation

[0051] This invention discloses a power amplification driving method for long-wave communication antennas, referring to... Figure 1 This includes steps S1-S5:

[0052] S1: Obtain the real-time voltage signal sequence and real-time current signal sequence of the power amplifier.

[0053] It should be noted that in long-wave, high-power transmission scenarios, the acquisition quality of high-frequency, high-voltage signals directly determines the accuracy of subsequent control. Distributed capacitance and mutual inductance effects in the physical circuitry can introduce common-mode interference into the original signal. This physical noise means that directly using the original waveform for calculations will introduce significant random errors. Therefore, the core objective of this step is to obtain clean, time-synchronized digital sequences of voltage and current, providing a reliable data foundation for constructing a physical observation field.

[0054] Preferably, as an example, acquiring the real-time voltage signal sequence and real-time current signal sequence of the power amplifier includes:

[0055] First, a broadband Hall current sensor and a high-voltage divider probe are connected to the output of the resonant circuit of the power amplifier to collect analog voltage and analog current signals for each modulation cycle.

[0056] Next, a high-precision analog-to-digital converter is used to synchronously discretize the analog voltage signal and the analog current signal at a preset sampling frequency to obtain discrete voltage data and discrete current data.

[0057] Subsequently, a first-in-first-out (FIFO) buffer queue is used to perform time-series alignment and storage processing on the discrete voltage and discrete current data, resulting in a length of [length missing]. The real-time voltage signal sequence and the real-time current signal sequence.

[0058] For ease of description, the discrete voltage data at each position in the real-time voltage signal sequence will be denoted as the instantaneous voltage value of each sampling point, and the discrete current data at each position in the real-time current signal sequence will be denoted as the instantaneous current value of each sampling point.

[0059] S2: Perform orthogonal phase shifting on the real-time current signal sequence to obtain orthogonal current components, and combine them with the real-time voltage signal sequence to calculate the instantaneous reactive potential energy characterizing the loop oscillation.

[0060] It should be noted that the tuning circuit of a long-wave communication antenna has an extremely high quality factor (Q), meaning that the circuit is extremely sensitive to changes in frequency and phase. Under continuous wave high-power transmission conditions, the microscopic thermal effects of magnetic components cause the circuit to gradually deviate from its resonant point. This physical characteristic implies that conventional voltage and current phase detection has a significant hysteresis, because by the time a macroscopic phase difference is detected, a large amount of reactive oscillation energy has often accumulated inside the circuit. If this energy term is ignored, the system can easily operate under microscopic detuning, leading to an increase in the standing wave ratio (VSWR). Therefore, the core objective of this step is to establish an instantaneous reactive potential energy observation field to capture microsecond-level detuning precursors in the time domain.

[0061] Preferably, as an example, the real-time current signal sequence is orthogonally phase-shifted to obtain orthogonal current components, and the instantaneous reactive potential energy characterizing the loop oscillation is calculated in combination with the real-time voltage signal sequence, including:

[0062] First, the real-time current signal sequence is convolved and phase-shifted using a discrete Hilbert filter of a preset order, so that all frequency components are phase-lagging by 90 degrees, resulting in an orthogonal current component sequence.

[0063] Next, the instantaneous reactive potential energy is calculated based on the real-time voltage signal sequence and the orthogonal current component sequence. The instantaneous reactive potential energy satisfies the following relationship:

[0064]

[0065] In the formula, Characterizing the first The instantaneous reactive potential energy of each modulation cycle; Characterizes the total number of sampling points within a single period, for example, Take 1024; Characterizing the first The modulation cycle number The instantaneous voltage value at each sampling point; Characterization of the first The modulation cycle number The corresponding values ​​in the orthogonal current components are obtained after phase-shifting the instantaneous current values ​​at each sampling point; Characterized by a preset sampling time interval, for example, Take 1 microsecond.

[0066] It is understandable that the discrete dot product term in the relation... This reflects the instantaneous reactive power density stored within the circuit at every microscopic moment. When the system is in perfect resonance, the voltage and current are in phase, and the resulting orthogonal current components... The voltage waveform is strictly orthogonal in the time domain, causing their integrals to accumulate and cancel each other out over a complete cycle, thus... Approaching zero indicates that the loop energy is completely radiated or consumed, with no residual internal oscillations. However, when the core thermal effect causes microscopic detuning of the loop parameters, the phase relationship between voltage and current breaks down, and orthogonality is lost. This results in the positive and negative energies in the integral term not being completely canceled out, leading to an imbalance in the accumulated result. It exhibits a significantly non-zero value. The absolute value of the value directly reflects the state of ineffective oscillation energy that resides within the loop and has not been effectively converted.

[0067] In this way, the above operations accurately describe the microscopic energy oscillation phenomenon caused by the thermal effect of magnetic components, effectively eliminating the control blind zone hidden danger caused by the hysteresis of conventional phase detection.

[0068] S3: Extract the fundamental amplitude and third harmonic amplitude of the real-time current signal sequence, and use the magnetic-thermal coupling inversion model to calculate the fundamental amplitude, the third harmonic amplitude and the instantaneous reactive potential energy to obtain the thermal drift of the tuned inductor.

[0069] It should be noted that the inductance drift of a tuned inductor is not a simple result of linear thermal expansion, but involves a complex physical process involving the impeded movement of magnetic domains in the magnetic material. The third harmonic appearing in the current waveform is direct evidence of distortion in the hysteresis loop, while the accumulation of reactive potential energy reflects the deviation in overall impedance. This multi-physics coupling characteristic means that a single indicator cannot accurately reconstruct the true state of the magnetic core. Therefore, the core objective of this step is to establish a magnetothermal coupling inversion model, using observable electrical characteristics to infer the unobservable thermal drift of the inductor.

[0070] Preferably, as an example, the fundamental amplitude and third harmonic amplitude of the real-time current signal sequence are extracted, and the fundamental amplitude, the third harmonic amplitude, and the instantaneous reactive potential energy are calculated using a magnetic-thermal coupling inversion model to obtain the thermal drift of the tuned inductor, including:

[0071] First, the real-time current signal sequence is subjected to spectral analysis using a Fast Fourier Transform algorithm to obtain the fundamental amplitude at the fundamental frequency and the third harmonic amplitude at the third harmonic. Simultaneously, the initial nominal inductance value in the system memory is acquired.

[0072] Next, the fundamental wave amplitude, the third harmonic wave amplitude, and the instantaneous reactive potential energy are subjected to multi-dimensional feature fusion and weighted calculation using the magnetic-thermal coupling inversion formula to obtain the tuned inductor thermal drift, which characterizes the degree of deviation of the tuned core parameters, specifically satisfying the following formula:

[0073]

[0074] In the formula, Characterizing the first The amount of thermal drift of the tuned inductor per modulation cycle; Characterizes the initial nominal inductance value; Characterizing the first The amplitude of the third harmonic in each modulation cycle; Characterizing the first The fundamental amplitude value for each modulation period; Characterizing the first The instantaneous reactive potential energy of each modulation cycle; The baseline energy storage constant characterizing the system; and Both represent the preset weighted coupling coefficients; Characterizing the preset zero-prevention parameter, for example, Take 0.01, This is a magnetic-thermal coupling inversion model.

[0075] It is understandable that the ratio of the first term in the relation is... This reflects the degree of nonlinear distortion of the current waveform. This value increases significantly as the flipping of magnetic domains inside the core is hindered, characterizing defects at the magnetic level; the second ratio... This reflects the degree of imbalance in the loop energy; the value increases as the loop impedance deviates from the resonant point, characterizing defects at the impedance level. These two logical terms are fused through weighting coefficients and are interconnected, jointly characterizing the comprehensive degradation state of the magnetic core under thermal stress. When both ratios increase simultaneously, the negative sign at the front of the formula is used to determine the relationship between them. The product directly affects the drift of the output. The value shows a negative increase. This trend indicates that the inductance is undergoing substantial thermal decay.

[0076] In this way, the above operations accurately describe the parameter drift phenomenon caused by the coupling of complex physical fields, and map the invisible microscopic physical changes into visible digital indicators, effectively solving the technical problem that a single indicator cannot accurately reproduce the true state of the magnetic core.

[0077] It should be noted that the important preset parameter is the magnetic-thermal coupling weighting coefficient. and Methods for obtaining [the information] include:

[0078] A standard tuned magnetic core coil was subjected to variable-temperature testing using a temperature-rising impedance scanning test platform to obtain the true inductance rate of change, harmonic ratio, and energy deviation ratio at different temperatures. The test data were then fitted using a binary linear regression algorithm to obtain the coefficients. and .

[0079] It should also be noted that the methods for obtaining the benchmark energy storage constant include:

[0080]

[0081] In the formula, Characterizes the effective value of the rated maximum current of the system; This characterizes the initial nominal inductance value.

[0082] S4: Obtain the DC bus voltage, peak current, and pre-stored initial nominal inductance value; based on the principle of charge conservation, use the thermal drift of the tuned inductor to correct the initial nominal inductance value to obtain a real-time estimated inductance value; calculate the update dead time that satisfies the zero-voltage turn-on condition based on the real-time estimated inductance value, the DC bus voltage, and the peak current.

[0083] It's important to note that the physical essence of achieving zero-voltage switching (ZVS) is based on the law of charge conservation. When the inductance decreases due to thermal drift, the characteristic impedance of the circuit decreases, and the rate of current rise increases. However, the inductor's energy density decreases, altering its ability to draw charge from the junction capacitance of the switching transistor. Maintaining a fixed dead time would result in incomplete or excessive charge extraction. Therefore, the core objective of this step is to dynamically calculate the optimal time window for charge extraction based on the real-time changes in the inductance value, according to the principle of charge conservation.

[0084] Preferably, as an example, the DC bus voltage, peak current, and pre-stored initial nominal inductance value are obtained; based on the principle of charge conservation, the initial nominal inductance value is corrected using the thermal drift of the tuned inductor to obtain a real-time estimated inductance value; and the update dead time to meet the zero-voltage turn-on condition is calculated based on the real-time estimated inductance value, the DC bus voltage, and the peak current, including:

[0085] First, preset hardware parameters are retrieved from non-volatile memory, including the initial nominal inductance value and the equivalent output junction capacitance of the switching transistor, and the peak DC bus voltage and current are collected using sensors.

[0086] Next, based on the initial nominal inductance value, the equivalent output junction capacitance of the switching transistor, the DC bus voltage, and the peak current, a dynamic mapping process is performed to obtain the updated dead time that satisfies the zero-voltage turn-on condition. The specific mapping relationship is as follows:

[0087]

[0088] In the formula, Characterizing the first The update dead time for each modulation cycle; Characterizes the preset equivalent output junction capacitance of the switching transistor; Characterizing the first The DC bus voltage for each modulation cycle; Characterizing the first The peak current for each modulation cycle; Characterizes the initial nominal inductance value; Characterizing the first The amount of thermal drift of the tuned inductor per modulation cycle; Characterizes the preset minimum safety timing margin of the system. Take 100 nanoseconds; Characterizes the preset zero-prevention parameter. Indicates the first Real-time estimated inductance value for each modulation cycle.

[0089] It is understandable that the baseline term in the first part of the relation is... This reflects the fundamental physical requirements for achieving zero-voltage switching, among which, It represents the total amount of charge stored on the parasitic junction capacitance of the switching transistor at the turn-off moment, which must be extracted; while This characterizes the maximum current draw capability that the current-cycle inductor circuit can provide. Logically, when the DC bus voltage... As the voltage increases, the amount of charge stored in the junction capacitance increases, or when the loop current... When the voltage decreases, the charge extraction rate slows down. Both of these situations lead to an increase in the reference term value, thereby driving the control system to extend the dead time. This allows sufficient time for the weak current to clear more charge, preventing forced switching on before the voltage drops to zero. (The square root term in the equation...) This constitutes an adaptive correction factor based on the evolution of physical parameters. When A negative value indicates that as the inductance decreases due to thermal effects, the denominator decreases, causing the correction factor to be greater than 1. This change reflects the weakening energy storage capacity of the circuit as the inductance decreases, resulting in a lower energy storage capacity at the same peak current. In this case, the inductor needs more time to complete the connection of the switching transistor junction capacitance. The charge extraction is affected. Therefore, increasing this correction factor directly drives the extension of the dead time. This adaptive adjustment ensures that the dead window remains sufficiently wide even under harsh operating conditions where inductor parameters deteriorate, guaranteeing that the junction capacitance charge is completely released.

[0090] Thus, the above operations accurately describe the adaptive update of the dead window corresponding to the decrease in charge extraction capability caused by inductor thermal drift, effectively eliminating the fatal hidden danger of hard switching failure caused by fixed dead time.

[0091] Finally, determine whether the update dead time exceeds the preset dead time limit.

[0092] When the update dead time exceeds the dead time limit, the update dead time is clamped to the boundary value of the dead time limit and a detuning alarm signal is issued.

[0093] S5: Use the updated dead time to control the operation of the switching transistor to achieve power amplification drive.

[0094] It is important to note that the calculated dead time must be precisely mapped to the hardware driver circuitry to be effective. Any timing jitter or delay will cause the aforementioned precise calculations to fail. Therefore, the core objective of this step is to seamlessly convert the digital domain calculation results into a pulse-width modulation signal in the physical domain.

[0095] Preferably, as an example, the operation of the switching transistor is controlled using the updated dead time to achieve power amplification drive, including:

[0096] First, the update dead time is discretized and quantized using a clock cycle conversion unit to obtain the corresponding dead time count value.

[0097] Next, the dead-time count value is loaded using the register write interface of the digital signal processor and written into the dead-time configuration register of the pulse width modulation module.

[0098] Subsequently, at the beginning of the next modulation cycle, the pulse width modulation generator is used to process the updated dead-time configuration register data to obtain a complementary gate drive signal with updated dead time.

[0099] Finally, the complementary gate drive signal is used to alternately turn on the upper and lower bridge arm switches of the power amplifier to achieve power amplification drive for the long-wave communication antenna.

[0100] To demonstrate the effectiveness of the solution, relevant experiments were conducted. Below are the images obtained from the experiments:

[0101] Figure 2The graph shows the inversion and tracking curves for the thermal drift of the tuning inductor. The horizontal dotted line represents the pre-stored initial nominal inductance value, which remains constant over time and serves as a reference. The solid line represents the calculated real-time estimated inductance value, which shows a monotonically decreasing trend over time. The dashed line represents the third harmonic amplitude extracted from the current signal, which shows an increasing trend over time. The double-headed arrows indicate the amount of thermal drift of the tuning inductor, visually displaying the difference between the two values.

[0102] The graph clearly shows that the solid line accurately depicts the decay of inductance with increasing continuous operating time, which perfectly matches the physical law that magnetic materials experience a decrease in permeability due to heating. The rising dotted line and the falling solid line exhibit a high negative correlation on the time axis. This proves the correctness of the magnetic-thermal coupling inversion model proposed in this invention, that is, it successfully utilizes observable harmonic distortion characteristics to inversely calculate the unobservable thermal drift of the inductance.

[0103] Figure 3 The waveform comparison chart shows the dynamic correction of dead time. The dashed waveform represents the drain-source voltage waveform of the switching transistor under the conventional fixed dead time strategy, while the solid waveform represents the drain-source voltage waveform of the switching transistor under the updated dead time strategy.

[0104] The image shows a distinct voltage oscillation spike at the beginning of the cycle in the dashed waveform. This indicates that when the inductance decreases due to thermal effects, the original fixed dead time becomes too short, causing the switching transistor to turn on prematurely before the charge on the junction capacitance is fully removed. This hard-switching phenomenon is a major cause of power transistor overheating and failure. The solid waveform shows that by adaptively extending the dead time based on the principle of charge conservation, the voltage waveform smoothly drops to zero potential before the switching transistor turns on. This proves that this solution can dynamically adjust the timing according to real-time inductance changes, ensuring that the ZVS condition is still met under harsh operating conditions, thus guaranteeing the long-term safe operation of the equipment.

[0105] This invention also discloses a power amplification driving system for a long-wave communication antenna, including a processor and a memory. The memory stores computer program instructions, which, when executed by the processor, implement a power amplification driving method for a long-wave communication antenna according to the present invention.

[0106] The system also includes other components well known to those skilled in the art, such as communication buses and communication interfaces, the settings and functions of which are known in the art and will not be described in detail here.

[0107] In this invention, the aforementioned memory can be any tangible medium containing or storing a program that can be used or combined with an instruction execution system, apparatus, or device. For example, a computer-readable storage medium can be any suitable magnetic or magneto-optical storage medium, such as resistive random access memory (DRAM), dynamic random access memory (DRAM), static random access memory (SRAM), enhanced dynamic random access memory (DRAM), high-bandwidth memory, hybrid memory cube, etc., or any other medium that can be used to store desired information and can be accessed by an application, module, or both. Any such computer storage medium can be part of a device or accessible to or connected to a device.

Claims

1. A power amplification driving method for a long-wave communication antenna, characterized in that, Including the following steps: Obtain the real-time voltage signal sequence and real-time current signal sequence of the power amplifier; The real-time current signal sequence is quadrature phase-shifted to obtain quadrature current components, and the instantaneous reactive potential energy characterizing the loop oscillation is calculated by combining the real-time voltage signal sequence. The fundamental amplitude and third harmonic amplitude of the real-time current signal sequence are extracted, and the fundamental amplitude, the third harmonic amplitude and the instantaneous reactive potential energy are calculated using a magnetic-thermal coupling inversion model to obtain the thermal drift of the tuned inductor. Obtain the DC bus voltage, peak current, and pre-stored initial nominal inductance value; Based on the principle of charge conservation, the initial nominal inductance value is corrected by the thermal drift of the tuned inductor to obtain a real-time estimated inductance value. The update dead time that satisfies the zero-voltage turn-on condition is calculated based on the real-time estimated inductance value, the DC bus voltage, and the peak current. The updated dead time is used to control the operation of the switching transistor to achieve power amplification drive.

2. The power amplification driving method for a long-wave communication antenna according to claim 1, characterized in that, The instantaneous reactive potential energy satisfies the following relationship: In the formula, Characterizing the first The instantaneous reactive potential energy of each modulation cycle; Characterizes the total number of sampling points within a single period; Characterizing the first The modulation cycle number The instantaneous voltage value at each sampling point; Characterization of the first The modulation cycle number The corresponding values ​​in the orthogonal current components are obtained after phase-shifting the instantaneous current values ​​at each sampling point; Characterizes the preset sampling time interval.

3. The power amplification driving method for a long-wave communication antenna according to claim 1, characterized in that, The quadrature phase shifting of the real-time current signal sequence includes: Construct a discrete Hilbert filter of a predetermined order; The real-time current signal sequence is input into the discrete Hilbert filter for convolution operation, which causes the real-time current signal sequence to have a 90-degree phase lag, thus obtaining the orthogonal current components.

4. The power amplification driving method for a long-wave communication antenna according to claim 1, characterized in that, The thermal drift of the tuned inductor satisfies the following relationship: In the formula, Characterizing the first The amount of thermal drift of the tuned inductor per modulation cycle; Characterizes the initial nominal inductance value; Characterizing the first The amplitude of the third harmonic in each modulation cycle; Characterizing the first The fundamental amplitude value for each modulation period; Characterizing the first The instantaneous reactive potential energy of each modulation cycle; The baseline energy storage constant characterizing the system; and Both represent the preset weighted coupling coefficients; Characterizes the preset zero-prevention parameter.

5. The power amplification driving method for a long-wave communication antenna according to claim 4, characterized in that, The method for determining the weighted coupling coefficient includes: Temperature rise impedance characteristics of the tuned coil were tested to obtain inductance rate of change, harmonic distortion and residual energy data at different temperatures; The weighted coupling coefficient is obtained by performing multiple linear regression analysis on the inductance rate of change data, the harmonic distortion data, and the residual energy data using the least squares method.

6. The power amplification driving method for a long-wave communication antenna according to claim 4, characterized in that, The reference energy storage constant satisfies the following relationship: In the formula, Characterizes the effective value of the rated maximum current of the system; Characterizes the initial nominal inductance value.

7. The power amplification driving method for a long-wave communication antenna according to claim 6, characterized in that, The update dead time satisfies the following relationship: In the formula, Characterizing the first The update dead time for each modulation cycle; Characterizes the preset equivalent output junction capacitance of the switching transistor; Characterizing the first The DC bus voltage for each modulation cycle; Characterizing the first The peak current for each modulation cycle; Characterizes the initial nominal inductance value; Characterizing the first The amount of thermal drift of the tuned inductor per modulation cycle; Characterizes the preset minimum safety timing margin of the system; Characterizes the preset zero-prevention parameter.

8. The power amplification driving method for a long-wave communication antenna according to claim 1, characterized in that, After calculating the update dead time that satisfies the zero-voltage turn-on condition based on the real-time estimated inductance value, the DC bus voltage, and the peak current, the method further includes: Determine whether the update dead time exceeds the preset dead time limit; When the update dead time exceeds the dead time limit, the update dead time is clamped to the boundary value of the dead time limit and a detuning alarm signal is issued.

9. The power amplification driving method for a long-wave communication antenna according to claim 1, characterized in that, The method of controlling the switching transistor's operation using the updated dead time includes: The updated dead time is loaded into the pulse width modulation register of the digital signal processor; At the start of the next modulation cycle, complementary gate drive signals are generated using the updated pulse width modulation register parameters.

10. A power amplifier drive system for a long-wave communication antenna, characterized in that, include: A processor and a memory, the memory storing computer program instructions that, when executed by the processor, implement a power amplification driving method for a long-wave communication antenna according to any one of claims 1-9.

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