Driving circuit and driving module for multiple bridge arms

Abstract

The invention provides a driving circuit and a driving module. The driving circuit comprises a driving stage circuit which is coupled between a first power supply end and a second power supply end, the driving stage circuit is further provided with an output end for providing an output signal, and the output signal is a first voltage or a second voltage relative to a reference ground end; and a negative voltage generation circuit coupled to the second power supply terminal and providing a second voltage, the negative voltage generation circuit comprising: a first switching device coupled between the reference ground terminal and the second power supply terminal; the bias circuit is used for providing bias current for the first switching device; the control circuit receives a feedback signal representing a second voltage and adjusts the second voltage to a preset value according to the feedback signal and the reference signal, and the second voltage is a negative value relative to the reference ground end.

Description

Technical Field

[0001] This application relates to the field of circuits, and more specifically, to a drive circuit and a drive module for multi-bridge arms. Background Technology

[0002] In high-voltage, high-current power systems, due to the parasitic inductance and Miller capacitance effect in the circuit, power switching devices such as insulated-gate bipolar transistors (IGBTs) or silicon carbide MOSFETs in a half-bridge topology will induce voltage spikes on their gates when turned off. These voltage spikes may cause the gate voltage to fluctuate around 0V or even briefly exceed the threshold voltage, resulting in "Miller conduction" of the power switching device, causing unexpected half-bridge shoot-through, i.e., false conduction, which can burn out the power switching device in severe cases.

[0003] Therefore, to reduce the risk of false turn-on, a negative voltage is often applied at the gate to turn off the power device, a technique known as negative voltage drive. Negative voltage drive can significantly suppress the risk of false turn-on in power switching devices and enhance the system's robustness over a wide temperature range and in electromagnetic interference environments. It is a core guarantee mechanism for ensuring the safe and reliable operation of power switching devices (e.g., in applications such as automotive electronics where functional safety levels and power density requirements are stringent).

[0004] Negative pressure drive technology has always been a research focus in this field. Summary of the Invention

[0005] According to one aspect of this application, a driving circuit is provided, comprising: a driving stage circuit coupled between a first power supply terminal and a second power supply terminal, the driving stage circuit further having an output terminal providing an output signal, wherein the output signal is a first voltage or a second voltage relative to a reference ground terminal; and a negative voltage generating circuit coupled to the second power supply terminal and providing the second voltage, the negative voltage generating circuit comprising: a first switching device coupled between the reference ground terminal and the second power supply terminal; a bias circuit providing a bias current to the first switching device; and a control circuit receiving a feedback signal characterizing the second voltage and adjusting the second voltage to a preset value according to the feedback signal and the reference signal, wherein the second voltage is negative relative to the reference ground terminal.

[0006] According to one aspect of this application, a drive module for a multi-arm bridge is provided, wherein each arm bridge includes two power switching devices, and each power switching device corresponds to a drive circuit as described above. Attached Figure Description

[0007] Figure 1 A circuit diagram is shown for one embodiment of using a flyback circuit to generate a negative voltage, according to some embodiments.

[0008] Figure 2 A circuit diagram of another embodiment of using a flyback circuit to generate a negative voltage is shown, according to some embodiments.

[0009] Figure 3 It shows the basis Figure 2 The diagram shows a three-phase bridge drive circuit.

[0010] Figure 4 A circuit diagram of a driving circuit according to an embodiment of this application is shown.

[0011] Figure 5 Examples of embodiments according to this application are shown. Figure 4 More details of the drive circuit shown.

[0012] Figure 6 Examples of embodiments according to this application are shown. Figure 5 The diagram shows the key signal waveforms of the drive circuit.

[0013] Figure 7A A schematic diagram of a drive circuit used to drive a three-phase bridge according to an embodiment of this application is shown.

[0014] Figure 7B A schematic diagram of a drive circuit used to drive a three-phase bridge according to an embodiment of this application is shown.

[0015] Figure 7C The diagram shows key signal waveforms of a drive circuit with transient current regulation function according to an embodiment of this application.

[0016] Figure 8A , Figure 8B and Figure 8C An example circuit diagram of a reference signal generation circuit in a drive circuit according to an embodiment of this application is shown. Detailed Implementation

[0017] The technical solutions of the embodiments of this disclosure will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some, not all, of the embodiments of this disclosure. All other embodiments obtained by those skilled in the art based on the embodiments of this disclosure without creative effort are within the scope of protection of this disclosure.

[0018] In some solutions in this field, negative pressure is generated by using a charge pump or a DC-DC converter circuit such as a buck-boost circuit or a fly-back circuit.

[0019] Figure 1A circuit diagram is shown for one embodiment of using a flyback circuit to generate a negative voltage, according to some embodiments.

[0020] For example, such as Figure 1 As shown, under the control of the flyback controller 10, a negative voltage is generated using the secondary coil tap TS of the flyback circuit 100. Figure 1 In this circuit, by disconnecting the reference ground terminal GND from the center tap TS of the secondary coil, stable positive voltage VCC0 and negative voltage VEE0 are obtained at the two ends of the coil, respectively. This circuit works in conjunction with the driver stage circuit DRV to provide a stable positive bias voltage VCC0 or negative bias voltage VEE0 to the gate G of the power switching device SW1.

[0021] However, the specific value of the positive bias voltage VCC0 or negative bias voltage VEE0 of the power switching device SW1 is determined by the turns ratio of the primary and secondary sides. To change the negative bias voltage VEE0, the primary input voltage VIN or the turns ratio of the coil can only be changed. When only the primary input voltage VIN is changed, the positive bias voltage VCC0 or the negative bias voltage VEE0 of the power switching device SW1 will change synchronously and proportionally. When only the turns ratio is changed, the value of the positive bias voltage VCC0 will also change with the adjustment of the negative bias voltage VEE0. Therefore, the negative bias voltage VEE0 generated by this technical solution is not easy to adjust.

[0022] Figure 2 A circuit diagram of another embodiment of using a flyback circuit to generate a negative voltage is shown, according to some embodiments.

[0023] Figure 2 The technical solution shown uses a Zener diode Z1 biased between the two output terminals of the secondary coil of the flyback circuit. The characteristic of the Zener diode Z1 is that it can maintain a stable voltage within a specific current range when operating under reverse bias. By using a quiescent current (such as the resistor R0 shown in the figure) to bias the Zener diode Z1, the voltage across the Zener diode Z1 can remain constant. By connecting the negative terminal of the Zener diode Z1 to the reference ground terminal GND, a stable negative bias voltage VEE0 can be obtained at the positive terminal of the Zener diode Z1.

[0024] However, this method has the following drawbacks:

[0025] 1. The operating current range of the Zener diode Z1 needs to be strictly controlled, usually through a series resistor (such as...). Figure 2 The voltage regulation is achieved using the resistor R0 shown in the diagram. Too small a current will result in poor voltage regulation, while too large a current may damage the Zener diode, further leading to damage to the power device.

[0026] 2. When Zener diode Z1 is used to generate negative bias voltage for the power switching devices on each lower arm of a multiphase bridge, such as a three-phase bridge, the reference ground terminals of each lower arm are connected together via a high-voltage bus. The negative bias voltage required to turn off the power switching devices of each lower arm is usually provided by the secondary coil of the same flyback circuit.

[0027] Figure 3 It shows the basis Figure 2 The diagram shows a three-phase bridge drive circuit.

[0028] like Figure 3 As shown, the three-phase bridge including switching devices SW1-SW6 is a three-phase inverter bridge used to drive motor 20; however, those skilled in the art will know that this three-phase bridge can be used for other purposes. The voltage drops from the reference ground terminals (shown as GND1, GND2, and GND3, respectively) of the power switching devices (SW2 / SW4 / SW6) on each lower arm of the three-phase bridge to the negative voltage output terminal (also denoted as VEE0) of the flyback circuit may differ. For example, the reference ground terminal GND1 may be connected to the reference ground of the entire system (e.g., the vehicle body), and due to the presence of large currents on the bus and unavoidable bus impedance, there will be a voltage difference between the reference ground terminals GND2 and GND3 relative to the reference ground terminal GND1. Similarly, the voltage drops seen by the Zener diodes connected to the respective reference ground terminals relative to the same negative voltage output terminal VEE0 will also differ. Furthermore, because the precision, current-voltage characteristic curves, and current capabilities of each Zener diode cannot be exactly the same due to manufacturing variations, even without considering the voltage difference between reference grounds GND1, GND2, and GND3, the current flowing through each Zener diode is uneven, resulting in different voltage drops observed in each Zener diode. Moreover, if the voltage difference between reference grounds GND1, GND2, and GND3 is taken into account, the current imbalance is further exacerbated. In practical applications, Zener diodes carrying more current will have significantly reduced lifespans, making them more prone to failure and safety risks.

[0029] Furthermore, the drive circuitry used to drive power switching devices can be packaged into a single driver chip, employing... Figure 2 or Figure 3 The technical solution requires more off-chip components or additional pins for the driver chip, thus increasing design complexity and cost. Furthermore, adopting... Figure 2 or Figure 3 The technical solution requires additional off-chip capacitors to generate negative bias voltage and cannot detect abnormalities such as negative voltage or undervoltage in a timely manner.

[0030] In addition, there are some negative voltage drive technologies, but besides the disadvantages mentioned above, they may also have problems such as complex circuits, many modules, large area, high cost and power consumption, and / or large output voltage ripple or electromagnetic interference (EMI) problems, etc.

[0031] Therefore, various embodiments of this application provide a drive circuit for switching devices that can provide a stable negative bias voltage for turning off power switching devices with low power consumption, and the negative bias voltage can be set more conveniently as needed. Furthermore, the drive circuit provided in this application can also be used for temperature regulation and transient current regulation when driving power switching devices on the lower arm of a multiphase bridge, so that the temperature / transient current between phases can be balanced. In addition, the drive circuit provided in this application can save on the cost of off-chip components, reduce size, and simplify board-level design.

[0032] The following will combine Figures 4 to 8C The relevant solutions for the driving circuit provided in this application are described in detail.

[0033] Figure 4 A circuit diagram of a drive circuit 400 according to an embodiment of this application is shown. The drive circuit 400 includes a drive stage circuit 410 and a negative voltage generating circuit 420. The drive stage circuit 410 is coupled between a first power supply terminal VCC and a second power supply terminal VEE, wherein the first power supply terminal VCC has a first voltage relative to the reference ground terminal GND, and the second power supply terminal VEE has a second voltage relative to the reference ground terminal GND. For simplicity, the first voltage will be referred to as the first voltage VCC and the second voltage as the second voltage VEE. The drive stage circuit 410 also has an output terminal OUT that provides an output signal VG, wherein the output signal VG has a first state and a second state. In one embodiment, when the output signal VG has the first state, that is, when the output signal VG has the first voltage VCC relative to the reference ground terminal GND, the power device M1 is turned on; when the output signal VG has the second state, that is, when the output signal VG has the second voltage VEE relative to the reference ground terminal GND, the power device M1 is turned off. In one embodiment, the power device M1 can be an N-type field-effect transistor, a P-type field-effect transistor, an insulated-gate bipolar transistor (IGBT), or a bipolar transistor. For example... Figure 4As shown, the drive stage circuit 410 may include a first transistor S1 and a second transistor S2 connected in series with a first power supply terminal VCC and a second power supply terminal VEE. When the first transistor S1 is turned on and the second transistor S2 is turned off, the output signal VG has a first voltage VCC relative to the reference ground terminal GND, and the power device M1 is turned on. When the second transistor S2 is turned on and the first transistor S1 is turned off, the output signal VG has a second voltage VEE relative to the reference ground terminal GND, and the power device M1 is turned on. In one embodiment, the drive circuit 400 further has an input terminal IN to receive a switch control signal PWM, and controls the turning on and off of the first transistor S1 and the second transistor S2 according to the switch control signal PWM. Figure 4 In the embodiment shown, when the switch control signal PWM is logic high, the output signal VG is the first voltage VCC, and the power device M1 is turned on. When the switch control signal PWM is logic low, the output signal VG is the second voltage VEE, and the power device M1 is turned off.

[0034] continue Figure 4 The description states that the negative voltage generating circuit 420 is coupled to the second power supply terminal VEE and provides a second voltage VEE. The negative voltage generating circuit 420 includes a first switching device M0, a bias circuit 422, and a control circuit 424. The first switching device M0 is coupled between the reference ground terminal GND and the second power supply terminal VEE. The bias circuit 422 provides a bias current IB to the first switching device M0. The control circuit 422 receives a feedback signal VFB characterizing the second voltage VEE and adjusts the second voltage VEE to a preset value VSET based on the feedback signal VFB and the reference signal VREF. VSET is set according to the turn-off characteristics of the power device M1. In one embodiment, the preset value VSET is -2V; in another embodiment, VSET is -5V.

[0035] exist Figure 4 In the embodiments, there are generally other inherent circuits between the first power supply terminal VCC and the reference ground terminal GND. These inherent circuits can be used as bias circuits. That is, the bias circuit that provides the bias current IB can be the inherent circuit of the drive circuit 400 between the first power supply terminal VCC and the reference ground terminal GND. Therefore, there is no need to add an additional bias circuit 422 to provide the bias current IB. Thus, the bias circuit 422 will not increase the additional power consumption of the drive circuit.

[0036] exist Figure 4In the driving circuit shown, the first switching device M0 can be operated in a constant current state, such as the saturation region of a MOSFET, to prevent the voltage across the first switching device M0, i.e., the second voltage VEE, from changing with the current flowing through the first switching device. When a change in the current second voltage VEE is detected, the control circuit 424 generates a current adjustment signal VR to the control terminal of the first switching device M0 to adjust the current flowing through the first switching device M0. Therefore, in Figure 4 In this circuit, the second voltage VEE can be stabilized near the preset value VSET by using the first switching device M0 with constant current characteristics and the control circuit 424.

[0037] Figure 5 Examples of embodiments according to this application are shown. Figure 4 More details of the drive circuit shown. Figure 5 and Figure 4 The same or similar reference numerals in the figures indicate the same devices.

[0038] The control circuit 524 in the drive circuit 500 may include a reference signal generation circuit 550 and an adjustment circuit 560. The reference signal generation circuit 550 generates a reference signal VREF. The first input terminal IN1 of the adjustment circuit 560 receives the reference signal VREF, and the second input terminal IN2 of the adjustment circuit 560 receives a feedback signal VFB characterizing the second voltage VEE. Based on the reference signal VREF and the feedback signal VFB, the adjustment circuit 560 outputs a current adjustment signal VR to the control terminal of the first switching device M0.

[0039] The control circuit 524 needs to control the current flowing through the first switching device M0 according to the change of the second voltage VEE. When the second voltage VEE changes, the feedback signal VFB changes accordingly, thus changing the current regulation signal VR. Under the control of the current regulation signal VR, the first switching device M0 adjusts the current flowing through it, so that the second voltage VEE stabilizes at a preset value VSET. In one embodiment, the preset value VSET is set by setting the value of the reference signal VREF. In one embodiment, the feedback signal VFB, representing the second voltage VEE, can be obtained by the feedback circuit 525 by acquiring the voltage difference between the reference ground terminal GND and the second power supply terminal VEE (i.e., the voltage difference across the capacitor CVEE). In one embodiment, the feedback circuit 525 can be a resistor voltage divider feedback circuit or a capacitor voltage divider circuit. In one embodiment, the preset value VSET is set by adjusting the value of the feedback voltage VFB. It should be noted that the feedback circuit can adopt other structures. This application does not limit the specific feedback circuit; any feedback circuit structure that can acquire the second voltage VEE in real time is included in this application.

[0040] exist Figure 5In the illustrated embodiment, the adjustment circuit 560 can be an operational amplifier, and one of the non-inverting and inverting inputs of the operational amplifier serves as one of the first input IN1 (receiving a reference signal) and the second input IN2 (receiving a feedback signal), and the other of the non-inverting and inverting inputs serves as the other of the first input IN1 (receiving a reference signal) and the second input IN2 (receiving a feedback signal).

[0041] The following describes the operation of control circuit 524 in stabilizing the second voltage VEE.

[0042] exist Figure 5 In this example, taking the first switching device M0 as an NMOS transistor operating in the constant current region, with the reference signal VREF provided to the inverting input of operational amplifier 560 and the feedback signal VFB provided to the non-inverting input of operational amplifier 560, when the feedback signal VFB of the second voltage VEE decreases, since both the feedback signal VFB and the reference signal VREF are relative to VEE, the absolute value of VEE decreases while its actual value increases, meaning the voltage difference across capacitor CVEE decreases. At this time, operational amplifier 560 compares the feedback signal VFB with the reference signal VREF, and the current adjustment signal VR decreases, thus the gate-source voltage Vgs of the first switching device M0 also decreases. As the gate-source voltage Vgs decreases, according to the volt-ampere characteristic curve of the NMOS transistor, the current flowing through the first switching device M0 also decreases, thereby causing the bias current IB of the bias circuit 522 to supply the second capacitor C. EE Charging causes the second capacitor C to... EE The voltage difference between the two ends increases, which in turn causes the absolute value of the second voltage VEE to increase and the actual value to decrease.

[0043] On the other hand, when the feedback signal VFB of the second voltage VEE increases (i.e., the absolute value of VEE increases while the actual value decreases) and the voltage difference across capacitor CVEE increases, the operational amplifier 560 compares the feedback signal VFB with the reference signal VREF, increasing the current adjustment signal VR. Consequently, the gate-source voltage Vgs of the first switching device M0 also increases. As the gate-source voltage Vgs increases, the current flowing through the first switching device M0 increases, causing the second capacitor C... EE The discharge current flows through the first switching device M0, causing the second capacitor C to... EE The voltage difference across the two ends decreases, which in turn causes the absolute value of the second voltage VEE to decrease and its actual value to increase.

[0044] This feedback control mechanism allows the second voltage VEE to remain stable at the preset value VSET.

[0045] Figure 5The reference signal generation circuit 550 generates a reference signal VREF. The reference signal generation circuit 550 may include a reference current generation circuit 552 and a first resistor 554. The reference current generation circuit 552 is connected to an internal power supply VDD and a first terminal of the first resistor 554, providing a reference current IREF to the first resistor 554. The second terminal of the first resistor 554 is connected to a second power supply terminal VEE, and the voltage at the first terminal of the first resistor 554 is the reference signal VREF. The reference current IREF passes through the first resistor 554 and generates a voltage drop across it, thereby generating a reference signal VREF relative to the second voltage VEE at the first terminal of the first resistor 554. The internal power supply VDD may be a power supply within the drive circuit, such as a switching power supply or a linear power supply that converts the first power supply voltage VCC, and has a power supply voltage less than or equal to the first power supply voltage VCC. In one embodiment, the internal power supply VDD = VEE + 5V, that is, the value of the internal power supply VDD is 5V greater than the second voltage.

[0046] exist Figure 5 In the illustrated embodiment, at least one of the reference current IREF or the resistance value of the first resistor can be preset by signals external to the drive circuit 500 (e.g., ext1 and ext2 shown), and remains constant after being set, thereby keeping the reference signal VREF constant to stabilize the second voltage VEE at the current preset value VSET until a new setting signal is received again. In other words, the second voltage VEE can be conveniently adjusted in real-time according to actual needs. This setting method can be achieved, for example, through the Serial Peripheral Interface (SPI) bus technology.

[0047] Therefore, in Figure 5 In this circuit, the control circuit 524 includes a simple reference signal generation circuit 510 and an adjustment circuit 560, thus the circuit is simple, occupies a small area, and is low in cost. Furthermore, the second voltage VEE provided by the drive circuit 500 can be instantly adjusted by changing the reference signal using an external signal.

[0048] Figure 6 Examples of embodiments according to this application are shown. Figure 5 The diagram shows waveforms of some key signals in the drive circuit 500. Note that... Figure 6 The waveforms of the various parameters shown are only for illustrating their changing trends, without specifying their specific values ​​or the relative values ​​between the parameters.

[0049] exist Figure 6 Before time t0, the drive circuit 500 is in a stable state. At time t0, transistor S1 of the drive stage circuit 510 is turned on, and the current I flowing through transistor S1... S1 The load capacitor C will be supplied between t0 and t1.LOAD (Not shown, usually refers to the input capacitor of power device M1) charging. After time t1, even though S1 remains on, the load capacitor C... LOAD The charging is complete, therefore the current I is... S1 The current I flowing through transistor S1 between t0 and t1 is 0. S1 Through load capacitor C LOAD Then they pass through the second capacitor C. EE The first switching device M0 reaches the second power supply terminal VEE, so the load capacitance C LOAD The voltage across the terminals is V OUT It will gradually increase until the load capacitance C is reached. LOAD Charging complete. From load capacitor C LOAD Outflowing current I LOAD Divided into flows through the second capacitor C EE Current I EE Part of the current I flowing through the first switching device M0 M0 Part (I) M0 It may also include a bias current IB, but the bias current IB is very small compared to the charging current. Because the current I... EE Give the second capacitor C EE Charging will therefore cause the second capacitor C to... EE The voltage difference across the two terminals increases rapidly, causing VEE to decrease rapidly accordingly. Additionally, the current I flowing through the first switching device M0... M0 Due to the current I LOAD The increase is due to the portion of the load current I. During this process, although VEE continuously decreases, even below the preset value VSET, the increase is due to the load current I... LOAD The dominant current can affect the feedback regulation process, so VEE failed to stabilize to the preset value VSET in time.

[0050] At time t1, the load capacitance C LOAD Once charging is complete, the current I flowing through transistor S1 will... S1 and load current I LOAD The load capacitance C is 0. LOAD The voltage across the terminals is V OUT It will stabilize. At this point, the second voltage VEE can be stabilized through feedback control, since VEE is lower than the preset value, i.e., the second capacitor C. EE The voltage difference between the two ends is too large, so it can be addressed by adjusting the second capacitor C. EE This is achieved by discharging the first switching device M0, thus flowing through the second capacitor C. EE Current I EECompared to the previous charging stage, the current direction is reversed and the current value gradually decreases. Correspondingly, the discharge current flows through the first switching device M0. The discharge current at this time is less than the previous load current I. LOAD Part of the current, therefore current I M0 The current is gradually decreased until it enters the constant current region and remains constant. This is achieved by adjusting the second capacitor C. EE During discharge, the second voltage VEE gradually rises toward the preset value and eventually remains near the preset value.

[0051] On the other hand, when it is necessary to turn off the external power device M1 at time t3, transistor S2 is turned on, and the second capacitor C... EE and load capacitance C LOAD The discharge occurs via transistor S2 until it is complete at time t4. During this process, the load capacitance C... LOAD The voltage across the terminals is V OUT It will gradually decrease until V OUT It is pulled down to the second voltage VEE. During this process, the second capacitor C EE Discharge occurs, therefore the second voltage VEE will be higher than the preset value VSET. According to the principle of feedback control, a portion of the bias current from the bias circuit needs to be discharged to the second capacitor C. EE The charging process aims to reduce the current VEE, but due to the very large discharge current, the second capacitor C is ultimately reduced after the discharge current is equalized. EE The discharge current flows almost completely through the load capacitor C. LOAD Because of transistor S2, the second voltage cannot be stabilized in time through the feedback control process, so the second voltage VEE will rise rapidly during t3-t4.

[0052] The second capacitor C at time t4 EE and load capacitance C LOAD After discharge is complete, the second voltage VEE can be stabilized through feedback control. Since VEE is higher than the preset value at this point, i.e., the second capacitor C... EE The voltage difference between the two ends is too small, therefore a feedback control process can be used to control the second capacitor C. EE Charging is performed so that the second voltage VEE gradually decreases to approach the preset value.

[0053] Figure 4 The described drive circuit 400 and Figure 5 The described drive circuit 500 can be used to reliably turn off power devices. In some embodiments of a multi-half-bridge drive module, each half-bridge includes two power switching devices, wherein each power switching device is driven by a drive circuit 400 or drive circuit 500.

[0054] Figure 7A A schematic diagram of a drive circuit 400 or 500 according to an embodiment of this application is shown for driving power switching devices on each arm of a three-phase bridge. Figure 7A In the middle, the first power supply terminals of all the lower bridge arm drive circuits, namely VCC2, VCC4 and VCC6, adopt an independent power supply structure (such as flyback power supply) just like the upper bridge arm. They are independent and not connected together.

[0055] Figure 7B A schematic diagram is shown illustrating a drive circuit 400 or 500 according to an embodiment of this application used to drive various power switching devices on the respective arms of a three-phase bridge. Figure 7B In the illustrated embodiment, to save costs, a single power supply is used. The first power supply terminals (VCC2, VCC4, and VCC6) of the drive circuits corresponding to the power switching devices (SW2, SW4, and SW6 as shown in Figure 7) on all lower arms of the three-phase bridge need to be connected together, and the second power supply terminals (VEE2, VEE4, and VEE56) of the drive circuits corresponding to all power switching devices (SW2, SW4, and SW6 as shown in Figure 7) on all lower arms need to be connected together. Figure 7B Although this connection method is relatively low-cost, the parasitic voltage caused by parasitic resistance at each reference ground terminal (GND2, GND4, and GND6) to the drive system ground VSS can result in different voltages across capacitors CEE2, CEE4, and CEE4. This can easily lead to an imbalance in the average current of each of the three half-bridges, causing one bridge to overheat and making it more prone to failure and safety risks.

[0056] To solve the above problems, Figure 7B The drive circuit 400 can be equipped with a temperature regulation function. Specifically, in some embodiments, the temperature at the first switching device M0 in each drive circuit 400 can be detected, and when the temperature of one or more first switching devices M0 is sufficiently high, measures can be taken to reduce the temperature, such as reducing the current flowing through these first switching devices M0. In this way, the temperature, and thus the average current, in each drive circuit can be made substantially balanced.

[0057] As mentioned earlier, the current flowing through the first switching device M0 is controlled by the current adjustment signal VR output by the regulating circuit based on the reference signal VREF and the feedback signal VFB. Therefore, the current of the first switching device M0 can be changed by altering the reference signal VREF. For example, if the temperature of a certain phase exceeds the temperature threshold TH1, adjusting the value of the reference signal VREF for that phase reduces the current flowing through the first switching device M0, diverting some of the current to other phases.

[0058] Therefore, in some embodiments, it is combined again. Figure 5 The reference signal generation circuit 550 in the drive circuit 500 (also the drive circuit 400 in Figure 7) maintains the value of the reference signal VREF when the temperature of the first switching device M0 is less than the temperature threshold TH; when the temperature of the first switching device M0 is greater than the temperature threshold TH, the value of the reference signal VREF increases or decreases to control the current flowing through the first switching device M0 to decrease. By reducing the current flowing through the first switching device M0, the temperature of the first switching device M0 can be prevented from rising further, and the temperature between each phase can be made relatively uniform, thus realizing the temperature regulation function.

[0059] In addition to Figure 7B Circuits using the same flyback power supply in the lower bridge arm also have the problem of transient current leakage. For example, when the first transistor S1 in the first phase drive circuit is turned on, the first transistor S1 will generate a large transient current in a short period of time. This large transient current can leak to other phases, and in severe cases, it can also cause the temperature of the first switching device M0 in other phases to rise.

[0060] To solve the above problems, Figure 7B The drive circuit 400 can be configured with a transient current regulation function. This transient current regulation function can be achieved by changing the reference signal VREF to control the current flowing through the first switching device M0 during the first time period after the first transistor S1 is turned on. For example, the reference signal generation circuit 550 in the drive circuit 500 increases or decreases the reference signal VREF during the first time period after the first transistor S1 is turned on, thereby increasing the current flowing through the first switching device M0.

[0061] Figure 7C Waveforms of various signals in a drive circuit with transient regulation are presented. Figure 6 The waveform shown is different in that... Figure 6 In this context, the value of the reference signal VREF remains unchanged after being set. Figure 7C During the first time period when the first transistor S1 is turned on, the reference signal VREF decreases to increase the current flowing through the first switching device M0. Figure 7C In this structure, the first time period during which the first transistor S1 is turned on is the ta-tb time period. During this first time period, the current flowing through the first switching device M0 increases, which allows the instantaneous current during the first time period to flow through the first switching device M0 as much as possible, preventing leakage through the ground line VSS to the first switching devices M0 of other phases. Therefore, the transient current regulation function of the multi-bridge arm structure is realized.

[0062] It should be noted that the temperature imbalance and transient current imbalance issues described above apply to the multiple drive circuits 400 or 500 used in the lower arm of a multiphase bridge. When drive circuits 400 or 500 are used only for single-phase power switching devices, since there is no current imbalance flowing through the first switching device M0 or transient current leakage to other phases, the aforementioned temperature regulation and transient current regulation functions can be disabled. In other words, an external enable signal can be used to control whether to activate the change of reference signal to achieve temperature regulation and / or transient current regulation.

[0063] It is important to note that when drive circuits 400 or 500 are used only for single-phase power switching devices, transient current regulation can be enabled to reduce the output ripple of the single-phase output voltage VEE. For example, in drive circuit 500, the reference signal generation circuit 550 increases or decreases the reference signal VREF during a first time period after the first transistor S1 is turned on, thereby increasing the current flowing through the first switching device M0. This first time period can be as described above. Figure 7C The ta-tb time period is described. Thus, during the first time period after the first transistor S1 is turned on, the current flowing through the first switching device M0 increases, which can reduce the output ripple of the second voltage VEE.

[0064] Figure 8A An example circuit diagram of a reference signal generation circuit according to an embodiment of this application is shown. Figure 8A In this circuit, the reference signal generation circuit 850 may include a first resistor 854 and a current generation circuit 852. A first terminal of the first resistor 854 is connected to a first input terminal IN1 of the adjustment circuit 860 to provide the reference signal VREF, and a second terminal is connected to a second power supply terminal VEE. The current generation circuit 852 is connected to the first terminal of the first resistor 854 and provides a reference current IREF. Since the reference signal VREF can be set according to a command signal, at least one of the resistance value of the first resistor 854 or the reference current IREF is variable. In one embodiment, the value of the reference current IREF is changed by a first command signal ext1, or the resistance value of the first resistor 854 is changed by a second command signal ext2.

[0065] When the drive circuit 400 or 500 has a temperature regulation function Figure 8A The reference signal generation circuit shown is still applicable. Its specific working principle is as follows: when the temperature value of the first switching device M0 is less than the temperature threshold TH, the reference current IREF remains unchanged, and the first reference signal VREF remains unchanged; when the temperature value of the first switching device M0 is greater than the temperature threshold TH, the reference current IREF changes with the temperature, and the reference signal VREF changes with the temperature.

[0066] When the drive circuit 400 or 500 has a transient current regulation function, it corresponds to... Figure 8A The reference signal generation circuit shown is still applicable. Its specific working principle is as follows: during the first time period when the first transistor S1 is turned on, the reference current IREF changes, and the corresponding reference voltage VREF changes.

[0067] Figure 8B A circuit diagram of a reference signal generation circuit according to another embodiment of the present invention is provided. Figure 8B In the reference signal generation circuit, a controlled current generation circuit 852 may be included. The controlled current generation circuit 852 generates different reference currents IREF according to the control signal. The reference current IREF flows through the first resistor 854 to obtain the reference signal VREF. Figure 8B The reference signal generation circuit shown can be applied to drive circuits with temperature regulation and / or transient current regulation functions.

[0068] Figure 8C A circuit diagram of a reference signal generation circuit according to yet another embodiment of the present invention is provided. Figure 8C In the illustrated embodiment, the drive circuit enables both temperature regulation and transient current regulation functions. The current generation circuit 850 may include a first controlled current generation circuit I1, a second controlled current generation circuit I2, and a third controlled current generation circuit I3.

[0069] The first controlled current generating circuit I1 is used to generate the first current IC1 (same as above). Figure 8A The reference current IREF is controlled by a signal from outside the drive circuit. It is used to set the current value of the reference current, and once set, the current value of the reference current remains constant to generate the reference signal VREF.

[0070] The second controlled current generating circuit I2 operates as follows: when the temperature of the first switching device M0 exceeds the temperature threshold TH, a temperature regulating current IC2 is provided to the reference signal generating circuit based on the temperature value. At this time, the temperature regulating current IC2 provided by the temperature regulating circuit can be superimposed with the reference current IREF, forming a voltage drop across the first resistor 854, thereby regulating the reference signal VREF and consequently regulating the current flowing through the first switching device M0. Depending on the type of the first switching device M0 and its connection to the first input terminal IN1 and the second input terminal IN2 of the regulating circuit, the directions of the temperature regulating current IC2 and the reference current IREF can be in the same direction or opposite directions. The control signal required by the second controlled current generating circuit I2 can come from outside the drive circuit or be generated based on the temperature value through the control logic inside the drive circuit. The second controlled current generating circuit I2 is integrated with the inherent over-temperature protection circuit of the drive circuit and can be associated with multiple temperature sensors.

[0071] The third controlled current generating circuit I3 operates by generating a transient regulating current IC3 during the first time period after the first transistor S1 is turned on. This transient regulating current IC3, combined with the reference current IREF and the optional temperature regulating current IC2, forms a voltage drop across the first resistor 854, thereby regulating the reference signal VREF and consequently the current flowing through the first switching device M0. The control signal required by the third controlled current generating circuit I2 can come from outside the drive circuit or be generated by the control logic within the drive circuit based on the switching control signal PWM of the first transistor S1 used in the drive stage circuit.

[0072] In summary, the driving circuit according to the embodiments of this application, since the circuit components included in the negative voltage generation circuit are all common components found in existing driving circuits, has a simple circuit design, low complexity, low cost, and reduced size, thus simplifying board-level design. Furthermore, the first switching device operates in the constant current region, and its inherent circuitry can be used as a bias circuit to provide bias current. The current of the first switching device is automatically adjusted through a feedback control mechanism, thereby regulating the negative bias voltage. Therefore, a relatively stable negative bias voltage can be provided with lower power consumption. Additionally, when this driving circuit is used to drive the power switching devices on the lower arm of a multiphase bridge, the average current flowing through the first switching device (or the temperature of the first switching device) across multiple phases can be balanced by changing the reference signal, and leakage of the driving current of this phase to other phases can be prevented. Optionally, a protection circuit can also be provided to more promptly detect abnormal conditions such as overvoltage, undervoltage, and overstability, improving the reliability of the driving circuit.

[0073] Unless otherwise defined, all terms used herein (including technical and scientific terms) shall have the same meaning as commonly understood by one of ordinary skill in the art to which this invention pertains. It should also be understood that terms such as those defined in a common dictionary shall be interpreted as having a meaning consistent with their meaning in the context of the relevant art, and not as being interpreted in an idealized or highly formalized sense, unless expressly defined herein.

[0074] In this disclosure, the terms "first" and "second," etc., are used only to distinguish various elements and do not indicate any order, importance, or priority.

[0075] Even if a particular combination of features is defined in the claims and / or disclosed in the specification, such combinations are not intended to limit the disclosure of aspects. In fact, many of these features can be combined in ways that are not defined in the claims and / or disclosed in the specification. Although each dependent claim listed below may directly depend on only one claim, the disclosure of aspects includes every dependent claim in combination with every other claim in the claim set.

[0076] The embodiments of this disclosure described above are merely illustrative and not restrictive. Those skilled in the art will understand that various modifications and combinations can be made to these embodiments or their features without departing from the principles and spirit of this disclosure, and such modifications should fall within the scope of this disclosure.

Claims

1. A driving circuit, comprising: The driver stage circuit is coupled between the first power supply terminal and the second power supply terminal. The driver stage circuit also has an output terminal that provides an output signal, wherein the output signal is a first voltage or a second voltage relative to the reference ground terminal. as well as A negative voltage generating circuit is coupled to a second power supply terminal and provides the second voltage. The negative voltage generating circuit includes: A first switching device is coupled between the reference ground terminal and the second power supply terminal; The bias circuit provides bias current to the first switching device; and The control circuit receives a feedback signal characterizing the second voltage and adjusts the second voltage to a preset value based on the feedback signal and a reference signal, wherein the second voltage is negative relative to the reference ground terminal.

2. The driving circuit according to claim 1, wherein the first switching device is an N-type field-effect transistor, a P-type field-effect transistor, an insulated-gate bipolar transistor, or a bipolar transistor.

3. The driving circuit according to claim 1, wherein the driving stage circuit includes a first transistor and a second transistor connected in series between the first power supply terminal and the second power supply terminal, wherein when the first transistor is turned on and the second transistor is turned off, the output signal is a first voltage relative to the reference ground terminal, and when the first transistor is turned off and the second transistor is turned on, the output signal is a second voltage relative to the reference ground terminal.

4. The driving circuit according to claim 3 further includes an input terminal for receiving a switch control signal and controlling the first transistor and the second transistor to turn on and off according to the switch control signal.

5. The driving circuit according to claim 1, wherein the preset value is set by setting the value of the reference signal or the value of the feedback signal.

6. The driving circuit according to claim 1, wherein the bias circuit is an inherent circuit of the driving circuit between the first power supply terminal and the reference ground terminal.

7. The driving circuit according to claim 1 further includes: A feedback circuit is coupled between the reference ground terminal and the second power supply terminal, and samples the second voltage to obtain a feedback signal characterizing the second voltage.

8. The driving circuit according to claim 1, wherein the control circuit comprises: The reference signal generation circuit provides a reference signal; as well as The regulating circuit has a first input terminal for receiving the reference signal, a second input terminal for receiving the feedback signal, and generates a current regulating signal to a first switching device based on the reference signal and the feedback signal to regulate the current flowing through the first switching device.

9. The driving circuit according to claim 8, wherein the value of the reference signal is fixed.

10. The driving circuit according to claim 8, wherein, The reference signal generation circuit includes: A first resistor, the first end of which is coupled to the first input terminal of the regulating circuit, and the second end of which is coupled to the second power supply terminal; and A reference current generating circuit is connected between the internal power supply and the first terminal of the first resistor to provide a reference current for the first resistor. The reference current value or the resistance value of the first resistor is set by an external signal and remains constant after being set.

11. The driving circuit according to claim 1, wherein during a first time period during which the driving stage circuit turns on the first power supply terminal and the reference ground terminal, the reference signal increases or decreases to increase the current flowing through the first switching device.

12. The driving circuit according to claim 11, wherein, The reference signal generation circuit includes: A first resistor, the first end of which is coupled to the first input terminal of the regulating circuit, and the second end of which is connected to the second power supply terminal; and A current generating circuit is coupled to the first end of the first resistor to provide a reference current for the first resistor; During the first time period when the driving stage circuit turns on the first power supply terminal and the reference ground terminal, the resistance value of the first resistor or the value of the reference current changes to change the reference signal.

13. A drive module for a multi-arm bridge, wherein each arm includes two power switching devices, wherein, Each power switching device corresponds to a drive circuit as described in any one of claims 1-12.

14. The driving module according to claim 13, wherein When the temperature of the first switching device is less than the temperature threshold, the value of the reference signal remains unchanged; and When the temperature value of the first switching device is greater than the temperature threshold, the value of the reference signal increases or decreases to reduce the current flowing through the first switching device.

15. The driving module according to claim 14, wherein, The reference signal generation circuit includes: A first resistor, the first end of which is coupled to the first input terminal of the regulating circuit, and the second end of which is coupled to the second power supply terminal; and A current generating circuit is coupled to the first end of the first resistor and provides a reference current for the first resistor; Wherein, at least one of the resistance value of the first resistor or the value of the reference current is variable.

Images