A small profile substrate integrated waveguide filter antenna for W-band

Abstract

This invention discloses a small, low-profile substrate-integrated waveguide filter antenna for the W-band, relating to antenna technology and addressing problems in existing technologies. A rectangular resonant cavity is formed by metal pillars penetrating from the upper copper layer to the bottom copper layer; one end of the rectangular resonant cavity is open as a signal input terminal; four circular holes are arranged in a row and column pattern within the rectangular resonant cavity on the upper copper layer, with a metal circular patch within each hole serving as the corresponding resonator; the four circular holes have equal diameters; the rectangular resonant cavity with the four circular holes is used to excite the TE12 mode; the TE12 mode further excites the TM11 mode on the four metal circular patches (resonators); two of the four resonators are set to diameter D1, and the other two are set to diameter D2; two radiation nulls are generated based on an impedance mismatch method. The advantages are that it achieves a fusion design of the antenna and filter, the entire filter antenna design is compact, it generates two radiation nulls, and the gain reaches 11 dBi.

Description

Technical Field

[0001] This invention relates to integrated antennas in the millimeter-wave band, and more particularly to a small, low-profile substrate-integrated waveguide filter antenna for the W band. Background Technology

[0002] With the increasing prevalence of wireless communication applications, frequency band resources are becoming increasingly crowded and scarce. Traditional 4G LTE and 5G NR bands offer insufficient bandwidth; even the 4.9GHz 5G NR band only achieves a maximum bandwidth of 300MHz per channel. Furthermore, interference exists between frequency bands, degrading the receiver's signal-to-noise ratio. Improving signal quality requires adding sufficient filtering components to the RF front-end, further complicating the design. For these reasons, millimeter-wave bands are being considered.

[0003] Starting from 30GHz, the millimeter-wave band is reached. Currently, radars increasingly used in unmanned driving systems operate in the E-band (71GHz~75GHz, 81GHz~85GHz). The W-band (75GHz~110GHz) also has broad applications, including satellite communication, missile homing, airborne detection radar, and foreign object detection (FOD) at airports. Of course, the extremely wide bandwidth of the W-band can also be used to transmit signals from fiber optics, Wi-Fi, macrocells, etc.

[0004] To ensure the communication quality of the aforementioned devices, designing filtering modules for millimeter-wave band communication equipment is necessary. However, due to factors such as high insertion loss, there are currently few implementations and applications of millimeter-wave band filters. Therefore, we consider integrating the filter and antenna into a single design, enabling the antenna to not only improve gain but also possess frequency selectivity.

[0005] Currently, a small number of designs also involve this type of filtered antenna. For example, patent number CN202610036699 uses a method of radiating patch stacking and introducing parasitic stubs to realize a Ka-band filtered antenna. It uses a stacking method of two dielectric layers and a metal layer to achieve a radiation band with a passband of 27 GHz to 31 GHz. Furthermore, parasitic stubs are designed in the first metal layer by metal slotting. These metal slots and stubs generate anti-phase currents at certain specific frequencies, do not radiate outwards, and generate two radiation nulls, namely 18 GHz and 20 GHz. The resulting stopband is 17.7 GHz to 21 GHz. This invention achieves a suppression level of 5 dBi in the passband and <-15 dBi in the stopband.

[0006] Patent number CN202510640503 utilizes a non-contact electromagnetic bandgap structure to achieve a filter antenna with a passband of 26 GHz to 32 GHz, a stopband of 22 GHz to 25 GHz, and a stopband of 33 GHz to 36 GHz. The highest in-band gain is approximately 10 dBi. Multiple resonant irises are placed within the waveguide cavity, forming multiple resonant cavities with the cavity structure. Simultaneously, a central perturbation pillar is introduced within some resonant cavities to increase additional coupling paths and generate transmission nulls. This filter antenna is manufactured using 3D printing technology.

[0007] Patent number CN202511114586 utilizes a defective ground structure and a metasurface to realize a filtered antenna with a passband of 27G~29.5G, a stopband of 24G~25.8G, and a stopband of 31.5G~32G, with a maximum in-band gain of 10dBi. It employs a Rogers R04450F PCB process with three metal layers and two dielectric layers. The first metal layer is the feed layer, and the second metal layer is the defective ground layer. Utilizing the defective ground structure as an equivalent LC resonant circuit, it exhibits inductive or capacitive behavior at specific frequencies, perturbing the current path and generating a high-sideband 30.7G radiation null. Furthermore, an "I"-shaped stub is introduced into the feed circuit to increase low-frequency reflection by extending the current path, thus introducing a radiation null near the low-sideband of 25.2G. The third metal layer is a set of 2x2 metal square patches used for signal radiation.

[0008] Patent number CN202511144589 describes a PCB process using three metal layers and two dielectric layers. It introduces a radiation null by adding a Π-shaped stub at the feed end and designs a 3x3 metasurface on the top metal layer. This invention achieves a passband of 3.26 GHz to 4.05 GHz, a maximum in-band gain of 10 dBi, and generates two radiation nulls outside the band at 2.8 GHz and 4.6 GHz, with a suppression depth >20 dBi.

[0009] Patent number WO 2024US61261 is a design scheme for a surface acoustic wave (SAW) filter antenna released by Qualcomm. It achieves a co-design of the SAW filter and antenna, using the antenna's resonant frequency as the first resonant frequency and the SAW filter's resonant frequency as the second resonant frequency, and coupling them together. This achieves wide in-band bandwidth and the presence of SAW filter nulls near the passband end out of band. Furthermore, this design can be extended to implement multi-frequency SAW filter antennas by connecting multiple SAW filters in parallel.

[0010] Patent CN201810973241 describes a magnetic pole filter antenna and its array. It is based on a substrate integrated waveguide design, featuring two metal radiating plates above a dielectric layer. These plates are surrounded on three sides by metal pillars to prevent signal leakage, with only one side being open-circuited, and a gap is maintained between the two plates. When a signal is fed into the first radiating plate, strong coupling causes the signal to couple to the second plate, where electrons are accelerated at the gap, radiating energy outwards. Furthermore, the strong coupling creates two radiation nulls at the two ends of the frequency band. This design achieves a frequency band of 4.4 GHz to 4.8 GHz.

[0011] Patent CN202122519482 describes a substrate-integrated waveguide filter antenna array. It incorporates bends in the waveguide, allowing the dominant mode to excite its degenerate modes, thus creating a bandwidth effect and providing out-of-band suppression. Its radiation mechanism primarily relies on microstrip line stubs. Open-circuit microstrip lines are placed at regular electrical intervals, allowing the signal to form a standing wave effect at the open-circuit stubs. Furthermore, a two-way power divider is implemented using microstrip lines, radiating the signal outwards through two radiating elements.

[0012] Patent number US15925180 describes an array antenna with filtering characteristics. Each group of array elements is placed half a wavelength apart. Each element consists of a substrate-integrated waveguide and a resonator, which is a notched metal radiating patch with specific frequency selectivity. This filtered antenna operates at 3.5 GHz.

[0013] A review of the literature on the aforementioned filtered antennas reveals that the methods employed generally focus on several aspects. One approach involves adding parasitic stubs at the signal feed end. By carefully controlling the physical length of these stubs, currents at specific frequencies can generate opposite phases on these stubs, thus creating radiation nulls. While this method can achieve radiation nulls on both sides of the antenna passband, it affects the in-band gain and impedance bandwidth. Furthermore, this method typically only introduces one null, not a suppression band. Another approach involves introducing a multi-cavity cascade at the signal feed end, or introducing a resonator capable of exciting higher-order modes. These resonators provide nulls, and the final resonator stage is used as the antenna to radiate the signal through etched slots. However, these methods result in significant insertion loss in the filter and lead to a less compact and larger design. In addition, the filtered antennas discussed in the published literature generally suffer from low implementation frequencies and insufficient in-band impedance bandwidth. With the increasing prevalence of millimeter-wave applications, improving the gain, increasing the impedance bandwidth, and enhancing the signal transmission stability of millimeter-wave devices are crucial. Summary of the Invention

[0014] The purpose of this invention is to provide a small, low-profile substrate-integrated waveguide filter antenna for the W-band, in order to solve the problems existing in the prior art.

[0015] The small, low-profile substrate-integrated waveguide filter antenna for the W-band described in this invention is composed of an upper copper layer, a dielectric layer, and a bottom copper layer stacked sequentially; and a rectangular resonant cavity is formed by metal pillars that penetrate and connect the upper copper layer to the bottom copper layer.

[0016] One end of the rectangular resonant cavity is open and serves as the signal input terminal; the upper copper layer has four circular holes arranged in a row and column pattern inside the rectangular resonant cavity, and the metal circular patch in each circular hole serves as the corresponding resonator. The four circular holes have the same diameter; a rectangular resonant cavity with four circular holes is used to excite the TE12 mode; the TE12 mode is then used to excite the TM11 mode on four metal circular patches; Furthermore, two of the four circular metal patches are set to diameter D1, and the other two are set to diameter D2; diameter D1 ≠ D2, based on the impedance mismatch method to generate two radiation null points.

[0017] This invention discloses a small, low-profile substrate integrated waveguide filter antenna for the W-band. Its advantages lie in its integrated antenna and filter design. The entire filter antenna device consists of only a rectangular SIW resonant cavity and four resonators with nested circular metal patches within it. There are no additional microstrip stubs or redundant resonant cavities for frequency selection, nor the bulky structure of cascaded multi-cavity systems for filtering. The filtering characteristics of this antenna utilize the principle of generating a radiation null point through impedance mismatch. Its radiation characteristics utilize the gap between the circular aperture and the circular metal patches on the integrated waveguide; when the gap is polarized, electrons gain additional energy and are accelerated, ultimately radiating outwards. The entire filter antenna design is compact, with core dimensions of only 5mm * 2.9mm * 0.7mm (length, width, height). In the 91GHz~98GHz frequency band, the highest gain reaches 11dBi. A radiation null was generated in the low-sideband and high-sideband respectively, with the low-sideband radiation null reaching -15 dBi and the high-sideband radiation null reaching -20 dBi. Attached Figure Description

[0018] Figure 1 This is a schematic diagram of the planar structure of the substrate-integrated waveguide filter antenna described in this invention.

[0019] Figure 2 This is a schematic diagram of the interlayer structure of the substrate-integrated waveguide filter antenna described in this invention.

[0020] Figure 3This is a return loss curve of the substrate-integrated waveguide filter antenna described in this invention.

[0021] Figure 4 This is a gain curve of the substrate-integrated waveguide filter antenna described in this invention.

[0022] Figure 5 This is the E-plane radiation pattern of the substrate-integrated waveguide filter antenna described in this invention.

[0023] Figure 6 This is the H-plane radiation pattern of the substrate-integrated waveguide filter antenna described in this invention.

[0024] Figure 7 This is the equivalent circuit diagram of the substrate-integrated waveguide filter antenna described in this invention.

[0025] Figure label: 1-Upper copper layer, 2-Metal pillar, 3-First circular hole, 4-Second circular hole, 5-Third circular hole, 6-Fourth circular hole, 7-First metal circular patch, 8-Second metal circular patch, 9-Third metal circular patch, 10-Fourth metal circular patch, 11-Signal input terminal, 12-Dielectric layer, 13-Bottom copper layer; W - Width of the rectangular resonant cavity, W1 - Row spacing of the center of the metal circular patch, L1 - Column spacing of the center of the metal circular patch, L2 - Distance from the center of the second column of metal circular patches to the terminal metal pillar of the rectangular resonant cavity, D1 - First diameter length, D2 - Second diameter length, H1 - Thickness of the upper copper layer, H2 - Thickness of the dielectric layer, H3 - Thickness of the bottom copper layer. Ind1 - Equivalent inductance of the first metal circular patch, C1 - Equivalent capacitance of the first metal circular patch; Ind2 - Equivalent inductance of the second metal circular patch, C2 - Equivalent capacitance of the second metal circular patch; Ind3 - Equivalent inductance of the third metal circular patch, C3 - Equivalent capacitance of the third metal circular patch; Ind4 - Equivalent inductance of the fourth metal circular patch, C4 - Equivalent capacitance of the fourth metal circular patch; Lcouple1 - the coupling inductance between the first and third metal circular patches; Ccouple1 - the coupling capacitance between the first and third metal circular patches; Lcouple2 - the coupling inductance between the second and fourth metal circular patches; Ccouple2 - the coupling capacitance between the second and fourth metal circular patches; TL1 - Equivalent microstrip line representing the distance between the two rows of circular metal patches; TL2 - Equivalent microstrip line representing the distance from the second row of circular metal patches to the terminal metal pillar of the rectangular resonator. Detailed Implementation

[0026] like Figure 1 and Figure 2 As shown, the main body of the small, low-profile substrate integrated waveguide filter antenna for the W-band described in this invention is a rectangular substrate integrated waveguide with core dimensions of only 5mm * 2.9mm * 0.7mm (length, width, height). The top and bottom layers are copper layers, with a dielectric layer in the middle, typically made of Rogers 5880 material. Except for one open-circuit side, the other three sides of the integrated waveguide are surrounded by metal pillars. The open-circuit side serves as the signal input terminal and is used to connect external test components during testing. The remaining three sides are secured with equally spaced metal pillars. The metal pillars extend from the top copper layer to the bottom copper layer, primarily to prevent signal leakage. The entire integrated waveguide is rectangular, forming a rectangular resonant cavity.

[0027] Specifically, four circular metal elements are etched away from the upper copper layer within the rectangular resonant cavity, creating four circular holes that expose the intermediate dielectric layer. These four removed circular holes are of equal diameter. A metal circular patch is placed within each of the four holes, serving as the corresponding resonator. The outer diameter of the metal circular patch (resonator) is smaller than the inner diameter of the corresponding circular hole, thus forming a slit for radiation. The four metal circular patches consist of two sizes, D1 and D2, with each pair of patches having equal diameters to generate two distinct zero-points of reflection. Without loss of generality, D1 is designed to be slightly larger, and D2 relatively smaller, but the diameters of these two sets of metal circular patches are not significantly different, ranging from approximately 1.1 to 1.3 times.

[0028] In one embodiment, the arrangement and corresponding diameters of the four metal circular patches are as follows: The diameter of the first circular metal patch located in the first row and first column is D2; The diameter of the second metal circular patch located in the first row and second column is D1; The diameter of the third metal circular patch located in the second row and first column is D1; The diameter of the fourth metal circular patch located in the second row and second column is D2; The direction from the signal input end of the rectangular resonant cavity to the terminal end is the row arrangement direction; The column arrangement direction is perpendicular to the row arrangement direction in the plane of the upper copper layer.

[0029] This staggered arrangement of two rows and two columns of different sizes is because if the diameters of the metal circular patches in the same column or row were the same, the radiation gap widths in the first and second columns, or the first and second rows, of the entire rectangular cavity would be inconsistent, resulting in asymmetrical longitudinal or lateral distortion of the radiation pattern. To solve this problem, two metal circular patches of the same diameter are placed diagonally opposite each other in the row and column arrangement. This arrangement ultimately makes the radiation pattern formed at the far end approximately symmetrical. The reason for choosing two different diameters instead of four identical diameters is as mentioned earlier: diameter D2 is used to determine one reflection null point of the antenna; while diameter D1 is used to determine the second reflection null point of the antenna. If the diameters of the four metal circular patches are set to be the same, the antenna can only obtain one reflection null point, and the broadband impedance effect cannot be formed.

[0030] In the same row, two circular metal patches are placed with their centers approximately one waveguide wavelength L1 apart. Two rows are placed approximately half a wavelength apart, W1. Those skilled in the art, based on the technical solution disclosed in this invention, can readily understand without inventive effort that the corresponding frequency band or other electrical characteristics can be adjusted by modifying the parameters in the antenna. For example, modifying the diameters D1 and D2 can obtain reflection nulls in other frequency bands; modifying the distance L1 can adjust the maximum gain of the antenna; modifying the distance W1 can adjust the bandwidth impedance of the desired frequency band; and modifying the distance L2 can adjust the required transmission null to change the filter response, etc.

[0031] The principle analysis of the small, low-profile substrate integrated waveguide filter antenna for the W-band described in this invention is as follows: The dominant mode resonant frequency of the rectangular resonant cavity formed by the substrate-integrated waveguide is determined by the following formula: ; Among them, f mn Here, the resonant frequencies are of orders m and n, C is the speed of light, u is the magnetic permeability constant, ε is the dielectric constant, m and n are the orders of the resonant modes, and L is the resonant frequency. ef and W ef It refers to the effective length and width corresponding to the mode excited by the rectangular resonant cavity.

[0032] The resonant frequencies of the four circular metal patches on the top layer of the substrate-integrated waveguide are determined by the following formula: ; Among them, f mn The resonant frequencies are of order m and n, where u is the magnetic permeability constant and ε is the resonant frequency. r It is the dielectric constant, r is the radius of the harmonic oscillator, and k is the dielectric constant. mn It is the root when the current is 0 in the m and n modes, and C is the speed of light.

[0033] After the signal enters the cavity from the port, the four circular holes generate a higher-order mode TE12 within the rectangular cavity. Simultaneously, this higher-order mode TE12 excites a TM11 mode on the corresponding metal circular patch. The zero-point reflection effect of D1 and D2 is as follows: Figure 3 As shown, adjusting D1 and D2 adjusts the position of the reflection null, ultimately ensuring it falls within the desired frequency band. Simultaneously, adjusting the vertical distance W1 between the two rows brings the two reflection nulls closer together, ultimately forming... Figure 3 The broadband impedance effect.

[0034] By adjusting the distance L2, the frequencies of the two radiation nulls can be adjusted. The impedance of the entire filter antenna, Z, is determined by the impedance of the four circular metal patches at resonance. mat The distance L2 represents the distance from the original matching impedance Z. mat Based on this, an impedance jZ0tgθ is superimposed. Here, Z0 represents the characteristic impedance, and θ represents the electrical length between the center of the second column of circular metal patches and the terminal metal post of the rectangular cavity. At this point, the input impedance Z seen from the signal input port into the rectangular cavity is... in Determined by the following formula: Z in =Z mat + jZ0tgθ.

[0035] When the electric length θ is equal to a positive integer multiple of Π / 2, Z in =∞, at this point, the entire antenna impedance is mismatched. The mode corresponding to the mismatched impedance forms a standing wave within the rectangular cavity, which cannot radiate outward through the gap between the metal circular patch and the rectangular cavity, thus creating a radiation null. The low-sideband radiation null is at 72 GHz with a suppression depth of 26 dB, and the high-sideband radiation null is at 109.2 GHz with a suppression depth of 30 dB. Due to the introduction of the radiation null, the antenna gain is below -5 dBi in the frequency range of 70 GHz to 73 GHz and in the frequency range of 107.5 GHz to 110 GHz. Therefore, a bandpass filter response is generated in the W-band of 70 GHz to 110 GHz.

[0036] The equivalent circuit of the filter antenna for the rectangular resonant cavity with four embedded metal circular patches is as follows: Figure 7 As shown. The equivalent LC circuit of each metal circular patch corresponds to Figure 7 The resonant circuit formed by capacitor C and inductor L. The electromagnetic coupling effect between the various circular metal patches is equivalent to... Figure 7 The capacitor Ccouple and the inductor Lcouple. Additionally, the distance L1 is equivalent to... Figure 7 The microstrip line TL1, and the distance L2 are equivalent to Figure 7 The microstrip line TL2.

[0037] The electrical characteristics of the antenna are as follows Figures 3 to 6 As shown. In Figure 3 In the model, two reflection nulls are located at approximately 92.5 GHz and 97 GHz, forming a wide -10 dB impedance bandwidth. The return loss is -11 dB at 91 GHz and -11 dB at 98 GHz. Figure 4 In the 91G range, the gain is 11.4 dBi, and the gain is 11.5 dBi for the 98G range. A radiation null is generated at the low sideband position of 72G, with a suppression level of approximately -15 dBi; another radiation null is generated at the high sideband position of 109.2G, with a suppression level of -22 dBi. Figure 5 This is the E-plane radiation pattern of the filtered antenna, where the gain is 11 dBi and the beamwidth is approximately ±16 degrees when Theta is 0 degrees. Figure 6 This is the H-plane radiation pattern of the filtered antenna, with a 3dB beamwidth of approximately ±25 degrees.

[0038] For those skilled in the art, various other corresponding changes and modifications can be made based on the technical solutions and concepts described above, and all such changes and modifications should fall within the protection scope of the claims of this invention.

Claims

1. A small, low-profile substrate-integrated waveguide filter antenna for the W-band, comprising an upper copper layer, a dielectric layer, and a bottom copper layer stacked sequentially; and a rectangular resonant cavity formed by metal pillars penetrating and connecting the upper copper layer to the bottom copper layer; Its features are, One end of the rectangular resonant cavity is open and serves as the signal input terminal; the upper copper layer has four circular holes arranged in a row and column pattern inside the rectangular resonant cavity, and the metal circular patch in each circular hole serves as the corresponding resonator. The four circular holes have the same diameter; a rectangular resonant cavity with four circular holes is used to excite the TE12 mode; the TE12 mode is then used to excite the TM11 mode on four metal circular patches; Furthermore, two of the four circular metal patches are set to diameter D1, and the other two are set to diameter D2; diameter D1 ≠ D2; A method based on impedance mismatch is used to generate two radiation nulls.

2. The small, low-profile substrate-integrated waveguide filter antenna for the W-band according to claim 1, characterized in that, Two metal circular patches of the same diameter are located at opposite corners of the matrix.

3. The small, low-profile substrate-integrated waveguide filter antenna for the W-band according to claim 2, characterized in that, The diameter of the first circular metal patch located in the first row and first column is D2; The diameter of the second metal circular patch located in the first row and second column is D1; The diameter of the third metal circular patch located in the second row and first column is D1; The diameter of the fourth metal circular patch located in the second row and second column is D2; The direction from the signal input end of the rectangular resonant cavity to the terminal end is the row arrangement direction; The column arrangement direction is perpendicular to the row arrangement direction in the plane of the upper copper layer.

4. The small, low-profile substrate-integrated waveguide filter antenna for the W-band according to claim 3, characterized in that, The diameter of the circular metal patch is D1 > D2.

5. The small, low-profile substrate-integrated waveguide filter antenna for the W-band according to claim 4, characterized in that, The diameter D1 of the metal circular patch is 1.1 to 1.3 times that of D2.

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