Resonant LLC switching power supply converter and zero-voltage switching control circuit thereof

Abstract

The utility model discloses a resonant LLC switching power supply converter and a zero-voltage switch control circuit thereof, and the converter comprises a first switch module, a second switch module, a third switch module, a fourth switch module, a first capacitor module, a second capacitor module, a third capacitor module, an inductor module, and an isolation module. The zero-voltage switch control circuit comprises a reference voltage module, a current detection module, a first control module and a second control module, and the reference voltage module, the current detection module, the first control module and the second control module work cooperatively. The switching loss of the first switching module and the second switching module of the resonant LLC switching power supply converter is dynamically controlled and greatly reduced.

Description

Technical Field

[0001] This utility model relates to the field of switching power supply technology, and in particular to a resonant LLC switching power supply converter and its zero-voltage switching control circuit. Background Technology

[0002] In existing technologies, resonant half-bridge LLC switching power converters are widely used in power conversion fields due to their high efficiency and good load adaptability. The bridge arm of this converter typically consists of a first MOSFET and a second MOSFET, which achieve current path switching in the circuit through complementary turn-on and turn-off. The main advantage of LLC converters lies in their use of the resonant principle, enabling soft turn-on during operation, thereby reducing switching losses and improving conversion efficiency. However, under high-power operating conditions, when the current in the circuit is large, the turn-off loss of the MOSFETs becomes more significant. Specifically, during the complementary turn-on or turn-off process of the first and second MOSFETs, sudden current changes lead to additional losses during switching. These losses mainly manifest as conduction losses and crossover losses generated by the MOSFETs during hard switching, especially at high currents, where these losses increase exponentially with the current, thus affecting the overall efficiency and thermal performance of the power converter. Furthermore, with excessively high currents, the switching speed and control waveform of the MOSFETs may also be affected, leading to significant electromagnetic interference (EMI) and additional heat in the circuit, thereby reducing the stability and reliability of the system. Therefore, effectively reducing switching losses and optimizing the switching characteristics of MOSFETs, especially under high current operating conditions, has become a significant technical challenge in current LLC switching power supply design. Utility Model Content

[0003] This utility model provides a resonant LLC switching power converter and its zero-voltage switching control circuit to solve the above-mentioned technical problems.

[0004] A first aspect of this utility model provides a zero-voltage switching control circuit for a resonant LLC switching power converter. The resonant LLC switching power converter includes a first switching module, a second switching module, an inductor module, a first capacitor module, and an isolation module. The first switching module receives a first control signal, and the second switching module receives a second control signal, such that the first switching module and the second switching module respectively form a resonant circuit with the inductor module, the first capacitor module, and the isolation module. The zero-voltage switching control circuit includes:

[0005] The third switch module and the second capacitor module are connected in series and then connected in parallel with the first switch module;

[0006] The fourth switch module and the third capacitor module are connected in series and then connected in parallel with the second switch module;

[0007] The reference voltage module is used to output a reference voltage.

[0008] The current detection module has a first input terminal that collects the primary or secondary current flowing through the transformer in the resonant LLC switching power converter, and a second input terminal that is connected to the output terminal of the reference voltage module. It is used to convert the primary or secondary current into a sampled voltage and compare the sampled voltage with the reference voltage to output a first comparison signal.

[0009] The first control module receives the first control signal at its first input terminal, connects its second input terminal to the output terminal of the reference voltage module, connects its third input terminal to the output terminal of the current detection module, and connects its output terminal to the control terminal of the third switch module. It outputs a first on / off control signal to the third switch module according to the first control signal, the reference voltage, and the first comparison signal.

[0010] The second control module receives the second control signal at its first input terminal, connects its second input terminal to the output terminal of the reference voltage module, connects its third input terminal to the output terminal of the current detection module, and connects its output terminal to the control terminal of the fourth switch module. Based on the second control signal, the reference voltage, and the first comparison signal, it outputs a second on / off control signal to the fourth switch module.

[0011] Optionally, the first control module includes:

[0012] The first delay trigger module receives the first control signal at its first input terminal, the reference voltage at its second input terminal, and the first comparison signal at its third input terminal, and is used to output a first weak current on / off control signal according to the first control signal, the reference voltage, and the first comparison signal.

[0013] The first driving module has its input terminal connected to the output terminal of the first delay trigger module, and is used to amplify and isolate the first weak current on / off control signal before outputting the first on / off control signal.

[0014] Optionally, the first delay triggering module includes:

[0015] The first comparison module receives the first control signal at its first input terminal and the reference voltage at its second input terminal. It is used to convert the first control signal into a first acquisition voltage after a delay and to output a first trigger signal when the first acquisition voltage is greater than the reference voltage.

[0016] The first trigger module has a first input terminal that receives the first comparison signal and a second input terminal that is connected to the output terminal of the first comparison module, and is used to output the first weak current on / off control signal according to the first trigger signal.

[0017] Optionally, the second control module includes:

[0018] The second delay trigger module receives the second control signal at its first input terminal, the reference voltage at its second input terminal, and the first comparison signal at its third input terminal, and is used to output a second weak current on / off control signal according to the second control signal, the reference voltage and the first comparison signal;

[0019] The second driving module has its input terminal connected to the output terminal of the second delay trigger module, and is used to amplify and isolate the output second weak current on / off control signal before outputting the second on / off control signal.

[0020] Optionally, the second delay trigger module includes:

[0021] The second comparison module receives the second control signal at its first input terminal and the reference voltage at its second input terminal. It is used to delay the second control signal into a second acquisition voltage and output a second trigger signal when the second acquisition voltage is greater than the reference voltage.

[0022] The second trigger module has a first input terminal that receives the first comparison signal and a second input terminal that is connected to the output terminal of the second comparison module, and is used to output the second weak current on / off control signal according to the second trigger signal.

[0023] Optionally, the current detection module includes a first resistor, a second resistor, a third resistor, a fourth resistor, a fifth resistor, a sixth resistor, a first capacitor, a second capacitor, a third capacitor, a first comparator, and a first diode;

[0024] One end of the first capacitor is the first input terminal of the current detection module. The other end of the first capacitor is connected to one end of the first resistor, one end of the second resistor, one end of the third resistor, and one end of the second capacitor. The other end of the first resistor is connected to one end of the fourth resistor and the non-inverting input terminal of the first comparator. The other end of the second resistor, the other end of the second capacitor, one end of the sixth resistor, the ground terminal of the first comparator, and one end of the third capacitor are all connected to ground. The other end of the third resistor and one end of the fifth resistor are connected to the second input terminal of the current detection module. The collector of the sixth transistor is connected to one end of the fifteenth resistor. The other end of the fifteenth resistor is connected to the other end of the sixth resistor and the inverting input terminal of the first comparator. The other end of the fourth resistor is connected to the cathode of the first diode. The anode of the first diode and the output terminal of the first comparator are connected to the output terminal of the current detection module. The power supply terminal of the first comparator and the other end of the third capacitor are connected together to receive the first power supply.

[0025] Optionally, the reference voltage module includes a seventh resistor, a fourth capacitor, and a three-terminal Zener diode. One end of the seventh resistor receives a first power supply, and the other end of the seventh resistor, one end of the fourth capacitor, the cathode of the three-terminal Zener diode, and the power supply terminal of the three-terminal Zener diode are all connected to the output terminal of the reference voltage module. The anode of the three-terminal Zener diode and the other end of the fourth capacitor are all connected to ground.

[0026] Optionally, the first comparison module includes an eighth resistor, a ninth resistor, a tenth resistor, a fifth capacitor, a sixth capacitor, a second diode, and a second comparator; the first trigger module includes a first trigger and a seventh capacitor.

[0027] One end of the fifth capacitor and one end of the ninth resistor are connected to ground. One end of the eighth resistor receives the first control signal. The other end of the eighth resistor is connected to the other end of the ninth resistor, the other end of the fifth capacitor, one end of the tenth resistor, and the non-inverting input of the second comparator. The inverting input of the second comparator receives the reference voltage. The power supply terminal of the second comparator and one end of the sixth capacitor are connected to the first power supply. The ground terminal of the second comparator and the other end of the sixth capacitor are connected to ground. The other end of the tenth resistor is connected to the cathode of the second diode. The anode of the second diode is connected to the output terminal of the second comparator and the C pin of the first flip-flop. The D pin of the first flip-flop receives the first comparison signal. The R pin, ground terminal, S pin of the first flip-flop, and one end of the seventh capacitor are connected to ground. The power supply terminal of the first flip-flop and the other end of the seventh capacitor are connected to the first power supply. The Q pin of the first flip-flop is the output terminal of the first trigger module.

[0028] The first driving module includes a third transistor, a fourth transistor, a fifth transistor, a sixth transistor, an eleventh resistor, a twelfth resistor, a thirteenth resistor, a fourteenth resistor, a fifteenth resistor, a sixteenth resistor, a seventeenth resistor, and an eighth capacitor. One end of the eleventh resistor is the input terminal of the first driving module. The other end of the eleventh resistor is connected to one end of the twelfth resistor and the base of the fifth transistor. The collector of the fifth transistor is connected to one end of the thirteenth resistor. The emitter of the fifth transistor and the other end of the twelfth resistor are both connected to ground. The other end of the thirteenth resistor is connected to one end of the fourteenth resistor and the sixth capacitor. The base of the transistor, the other end of the fourteenth resistor, the emitter of the sixth transistor, the collector of the third transistor, and one end of the eighth capacitor are all connected to receive the second power supply. The other end of the fifth resistor is connected to the base of the third transistor, the base of the fourth transistor, and one end of the sixteenth resistor. The emitter of the third transistor is connected to one end of the seventeenth resistor and the emitter of the fourth transistor. The other end of the seventeenth resistor is the first output terminal of the first driving module. The collector of the fourth transistor, the other end of the sixteenth resistor, and the other end of the eighth capacitor are all connected to form the second output terminal of the first driving module.

[0029] Optionally, the second comparison module includes an eighteenth resistor, a nineteenth resistor, a twentieth resistor, a ninth capacitor, a tenth capacitor, a third diode, and a third comparator; the second trigger module includes a second trigger and an eleventh capacitor.

[0030] One end of the ninth capacitor and one end of the nineteenth resistor are connected to ground. One end of the eighteenth resistor receives the second control signal. The other end of the eighteenth resistor is connected to the other end of the nineteenth resistor, the other end of the ninth capacitor, one end of the twentieth resistor, and the non-inverting input of the third comparator. The inverting input of the third comparator receives the reference voltage. The power supply terminal of the third comparator and one end of the tenth capacitor are connected to the first power supply. The ground terminal of the third comparator and the other end of the tenth capacitor are connected to ground. The other end of the twentieth resistor is connected to the cathode of the third diode. The anode of the third diode is connected to the output terminal of the third comparator and the C pin of the second flip-flop. The D pin of the second flip-flop receives the first comparison signal. The R pin, ground terminal, S pin of the second flip-flop, and one end of the eleventh capacitor are connected to ground. The power supply terminal of the second flip-flop and the other end of the eleventh capacitor are connected to the first power supply. The Q pin of the second flip-flop is the output terminal of the second trigger module.

[0031] The second driving module includes a first transistor, a second transistor, a twenty-first resistor, a twenty-second resistor, a twenty-third resistor, and a twelfth capacitor. One end of the twenty-first resistor is the input terminal of the second driving module. The other end of the twenty-first resistor is connected to the base of the first transistor, the base of the second transistor, and one end of the twenty-second resistor. The collector of the first transistor and one end of the twelfth capacitor are connected together to receive a first power supply. The emitter of the first transistor is connected to the emitter of the second transistor and one end of the twenty-third resistor. The other end of the twenty-third resistor is the first output terminal of the second driving module. The collector of the second transistor and the other end of the twelfth capacitor are connected together to form the second output terminal of the second driving module.

[0032] A second aspect of this utility model provides a resonant LLC switching power converter, including the zero-voltage switching control circuit described in the first aspect.

[0033] The technical advantages of this invention are as follows: Through the coordinated operation of the reference voltage module, current detection module, first control module, and second control module, dynamic control and significant reduction of switching losses of the first and second switching modules of the resonant LLC switching power converter are achieved. When the load condition changes, the first and second control modules can intelligently adjust the on / off states of the third and fourth switching modules to ensure zero-voltage turn-off or turn-on under high-current loads, while disconnecting unnecessary capacitor modules under low-current loads to reduce power loss. This design effectively avoids the efficiency degradation problem caused by increased hard-turn-off losses of the first and second switching modules under high load currents in the prior art, as well as the problem of increased hard-turn-on losses due to the intermittent pulse groups and skipped-cycle states of the control circuit output in standby and low-load conditions. The circuit of this invention significantly reduces the hard-turn-on and hard-turn-off losses of the first and second switching modules, maintaining low standby power consumption and high-efficiency operation, and significantly improving the overall efficiency and adaptability of the power converter. Attached Figure Description

[0034] To more clearly illustrate the technical solutions of the embodiments of this utility model, the drawings used in the description of the embodiments of this utility model will be briefly introduced below. Obviously, the drawings described below are only some embodiments of this utility model. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0035] Figure 1 This is a schematic diagram of the zero-voltage switching control circuit of a resonant LLC switching power converter provided in Embodiment 1 of this utility model;

[0036] Figure 2 This is a schematic diagram of the structure of the first control module in the zero-voltage switching control circuit of a resonant LLC switching power converter provided in Embodiment 1 of this utility model;

[0037] Figure 3 This is a schematic diagram of the structure of the first delay trigger module in the first control module of the zero-voltage switching control circuit in a resonant LLC switching power converter according to Embodiment 1 of this utility model;

[0038] Figure 4 This is a schematic diagram of the structure of the second control module in the zero-voltage switching control circuit of a resonant LLC switching power converter provided in Embodiment 1 of this utility model;

[0039] Figure 5This is a schematic diagram of the structure of the second delay trigger module in the second control module of the zero-voltage switching control circuit in a resonant LLC switching power converter provided in Embodiment 1 of this utility model;

[0040] Figure 6 This is a circuit diagram of a resonant LLC switching power converter provided in Embodiment 1 of this utility model;

[0041] Figure 7 This is a timing diagram of the transistor drive voltage waveform output by the first control module and the second control module of a resonant LLC switching power converter provided in Embodiment 1 of this utility model;

[0042] Figure 8 This is another structural schematic diagram of the zero-voltage switching control circuit in a resonant LLC switching power converter provided in Embodiment 1 of this utility model;

[0043] Figure 9 This is a circuit diagram of a zero-voltage switching control circuit in a resonant LLC switching power converter provided in Embodiment 1 of this utility model;

[0044] Figure 10 This is a time-axis comparison diagram of the driving voltage waveforms of the main switching transistors Q1 and Q2 and the current waveforms of the main components in a resonant LLC switching power converter provided in Embodiment 1 of this utility model.

[0045] Figure 11 This is a time-axis comparison diagram of the driving voltage waveforms of the main switching transistors Q1 and Q2 in a resonant LLC switching power converter provided in Embodiment 1 of this utility model with the voltage waveforms of the main components.

[0046] Figure 12 yes Figure 10 , Figure 11 The detailed unfolded diagram of the voltage and current waveforms of the four operating modes marked in the figure, with the same time axis comparison;

[0047] In the diagram: 101, First switch module; 102, Second switch module; 103, Third switch module; 104, Fourth switch module; 105, First capacitor module; 106, Second capacitor module; 107, Third capacitor module; 108, Energy storage and isolation module; 10, Main control module; 20, Zero-voltage switch control circuit; 201, Current detection module; 202, Reference voltage module; 203, First delay trigger module; 204, First drive module; 205, Second delay trigger module; 206, Second drive module; 211, First control module; 212, Second control module; 241, First comparison module; 242, First trigger module; 243, Second comparison module; 244, Second trigger module. Detailed Implementation

[0048] The technical solutions of the present utility model will be clearly and completely described below with reference to the accompanying drawings of the embodiments. Obviously, the described embodiments are only some embodiments of the present utility model, not all embodiments. Based on the embodiments of the present utility model, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present utility model.

[0049] It should be understood that this invention can be embodied in various forms and should not be construed as being limited to the embodiments set forth herein. Rather, providing these embodiments will make the disclosure thorough and complete, and will fully convey the scope of this invention to those skilled in the art. In the drawings, for clarity, the dimensions of layers and regions, as well as their relative dimensions, may be exaggerated. The same reference numerals denote the same elements throughout.

[0050] It should be understood that when an element or layer is referred to as "on," "adjacent to," "connected to," or "coupled to" other elements or layers, it may be directly on, adjacent to, connected to, or coupled to other elements or layers, or there may be intervening elements or layers. Conversely, when an element is referred to as "directly on," "directly adjacent to," "directly connected to," or "directly coupled to" other elements or layers, there are no intervening elements or layers. It should be understood that although the terms first, second, third, etc., may be used to describe various elements, components, areas, layers, and / or portions, these elements, components, areas, layers, and / or portions should not be limited by these terms. These terms are only used to distinguish one element, component, area, layer, or portion from another element, component, area, layer, or portion. Therefore, without departing from the teachings of this utility model, the first element, component, area, layer, or portion discussed below may be referred to as the second element, component, area, layer, or portion.

[0051] To fully understand this utility model, detailed structures and steps will be presented in the following description to illustrate the technical solution proposed by this utility model. Preferred embodiments of this utility model are described in detail below; however, in addition to these detailed descriptions, this utility model may have other embodiments.

[0052] Example 1

[0053] This embodiment provides a resonant LLC switching power converter, such as... Figure 1As shown, the resonant LLC switching power converter includes a first switching module 101, a second switching module 102, an inductor module 109, a first capacitor module 105, and an isolation module 108. The first switching module 101 receives a first control signal, and the second switching module 102 receives a second control signal, so that the first switching module 101 and the second switching module 102 form a resonant circuit with the inductor module, the first capacitor module 105, and the isolation module 108, respectively. The resonant LLC switching power converter also includes:

[0054] The third switch module 103 and the second capacitor module 106 are connected in series and then connected in parallel with the first switch module 101.

[0055] The fourth switch module 104 and the third capacitor module 107 are connected in series and then connected in parallel with the second switch module 102.

[0056] Zero-voltage switching control circuit 20 includes:

[0057] Reference voltage module 202 is used to output a reference voltage;

[0058] The current detection module 201 has a first input terminal that collects the primary or secondary current flowing through the isolation module 108 in the resonant LLC switching power converter, and its second input terminal is connected to the output terminal of the reference voltage module 202. It is used to convert the primary or secondary current into a collected voltage and compare the collected voltage with the reference voltage to output a first comparison signal.

[0059] The first control module 211 receives a first control signal at its first input terminal, connects its second input terminal to the output terminal of the reference voltage module 202, connects its third input terminal to the output terminal of the current detection module 201, and connects its output terminal to the control terminal of the third switch module 103. It outputs a first on / off control signal to the third switch module 103 according to the first control signal, the reference voltage, and the first comparison signal.

[0060] The second control module 212 receives a second control signal at its first input terminal, is connected to the output terminal of the reference voltage module 202 at its second input terminal, is connected to the output terminal of the current detection module 201 at its third input terminal, and is connected to the control terminal of the fourth switch module 104 at its output terminal. It outputs a second on / off control signal to the fourth switch module 104 according to the second control signal, the reference voltage, and the first comparison signal.

[0061] The first switching module 101 is controlled by a first control signal V1. When it receives an on control signal, the first switching module 101 turns on, allowing current to flow to the inductor module 109 and the isolation module 108. When it receives a off control signal, the first switching module 101 turns off, interrupting the current path. The second switching module 102 works in conjunction with the first switching module 101 to provide a switching signal of another phase, forming a complete bridge topology. The second switching module 102 is controlled by a second control signal V2. When the first switching module 101 is off, the second switching module 102 turns on to ensure current continuity. The third switching module 103 and the fourth switching module 104 are dynamically adjusted according to the output current state to achieve zero-voltage soft turn-on and soft turn-off, reducing losses and interference. The on and off states of the third switching module 103 and the fourth switching module 104 are strictly controlled by the load current and the main switch state to ensure no interference to the main circuit and maintain high-efficiency operation. When the circuit is turned on, the first capacitor module 105, together with the inductor module 109 and the isolation module 108, forms an oscillation circuit. The second capacitor module 106, as part of the resonant circuit, provides the resonant capacitance required by the circuit during the switching transition of the first switch module 101. It forms a resonant circuit with the inductor module 109, the isolation module 108, and the first switch module 101 to generate a resonant signal. During the resonance process, it stores and releases energy, reducing the turn-off loss of the first switch module 101. The third capacitor module 107, as part of the resonant circuit, provides the resonant capacitance required by the circuit during the switching transition of the second switch module 102. It forms a resonant circuit with the inductor module 109, the isolation module 108, and the second switch module 102 to generate a resonant signal. During the resonance process, it stores and releases energy, reducing the turn-off loss of the second switch module 102.

[0062] The reference voltage module 202 outputs a stable reference voltage, which serves as a reference value for the current detection module 201, the first control module 211, and the second control module 212, determining the criteria for current control and comparison in the resonant LLC switching power converter. The reference voltage module 202 generates and outputs a fixed reference voltage, which is provided to the current detection module 201 for comparison with the acquired voltage. It is also provided to the first and second control modules 211 to participate in generating the on / off control signal. The current detection module 201 acquires the output current signal of the resonant LLC switching power converter, converts the acquired output current into a corresponding acquired voltage, compares the acquired voltage with the reference voltage, and generates a first comparison signal as the basis for subsequent control. The current detection module 201 receives the output current signal from the second terminal of the first capacitor module 105, converts the output current signal into an acquired voltage through an internal conversion circuit, and compares the magnitude of the acquired voltage with the reference voltage in real time. When the acquired voltage exceeds the reference voltage, the first comparison signal is output, indicating whether the current current state meets the trigger condition. The first control module 211 generates a first on / off control signal based on a first control signal, a reference voltage, and a first comparison signal. This signal controls the on / off operation of the third switch module 103 and participates in the dynamic adjustment of the resonant circuit during the switching transition of the first switch module 101. The first control module 211 receives a first control signal (indicating the on / off logic start point), a reference voltage (as a reference value), and a first comparison signal (indicating the current current state). It determines whether the first control signal meets the trigger condition and performs logical processing on the reference voltage and the first comparison signal. Based on the comprehensive judgment, it generates the first on / off control signal. This signal is output to the control terminal of the third switch module 103, triggering the on / off operation of the third switch module 103, thus significantly reducing the switching losses of the first switch module 101 in different operating states. The second control module 212 generates a second on / off control signal based on a second control signal, a reference voltage, and a first comparison signal. This signal controls the on / off operation of the fourth switch module 104 and participates in the dynamic adjustment of the resonant circuit during the switching transition of the second switch module 102. The second control module 212 receives the second control signal, the reference voltage, and the first comparison signal, determines whether the second control signal meets the trigger condition, and performs logical processing on the reference voltage and the first comparison signal. Based on the comprehensive determination, a second on / off control signal is generated and output to the control terminal of the fourth switch module 104 to trigger the on / off action of the fourth switch module 104, thereby significantly reducing the switching losses of the first switch module 101 in different operating states.

[0063] The technical advantages of the solution provided in this embodiment are as follows: Through the coordinated operation of the reference voltage module, current detection module, first control module, and second control module, dynamic control and significant reduction of switching losses of the first and second switching modules of the resonant LLC switching power converter are achieved. When the load state changes, the first and second control modules can intelligently adjust the on / off states of the third and fourth switching modules to ensure zero-voltage turn-off or turn-on under high current loads, while disconnecting unnecessary capacitor modules under low current loads to reduce power loss. This design effectively avoids the efficiency reduction problem caused by increased hard turn-off losses of the first and second switching modules under high load currents in the prior art, as well as the problem of increased hard turn-on losses due to the intermittent pulse groups and skipped cycle states of the drive waveform output by the control circuit under standby and low load conditions. The circuit of this utility model significantly reduces the hard turn-on and hard turn-off losses of the first and second switching modules, maintains low standby power consumption and high efficiency operation, and significantly improves the overall efficiency and adaptability of the power converter.

[0064] As one implementation method, such as Figure 2 As shown, the first control module 211 includes:

[0065] The first delay trigger module 203 receives a first control signal at its first input terminal, a reference voltage at its second input terminal, and a first comparison signal at its third input terminal. It is used to output a first weak current on / off control signal based on the first control signal, the reference voltage, and the first comparison signal.

[0066] The first driving module 204 has its input terminal connected to the output terminal of the first delay trigger module 203, and is used to amplify and isolate the first weak current on / off control signal before outputting the first on / off control signal.

[0067] The first delay trigger module 203 delays the first control signal and, combined with the reference voltage and the first comparison signal, outputs a first weak current on / off control signal to achieve precise on / off control of the output weak current. The first delay trigger module 203 receives three input signals: the first control signal serves as the core control basis, indicating the starting point of the trigger logic; the reference voltage is compared with an internally generated signal to determine the trigger condition; the first comparison signal assists in determining whether the trigger condition is met; it delays the first control signal to generate a first acquisition voltage and monitors the relationship between the first acquisition voltage and the reference voltage in real time. When the first acquisition voltage exceeds the reference voltage and the first comparison signal meets the trigger condition, it outputs the first weak current on / off control signal to control the operation of subsequent circuit modules. If the condition is not met, no signal is output to ensure safe circuit operation. The first drive module 204 receives the first weak current on / off control signal output by the first delay trigger module 203, amplifies and electrically isolates the first weak current on / off control signal, and then outputs a first on / off control signal to drive subsequent circuits. The control signal is electrically isolated from the power circuit by means of an isolation circuit, so as to avoid interference from high voltage or high current to the control loop. The first on / off control signal is output to directly drive the on / off operation of the power device and complete the control objective.

[0068] The technical advantages of this implementation are as follows: Through the coordinated operation of the first delay trigger module and the first drive module, precise on / off control and safe driving of the third switch module are achieved. The first delay trigger module, based on delay processing and a multi-signal judgment mechanism, outputs a reliable weak current on / off control signal only when the triggering conditions are met, ensuring the accuracy and safety of the control. The first drive module amplifies and electrically isolates the weak current signal, converting it into a high current or high voltage signal capable of driving power devices, while avoiding interference from high voltage or large current to the control circuit. This design effectively improves the driving capability, anti-interference ability, and system stability of the control signal, achieving efficient and safe circuit operation.

[0069] As one implementation method, such as Figure 3 As shown, the first delay trigger module 203 includes:

[0070] The first comparison module 241 receives a first control signal at its first input terminal and a reference voltage at its second input terminal. It is used to delay the first control signal into a first acquisition voltage and output a first trigger signal when the first acquisition voltage is greater than the reference voltage.

[0071] The first trigger module 242 receives a first comparison signal at its first input terminal and its second input terminal is connected to the output terminal of the first comparison module 241, and is used to output a first weak current on / off control signal according to the first trigger signal.

[0072] The first comparison module 241 delays the first control signal to generate a first sampled voltage and monitors the relationship between the first sampled voltage and the reference voltage in real time. When the first sampled voltage exceeds the reference voltage, the first comparison module 241 outputs a first trigger signal. The first trigger module 242 detects the status of the first trigger signal and activates the trigger logic when a valid first trigger signal is received. According to the trigger logic, a first weak current on / off control signal is output, instructing subsequent circuit modules to perform weak current on / off control. When the trigger signal is invalid or does not meet the conditions, no control signal is output to ensure that the circuit is in a safe state.

[0073] The technical advantages of this embodiment are as follows: through the coordinated operation of the first comparison module and the first trigger module, delayed processing and precise trigger control of the first control signal are achieved. The first comparison module converts the first control signal into a first sampled voltage and monitors its magnitude relationship with the reference voltage in real time, outputting the first trigger signal only when the trigger condition is met. The first trigger module generates a first weak current on / off control signal based on the trigger signal state, ensuring that subsequent circuit modules perform weak current on / off operations at appropriate times. This design possesses technical advantages such as precise delay, rapid response, and high safety, effectively avoiding false triggering and ensuring the stability and reliability of the circuit.

[0074] As one implementation method, such as Figure 4 As shown, the second control module 212 includes:

[0075] The second delay trigger module 205 has a first input terminal receiving a second control signal, a second input terminal receiving a reference voltage, and a third input terminal receiving a first comparison signal. It is used to output a second weak current on / off control signal according to the second control signal, the reference voltage, and the first comparison signal.

[0076] The second drive module 206 has its input terminal connected to the output terminal of the second delay trigger module, and is used to amplify and isolate the output second weak current on / off control signal before outputting the second on / off control signal.

[0077] The main function of the second delay trigger module 205 is to output a second weak current on / off control signal at an appropriate time based on the input second control signal, reference voltage, and first comparison signal. The second control signal indicates the trigger logic, typically triggered when the level changes. The reference voltage is used as a reference voltage and compared with an internally generated voltage to determine the trigger condition. The first comparison signal assists in determining whether the trigger condition is met, delays the second control signal to generate a second acquisition voltage, and monitors the relationship between the second acquisition voltage and the reference voltage in real time. When the second acquisition voltage exceeds the reference voltage and the first comparison signal meets the trigger condition, the second delay trigger module 205 outputs the second weak current on / off control signal to control the operation of subsequent circuit modules. The main function of the second drive module 206 is to receive the second weak current on / off control signal output by the second delay trigger module 205, amplify and electrically isolate the signal, and output a second on / off control signal to drive subsequent circuits. The second drive module 206 outputs the current-amplified second on / off control signal to control the on / off operation of the fourth switch module 104.

[0078] The technical advantages of this implementation are as follows: Through the coordinated operation of the second delay trigger module and the second drive module, precise on / off control and safe driving of the fourth switch module are achieved. The second delay trigger module, based on delay processing and a multi-signal judgment mechanism, outputs a reliable weak current on / off control signal only when the triggering conditions are met, ensuring the accuracy and safety of the control. The second drive module amplifies and electrically isolates the weak current signal, converting it into a high current or high voltage signal capable of driving power devices, while avoiding interference from high voltage or large current to the control circuit. This design effectively improves the driving capability, anti-interference ability, and system stability of the control signal, achieving efficient and safe circuit operation.

[0079] As one implementation method, such as Figure 5 As shown, the second delay trigger module 205 includes:

[0080] The second comparison module 243 receives a second control signal at its first input terminal and a reference voltage at its second input terminal. It is used to convert the second control signal into a second acquisition voltage after a delay and to output a second trigger signal when the second acquisition voltage is greater than the reference voltage.

[0081] The second trigger module 244 has a first input terminal that receives a first comparison signal and a second input terminal that is connected to the output terminal of the second comparison module, and is used to output a second weak current on / off control signal according to the second trigger signal.

[0082] The main function of the second comparison module 243 is to compare the input second control signal with a reference voltage and generate a second acquisition voltage based on the comparison result, which is used to trigger subsequent control signal output. The second control signal, as the input signal controlled by the second switch module 102, indicates whether a delayed trigger is needed, typically triggered when the level changes. The reference voltage provides a reference voltage for comparison with the voltage converted from the second control signal. The second comparison module 243 converts the received second control signal into a second acquisition voltage through a delay process. This delay process ensures accurate system timing and avoids control instability due to rapid signal changes. The converted second acquisition voltage is compared with the reference voltage. If the second acquisition voltage is greater than the reference voltage, the system is considered to meet the trigger condition, and the module outputs a second trigger signal. If the condition is not met, subsequent operations are not triggered. The main function of the second trigger module 244 is to further output a second weak current on / off control signal for controlling the subsequent circuit based on the second trigger signal output by the second comparison module 243 and the first comparison signal, used to control the on / off state of the fourth switch module 104 in the subsequent circuit.

[0083] The technical advantages of this implementation are as follows: through the coordinated operation of the second comparison module and the second trigger module, delayed processing and precise trigger control of the second control signal are achieved. The second comparison module converts the second control signal into a second sampled voltage and monitors its relationship with the reference voltage in real time, outputting the second trigger signal only when the trigger condition is met; the second trigger module generates a second weak current on / off control signal based on the trigger signal state, ensuring that subsequent circuit modules perform weak current on / off operations at appropriate times. This design possesses technical advantages such as precise delay, rapid response, and high safety, effectively avoiding false triggering and ensuring the stability and reliability of the circuit.

[0084] In one implementation, the current detection module 201 includes a first resistor, a second resistor, a third resistor, a fourth resistor, a fifth resistor, a sixth resistor, a first capacitor, a second capacitor, a third capacitor, a first comparator, and a first diode;

[0085] One end of the first capacitor is the first input terminal of the current detection module 201. The other end of the first capacitor is connected to one end of the first resistor, one end of the second resistor, one end of the third resistor, and one end of the second capacitor. The other end of the first resistor is connected to one end of the fourth resistor and the non-inverting input terminal of the first comparator. The other end of the second resistor, the other end of the second capacitor, one end of the sixth resistor, the ground terminal of the first comparator, and one end of the third capacitor are all connected to ground. The other end of the third resistor and one end of the fifth resistor are connected to the second input terminal of the current detection module 201. The other end of the fifth resistor is connected to the other end of the sixth resistor and the inverting input terminal of the first comparator. The other end of the fourth resistor is connected to the cathode of the first diode. The anode of the first diode and the output terminal of the first comparator are connected to the output terminal of the current detection module 201. The power supply terminal of the first comparator and the other end of the third capacitor are connected together to receive the first power supply.

[0086] In one implementation, the reference voltage module 202 includes a seventh resistor, a fourth capacitor, and a three-terminal Zener diode. One end of the seventh resistor receives a first power supply, and the other end of the seventh resistor, one end of the fourth capacitor, the cathode of the three-terminal Zener diode, and the power supply terminal of the three-terminal Zener diode are all connected to the output terminal of the reference voltage module 202. The anode of the three-terminal Zener diode and the other end of the fourth capacitor are all connected to ground.

[0087] In one implementation, the first comparison module 241 includes an eighth resistor, a ninth resistor, a tenth resistor, a fifth capacitor, a sixth capacitor, a second diode, and a second comparator; the first trigger module 242 includes a first trigger and a seventh capacitor.

[0088] One end of the fifth capacitor and one end of the ninth resistor are connected to ground. One end of the eighth resistor receives the first control signal. The other end of the eighth resistor is connected to the other end of the ninth resistor, the other end of the fifth capacitor, one end of the tenth resistor, and the non-inverting input of the second comparator. The inverting input of the second comparator receives the reference voltage. The power supply terminal of the second comparator and one end of the sixth capacitor are connected to the first power supply. The ground terminal of the second comparator and the other end of the sixth capacitor are connected to ground. The other end of the tenth resistor is connected to the cathode of the second diode. The anode of the second diode is connected to the output terminal of the second comparator and the C pin of the first flip-flop. The D pin of the first flip-flop receives the first comparison signal. The R pin, ground terminal, S pin, and one end of the seventh capacitor of the first flip-flop are connected to ground. The power supply terminal of the first flip-flop and the other end of the seventh capacitor are connected to the first power supply. The Q pin of the first flip-flop is the output terminal of the first trigger module 242.

[0089] The first driving module 204 includes a third transistor, a fourth transistor, a fifth transistor, a sixth transistor, an eleventh resistor, a twelfth resistor, a thirteenth resistor, a fourteenth resistor, a fifteenth resistor, a sixteenth resistor, a seventeenth resistor, and an eighth capacitor. One end of the eleventh resistor is the input terminal of the first driving module 204. The other end of the eleventh resistor is connected to one end of the twelfth resistor and the base of the fifth transistor. The collector of the fifth transistor is connected to one end of the thirteenth resistor. The emitter of the fifth transistor and the other end of the twelfth resistor are both connected to ground. The other end of the thirteenth resistor is connected to one end of the fourteenth resistor and the base of the sixth transistor. The other end of the fourteenth resistor, the emitter of the sixth transistor, the collector of the third transistor, and one end of the eighth capacitor are connected together to receive the second power supply. The collector of the sixth transistor is connected to one end of the fifteenth resistor. The other end of the fifteenth resistor is connected to the base of the third transistor, the base of the fourth transistor, and one end of the sixteenth resistor. The emitter of the third transistor is connected to one end of the seventeenth resistor and the emitter of the fourth transistor. The other end of the seventeenth resistor is the first output terminal of the first driving module 204. The collector of the fourth transistor, the other end of the sixteenth resistor, and the other end of the eighth capacitor are connected together to form the second output terminal of the first driving module 204.

[0090] In one implementation, the second comparison module 243 includes an eighteenth resistor, a nineteenth resistor, a twentieth resistor, a ninth capacitor, a tenth capacitor, a third diode, and a third comparator; the second trigger module 244 includes a second trigger and an eleventh capacitor.

[0091] One end of the ninth capacitor and one end of the nineteenth resistor are connected to ground. One end of the eighteenth resistor receives the second control signal. The other end of the eighteenth resistor is connected to the other end of the nineteenth resistor, the other end of the ninth capacitor, one end of the twentieth resistor, and the non-inverting input of the third comparator. The inverting input of the third comparator receives the reference voltage. The power supply terminal of the third comparator and one end of the tenth capacitor are connected to the first power supply. The ground terminal of the third comparator and the other end of the tenth capacitor are connected to ground. The other end of the twentieth resistor is connected to the cathode of the third diode. The anode of the third diode is connected to the output of the third comparator and the C pin of the second flip-flop. The D pin of the second flip-flop receives the first comparison signal. The R pin, ground terminal, S pin of the second flip-flop, and one end of the eleventh capacitor are connected to ground. The power supply terminal of the second flip-flop and the other end of the eleventh capacitor are connected to the first power supply. The Q pin of the second flip-flop is the output of the second trigger module 244.

[0092] The second driving module 206 includes a first transistor, a second transistor, a twenty-first resistor, a twenty-second resistor, a twenty-third resistor, and a twelfth capacitor. One end of the twenty-first resistor is the input terminal of the second driving module 206. The other end of the twenty-first resistor is connected to the base of the first transistor, the base of the second transistor, and one end of the twenty-second resistor. The collector of the first transistor and one end of the twelfth capacitor are connected together to receive the first power supply. The emitter of the first transistor is connected to the emitter of the second transistor and one end of the twenty-third resistor. The other end of the twenty-third resistor is the first output terminal of the second driving module 206. The collector of the second transistor and the other end of the twelfth capacitor are connected together to form the second output terminal of the second driving module 206.

[0093] The embodiments of this utility model will be described in detail below through specific circuit structures:

[0094] like Figure 6 As shown, the first switch module 101 is the main switch transistor Q1, the second switch module 102 is the main switch transistor Q2, the third switch module 103 is the auxiliary switch transistor Q3, the fourth switch module 104 is the auxiliary switch transistor Q4, the first capacitor module 105 is the resonant capacitor C2, the second capacitor module 106 is the auxiliary capacitor C4, the third capacitor module 107 is the auxiliary capacitor C5, the energy storage isolation module includes an inductor L1 and a transformer T1, and the resonant LLC switching power converter also includes: resistors R1, R2, R3, R4, R5, R6, R7, R8, capacitors C1 and C3, diode D1, and diode D2.

[0095] One end of the auxiliary capacitor C4 is connected to the drain D of the auxiliary switching transistor Q3. The source S of the auxiliary switching transistor Q3, the source S of the main switching transistor Q1, the drain D of the main switching transistor Q2, the positive terminal of the auxiliary capacitor C5, one end of the resistor R2, one end of the resistor R6, one end of the inductor L1, the second output terminal of the main control module 10 (output drive signal VS-Q1), the second input terminal of the zero-voltage switching control circuit 20 (voltage sampling terminal of drive signal VS-Q1), the negative terminal VCC2-GND of the second auxiliary DC power supply, the second output terminal of the zero-voltage switching control circuit 20 (output drive signal VS-Q3), and the seventh input terminal of the zero-voltage switching control circuit 20 (second power supply negative terminal) are all connected together. The negative terminal of auxiliary capacitor C5 is connected to the drain (D) of auxiliary switching transistor Q4. The other end of resistor R2 is connected to the gate (G) of switching transistor Q1 and one end of resistor R1. The other end of resistor R6 is connected to the gate (G) of auxiliary switching transistor Q3 and one end of resistor R5. The other end of resistor R1 is connected to the first output terminal (output drive signal VG-Q1) of the main control module 10. The other end of resistor R5 is connected to the first output terminal (output drive signal VG-Q3) of the zero-voltage switching control circuit 20. The other end of inductor L1 is connected to one end of the primary coil of transformer T1. The other end of the primary coil of transformer T1, one end of resonant capacitor C2, the second input terminal (IS current sampling terminal) of the main control module 10, and the first input terminal (IS current sampling terminal) of the zero-voltage switching control circuit 20 are connected together. The other end of resonant capacitor C2, the source (S) of main switching transistor Q2, the source (S) of auxiliary switching transistor Q4, one end of resistor R4, one end of resistor R8, the negative terminal of input capacitor C1, and the input DC current are connected together. The negative terminal VI-GND, the negative terminal VCC1-GND of the first auxiliary DC power supply, the fourth output terminal (output drive signal VS-Q2) of the main control module 10, the fourth output terminal (output drive signal VS-Q4) of the zero-voltage switch control circuit 20, the fifth input terminal (first power supply negative terminal) of the main control module 10, the fourth input terminal (first power supply negative terminal) of the main control module 10, and the fifth input terminal (first power supply negative terminal) of the zero-voltage switch control circuit 20 are connected together at thirteen points; the other end of resistor R4 is connected to the gate G of the main switch transistor Q2 and one end of resistor R3, the other end of resistor R8 is connected to the gate G of the auxiliary switch transistor Q4 and one end of resistor R7; the other end of resistor R3 is connected together at three points: the third output terminal VG-Q2 of the main control module 10 and the third input terminal (VG-Q2 voltage sampling terminal) of the zero-voltage switch control circuit 20, and the other end of resistor R7 is connected to the third output terminal (output drive signal VG-Q4) of the zero-voltage switch control circuit 20. The first auxiliary DC power supply positive terminal VCC1, the third input terminal of the main control module 10 (the first auxiliary DC power supply positive terminal VCC1), and the fourth input terminal of the zero-voltage switch control circuit 20 (the first power supply positive terminal) are connected together.The positive terminal VCC2 of the second auxiliary DC power supply is connected to the sixth input terminal (positive terminal of the second power supply) of the zero-voltage switch control circuit 20. The first end of the secondary coil of transformer T1 is connected to the positive terminal of diode D1, the negative terminal of diode D1, the negative terminal of diode D2, the positive terminal of output capacitor C3, the positive terminal VO of the switching power converter output, and the fifth input terminal (positive terminal of output voltage sampling) of the main control module 10. The second end of the secondary coil of transformer T1 is connected to the positive terminal of diode D2, the negative terminal of output capacitor C3, the third end of the secondary coil of transformer T1 (coil center tap), the negative terminal VO-GND of the switching power converter output, and the sixth input terminal (negative terminal of output voltage sampling) of the main control module 10.

[0096] in, Figure 6 and Figure 7The signals in the diagram have the following meanings: U(GS-Q*) is the voltage difference waveform between the gate G and source S of the switching transistor Q*, where * indicates a numerical label; U(VS-Q1) is the voltage difference waveform between the source S circuit node A of the switching transistor Q1 (the signal received by this node is the drive signal VS-Q1) and VCC1-GND (0-point reference point); VG-Q1 is the first positive drive signal output by the main control module 10, connected to the gate G of the main switching transistor Q1 through resistor R1; VS-Q1 is the first negative drive signal output by the main control module 10, connected to the source S of the main switching transistor Q1, and has the same potential as the negative terminal VCC2-GND of the second auxiliary power supply; VG-Q2 is the second positive drive signal output by the main control module 10, connected to the gate G of the main switching transistor Q2 through resistor R3; VS-Q2 is the second negative drive signal output by the main control module 10, connected to the source S of the main switching transistor Q2, and has the same potential as the negative terminal VCC1-GND of the first auxiliary power supply. VG-Q3 is the third positive drive signal output by the zero-voltage switching control circuit 20, connected to the gate G of the auxiliary switching transistor Q3 through resistor R5; VS-Q3 is the third negative drive signal output by the zero-voltage switching control circuit 20, connected to the source S of the auxiliary switching transistor Q3, and its potential is the same as the first negative drive signal VS-Q1 and the second auxiliary power supply negative terminal VCC2-GND; VG-Q4 is the fourth positive drive signal output by the zero-voltage switching control circuit 20, connected to the gate G of the auxiliary switching transistor Q4 through resistor R7; VS-Q4 is the fourth negative drive signal output by the zero-voltage switching control circuit 20, connected to the source S of the auxiliary switching transistor Q4, and its potential is the same as the first auxiliary power supply negative terminal VCC1-GND; I(VO) is the current value flowing out of the output terminal VO of the switching power converter; VP is the output voltage signal waveform of the current detection module; IH represents a larger current value, at which time the drive pulse signal output by the switching power converter is a continuous pulse; IS represents a set current value. When the output current value I(VO) of the innovative resonant LLC switching power converter of this utility model is less than the set value IS, it operates in low current load state 1. At this time, the drive pulse signal output by the main control module 10 is a high-frequency continuous pulse or discontinuous pulse group. At this time, the voltage difference between the gate and source of the auxiliary switching transistors Q3 and Q4, U(GS-Q3) and U(GS-Q3), are both low level (close to 0V). When the output current value I(VO) is greater than the set value IS, it operates in high current load state 2. The drive pulse signal output by the main control module 10 is a low-frequency continuous pulse. At this time, the voltage difference between the gate and source of the auxiliary switching transistors Q3 and Q4, U(GS-Q3) and U(GS-Q3), are both high level (close to their auxiliary power supply voltage value).IL represents a smaller current value, at which point the drive pulse signal output by the main control module 10 is a discontinuous pulse group or a continuous pulse with a higher frequency. VH represents a high signal level. Ton is the high-level period of the drive signal voltage within one switching cycle T, i.e., the on-time period of the switching transistor within one cycle T. Toff is the low-level period of the drive signal voltage within one switching cycle T, i.e., the off-time period of the switching transistor within one cycle T. 180° is the phase difference between the waveforms of the first drive signal voltage difference U(GS-Q1) and the second drive signal voltage difference U(GS-Q2) output by the main control module 10.

[0097] like Figure 8 As shown, the zero-voltage switching control circuit 20 includes a current detection module 201, a reference voltage module 202, a first delay trigger module 203, a second delay trigger module 205, a first drive module 204, and a second drive module 206. The input and output signals of each module are as follows:

[0098] The input signals of the current detection module 201 include: the voltage waveform IS across the resonant capacitor C2 (the amplitude of the AC component in this voltage waveform is proportional to the maximum current flowing through the resonant capacitor C2) and the reference voltage V-REF. Both signals use the negative terminal VCC1-GND of the first auxiliary DC power supply as the reference zero point. Its output signal is the switching power converter output power status signal VP, also using the negative terminal VCC1-GND of the first auxiliary DC power supply as the reference zero point.

[0099] The output signal of the reference voltage module 202 includes: a reference voltage source V-REF generated by the positive terminal VCC1 of the first auxiliary DC power supply and regulated, with the negative terminal VCC1-GND of the first auxiliary DC power supply as the reference 0 point.

[0100] The input signals of the first delay trigger module 203 include: reference voltage V-REF, voltage difference between the source S circuit node A signal VS-Q1 of the main switching transistor Q1 and the negative terminal VCC1-GND of the first auxiliary DC power supply, and power status signal VP output by the current detection module 201. Both signal VS-Q1 and signal VP take the negative terminal VCC1-GND of the auxiliary DC power supply as the reference 0 point. Its output signal is the small current drive level signal V-Q3 of the auxiliary switching transistor Q3, which takes the negative terminal VCC1-GND of the first auxiliary DC power supply as the reference 0 point. Its working power supply positive terminal is VCC1 and its working power supply negative terminal is VCC1-GND.

[0101] The input signals of the second delay trigger module 205 include: a reference voltage V-REF, a drive voltage signal VG-Q2 for the main switching transistor Q2, and an output power status signal VP from the current detection module 201. Both signals VG-Q2 and VP use the negative terminal VCC1-GND of the first auxiliary DC power supply as their reference zero point. Its output signal is a small current drive level signal V-Q4 for the auxiliary switching transistor Q4, also using the negative terminal VCC1-GND of the first auxiliary DC power supply as its reference zero point. Its operating power supply positive terminal is VCC1, and its operating power supply negative terminal is VCC1-GND.

[0102] The input signal of the first driving module 204 is the small current driving level signal V-Q3 of the auxiliary switching transistor Q3, and its output signal is the large current driving voltage signal VG-Q3 of the auxiliary switching transistor Q3. The negative terminal VCC2-GND of the second auxiliary DC power supply (or circuit nodes VS-Q3, VS-Q1) is used as the 0 point reference point.

[0103] The second drive module 206 receives a low-current drive signal from the auxiliary switching transistor Q4. Its output signal is a high-current drive voltage signal VG-Q4 from the auxiliary switching transistor Q4, with the negative terminal of the first auxiliary DC power supply, VCC1-GND, as the zero-point reference.

[0104] Wherein, VCC1 is the positive terminal of the first auxiliary DC power supply; VCC1-GND is the negative terminal of the first auxiliary DC power supply; VCC2 is the positive terminal of the second auxiliary DC power supply; and VCC2-GND is the negative terminal of the second auxiliary DC power supply.

[0105] exist Figure 6 circuits and Figure 8 In the circuit block diagram shown, the positive terminal of the resonant capacitor C2 is connected to the first input terminal of the current detection module 201 and inputs the current signal IS. The output terminal of the current detection module 201 is connected to the first input terminal of the first delay trigger module 203 and the first input terminal of the second delay trigger module 205 at the same three points and outputs the voltage signal waveform VP.

[0106] The second input terminal of the first delay trigger module 203 is connected to the source S circuit node of the main switching transistor Q1, and receives the first negative drive signal VS-Q1; the third input terminal of the first delay trigger module 203, the third input terminal of the second delay trigger module 205, and the output terminal of the reference voltage module 202 are connected together and receive the reference voltage signal V-REF; the output terminal of the first delay trigger module 203 is connected to the first input terminal of the first drive module 204 and outputs a small current drive level signal V-Q3; the first output terminal of the first drive module 204 is connected to one end of resistor R5, and the other end of resistor R5 is connected to the gate G of the auxiliary switching transistor Q3, and outputs the third positive drive signal VG-Q3; the second output terminal of the first drive module 204 is connected to the source S of the auxiliary switching transistor Q3, and outputs the third negative drive signal VS-Q3; the positive terminal of the second auxiliary power supply is connected to the second input terminal of the first drive module 204, and outputs the power signal VCC2; the negative terminal of the second auxiliary power supply VCC2-GND is connected to the third input terminal of the first drive module 204.

[0107] The second input terminal of the second delay trigger module 205 is connected to the drive voltage output terminal of the main switching transistor Q2 of the main control module 10, and receives the second positive drive signal VG-Q2; the output terminal of the second delay trigger module 205 is connected to the first input terminal of the second drive module 206, and outputs a small current drive level signal V-Q4; the first output terminal of the second drive module 206 is connected to one end of the resistor R7, and outputs the fourth positive drive signal VG-Q4; the other end of the resistor R7 is connected to the gate G of the auxiliary switching transistor Q4; the second output terminal of the second drive module 206 is connected to the negative terminal of the input DC voltage VI-GND, the negative terminal of the auxiliary power supply VCC1-GND, and the source S of the auxiliary switching transistor Q4, and receives the fourth negative drive signal VS-Q4.

[0108] The positive terminal VG-Q1 of the first auxiliary power supply is connected to the second input terminal of the current detection module 201, the first input terminal of the reference voltage module 202, the fourth input terminal of the first delay trigger module 203, the fourth input terminal of the second delay trigger module 205, the fourth input terminal of the first drive module 204, and the second input terminal of the second drive module 206, and outputs a power signal.

[0109] The negative terminal VCC-GND of the first auxiliary power supply is connected to the third input terminal of the current detection module 201, the second input terminal of the reference voltage module 202, the fifth input terminal of the first delay trigger module 203, the fifth input terminal of the second delay trigger module 205, the fifth input terminal of the first drive module 204, and the third input terminal of the second drive module 206.

[0110] The component correspondence in the current detection module 201 is as follows: the first comparator is comparator IC201, the first resistor is resistor R201, the second resistor is resistor R202, the third resistor is resistor R203, the fourth resistor is resistor R204, the fifth resistor is resistor R205, the sixth resistor is resistor R206, the first capacitor is capacitor C201, the second capacitor is capacitor C202, and the third capacitor is capacitor C203. The connection method is as follows: one end of capacitor C201 is connected to the current signal IS; the other end of capacitor C201 is connected to one end of resistor R201, one end of resistor R202, one end of resistor R203, and one end of capacitor C202; the other end of resistor R201 is connected to one end of resistor R204 and the non-inverting input of comparator IC201; the other end of resistor R202 is connected to the negative terminal of the first auxiliary power supply VCC1-GND (reference 0 potential point); the other end of resistor R203 is connected to one end of resistor R205 and the reference voltage V-REF; the other end of resistor R205 is connected to one end of resistor R206 and the inverting input of comparator IC201; the other end of resistor R204 is connected to the cathode of diode D201; the anode of diode D201 is connected to the output of comparator IC201 and outputs the signal VP. The positive power supply terminal of comparator IC201 is connected to one end of capacitor C203 and is connected to the positive terminal of the first auxiliary power supply, and receives the power supply signal VCC1. The negative power supply terminal of comparator IC201 is connected to the other end of capacitor C203 and is connected to the negative terminal of the first auxiliary power supply VCC1-GND.

[0111] The component correspondence in the reference voltage module 202 is as follows: the seventh resistor is resistor R207, the three-terminal Zener diode IC202, and the fourth capacitor is capacitor C204. The connection method is as follows: one end of resistor R207 is connected to the positive terminal of the first auxiliary power supply to receive the power signal VCC1; the other end of resistor R207, the cathode of the three-terminal Zener diode IC202, and one end of capacitor C204 are connected together to output the reference voltage signal V-REF; the other end of capacitor C204 and the anode of IC202 are connected together to the negative terminal of the first auxiliary power supply VCC1-GND.

[0112] The component correspondence in the first delay trigger module 203 is as follows: the second comparator is comparator IC204, the first flip-flop is D flip-flop IC203, the eighth resistor is resistor R208, the ninth resistor is resistor R209, the tenth resistor is resistor R210, the fifth capacitor is capacitor C205, the sixth capacitor is capacitor C206, and the seventh capacitor is capacitor C207. The connection method is as follows: One end of resistor R208 receives the first negative drive signal VS-Q1. The other end of resistor R208 is connected to five points: resistor R209, resistor R210, one end of capacitor C205, and the non-inverting input of comparator IC204. The other ends of resistor R209 and capacitor C205, the reset input R and reset input S of trigger IC203, the negative power supply of IC203, one end of capacitor C206, one end of capacitor C207, and the negative power supply of comparator IC204 are connected to the first auxiliary power supply negative terminal VCC1-GND. The other end of resistor R210 is connected to diode D2. The cathode of 02, the anode of diode D202, the output of comparator IC204, and the clock input C of D flip-flop IC203 are connected together to receive signal V-T1. The data input D of D flip-flop IC203 is connected to the output of comparator IC201 in current detection module 201 to receive signal VP. The other ends of capacitors C206 and C207, as well as the positive power supply terminals of comparator IC204 and D flip-flop IC203, are connected together to the positive terminal of the first auxiliary power supply to receive power signal VCC1. The data output Q of D flip-flop IC203 is connected to the output small current drive level signal V-Q3.

[0113] The component correspondence in the first drive module 204 is as follows: the eleventh resistor is resistor R211, the twelfth resistor is resistor R212, the thirteenth resistor is resistor R213, the fourteenth resistor is resistor R214, the fifteenth resistor is resistor R215, the sixteenth resistor is resistor R216, the seventeenth resistor is resistor R217, the third transistor is NPN type switching transistor Q203, the fourth transistor is NPN type switching transistor Q205, the fifth transistor is PNP type switching transistor Q204, the sixth transistor is PNP type switching transistor Q206, and the eighth capacitor is capacitor C208. The connection method is as follows: One end of resistor R211 receives the small current drive level signal V-Q3; the other end of resistor R211 is connected to one end of resistor R212 and the base B of NPN switching transistor Q205; the emitter E of NPN switching transistor Q205 and the other end of resistor R211 are connected to the negative terminal VCC1-GND of the first auxiliary power supply; the collector C of NPN switching transistor Q205 is connected to one end of resistor R213; the other end of resistor R213 is connected to the base B of PNP switching transistor Q206 and one end of resistor R214; the other end of resistor R214, the emitter E of PNP switching transistor Q206, the collector C of NPN switching transistor Q203, and one end of capacitor C208 are connected together. The PNP transistor Q206 is connected to the positive terminal of the second auxiliary power supply, receiving the power signal VCC2. The collector C of the PNP transistor Q206 is connected to one end of the resistor R215. The other end of the resistor R215, the base B of the PNP transistor Q204, the base B of the NPN transistor Q203, and one end of the resistor R216 are connected together. The emitters E of the PNP transistor Q204 and the NPN transistor Q203 are connected to one end of the resistor R217. The other end of the resistor R217 is the output terminal, outputting the third positive drive signal VG-Q3. The collector C of the PNP transistor Q204 and the other end of the resistor R216 are connected to the negative terminal VCC2-GND of the second auxiliary power supply, and outputting the third negative drive signal VS-Q3.

[0114] The component correspondence in the second delay trigger module 205 is as follows: the third comparator is comparator IC205, the second flip-flop is D flip-flop IC206, the eighteenth resistor is resistor R218, the nineteenth resistor is resistor R219, the twentieth resistor is resistor R220, the ninth capacitor is capacitor C209, the tenth capacitor is capacitor C210, and the eleventh capacitor is capacitor C211. The connection method is as follows: One end of resistor R218 receives the second positive drive signal VG-Q2. The other end of resistor R218 is connected to one end of resistor R219, one end of resistor R220, one end of capacitor C209, and the non-inverting input of comparator IC205. The other end of resistor R219, the other end of capacitor C209, one end of capacitors C210 and C211, the negative power supply of comparator IC205, the reset input R and reset input S of D flip-flop IC206, and the negative power supply of IC206 are connected to the negative terminal of the first auxiliary power supply VCC1-GND. The other end of resistor R220 is connected to... The cathode of diode D203, the anode of diode D203, the output terminal of comparator IC205, and the clock input terminal C of D flip-flop IC206 are connected together to output signal V-T2. The data input terminal D of D flip-flop IC206 is connected to the output terminal of comparator IC201 in current detection module 201 to receive signal VP. The other ends of capacitors C210 and C211, the positive terminals of comparator IC205 and D flip-flop IC206 are connected together to the positive terminal of the first auxiliary power supply to receive power signal VCC1. The data output terminal Q of D flip-flop IC206 outputs a small current drive level signal V-Q4.

[0115] The component correspondence in the second drive module 206 is as follows: the twenty-first resistor is resistor R221, the twenty-second resistor is resistor R222, the twenty-third resistor is resistor R223, the first transistor is NPN type switching transistor Q201, the second transistor is PNP type switching transistor Q202, and the twelfth capacitor is capacitor C212. The connection method is as follows: One end of resistor R221 receives a small current drive level signal V-Q4. The other end of resistor R221, the base B of PNP switching transistor Q202, the base B of NPN switching transistor Q201, and one end of resistor R222 are connected together. The emitter E of NPN switching transistor Q201, the emitter E of PNP switching transistor Q202, and one end of resistor R223 are connected together. The other end of resistor R223 is connected to output the fourth positive drive signal VG-Q4. The collector C of NPN switching transistor Q201 is connected to the positive terminal of the first auxiliary power supply and receives the power supply signal VCC1. The collector C of PNP switching transistor Q202 and one end of resistor R216 are connected to the negative terminal VCC1-GND of the first auxiliary power supply and output the fourth negative drive signal VS-Q4.

[0116] In the innovative resonant LLC switching power converter of this invention, the main control module 10 outputs drive signals for two main switching transistors Q1 and Q2, with different reference 0 potentials for these two main drive signals. The first positive drive signal is VG-Q1, and the first negative drive signal is VS-Q1, both having the same potential as the negative terminal VCC2-GND of the second auxiliary DC power supply. The second positive drive signal is VG-Q2, and the second negative drive signal is VS-Q2, both having the same potential as the negative terminal VCC1-GND of the first auxiliary DC power supply.

[0117] In this invention, a novel zero-voltage switching control circuit 20 outputs drive signals for two auxiliary switching transistors Q3 and Q4. These two auxiliary drive signals have different reference zero potentials. The third positive drive signal is VG-Q3, and the third negative drive signal is VS-Q3, which has the same potential as the first negative drive signal VS-Q1 and the second auxiliary DC power supply negative terminal VCC2-GND. The fourth positive drive signal is VG-Q4, and the fourth negative drive signal is VS-Q4, which has the same potential as the second auxiliary DC power supply negative terminal VCC1-GND.

[0118] like Figure 7 As shown, the first drive signal waveform U(GS-Q1) (i.e., the voltage difference between the first positive drive signal VG-Q1 and the first negative drive signal VS-Q1) output by the main control module 10 and the second drive signal waveform U(GS-Q2) (i.e., the voltage difference between the second positive drive signal VG-Q2 and the second negative drive signal VS-Q2) have a phase difference of 180° within one switching cycle.

[0119] like Figure 6 , Figure 7 As shown, the third drive signal waveform U(GS-Q3) output by the zero-voltage switching control circuit 20 changes from low level to high level, and the auxiliary switch Q3 is triggered to conduct at the same time. Two necessary conditions are required:

[0120] First condition: The output current value I(VO) of the LLC power converter is a large current value IH, such as Figure 7 As shown, the current value IH must meet the condition: IH is greater than the set current value IS, that is... At this time, the drive signal waveforms U(GS-Q1) and U(GS-Q2) output by the main control module 10 are continuous square waves.

[0121] Second condition: The first drive signal waveform U(GS-Q1) is at a high level or the source drive signal VS-Q1 of the main switch transistor Q1 is at a high voltage (reference 0 potential is VCC1-GND). This condition is only used as the trigger condition for the start of the transition of U(GS-Q3) from low level to high level. When U(GS-Q1) remains at a high level, this second condition is no longer used as a basis for judgment.

[0122] Same as above, such as Figure 7 As shown, the fourth drive signal waveform U(GS-Q4) output by the zero-voltage switching control circuit 20 changes from low level to high level, and the auxiliary switch Q4 is triggered to conduct at the same time. Two necessary conditions are required:

[0123] First condition: The converter output current value I(VO) is a large current value IH, such as Figure 7 As shown, the current value IH must meet the condition: IH is greater than the set current value IS, that is... At this time, the drive signal waveforms U(GS-Q1) and U(GS-Q2) output by the main control module 10 are continuous square wave waveforms.

[0124] Second condition: The second drive signal waveform U(GS-Q2) is at a high level. This condition is only used as the trigger condition for the transition of U(GS-Q4) from low level to high level. When U(GS-Q4) remains at a high level, this second condition is no longer used as a basis for judgment.

[0125] As described above, when the power converter transitions from a low-current load state 1 to a high-current load state 2, auxiliary switching transistors Q3 and Q4 cannot be immediately turned on. At this time, the zero-voltage switching control circuit 20 still needs to determine the states of the main switching transistors Q1 and Q2, waiting for them to enter a stable on-state before immediately triggering the auxiliary switching transistors Q3 and Q4 to turn on. Once the auxiliary switching transistors Q3 and Q4 are triggered to turn on, they are locked in the on-state. Only when the following conditions are met to trigger the turn-off of the auxiliary switching transistors Q3 and Q4 can they switch to the off-state. Therefore, both auxiliary switching transistors Q3 and Q4 achieve highly efficient zero-voltage soft-turn-on functionality, and their turn-on does not interfere with the main circuit.

[0126] like Figure 7 As shown, the third drive signal waveform U(GS-Q3) output by the zero-voltage switching control circuit 20 changes from high level to low level, and the auxiliary switch Q3 is triggered to turn off, which requires two necessary conditions:

[0127] First condition: The output current value I(VO) of the power converter is a small current value IL, such as Figure 7As shown, the current value IL must meet the condition: IL is less than the set current value IS, that is... At this time, the drive signal waveforms U(GS-Q1) and U(GS-Q2) output by the main control module 10 are either discontinuous square wave groups (i.e., the time interval between repeated low-level occurrences in the square wave exceeds one square wave period) or continuous square waves with higher frequency.

[0128] Second condition: The first drive signal waveform U(GS-Q1) is at a high level or the source potential VS-Q1 of the switching transistor Q1 is at a high voltage (reference 0 potential is VCC1-GND). This condition is only used as the trigger condition for the start of the transition of U(GS-Q3) from high level to low level. When U(GS-Q1) remains at a low level, this second condition is no longer used as the basis for judgment.

[0129] Same as above, such as Figure 7 As shown, the fourth drive signal waveform U(GS-Q4) output by the zero-voltage switching control circuit 20 changes from high level to low level, and the auxiliary switching transistor Q4 is triggered to turn off, which requires two necessary conditions:

[0130] First condition: The output current value I(VO) of the power converter is a small current value IL, such as Figure 7 As shown, the current value IL must meet the condition: IL is less than the set current value IS, that is... At this time, the drive signal waveforms U(GS-Q1) and U(GS-Q2) output by the main control module 10 are either discontinuous square wave groups (i.e., the time interval between repeated low-level occurrences in the square wave exceeds one square wave period) or continuous square waves with higher frequency.

[0131] Second condition: The second drive signal waveform U(GS-Q2) is at a high level. This condition is only used as the trigger condition for the transition of U(GS-Q4) from high level to low level. When U(GS-Q4) remains at a low level, this second condition is no longer used as a basis for judgment.

[0132] As described above, when the power converter transitions from "high current load state 2" to "low current load state 1," the auxiliary switching transistors Q3 and Q4 cannot be immediately turned off. At this time, the zero-voltage switching control circuit 20 still needs to determine the states of the main switching transistors Q1 and Q2, waiting for them to enter a stable on-state before immediately triggering the auxiliary switching transistors Q3 and Q4 to turn off. Once the auxiliary switching transistors Q3 and Q4 are triggered to turn off, they are locked in the off state. Only when the aforementioned conditions for triggering the conduction of auxiliary switching transistors Q3 and Q4 are met can they switch to the on state. Therefore, both auxiliary switching transistors Q3 and Q4 achieve highly efficient zero-voltage soft-turn-off functionality, and the turn-off of auxiliary switching transistors Q3 and Q4 does not cause any interference to the main circuit.

[0133] The component symbols in the above schematic diagrams include the meanings of individual physical components and series or parallel combinations of physical components of the same nature.

[0134] Some components in the schematic can be selected using extreme values, for example:

[0135] a. The resistance values ​​of resistors R2, R4, R6, R8, R204, R206, R209, R210, R212, R214, R216, R219, R220, and R222 can be infinite, which is equivalent to an open circuit and can be omitted. The resistor symbols do not need to be drawn in the schematic diagram.

[0136] b. The resistance values ​​of resistors R1, R3, R5, R7, R201, R205, R208, R215, R217, R218, and R223 can be 0Ω. In this case, the symbol of the resistor can be replaced by a short circuit of the connecting wire.

[0137] c. Diodes D201, D202, and D203 can be removed, as they are equivalent to being short-circuited, and can be replaced by short-circuiting the connecting wires.

[0138] d. The capacitance values ​​of capacitors C203, C204, C205, C206, C207, C208, C209, C210, C211, and C212 can be 0F. In this case, the capacitors are equivalent to an open circuit, and the capacitor symbols do not need to be drawn in the schematic diagram.

[0139] This utility model presents an innovative resonant LLC switching power supply converter, such as... Figure 9 , Figure 11 , Figure 12 The working principle of the circuit shown is explained below:

[0140] As shown in Figure 9, this utility model is based on Figure 11 The principle analysis of a detailed implementation circuit of the zero-voltage switch control circuit 20 shown in the block diagram is as follows:

[0141] (1) The working principle of the current detection module 201 is explained as follows:

[0142] Since the effective value of the AC component in the voltage waveform across the resonant capacitor C2 of the resonant LLC switching power converter is proportional to the effective value of the current flowing through capacitor C2, the current value of the resonant cavity of the LLC switching power converter can be represented by detecting the magnitude of the effective value of the AC component of the voltage across capacitor C2, as shown in Formula 1:

[0143] When U(GS-Q1) is high and U(GS-Q2) is low:

[0144] Formula 1

[0145] When U(GS-Q1) is low and U(GS-Q2) is high:

[0146] Formula 1

[0147] In Formula 1, the symbols mean: C* is the capacitance value of capacitor C*; * is any positive integer or letter symbol. When the voltage U(GS-Q1) is high and the main switching transistor Q1 is turned on, the instantaneous value of the current flowing through the drain D and source S of the main switching transistor Q1 is taken as the positive direction, with the direction of the current flowing into the drain D as the positive direction. When the voltage U(GS-Q2) is high and the main switching transistor Q2 is turned on, the instantaneous value of the current flowing through the drain D and source S of the main switching transistor Q2 is taken as the positive direction, with the direction of the current flowing into the drain D as the positive direction. The instantaneous value of the current flowing through inductor L1 is defined as the direction of the current flowing out from one end of inductor L1 and into one end of the primary coil of transformer T1, which is taken as the positive direction. The instantaneous value of the current flowing through the primary coil of transformer T1 is defined as the positive direction of the current flowing out from one end of inductor L1 and into one end of the primary coil of transformer T1. The instantaneous value of the current flowing through capacitor C2 is defined as the direction of the current flowing out from one end of the primary coil of transformer T1 and into the positive voltage end of capacitor C2. Let be the rate of change of the instantaneous voltage across capacitor C2. An increase in voltage is a positive rate of change, and a decrease in voltage is a negative rate of change.

[0148] Applying Fourier analysis to Formula 1 above, the resonant cavity loop current... It can be approximated as a sine wave. Substituting this into Formula 1, we can obtain an approximate expression for the AC component waveform of the voltage across capacitor C2, as shown in Formula 2 below:

[0149] Formula 2

[0150] As shown in Formula 2, the maximum voltage amplitude of the AC component in the voltage waveform across capacitor C2 is... The maximum current amplitude of the current waveform in the resonant cavity circuit It is a fixed proportional relationship.

[0151] like Figure 9As shown, capacitors C201, C202, and resistor R202 in the current detection module 201 form an AC voltage divider circuit, wherein the capacitive reactances of capacitors C201 and C202 are respectively... , This voltage divider circuit increases the amplitude of the AC voltage generated across capacitor C202. The amplitude of the AC component in the voltage waveform across capacitor C2 It is a fixed proportional relationship.

[0152] like Figure 9 As shown, resistors R215 and R202 form a DC reference voltage divider circuit. Since resistor R202 is connected in parallel with capacitor C202, this DC voltage is the DC component of the voltage waveform across capacitor C202.

[0153] Therefore, as shown above, the voltage across capacitor C202 (resistor R202) is composed of the DC and AC components, and the resulting voltage is calculated as shown in Formula 3 below:

[0154] Formula 3

[0155] Meaning of symbols in Formula 2-3: For capacitor C201 at the operating switching frequency The capacitive reactance value; For capacitor C202 at the operating switching frequency The capacitive reactance value; This represents the operating switching frequency of the resonant LLC switching power converter. This represents the instantaneous value of the AC component of the voltage waveform across capacitor C2; This represents the maximum amplitude of the AC component of the voltage waveform across capacitor C2. This represents the maximum amplitude of the voltage waveform across capacitor C202. This represents the DC component voltage value of the voltage waveform across capacitor C202. The maximum voltage amplitude of the AC component of the voltage waveform across capacitor C202; This represents the maximum amplitude of the voltage waveform across capacitor C202. This refers to the current flowing through capacitor C2, which is the maximum amplitude of the resonant cavity current waveform of the resonant LLC switching power converter. This is the voltage difference between the positive terminal of the reference voltage V-REF and the reference zero-point potential voltage VCC1-GND.

[0156] The aforementioned current detection method, because it does not involve any current sensing element connected in series in the main resonant cavity circuit, does not significantly increase power consumption and is smaller in size, making PCB layout easier. This has led to its widespread application. Other existing current detection methods, such as direct current detection using series resistors and isolation detection using current transformers, can also be applied to the zero-voltage switching control circuit 20 of the innovative resonant LLC switching power converter of this invention, and will not be elaborated upon here.

[0157] The working principle of the reference voltage module 202 is explained as follows: The positive terminal VCC of the auxiliary power supply supplies power to the three-terminal regulator IC202 through resistor R207. The reference terminal R of the three-terminal regulator IC202 is connected to the cathode K. The anode A of the three-terminal regulator IC202 is connected to the negative terminal VCC1-GND (reference 0 potential) of the auxiliary power supply, so that the cathode K of the three-terminal regulator IC202 generates a regulated reference voltage V-REF relative to the reference 0 potential VCC1-GND.

[0158] The working principle of the first delay trigger module 203 is explained as follows: Figure 8 , Figure 9 As shown, the high-voltage input signal between node VS-Q1 and the 0-potential reference point VCC1-GND in the innovative resonant LLC switching power converter circuit of this utility model... After voltage division by resistors R208 and R209, a low-voltage waveform is formed proportionally, as calculated in Formula 4:

[0159] Formula 4

[0160] like Figure 7 and Figure 9 As shown, when the first drive signal waveform U(GS-Q1) output by the main control module 10 changes from low level to high level, the main switch transistor Q1 changes from the off state to the on state. By selecting appropriate resistance values ​​for resistors R208 and R209, the voltage across resistor R209 can be made to meet the following condition: At this time, the output voltage V-T1 of comparator IC204 changes from low level to high level. Trigger IC203 starts to trigger on the rising edge of the V-T1 voltage waveform, transmitting the signal VP level state input at its input terminal to the output terminal and outputting a small current drive level signal V-Q3.

[0161] When signal VP is low, the low-current drive level signal V-Q3 also goes low, indicating that the power converter is in low-current load state 1. When signal VP is high, the low-current drive level signal V-Q3 also goes high, indicating that the converter is in high-current load state 2. Furthermore, the current output state of the low-current drive level signal V-Q3 will be locked until the next rising edge of the V-T1 signal occurs.

[0162] When the first drive signal waveform U(GS-Q1) output by the main control module 10 changes from high level to low level, the main switch transistor Q1 changes from the on state to the off state. , At this time, the voltage V-T1 output by comparator IC204 changes from high level to low level, while trigger IC203 still locks the current output state of the small current drive level signal V-Q3.

[0163] Resistor R210 and diode D202 form the positive feedback hysteresis circuit of comparator IC204. When the output signal V-T1 of IC204 is high, current flows through the positive and negative terminals of diode D202 and across resistor R210 into resistor R209, causing the voltage across resistor R209 to... The increase in voltage level ensures that the comparator IC204 can maintain a stable high-level state for its output signal V-T1 even when subjected to interference.

[0164] The working principle of the first driving module 204 is explained as follows: Figure 6 , Figure 9 As shown, when the small current drive level signal V-Q3 changes from low to high, current flows out from the output pin of the trigger IC203 in the first delay trigger module 203, and then splits into two paths after passing through resistor R211. One path flows into the base B of the NPN switching transistor Q205, and then into the negative terminal VCC1-GND of the first auxiliary power supply through the collector C; the other path flows into the negative terminal VCC1-GND of the first auxiliary power supply through resistor R212. At this time, the NPN switching transistor Q205 changes from the off state to the on state.

[0165] As described above, due to the current amplification function of the NPN switching transistor Q205, a current larger than that flowing into its base B flows out from the positive terminal of the second auxiliary power supply and splits into two paths. One path flows into the emitter E of the PNP switching transistor Q206, and then through its base B into resistor R213; the other path flows through resistor R214 into resistor R213. This current flows through resistor R213 into the collector C of the NPN switching transistor Q205, and then through its emitter E into the negative terminal VCC1-GND of the first auxiliary power supply. At this time, the PNP switching transistor Q206 changes from the off state to the on state.

[0166] As described above, due to the current amplification function of the PNP switching transistor Q206, a current larger than that flowing into its base B flows out from the positive terminal VCC2 of the second auxiliary power supply, sequentially through the emitter E, collector C, and resistor R215 into the base B of the NPN switching transistor Q203. It then splits into two paths: one path flows through the emitter E into resistor R217, and the other path flows through resistor R216 into the negative terminal VCC2-GND of the second auxiliary power supply. At this time, the NPN switching transistor Q203 switches from the off state to the on state. Due to the current amplification function of the NPN switching transistor Q203, a current larger than that flowing into its base B flows out from the positive terminal VCC2 of the second auxiliary power supply, flowing through the collector C and emitter E into resistor R217. The third positive drive signal VG-Q3 is then output through resistor R217.

[0167] As described above, the third positive drive signal VG-Q3 flows through resistor R5 into the gate G of the NMOSFET auxiliary switching transistor Q3 and one end of resistor R6, then through the source S of the auxiliary switching transistor Q3 and the other end of resistor R6 to the common connection point between the output terminal VS-Q3 and the negative terminal VCC2-GND of the second auxiliary power supply. When the charging voltage of the parasitic capacitance between the gate G and the source S of Q3 exceeds the turn-on threshold voltage Vth of Q3, the auxiliary switching transistor Q3 switches from the off state to the on state. When the state transition is stable, since its input voltage V-Q3 remains at a high level, the voltage difference U(GS-Q3) between its output positive drive signal VG-Q3 and negative drive signal VS-Q3 will also remain at a high level, and the auxiliary switching transistor Q3 will remain in the on state.

[0168] like Figure 6 , Figure 9 As shown, when the low-current drive level signal V-Q3 transitions from high to low, the current flowing into the base B of the NPN switching transistor Q205 drops to 0A, at which point the NPN switching transistor Q205 transitions from the on state to the off state. Therefore, the current flowing through the base B of the PNP switching transistor Q206 drops to approximately 0A, and the PNP switching transistor Q206 also transitions from the on state to the off state. Consequently, the current flowing into the base B of the NPN switching transistor Q203 drops to approximately 0A, and the NPN switching transistor Q203 also transitions from the on state to the off state.

[0169] As shown above, since the NPN switching transistor Q203 is in the off state, the high-level drive voltage it provides between the gate G and source S of the auxiliary switching transistor Q3 through resistors R217 and R5 disappears. At this time, the charge stored in the parasitic capacitance between the gate G and source S of the auxiliary switching transistor Q3 flows out from its gate G and splits into two paths. One path flows back to its source S (i.e., circuit node VS-Q3) through resistor R6, and the other path flows sequentially through resistors R5 and R217 into the emitter E of the PNP switching transistor Q204, flows out from its base B, and then flows back to the source S of the auxiliary switching transistor Q3 (i.e., circuit node VS-Q3) through resistor R216. At this time, due to the current amplification function of the PNP switching transistor Q204, a larger current will flow from its collector C than from its base B and flow back to the source S of the auxiliary switching transistor Q3 (i.e., circuit node VS-Q3). When the discharge voltage of the parasitic capacitance between the gate (G) and source (S) of the auxiliary switching transistor Q3 is lower than the turn-on threshold voltage (Vth) of Q3, Q3 transitions from the on state to the off state. When the state transition stabilizes, since its input voltage V-Q3 remains low, the voltage difference U(GS-Q3) between its output positive drive signal VG-Q3 and negative drive signal VS-Q3 will also remain low, and Q3 will remain in the off state.

[0170] The working principle of the second delay trigger module 205 is explained as follows: The circuit structure and internal working principle of the second delay trigger module 205 are exactly the same as those of the first drive module 204. The input reference voltage signal V-REF and current status signal VP of the second delay trigger module 205 are the same as the corresponding input signals of the first drive module 204. The positive terminal VCC1 and the negative terminal VCC1-GND of the operating power supply of the second delay trigger module 205 are exactly the same as those of the first drive module 204.

[0171] The input voltage signal VG-Q2 of the second delay trigger module 205 is the positive terminal of the drive voltage signal of the main switching transistor Q2, and its circuit node is different from that of the corresponding input voltage signal VS-Q3 of the first drive module 204. The signal output terminal connection point VS-Q4 of the second delay trigger module 205 is different from that of the signal output terminal connection point V-Q3 of the first drive module 204.

[0172] As mentioned above, the working principle analysis process of the second delay trigger module 205 is similar to that of the first drive module 204, and will not be repeated here.

[0173] The working principle of the second drive module 206 is explained as follows: Figure 6 , Figure 9As shown, when the low-current drive level signal V-Q4 changes from low level to high level, the current flows out from the output terminal of the trigger IC206 in the second delay trigger module 205. After passing through resistor R221, the current is divided into two paths. One path flows into the base B of the NPN switching transistor Q201, and then into resistor R223 through its emitter E. The other path flows into the negative terminal VCC1-GND of the first auxiliary power supply through resistor R222. At this time, NPN switching transistor Q201 switches from the off state to the on state. Due to the current amplification function of NPN switching transistor Q201, a larger current than that flowing into its base B flows out from the positive terminal VCC1 of the first auxiliary power supply, through the collector C and emitter E of NPN switching transistor Q201, into resistor R223, through resistor R223 to the output terminal, then through resistor R7, into the gate G of auxiliary switching transistor Q4 and one end of resistor R8, and through the source S of auxiliary switching transistor Q4 and the other end of resistor R8 into the common connection point of the output terminal and the negative terminal VCC1-GND of the first auxiliary power supply. When the charging voltage of the parasitic capacitance between the gate G and the source S of auxiliary switching transistor Q4 exceeds the turn-on threshold voltage Vth of auxiliary switching transistor Q4, auxiliary switching transistor Q4 switches from the off state to the on state. When the state transition is stable, since its small current drive level signal V-Q4 is maintained at a high level, the voltage difference U(GS-Q4) between its output positive drive signal VG-Q4 and negative drive signal VS-Q4 will remain at a high level, and the auxiliary switching transistor Q4 will remain in the on state.

[0174] like Figure 6 , Figure 9 As shown, when the low-current drive level signal V-Q4 changes from high to low, the current flowing into the base B of the NPN switching transistor Q201 drops to 0A. At this time, the NPN switching transistor Q201 changes from the on state to the off state.

[0175] As shown above, since the NPN switching transistor Q201 is in the off state, the high-level drive voltage provided by it between the gate G and source S of the auxiliary switching transistor Q4 through resistors R223 and R7 disappears. At this time, the charge stored in the parasitic capacitance between the gate G and source S of the auxiliary switching transistor Q4 flows out from its gate G and is divided into two paths. One path flows back to its source S (i.e., circuit node VS-Q4) through resistor R8, and the other path flows sequentially through resistors R7 and R223 into the emitter E of the PNP switching transistor Q202, flows out from its base B, and then flows back to the source S of the auxiliary switching transistor Q4 (i.e., circuit node B, where the received signal is the drive signal VS-Q4). At this time, due to the current amplification function of the PNP switching transistor Q202, a larger current will flow from its collector C than from its base B and flow back to the source S of the auxiliary switching transistor Q4 (i.e., circuit node B). When the discharge voltage of the parasitic capacitance between the gate (G) and source (S) of the auxiliary switching transistor Q4 is lower than the turn-on threshold voltage (Vth) of Q4, Q4 transitions from the on state to the off state. When the state transition stabilizes, because its input voltage V-Q4 remains low, the voltage difference U(GS-Q4) between its output positive drive signal VG-Q4 and negative drive signal VS-Q4 remains low, and Q4 remains in the off state.

[0176] like Figure 7 , Figure 8 , Figure 9 As shown, the innovative resonant LLC switching power converter of this invention distinguishes between two operating states—low current load state 1 and high current load state—by detecting the magnitude of the resonant cavity current. In low current load state 1, the current detection module 201 outputs a low-level signal VP, while in high current load state 2, the current detection module 201 outputs a high-level signal VP. Each operating state has four operating modes.

[0177] The working process and principle analysis of the innovative resonant LLC switching power supply converter of this utility model under different load conditions and different operating modes are as follows:

[0178] (1) Low current load state 1:

[0179] like Figure 7 As shown, when the innovative resonant LLC switching power converter of this utility model switches from a high current load state 2 to a low current load state 1, the output signal VP of the current detection module 201 will switch from a high level to a low level.

[0180] At this point, based on the above analysis of the first driving module 204, the voltage difference U(GS-Q3) between the driving signal VG-Q3 output from the positive output terminal and the driving signal VS-Q3 output from the negative output terminal of the first driving module 204 will change from a high level to a low level, and the auxiliary switching transistor Q3 will change from the on state to the off state. When the state transition stabilizes, the voltage difference U(GS-Q3) will remain at a low level, and the auxiliary switching transistor Q3 will remain in the off state.

[0181] At this point, based on the above analysis of the second driving module 206, the voltage difference U(GS-Q4) between the driving signal VG-Q4 output from the positive output terminal and the driving signal VS-Q4 output from the negative output terminal of the second driving module 206 will change from a high level to a low level, and the auxiliary switching transistor Q4 will change from the on state to the off state. When the state transition stabilizes, the voltage difference U(GS-Q4) will remain at a low level, and Q4 will remain in the off state.

[0182] As described above, in low current load state 1, the auxiliary switching transistors Q3 and Q4 in the innovative resonant LLC switching power converter of this invention are both kept in the off state. At this time, its operating mode is the same as that of the existing LLC switching power converter, which will not be described again here.

[0183] (2) High current load condition 2:

[0184] like Figure 7 As shown, when the power converter switches from a low current load state 1 to a high current load state 2, the output signal VP of the current detection module 201 will switch from a low level to a high level.

[0185] At this point, based on the analysis of the first driving module 204 above, the voltage difference U(GS-Q3) between the third positive driving signal VG-Q3 and the third negative driving signal VS-Q3 of the first driving module 204 will change from low level to high level within a high-frequency switching cycle T, and the auxiliary switching transistor Q3 will change from the off state to the on state. When the switching state stabilizes, the voltage difference U(GS-Q3) will remain at a high level, and the auxiliary switching transistor Q3 will remain in the on state.

[0186] At this point, based on the analysis of the second driving module 206 above, the voltage difference U(GS-Q4) between the fourth positive driving signal VG-Q4 and the fourth negative driving signal VS-Q4 output by the second driving module 206 will change from low level to high level within a high-frequency switching cycle T, and the auxiliary switching transistor Q4 will change from the off state to the on state. When the switching state stabilizes, the voltage difference U(GS-Q4) will remain at a high level, and the auxiliary switching transistor Q4 will remain in the on state.

[0187] As described above, under high current load condition 2, both auxiliary switching transistors Q3 and Q4 in the innovative resonant LLC switching power converter of this invention remain in the on state. At this time, the power converter of this invention has the following four operating modes, the principles and processes of which are analyzed below:

[0188] Working Mode 1:

[0189] like Figure 7 As shown, during mode 1 of the high-current load state 2, the first output drive voltage U(GS-Q1) of the main control module 10 changes from low to high, while the second output drive voltage U(GS-Q2) remains low. The main switching transistor Q1 changes from off to on, and the main switching transistor Q2 turns off. The output drive voltages U(GS-Q3) and U(GS-Q4) of the zero-voltage switching control circuit 20 remain high, and the auxiliary switching transistors Q3 and Q4 remain on. As described in operating mode 4, the turn-on process of the main switching transistor Q1 is zero-voltage turn-on, therefore the turn-on power loss is close to 0W.

[0190] like Figure 7 As shown, since the on-resistance Rds(on) of auxiliary switching transistors Q3 and Q4 is extremely low, one end of the auxiliary capacitor C4 is effectively connected to the source S of the main switching transistor Q1, while the other end of the auxiliary capacitor C4 is directly connected to the drain D of the main switching transistor Q1. Therefore, the auxiliary capacitor C4 is effectively connected in parallel between the drain D and the source S of the main switching transistor Q1.

[0191] Similarly, one end of the auxiliary capacitor C5 is connected to the source S of the main switching transistor Q2, while the other end of the auxiliary capacitor C5 is directly connected to the drain D of the main switching transistor Q2. Therefore, the auxiliary capacitor C5 is equivalent to being connected in parallel between the drain D and the source S of the main switching transistor Q1.

[0192] At this time, current flows out from the positive terminal VI of the input DC voltage, through the drain D of the N-channel insulated-gate field-effect transistor Q1, the source S of the main switching transistor Q1, across inductor L1, across the primary coil of transformer T1, across capacitor C2, and into the negative terminal VI-GND of the input DC voltage. At this time, capacitor C2 charges; the energy stored in inductor L1 increases in the forward direction (defined here as current flowing from inductor L1 into transformer T1). At this time, the voltage across auxiliary capacitor C4... The voltage across the auxiliary capacitor C5 .

[0193] At this time, the output current from the positive terminal of the induced voltage of the secondary coil of transformer T1, after passing through the positive and negative terminals of diode D1, is divided into two paths. One path flows through the positive and negative terminals of capacitor C3 back to the negative terminal of the secondary coil voltage of T1. The other path flows through the positive terminal VO of the output DC voltage, the positive and negative terminals of the load, and the negative terminal VO-GND of the output DC voltage back to the negative terminal of the secondary coil voltage of T1. At this time, capacitor C3 is charging; diode D2 is turned off in reverse.

[0194] Working Mode 2:

[0195] like Figure 7 As shown, during mode 2 of the high-current load state 2, the first output drive voltage U(GS-Q1) of the main control module 10 changes from high level to low level in operating mode 1, while its second output drive voltage U(GS-Q2) remains low. The main switching transistor Q1 changes from the on state to the off state, and the main switching transistor Q2 remains off. The output drive voltages U(GS-Q3) and U(GS-Q4) of the zero-voltage switching control circuit 20 remain high, the auxiliary switching transistor Q3 remains on, and the auxiliary switching transistor Q4 remains on.

[0196] At this point, after the current flows out from the positive terminal VI of the input DC voltage, it splits into two paths. One path passes through the drain D of the main switching transistor Q1, the semiconductor channel resistance between the drain D and the source S of the main switching transistor Q1, and the source S of the main switching transistor Q1, before converging into the circuit node VS-Q1. The other path passes through the positive terminal of the auxiliary capacitor C4, the negative terminal of C4, the drain D of the auxiliary switching transistor Q3, and the source S of the auxiliary switching transistor Q3, before also converging into the circuit node VS-Q1. Then, it passes through the two ends of the inductor L1, the two ends of the primary coil of the transformer T1, and the positive terminal of the capacitor C2, before converging into the common connection point between the negative terminal of the capacitor C2 and the source S of the main switching transistor Q2, and finally flows back to the negative terminal VI-GND of the DC voltage.

[0197] As described above, when the current flows through the common connection node between the drain D of the main switching transistor Q1 and the positive terminal of the auxiliary capacitor C4, it splits into two branches. One branch flows through the semiconductor channel resistance between the drain D and source S of the main switching transistor Q1, and merges into the circuit node VS-Q1. This resistance value changes rapidly from near zero ohms when the main switching transistor Q1 is turned on (the transition period is generally several hundred nanoseconds) to several megaohms when the main switching transistor Q1 is turned off. The other branch of the current flows into the positive terminal of the auxiliary capacitor C4, through the negative terminal of the auxiliary capacitor C4, the drain D of the auxiliary switching transistor Q3, and the source S of the auxiliary switching transistor Q3, and also merges into the circuit node VS-Q1. At this time, the auxiliary capacitor C4 is charged, and its voltage across it increases from near 0V in mode 1 to... .

[0198] When the main switching transistor Q1 completes the turn-off process and the auxiliary capacitor C4 completes the charging process, since the current in the inductor L1 cannot change abruptly, the freewheeling current flows out from the positive terminal of the self-induced voltage of the inductor L1 (the positive direction current defined above), through the two ends of the primary coil of the transformer T1, the two ends of the capacitor C2, the source S of the main switching transistor Q2, the positive and negative terminals of the parasitic reverse body diode between the source S and the drain D of the main switching transistor Q2, and the drain D of the main switching transistor Q2, and flows back to the negative terminal of the self-induced voltage of the inductor L1.

[0199] At this time, the voltage between the source (S) and drain (D) of the main switching transistor Q2 is approximately -1V, and the main switching transistor Q2 meets the zero-voltage turn-on condition.

[0200] By selecting appropriate specifications for inductor L1 and transformer T1, the energy stored in inductor L1 is rapidly released, and the current in the resonant cavity increases. The current drops rapidly, and capacitor C2 continues to charge. When the resonant cavity current... The excitation current of transformer T1 is less than or equal to At this time, diode D1 quickly switches from the conducting state to the reverse turning-off state. At this time, diode D2 is still in the reverse turning-off state, and the secondary coil of transformer T1 also quickly stops charging capacitor C3, and capacitor C3 quickly switches from the charging state to the discharging state.

[0201] At this time, the current flows out from the positive terminal of capacitor C3, through the positive terminal VO of the output DC voltage, the positive and negative terminals of the load, the negative terminal VO-GND of the output DC voltage, and back to the negative terminal of capacitor C3.

[0202] As described above, the current flowing from VI, after passing through the common node of the drain D of the main switching transistor Q1 and the positive terminal of capacitor C4, splits into two branches. By selecting an appropriate capacitance for capacitor C4 and increasing the turn-off speed of the main switching transistor Q1, during the turn-off process of the main switching transistor Q1 in operating mode 2, one branch has a very small current value and a short duration through the semiconductor channel between the drain D and source S of the main switching transistor Q1, while the other branch has a larger current value and a longer duration through the auxiliary capacitor C4. This allows for near-zero current soft turn-off of the main switching transistor Q1, thereby significantly reducing the turn-off loss of the main switching transistor Q1 and further improving the efficiency of the switching power supply converter to meet the growing demands for energy conservation and carbon reduction in today's socio-economic environment.

[0203] Working Mode 3:

[0204] like Figure 7As shown, during mode 3 of the high-current load state 2, the first output drive voltage U(GS-Q1) of the main control module 10 remains at a low level, while the second output drive voltage U(GS-Q2) changes from a low level in mode 2 to a high level. The main switch transistor Q1 remains off, and the main switch transistor Q2 changes from off to on. The output drive voltages U(GS-Q3) and U(GS-Q4) of the zero-voltage switching control circuit 20 remain at a high level, and the auxiliary switch transistor Q3 and auxiliary switch transistor Q4 remain on. As described in mode 2, the turn-on process of the main switch transistor Q2 is zero-voltage turn-on, therefore the turn-on power loss is close to 0W.

[0205] By appropriately selecting the inductance value of inductor L1 and the magnetizing inductance value of the primary coil of transformer T1, the forward energy stored in inductor L1 can be rapidly released during the initial brief period of operating modes 2 and 3. At this time, current flows out from the positive terminal of capacitor C2, through the two ends of the primary coil of transformer T1, the two ends of inductor L1, the drain D of main switching transistor Q2, the source S of main switching transistor Q1, and back to the negative terminal of capacitor C2. At this time, capacitor C2 discharges; the energy stored in inductor L1 increases in the reverse direction (defined here as current flowing from transformer T1 into inductor L1 being reverse current).

[0206] At this time, the output current from the positive terminal of the induced voltage of the secondary coil of transformer T1, after passing through the positive and negative terminals of diode D2, is split into two paths. One path flows through the positive and negative terminals of capacitor C3 back to the negative terminal of the secondary coil voltage of T1. The other path flows through the positive terminal VO of the output DC voltage, the positive and negative terminals of the load, and the negative terminal VO-GND of the output DC voltage back to the negative terminal of the secondary coil voltage of T1. At this time, capacitor C3 is charging; diode D1 is reverse-biased and turned off.

[0207] Working Mode 4:

[0208] like Figure 7 As shown, during mode 4 of the high-current load state 2, the first output drive voltage U(GS-Q1) of the main control module 10 remains at a low level, while the second output drive voltage U(GS-Q2) changes from a high level during operating mode 3 to a low level. The main switch transistor Q1 remains off, and the main switch transistor Q2 changes from the on state to the off state. The output drive voltages U(GS-Q3) and U(GS-Q4) of the zero-voltage switching control circuit 20 remain at a high level, while the auxiliary switch transistor Q3 remains on, and the auxiliary switch transistor Q4 remains on.

[0209] At this point, the current flows out from the positive terminal of capacitor C2, through the two ends of the primary coil of transformer T1 and the two ends of inductor L1, and after passing through the common connection node of the drain D of main switching transistor Q2 and the positive terminal of auxiliary capacitor C5, it splits into two branches. One branch passes through the drain D of main switching transistor Q2, the semiconductor channel resistance between the drain D and the source S of main switching transistor Q2, and the source S of main switching transistor Q2, and flows into the negative DC voltage VI-GND. The other branch passes through the positive terminal of auxiliary capacitor C5, the negative terminal of auxiliary capacitor C5, the drain D of auxiliary switching transistor Q3, and the source S of auxiliary switching transistor Q3, and also flows into the negative DC voltage VI-GND.

[0210] As described above, when the current flows through the common connection point between the drain (D) of the main switching transistor Q2 and the positive terminal of the auxiliary capacitor C5, it splits into two branches. One branch flows through the semiconductor channel resistance between the drain (D) and source (S) of the main switching transistor Q2, and merges into the negative terminal of the DC voltage VI-GND. This resistance value changes rapidly from near zero ohms when the main switching transistor Q2 is turned on (the transition period is generally several hundred nanoseconds) to several megaohms when the main switching transistor Q2 is turned off. The other branch of the current flows into the positive terminal of the auxiliary capacitor C5, through the negative terminal of the auxiliary capacitor C5, the drain (D) of the auxiliary switching transistor Q4, and the source (S) of the auxiliary switching transistor Q4, and merges into the negative terminal of the input DC voltage VI-GND. At this time, the auxiliary capacitor C5 is charged, and the voltage across its terminals increases from near 0V in mode 3 to its maximum value. Simultaneously, auxiliary capacitor C4 discharges, and the voltage across its terminals changes from that in mode 3. The minimum discharge value is close to 0V.

[0211] As mentioned above, during the charging process of auxiliary capacitor C5, when the voltage of auxiliary capacitor C5... Greater than the voltage across capacitor C2 When the voltage across inductor L1 reverses, the stored energy in inductor L1 will be gradually released. The self-induced electromotive force of inductor L1 will continue to maintain the direction of the current (the reverse current as defined above), but its value will continue to decrease. The current flows out from the positive terminal of the self-induced voltage of inductor L1, through the common connection point of the source S of the main switching transistor Q1 and the negative terminal of the auxiliary capacitor C4, the positive and negative terminals of the parasitic reverse body diode between the source S and the drain D of the main switching transistor Q1, the common connection point of the drain D of the main switching transistor Q1 and the positive terminal of the auxiliary capacitor C4, the positive terminal of capacitor C1, the negative terminal of C1, the negative terminal of capacitor C2, the positive terminal of C2, and the two ends of the primary coil of transformer T1, before flowing into the negative terminal of the self-induced voltage of inductor L1.

[0212] At this time, the voltage between the source S and drain D of the main switching transistor Q1 is about -1V. The main switching transistor Q1 meets the zero-voltage turn-on condition and will achieve zero-voltage soft turn-on in the working mode 1 of the next cycle.

[0213] At this moment, the energy stored in inductor L1 is rapidly released, capacitor C2 continues to discharge, and diode D2 quickly switches from the conducting state to the reverse off state. Meanwhile, diode D1 remains in the reverse off state, and the secondary coil of transformer T1 quickly stops charging capacitor C3, which then rapidly switches from the charging state to the discharging state.

[0214] At this time, the current flows out from the positive terminal of capacitor C3, through the positive terminal VO of the output DC voltage, the positive and negative terminals of the load, the negative terminal VO-GND of the output DC voltage, and back to the negative terminal of capacitor C3.

[0215] As described above, in the innovative resonant LLC switching power converter of this utility model, such as Figure 3 As shown in the diagram and the above operating mode analysis, in modes 2 and 4 under high current load condition 2, due to the current bypassing effect of auxiliary capacitors C4 and C5, the main switching transistors Q1 and Q2 achieve extremely low current soft turn-off during the turn-off process. Therefore, this invention further reduces the power loss of the LLC switching power converter and further improves the conversion efficiency using an innovative method.

[0216] In summary, the innovative resonant LLC switching power converter of this invention meets the economic and social development direction of energy saving, consumption reduction, and low carbon emissions, and has broad application prospects.

[0217] ④ An example of an implementation circuit for the innovative resonant LLC switching power supply converter of this utility model is as follows: Figure 6 As shown, the parameters of each component in the circuit in this example are selected as follows:

[0218] Resistance: R1=R3=R5=R7=10Ω, R2=R4=R6=R8=10KΩ;

[0219] Main switching transistors Q1 and Q2: N-channel MOSFET SPP20N60C3 (20A, 600V);

[0220] Auxiliary switching transistors Q1 and Q2: N-channel MOSFET SPP02N60C3 (2A, 600V);

[0221] Inductance L1 = 235uH; Capacitor C2 = 15nH; Magnetizing inductance LP1 of primary coil of transformer T1 = 1mH;

[0222] DC input voltage: U(VI) = 380V;

[0223] Steady-state DC output voltage: U(VO) = ​​24V; Output power: 230W;

[0224] Capacitors: C1 = 220uF, C3 = 4400uF;

[0225] Auxiliary power supply voltage: U(VCC1,VCC1-GNG)=12V;

[0226] The switching frequency of the switching converter at full output load is 83.3kHz.

[0227] The dead time of the first drive signal waveform U(GS-Q1) and the second drive signal waveform U(GS-Q2) output by the main control module 10 is 900ns.

[0228] When the aforementioned switching power supply converter, which utilizes the innovative technology of this utility model, operates at full load (belonging to the aforementioned high-current load state 2), the following measurements are obtained: Figure 6 In the circuit shown, a time-axis comparison diagram of the driving voltage waveforms of the main switching transistors Q1 and Q2 and the current waveforms of the main components is shown below. Figure 7 As shown above, the following can be measured: Figure 6 A time-axis comparison diagram of the drive voltage waveforms of the main switching transistors Q1 and Q2 in the circuit shown is provided. Figure 11 As shown above. Figure 10 , Figure 11 Detailed unfolded diagrams of the voltage and current waveforms of the four operating modes identified in the diagram, showing the same time axis comparison. Figure 12 As shown:

[0229] Figure 10-12 The symbols in the diagram represent the following: U(GS-Q*) is the voltage waveform between the gate (G) and source (S) of the NMOSFET transistor Q*, where * represents any number or letter; U(DS-Q*) is the voltage waveform between the drain (D) and source (S) of the NMOSFET transistor Q*; I(DS-Q*) is the current waveform flowing through the drain (D) and source (S) of the NMOSFET transistor Q*; I(C*) is the current waveform of the capacitor, with the direction of current flow from the positive terminal of the capacitor as the positive measurement direction; U(L1) is the voltage waveform across inductor L1, with the end of inductor L1 connected to the drain (D) of transistor Q2 as the positive measurement voltage terminal; I(L1) is the current waveform flowing through inductor L1, with the direction of current flowing into the positive measurement voltage terminal of L1 as the positive measurement direction; U(VO) is the voltage waveform at the output terminal VO of the switching power supply converter, with the direction of current flow from the positive terminal of L1 as the positive measurement direction; U(VO) is the voltage waveform at the output terminal VO of the switching power supply converter. Figure 6 In the circuit, VO-GND is referenced to the 0 level. U(VI) represents the voltage waveform at the input terminal VI of the switching power converter. Figure 6 The circuit node VI-GND is referenced to level 0. From Figure 6 , Figure 7As can be seen, when the innovative resonant LLC switching power converter of this invention operates under high current load condition 2, the driving voltages U(GS-Q3) and U(GS-Q4) of the auxiliary switching transistors Q3 and Q4 are both maintained at a high level, causing the auxiliary switching transistors Q3 and Q4 to conduct with extremely low on-resistance. At this time, the auxiliary capacitor C3 connected in series with the auxiliary switching transistor Q3 is equivalent to being connected in parallel across the drain D and source S of the main switching transistor Q1, and the auxiliary capacitor C4 connected in series with the auxiliary switching transistor Q4 is equivalent to being connected in parallel across the drain D and source S of the main switching transistor Q2.

[0230] like Figure 10-12 As shown, during the transition of the main switching transistor Q1 from the on state to the off state in operating mode 2, the current I(DS-Q1) flowing through the drain (D) and source (S) of the main switching transistor Q1 immediately drops from approximately 500mA to less than 100mA. Since the main switching transistor Q1 is completely off at this time, this current of approximately 100mA is the current flowing through the parasitic capacitance between the drain (D) and source (S) of the main switching transistor Q1, and its loss is extremely low and can be ignored. At this time, the current I(C3) flowing through the auxiliary capacitor C3 immediately rises from approximately 50mA to a peak value of approximately 400mA.

[0231] like Figure 10-12 As shown, when the drive voltage U(GS-Q1) of the main switching transistor Q1 has dropped to near 0V, and the main switching transistor Q1 is completely turned off, the drain-source voltage U(DS-Q1) of the main switching transistor Q1 only rises from near 0V to about 380V at a relatively slow slope. Therefore, in operating mode 2, there is no crossover period where the drain-source current waveform I(DS-Q2) and voltage waveform U(GS-Q2) of Q2 are both significantly greater than 0.

[0232] In a half-bridge LLC switching power converter, the turn-off loss of the NMOSFET main switching transistor Q1 is calculated using the following formula (Formula 5):

[0233] Formula 5

[0234] Therefore, it can be seen that the innovative resonant LLC switching power converter of this utility model realizes the time-free switching of the resonant cavity current I(L1) path during the rapid turn-off process of the main switching transistor Q1. I(L1) switches from the semiconductor conduction channel between the drain D and source S of the main switching transistor Q1 (operating mode 1) to the branch flowing into the auxiliary capacitor C3 and the parasitic capacitance branch of the main switching transistor Q1 (operating mode 2) without delay, thereby causing the current I(DS-Q1) of the main switching transistor Q1 to drop significantly and immediately without delay during its turn-off process.

[0235] Formula 5 The parameter is the voltage between the drain (D) and source (S) of the main switching transistor Q1 during the turn-off process. With current value At the same time, the crossover time period with values ​​greater than zero, such as Figure 10-12 As shown and as mentioned above, it can be seen that Therefore, according to Formula 5, Therefore, the innovative resonant LLC switching power converter of this utility model makes the turn-off loss of the main switching transistor Q1 very small, realizing the zero-current soft turn-off function of the main switching transistor Q1.

[0236] like Figure 10-12 As shown, during the transition of the main switching transistor Q2 from the on state to the off state in operating mode 4, the current I(DS-Q2) flowing through the drain (D) and source (S) of the main switching transistor Q2 immediately drops from approximately 500mA to less than 100mA. Since the main switching transistor Q2 is completely off at this time, this current of approximately 100mA is the current flowing through the parasitic capacitance between the drain (D) and source (S) of the main switching transistor Q2, and its loss is extremely low and can be ignored. At this time, the current I(C3) flowing through the auxiliary capacitor C3 immediately rises from approximately 50mA to a peak value of approximately 400mA.

[0237] like Figure 10-12 As shown, when the drive voltage U(GS-Q2) of the main switching transistor Q2 has dropped to near 0V, and the main switching transistor Q2 is completely turned off, the drain-source voltage U(DS-Q2) of the main switching transistor Q2 only rises from near 0V to about 380V at a relatively slow slope. Therefore, in operating mode 4, there is no crossover period where the drain-source current waveform I(DS-Q2) and voltage waveform U(GS-Q2) of the main switching transistor Q2 are both significantly greater than 0.

[0238] The hard turn-off loss of the NMOSFET auxiliary switching transistor Q4 in the half-bridge LLC switching power converter is calculated using the following formula (Formula 6):

[0239] Formula 6

[0240] Meaning of symbols in formulas 5-6: Input the instantaneous voltage value between the negative terminal of the DC voltage VI-GND of the circuit node VS-Q1 and the reference point at position 0; This is the instantaneous voltage across resistor R*. The voltage between the drain (D) and source (S) of transistor Q* during the transition period from a fully on state to a fully off state. With current Simultaneously, the time period in which the value is significantly greater than 0; This refers to the operating switching frequencies of the main switching transistors Q1 and Q2 in the resonant LLC switching power converter. This represents the power loss during the transition period when the switching transistor Q* changes from a fully on state to a fully off state. This represents the instantaneous voltage between the drain (D) and source (S) of the NMOSFET switch Q*. The instantaneous value of the current flowing through the semiconductor channel between the drain D and source S of the NMOSFET switch Q*; The peak current flowing through the semiconductor channel between the drain (D) and source (S) of the NMOSFET switch Q*; * represents a number or letter designation.

[0241] Therefore, it can be seen that the innovative resonant LLC switching power converter of this utility model achieves a path-free switching of the resonant cavity current I(L1) during the rapid turn-off process of the main switching transistor Q2. I(L1) switches from the semiconductor conduction channel between the drain D and source S of the main switching transistor Q2 (operating mode 3) without delay to the branch flowing into the auxiliary capacitor C4 and the parasitic capacitance branch of the main switching transistor Q2 (operating mode 4), so that the current I(DS-Q2) of the main switching transistor Q2 drops significantly immediately without delay during its turn-off process.

[0242] Formula 6 The parameter is the voltage between the drain (D) and source (S) of the main switching transistor Q2 during its turn-off process. With current value At the same time, the crossover time period is greater than zero, such as Figure 10-12 As shown and as mentioned above, it can be seen that Therefore, according to Formula 6, Therefore, the innovative resonant LLC switching power converter of this utility model makes the turn-off loss of the main switching transistor Q2 very small, realizing the soft turn-off function of the main switching transistor Q2.

[0243] like Figure 10 As shown, when the voltage U(DS-Q1) between the drain D and source S of the main switching transistor Q1 drops to near 0V, its driving voltage U(GS-Q1) only rises from near 0V to about 12V. Therefore, the innovative resonant LLC switching power converter of this invention makes the turn-on loss of the main switching transistor Q1 very small, realizing the zero-voltage soft-turn-on function of the main switching transistor Q1.

[0244] like Figure 10As shown, when the voltage U(DS-Q2) between the drain D and source S of the main switching transistor Q2 drops to near 0V, its second driving voltage U(GS-Q2) only rises from near 0V to about 12V. Therefore, the innovative resonant LLC switching power converter of this invention makes the turn-on loss of the main switching transistor Q2 very small, realizing the zero-voltage soft-turn-on function of the main switching transistor Q2.

[0245] In summary, this innovative resonant LLC switching power converter achieves zero-voltage soft-turn-on and zero-current soft-turn-off functions for the main switching transistors Q1 and Q2, resulting in extremely low switching losses and further improving the efficiency of the power converter. Therefore, this invention meets the growing demands of today's socio-economic society for high efficiency, energy saving, and carbon reduction, and has the potential for large-scale application and broad socio-economic benefits.

[0246] like Figure 1-12 As shown, the half-bridge switching circuit in the innovative resonant LLC switching power converter of this utility model is innovatively composed of two main switching transistors Q1 and Q2, two auxiliary switching transistors Q3 and Q4, and two auxiliary capacitors C4 and C5, and its innovative connection method.

[0247] This innovative resonant LLC switching power converter achieves a zero-delay switching of the resonant cavity current I(L1) path during the high-current load state 2 operating mode 2 period. During the rapid turn-off process of the main switching transistor Q1, I(L1) switches from the semiconductor conduction channel between the drain D and source S of Q1 (operating mode 1) to the branch flowing into the auxiliary capacitor C3 and the parasitic capacitance branch of the main switching transistor Q1 (operating mode 2) without delay. This results in a significant and immediate drop in the main switching transistor Q1 current I(DS-Q1) during its turn-off process.

[0248] Formula 5 The parameter is the voltage between the drain (D) and source (S) of the main switching transistor Q1 during the turn-off process. With current value As mentioned above, the crossover time period with values ​​greater than zero indicates that in the innovative resonant LLC switching power supply converter of this utility model... Therefore, according to Formula 5, Therefore, the innovative resonant LLC switching power converter of this utility model makes the turn-off loss of its main switching transistor Q1 very small, realizing the zero-current soft turn-off function of the main switching transistor Q1.

[0249] In the high-current load state 2, the innovative resonant LLC switching power converter of this invention achieves a path-free switching of the resonant cavity current I(L1) during the fast turn-off process of the main switching transistor Q2 in the operating mode 4 period. I(L1) switches without delay from the semiconductor conduction channel between the drain D and source S of the main switching transistor Q2 (operating mode 3) to the branch flowing into the auxiliary capacitor C4 and the parasitic capacitance branch of the main switching transistor Q2 (operating mode 4), so that the current I(DS-Q2) of the main switching transistor Q2 drops significantly and immediately without delay during its turn-off process.

[0250] Formula 6 The parameter is the voltage between the drain (D) and source (S) of the main switching transistor Q2 during its turn-off process. With current value As mentioned above, the crossover time period is also greater than zero, indicating that in the innovative resonant LLC switching power supply converter of this utility model... Therefore, according to Formula 6, Therefore, the innovative resonant LLC switching power converter of this utility model makes the turn-off loss of the main switching transistor Q2 extremely small, realizing the zero-current soft turn-off function of the main switching transistor Q2.

[0251] like Figure 11 As shown, when the voltage U(DS-Q1) between the drain D and source S of the main switching transistor Q1 drops to near 0V, its driving voltage U(GS-Q1) only rises from near 0V to about 12V. Therefore, the innovative resonant LLC switching power converter of this invention makes the turn-on loss of Q1 very small, realizing the zero-voltage soft-turn-on function of the main switching transistor Q1.

[0252] like Figure 11 As shown, when the voltage U(DS-Q2) between the drain D and source S of the main switching transistor Q2 drops to near 0V, its second driving voltage U(GS-Q2) only rises from near 0V to about 12V. Therefore, the innovative resonant LLC switching power converter of this invention makes the turn-on loss of the main switching transistor Q2 very small, realizing the zero-voltage soft-turn-on function of the main switching transistor Q2.

[0253] In summary, compared with the high-current hard shutdown state of the switching module in the prior art, the innovative resonant LLC switching power converter of this utility model achieves zero-voltage soft turn-on and zero-current soft turn-off functions of the main switching transistors Q1 and Q2 under high-current load state 2, making their switching losses very small. Compared with the prior art, the technology of this utility model further improves the efficiency of the power converter.

[0254] like Figure 7As shown, in the low current load state 1, the output drive signals U(GS-Q3) and U(GS-Q4) of the zero-voltage switching control circuit 20 of the innovative resonant LLC switching power converter of this utility model are both at low level, and the auxiliary switching transistors Q3 and Q4 are both kept in the off state. At this time, the auxiliary capacitors C4 and C5 are disconnected from the half-bridge arm main switching transistors Q1 and Q2.

[0255] A current technology has the following problem: Because a capacitor is directly connected in parallel between the drain (D) and source (S) of the bridge arm switch, when the drive pulse is a discontinuous pulse group output, the switch is in a hard-turn-on state during the first pulse after the drive pulse interruption. In this case, the added auxiliary capacitor increases the switching losses. Under normal circumstances, when the resonant LLC switching power converter is in no-load standby and under very low load, the control module adjusts the operating frequency to reach the set maximum safe frequency value to stabilize the output voltage, but this still cannot achieve voltage regulation. At this time, the control module will enter a state of continuously interrupted drive pulse group output to stabilize the output voltage. Therefore, the above-mentioned method increases the losses and reduces the efficiency of the LLC switching power converter in standby and low load conditions.

[0256] As described above, the innovative resonant LLC switching power converter of this invention does not suffer from the problem of increased switching losses in the prior art. Therefore, this invention can not only achieve the zero-current turn-off and zero-voltage turn-on functions of the main switching transistor Q1 or Q2 in the high-current load state 2, but also meet the requirement of not increasing the turn-on power consumption of the main switching transistor Q1 or Q2 in the low-current load state 1, thus achieving the requirements of low standby power consumption and high efficiency under low load.

[0257] In summary, the innovative resonant LLC switching power converter of this invention can further reduce losses and improve efficiency across the entire load range and all operating states. Therefore, it can fully meet the socio-economic development trend that places higher demands on energy saving, carbon reduction, low standby power consumption, and high efficiency under small loads. This is conducive to the large-scale popularization and application of this technology in more emerging application scenarios and under higher requirements.

[0258] like Figure 7 As shown, the input-output control logic definition of the innovative zero-voltage switching control circuit 20 in the innovative resonant LLC switching power converter of this utility model is as follows: the output drive signal U(GS-Q3) needs two necessary conditions to change from low level to high level or from high level to low level.

[0259] like Figure 7As shown, when the power converter transitions from a low-current load state 1 to a high-current load state 2, the auxiliary switching transistors Q3 and Q4 cannot be immediately turned on. At this time, the zero-voltage switching control circuit 20 still needs to determine the states of the main switching transistors Q1 and Q2, waiting for them to enter a stable on-state before immediately triggering the auxiliary switching transistors Q3 and Q4 to turn on. Once the auxiliary switching transistors Q3 and Q4 are triggered to turn on, they are locked in the on-state. At this point, they can only switch to the off-state if the following conditions are met to trigger their turn-off. Therefore, both auxiliary switching transistors Q3 and Q4 achieve highly efficient zero-voltage soft-turn-on functionality, and their turn-on does not interfere with the main circuit.

[0260] like Figure 7 As shown, when the LLC power converter transitions from a high-current load state 2 to a low-current load state 1, the auxiliary switching transistors Q3 and Q4 cannot be immediately turned off. At this time, the zero-voltage switching control circuit 20 still needs to determine the states of the main switching transistors Q1 and Q2, waiting for them to enter a stable on-state before immediately triggering the auxiliary switching transistors Q3 and Q4 to turn off. Once the auxiliary switching transistors Q3 and Q4 are triggered to turn off, they are locked in the off state. Only when the aforementioned conditions for triggering the conduction of the auxiliary switching transistors Q3 and Q4 are met can they switch to the on state. Therefore, both auxiliary switching transistors Q3 and Q4 achieve highly efficient zero-voltage soft-turn-off functionality, and the turn-off of auxiliary switching transistors Q3 and Q4 does not cause any interference to the main circuit.

[0261] Example 2

[0262] This second embodiment provides a resonant LLC switching power converter, including the zero-voltage switching control circuit provided in the first embodiment.

[0263] This innovative resonant LLC switching power converter not only achieves high-efficiency zero-voltage soft turn-on and zero-current soft turn-off for the main switching transistors Q1 and Q2 under high-current load condition 2, but also achieves high-efficiency zero-voltage turn-on and zero-voltage turn-off for the auxiliary switching transistors Q3 and Q4 in all operating states.

[0264] The innovative zero-voltage switching control circuit 20, along with its circuit composition and connection method, in this innovative resonant LLC switching power converter are all newly added independent circuits that do not require modification of existing control circuits. The main control module 10 used in this invention is compatible with existing control modules, thus reducing the overall R&D and upgrade costs of the power system. Therefore, it is possible to upgrade the less efficient existing resonant LLC switching power converter to the innovative resonant LLC switching power converter of this invention at extremely low cost. This invention achieves high efficiency and energy saving requirements under any operating current condition.

[0265] The innovative half-bridge switching circuit of this invention consists of two main switching transistors Q1 and Q2, two auxiliary switching transistors Q3 and Q4, and two auxiliary capacitors C4 and C5, along with an innovative connection method. Two identical sets of this innovative half-bridge circuit can be applied to a full-bridge power converter to meet the needs of different application scenarios.

[0266] In summary, the innovative resonant LLC switching power converter technology of this utility model simultaneously meets the growing demands of today's socio-economic society for high efficiency under both large and small loads, as well as for energy saving, carbon reduction, and low standby power consumption. Furthermore, this utility model technology is highly versatile, has low update costs, and high reliability, meeting the technical needs of numerous traditional and emerging application fields. Therefore, this utility model technology has the prospect of large-scale application and considerable social and economic benefits.

[0267] The innovative features of this resonant LLC switching power converter can also be applied to LLC switching power supplies with a full-bridge topology. The innovative resonant full-bridge LLC switching power converter is composed of the half-bridge arm switching circuit innovatively described above.

[0268] The innovative features of the resonant LLC switching power converter described in this invention can also be applied to full-bridge switching power supplies with phase-shifting control. The innovative phase-shifting full-bridge switching power converter of this invention consists of two sets of half-bridge arm switching circuits with the innovative features described in this invention.

[0269] The innovative resonant LLC switching power supply converter, resonant full-bridge LLC switching power supply converter, and phase-shifted full-bridge switching power supply converter of this invention can be applied to high-frequency switching power supplies of any appearance and structural form.

[0270] The innovative resonant LLC switching power converter, resonant full-bridge LLC switching power converter, and phase-shifted full-bridge switching power converter of this utility model can be implemented in any printed circuit board layout and connection method.

[0271] The components in the innovative resonant LLC switching power converter, resonant full-bridge LLC switching power converter, and phase-shifted full-bridge switching power converter of this utility model can be any packaged components of the same principle and performance, or their series or parallel combinations.

[0272] The above embodiments are only used to illustrate the technical solutions of this utility model, and are not intended to limit it. Although this utility model has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of this utility model, and should all be included within the protection scope of this utility model.

Claims

1. A zero-voltage switching control circuit of a resonant LLC switching power converter, the resonant LLC switching power converter comprising a first switching module, a second switching module, an inductor module, a first capacitor module, and an isolation module, the first switching module receiving a first control signal, the second switching module receiving a second control signal, such that the first switching module and the second switching module form a resonant circuit with the inductor module, the first capacitor module, and the isolation module, respectively, characterized in that, The resonant LLC switching power supply converter further comprises: a third switch module and a second capacitor module, which are connected in series and then connected in parallel with the first switch module; a fourth switch module and a third capacitor module, which are connected in series and then connected in parallel with the second switch module; The zero-voltage switching control circuit comprises: a reference voltage module for outputting a reference voltage; a current detection module, whose first input end collects a primary end current or a secondary end current flowing through the isolation module in the resonant LLC switching power supply converter, and whose second input end is connected to the output end of the reference voltage module, for converting the primary end current or the secondary end current into a collected voltage and comparing the collected voltage with the reference voltage to output a first comparison signal; a first control module, whose first input end receives the first control signal, whose second input end is connected to the output end of the reference voltage module, and whose third input end is connected to the output end of the current detection module, and whose output end is connected to the control end of the third switch module, for outputting a first on-off control signal to the third switch module according to the first control signal, the reference voltage and the first comparison signal; a second control module, whose first input end receives the second control signal, whose second input end is connected to the output end of the reference voltage module, and whose third input end is connected to the output end of the current detection module, and whose output end is connected to the control end of the fourth switch module, for outputting a second on-off control signal to the fourth switch module according to the second control signal, the reference voltage and the first comparison signal.

2. The zero voltage switching control circuit of claim 1, wherein, The first control module comprises: a first delay trigger module, whose first input end receives the first control signal, whose second input end receives the reference voltage, and whose third input end receives the first comparison signal, for outputting a first weak current on-off control signal according to the first control signal, the reference voltage and the first comparison signal; a first drive module, whose input end is connected to the output end of the first delay trigger module, for amplifying and isolating the first weak current on-off control signal and then outputting a first on-off control signal.

3. The zero voltage switching control circuit of claim 2, wherein, The first delay trigger module comprises: a first comparison module, whose first input end receives the first control signal, and whose second input end receives the reference voltage, for converting the first control signal into a first collected voltage after time delay, and outputting a first trigger signal when the first collected voltage is greater than the reference voltage; a first trigger module, whose first input end receives the first comparison signal, and whose second input end is connected to the output end of the first comparison module, for outputting the first weak current on-off control signal according to the first trigger signal.

4. The zero voltage switching control circuit of claim 1, wherein, The second control module comprises: a second delay trigger module, whose first input end receives the second control signal, whose second input end receives the reference voltage, and whose third input end receives the first comparison signal, for outputting a second weak current on-off control signal according to the second control signal, the reference voltage and the first comparison signal; a second drive module, whose input end is connected to the output end of the second delay trigger module, for amplifying and isolating the second weak current on-off control signal and then outputting a second on-off control signal. The second driving module is connected with the output end of the second delay trigger module, and is used for amplifying and isolating the output second weak current on-off control signal to output a second on-off control signal.

5. The zero voltage switching control circuit of claim 4, wherein, The second delay trigger module comprises: The second comparison module receives the second control signal at a first input end and receives a reference voltage at a second input end, and is used for converting the second control signal into a second acquisition voltage through delay, and outputting a second trigger signal when the second acquisition voltage is greater than the reference voltage; The second trigger module receives the first comparison signal at a first input end and is connected with the output end of the second comparison module at a second input end, and is used for outputting the second weak current on-off control signal according to the second trigger signal.

6. The zero-voltage switching control circuit of claim 1, wherein, The current detection module comprises a first resistor, a second resistor, a third resistor, a fourth resistor, a fifth resistor, a sixth resistor, a first capacitor, a second capacitor, a third capacitor, a first comparator and a first diode. One end of the first capacitor is a first input end of the current detection module, and the other end of the one end of the first capacitor is connected with one end of the first resistor, one end of the second resistor, one end of the third resistor and one end of the second capacitor respectively, the other end of the first resistor is connected with one end of the fourth resistor and a non-inverting input end of the first comparator respectively, the other end of the second resistor, the other end of the second capacitor, one end of the sixth resistor, a ground end of the first comparator and one end of the third capacitor are connected with ground, the other end of the third resistor and one end of the fifth resistor are connected with each other and serve as a second input end of the current detection module, the other end of the fifth resistor is connected with the other end of the sixth resistor and an inverting input end of the first comparator respectively, the other end of the fourth resistor is connected with a cathode of the first diode, an anode of the first diode and an output end of the first comparator are connected with each other and serve as an output end of the current detection module, and a power supply end of the first comparator and the other end of the third capacitor are connected with each other and receive a first power supply.

7. The zero-voltage switching control circuit of claim 1, wherein, The reference voltage module comprises a seventh resistor, a fourth capacitor and a three-terminal voltage regulator, one end of the seventh resistor receives the first power supply, the other end of the seventh resistor, one end of the fourth capacitor, a cathode of the three-terminal voltage regulator and a power supply end of the three-terminal voltage regulator are connected with each other and serve as an output end of the reference voltage module, and an anode of the three-terminal voltage regulator and the other end of the fourth capacitor are connected with ground.

8. The zero voltage switching control circuit of claim 3, wherein, The first comparison module comprises an eighth resistor, a ninth resistor, a tenth resistor, a fifth capacitor, a sixth capacitor, a second diode and a second comparator, and the first trigger module comprises a first flip-flop and a seventh capacitor. One end of the fifth capacitor and one end of the ninth resistor are connected to ground, one end of the eighth resistor receives a first control signal, the other end of the eighth resistor is connected to the other end of the ninth resistor, the other end of the fifth capacitor, one end of the tenth resistor and the non-inverting input terminal of the second comparator respectively, the inverting input terminal of the second comparator receives the reference voltage, the power supply terminal of the second comparator and one end of the sixth capacitor are connected to receive the first power supply after being connected together, the ground terminal of the second comparator and the other end of the sixth capacitor are connected to ground, the other end of the tenth resistor is connected to the cathode of the second diode, the anode of the second diode is connected to the output terminal of the second comparator and the C pin of the first flip-flop respectively, the D pin of the first flip-flop receives the first comparison signal, the R pin, the ground terminal, the S pin of the first flip-flop and one end of the seventh capacitor are connected to ground, the power supply terminal of the first flip-flop and the other end of the seventh capacitor are connected to receive the first power supply after being connected together, and the Q pin of the first flip-flop is the output terminal of the first trigger module; The first driving module comprises a thirteenth transistor, a fourteenth transistor, a fifteenth transistor, a sixteenth transistor, an eleventh resistor, a twelfth resistor, a thirteenth resistor, a fourteenth resistor, a fifteenth resistor, a sixteenth resistor, a seventeenth resistor and an eighth capacitor, one end of the eleventh resistor is an input terminal of the first driving module, the other end of the eleventh resistor is connected to the base of the fifth transistor and one end of the twelfth resistor respectively, the collector of the fifth transistor is connected to one end of the thirteenth resistor, the emitter of the fifth transistor and the other end of the twelfth resistor are connected to ground, the other end of the thirteenth resistor is connected to one end of the fourteenth resistor and the base of the sixth transistor respectively, the other end of the fourteenth resistor, the emitter of the sixth transistor, the collector of the thirteenth transistor and one end of the eighth capacitor are connected together to receive the second power supply, the collector of the sixth transistor is connected to one end of the fifteenth resistor, the other end of the fifteenth resistor is connected to the base of the thirteenth transistor, the base of the fourteenth transistor and one end of the sixteenth resistor respectively, the emitter of the thirteenth transistor is connected to one end of the seventeenth resistor and the emitter of the fourteenth transistor respectively, the other end of the seventeenth resistor is a first output terminal of the first driving module, the collector of the fourteenth transistor, the other end of the sixteenth resistor and the other end of the eighth capacitor are connected together to be a second output terminal of the first driving module.

9. The zero voltage switching control circuit of claim 5, wherein, The second comparison module comprises an eighteenth resistor, a nineteenth resistor, a twentieth resistor, a ninth capacitor, a tenth capacitor, a third diode and a third comparator, and the second trigger module comprises a second flip-flop and an eleventh capacitor. One end of the ninth capacitor and one end of the nineteenth resistor are connected to ground, one end of the eighteenth resistor receives a second control signal, the other end of the eighteenth resistor is connected to the other end of the nineteenth resistor, the other end of the ninth capacitor, one end of the twentieth resistor and the non-inverting input terminal of the third comparator respectively, the inverting input terminal of the third comparator receives the reference voltage, the power supply terminal of the third comparator is connected to the first power supply after being connected to one end of the tenth capacitor, the ground terminal of the third comparator is connected to ground after being connected to the other end of the tenth capacitor, the other end of the twentieth resistor is connected to the cathode of the third diode, the anode of the third diode is connected to the output terminal of the third comparator and the C pin of the second flip-flop respectively, the D pin of the second flip-flop receives the first comparison signal, the R pin, the ground terminal, the S pin of the second flip-flop and one end of the eleventh capacitor are connected to ground, the power supply terminal of the second flip-flop is connected to receive the first power supply after being connected to the other end of the eleventh capacitor, and the Q pin of the second flip-flop is the output terminal of the second trigger module. The second drive module includes a first transistor, a second transistor, a twenty-first resistor, a twenty-second resistor, a twenty-third resistor and a twelfth capacitor, one end of the twenty-first resistor is an input terminal of the second drive module, the other end of the twenty-first resistor is connected to the base of the first transistor, the base of the second transistor and one end of the twenty-second resistor respectively, the collector of the first transistor is connected to receive the first power supply after being connected to one end of the twelfth capacitor, the emitter of the first transistor is connected to the emitter of the second transistor and one end of the twenty-third resistor respectively, the other end of the twenty-third resistor is a first output terminal of the second drive module, and the collector of the second transistor is connected to be a second output terminal of the second drive module after being connected to the other end of the twelfth capacitor.

10. A resonant LLC switching power supply converter, characterized by, The zero-voltage switching control circuit includes any one of claims 1 to 9. The zero-voltage switching control circuit includes any one of claims 1 to 9.

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