Amplifier circuit for pulse operation as well as system and method for amplifying a signal generated in pulse operation

Abstract

System for amplifying an RF signal generated in pulse operation, wherein the system comprises (300; 400): a first amplifier (313; 409), wherein the first amplifier (313; 409) is configured to amplify the RF signal (320) which is applied to an input (Vin; VINP, VINN) of the first amplifier (313; 409), a switchable impedance element (TR2; TR2P, TR2N) which is coupled to an output of the first amplifier (313; 409), wherein the switchable impedance element (TR2; TR2P, TR2N) is designed such that it includes a first impedance, when the RF signal (320) is active, and that it includes a second impedance when the RF signal (320) is not active, and a second amplifier (311; 411) which is coupled to the output of the first amplifier (313; 409), wherein the switchable impedance element comprises an LDMOS transistor (TR2; TR2P, TR2N).

Description

[0001] The present invention relates generally to RF communication systems and in particular to a system and a method for a burst-mode amplifier. A burst-mode amplifier is understood to be, in particular, an amplifier for burst operation, which amplifies a signal that is active at certain times and inactive at other times.

[0002] US 2006 / 0028269A1 describes a power amplifier with a switchable load. Depending on a control signal, the switchable load provides a first impedance to a first amplifier when that first amplifier is operating, and provides a second impedance to a second amplifier when that second amplifier is operating.

[0003] US patent 2005 / 0186919A1 discloses a power amplifier in which an output signal is selectively generated either by a single amplifier or by multiple amplifier stages.

[0004] US Patent 7,177,370 B2 describes an RF transmitter for both GSM and EDGE. It discloses a power amplifier comprising a power transistor, optionally another power transistor, and, in addition to an input matching circuit and an output matching circuit, a step matching circuit coupled between the collector terminal of the second power transistor and the gate terminal of the power transistor.

[0005] US 2007 / 0155347A1 describes an impedance matching network for a high-frequency signal, which includes a switchable impedance transformation circuit that in turn has a CMOS switch or transistor. Background of the invention

[0006] Third-generation and higher base station transmitters for applications such as mobile phones and other wireless communication technologies typically use bandwidth-optimized modulation schemes to transmit information. Bandwidth-optimized modulation schemes, such as CDMA, require a non-constant harmonic or envelope and must be able to handle a high peak-to-average ratio (PAR). Linear power amplifiers, such as AB amplifiers, are typically used in such systems due to their high linearity. However, AB amplifiers must be operated at a high underdrive level and therefore have low power efficiency.Furthermore, AB amplifiers must be designed and biased to handle high peak power levels, even though they are typically operated at a much lower average power. Currently, most state-of-the-art base station systems have a power efficiency of only 30%. This low efficiency results in high costs for setting up and operating base station systems.

[0007] The power efficiency of base station transmitting systems can be improved by using alternative RF amplifier systems and structures. In fact, power efficiency can be improved by using a switching amplifier, such as a Class F amplifier, an inverting Class F amplifier, or a Class D amplifier, instead of a conventional Class AB amplifier. A switching amplifier is more efficient than a Class AB amplifier because a larger portion of the amplifier's dynamic range is used during operation, and current and voltage peaks do not occur simultaneously. For systems requiring bandwidth-efficient modulation schemes, such as UMTS base stations, a special modulator must be designed to simultaneously modulate the time-dependent phase and amplitude information in the time domain.

[0008] Numerous systems have been proposed and developed to employ more power-efficient amplifier structures for bandwidth-efficient modulation schemes. One solution, known as EER (Envelope Elimination and Restoration), modulates the phase and amplitude of the desired output signal separately. EER generates a constant envelope of a phase-modulated signal along with a separate amplitude or envelope signal. The constant envelope of the phase-modulated signal is then used to drive the input of a power amplifier, and the envelope signal is used to adjust the bias or voltage supply to the output stage of the power amplifier accordingly. However, these bias or voltage control techniques offer only low power efficiency.Furthermore, EER becomes increasingly difficult to implement and exhibits even worse power efficiency as the modulation bandwidth increases. Broadband CDMA, for example, has a modulation bandwidth of approximately 50 MHz, which is generally too high for practical use of EER techniques.

[0009] Another solution to the power efficiency problems of bandwidth-efficient modulation is the use of pulse-width modulation with a switched-mode power amplifier. Amplitude modulation is achieved at the amplifier's output by varying the duty cycle of the pulse-width modulated signal applied to the amplifier's input. Pulse-width modulation can be applied to a signal using conventional pulse-width modulation (PWM) techniques or delta-sigma modulation techniques. However, delta-sigma modulation typically exhibits lower efficiency than comparable PWM, which can achieve power efficiencies of up to 60%.

[0010] Despite their advantages in terms of power efficiency, PWM techniques exhibit several problems in practical implementation. One such problem concerns the processing of RF pulses by a chain of RF amplifiers. Ideally, a PWM transmission system should comprise a system with a very high bandwidth RF signal path followed by a high-order transmit filter. In reality, matching networks and bias networks limit the system's bandwidth and introduce resonances that demodulate and attenuate the ideally shaped PWM signals along the signal path. Generally, the weaker the PWM signal and the more it deviates from its constant envelope property, the lower the system's power efficiency, thereby correspondingly degrading the system's signal integrity.

[0011] Therefore, in the field of RF communication systems, circuit system techniques are needed for the effective implementation of PWM systems, which implement bandwidth-efficient modulation schemes. Summary of the invention

[0012] According to the invention, this problem is solved by a system for amplifying an RF signal generated in pulse operation according to claim 1, an amplifier circuit for pulse operation according to claim 10, and by a method for amplifying a signal generated in pulse operation according to claim 18. The dependent claims define preferred and advantageous embodiments of the present invention.

[0013] According to one embodiment of the invention, a system is disclosed which amplifies an RF signal generated in pulsed operation. An RF signal generated in pulsed operation is understood to be, in particular, an RF signal (RF "Radio Frequency", i.e., a radio signal or a high-frequency signal) whose amplitude or envelope temporarily drops to essentially zero, so that the RF signal is virtually inactive during these times. The system comprises a first amplifier configured to amplify the RF signal provided at its input, a switchable impedance element coupled to an output of the first amplifier, and a second amplifier coupled to the output of the first amplifier. The switchable impedance element comprises a first impedance (i.e., has a first impedance value) when the RF signal is active (i.e.,The envelope has values ​​greater than 0), and a second impedance (i.e., it has a second impedance value) when the signal generated in pulse operation is inactive (i.e., the envelope is essentially equal to 0). The switchable impedance element comprises an LDMOS transistor.

[0014] Within the scope of the present invention, an amplifier circuit for pulse operation is also provided. The amplifier circuit comprises a first amplifier stage, a switchable impedance element, and an output stage. An input of the first amplifier stage is coupled to a switched RF signal source. A switched RF signal source is understood to be, in particular, a signal source that generates an RF signal in pulse operation such that the generated RF signal is intermittently active (i.e., the amplitude or envelope of the RF signal has values ​​greater than 0) and intermittently inactive (i.e., the amplitude or envelope of the RF signal is essentially equal to 0). The switchable impedance element is coupled to or connected to an output of the first amplifier stage and has a first impedance value when the switched RF signal source is active (i.e.,The output stage has a second impedance value when the switched RF signal source is inactive (i.e., generating a non-active RF signal). The output stage is coupled or connected to the output of the first amplifier stage. The switchable impedance element comprises an LDMOS transistor.

[0015] Finally, the present invention provides a method for amplifying a signal that is generated in pulse mode, i.e., is intermittently active and inactive. The method according to the invention comprises the following steps: • Applying the signal generated in pulse operation to a first amplifier. • Connecting a switchable impedance element to an output of the first amplifier. • Control the switchable impedance element such that the switchable impedance element assumes a first impedance value when the signal is active, and a second impedance value when the signal is not active. • Connecting a second amplifier to the output of the first amplifier.

[0016] The switchable impedance element includes an LDMOS transistor.

[0017] Features of the present invention are described in general terms below. The present invention is then described in detail below with reference to the figures and preferred embodiments of the invention. Brief description of the drawings Fig. 1A and Fig. 1b represent a system and a signal generator according to an embodiment of the invention. Fig. Figures 2a - 2c represent a state-of-the-art PWM amplifier circuit and its corresponding waveforms. Fig. Figures 3a - 3d represent a PWM amplifier circuit and its corresponding waveforms according to an embodiment of the invention. Fig. Figure 4 represents a PWM amplifier circuit according to a further embodiment of the invention.

[0018] Identical reference numerals and symbols in different figures generally refer to the same parts unless otherwise specified. The figures are designed to clearly illustrate relevant aspects of the embodiments according to the invention, and are not necessarily drawn to scale. To further clarify certain embodiments, a letter may follow a figure number, indicating variations of the same structure, material, or process step. Detailed description of the embodiments

[0019] The following section describes in detail the structure of the present invention with the aid of embodiments according to the invention. It should be noted, however, that the present invention provides numerous applicable inventive concepts which can be used in a large number of specific contexts. The specific embodiments described here only illustrate certain possibilities for carrying out and using the invention and therefore do not limit the scope of the present invention.

[0020] The present invention is described with reference to embodiments in a specific context, namely a system and a method for amplifying RF signals generated in pulse operation in a wireless communication system. However, the present invention can also be used in amplifier systems where the system has a time-varying amplifier characteristic or impedance indicator. Furthermore, the present invention can be used in pulse-width modulation systems or in oversampled modulation systems. Embodiments according to the invention can also be used in applications outside the field of wireless communication systems, for example, in wired communication systems, audio systems, or other systems that have a switching operation characteristic.

[0021] A System 100 for transmitting a bandwidth-efficient modulation scheme using a PWM signal is in Fig. Figure 1a shows the system 100, which comprises a signal generator 102, a switching amplifier 104, a bandpass filter 106, and an antenna 108. The signal generator 102 converts a signal S i (t) into a phase component S θ (t) and an amplitude or envelope component S env (t). The signal S i(t) is valid for every modulation scheme. In embodiments of the invention, modulation schemes are used that employ bandwidth-efficient modulation schemes, such as CMDA, WB CDMA, UMTS, LTE, and WiMAX. These bandwidth-efficient modulation schemes typically modulate the phase and amplitude of the signal, using PSK, BPSK, QPSK, M-stage PSK, or QAM. It should be noted that even if a modulation scheme with a constant envelope, such as QPSK, is initially used, the resulting signal does not have a constant envelope because signal processing with digital transmit filters is employed. In other embodiments of the invention, other modulation schemes, such as AM or FM, can be used.

[0022] In certain embodiments of the invention, the signal generator 102 is a pulse width modulator. A phase signal S env (t) with a constant envelope, which provides the phase information of the S i (t) is output by the signal generator 102 to an amplifier 104 operating in switching mode. S env (t), which is a pulse-width modulated waveform with a constant envelope and resembles a digital signal, is fed to the switching amplifier 104. The amplifier chain within the switching amplifier 104 can comprise at least one preamplifier and one power amplifier. In alternative embodiments of the invention, S env (t) comprise a multi-stage envelope signal and the switching amplifier 104 can comprise only one stage or more than two stages.

[0023] System 100 can be configured to operate as a transmitter within a UMTS base station. For example, it may be necessary for the switching amplifier 104 to output a power level between 45 dBm and 60 dBm. To achieve this level of output power, two or three amplifier stages of approximately 15 dB per stage are typically required within the switching amplifier 104. These amplifier stages can be implemented using LDMOS technology (LDMOS "lateral double-diffused MOSFET"). Of course, according to the invention, more or fewer amplifier stages using different amplifier profiles can also be employed.In addition, other components and technologies, such as MOS components, bipolar components or MESFET components (MESFET = metal-semiconductor field-effect transistor), using, for example, silicon technologies, SOI technologies or GaAs technologies, can be used.

[0024] The output of the switching amplifier 104 S a (t) drives a bandpass filter 106. The bandpass filter 106 can be a high-order bandpass filter, since pulse-width modulated signals generally have a much wider spectrum than the transmitted signal S. o (t) exhibit. In embodiments employing WB CDMA, the bandpass filter 106 is typically a cavity filter with a bandwidth of approximately 50 MHz. According to the invention, other bandwidths and other filter architectures can also be used, and the bandpass filter 106 may also be omitted.

[0025] In the wireless radio system or RF system 100, the antenna 108 transmits the signal S o (t) over the air. An exit S o (t) of the bandpass filter 106 can be directly coupled to the antenna 108, or the output S o (t) is indirectly coupled to antenna 108 via a matching network, a coupling network, or another circuit. The signal S o (t) can also be fed to a further amplifier stage or the signal S o (t) can be used to control a different transmission medium, for example a transmission line, a waveguide or an optical line.

[0026] With reference to Fig. Figure 1b shows the signal generator 102 according to the embodiment of the invention. The signal generator 102 comprises a baseband processor 128, a modulator 130, a quadrature modulator 150, and a switch 160. The baseband processor 128 extracts an amplitude information a(t) and a separate phase information j(t) from the input signal S. i (t). Modulator 130 outputs an envelope signal S env (t), which is based on the extracted amplitude a(t). The modulator 130 can comprise a pulse-width modulator, which, for example, implements state-of-the-art pulse-width-based modulation techniques using digital methods. Alternatively, the modulator 130 can comprise other modulator architectures, such as a delta-sigma modulator, a multi-stage pulse-width modulator, or another architecture that implements pulse-width-based modulation techniques.

[0027] The envelope signal S env (t) controls the activation of switch 160, thereby enabling the phase signal S θ1 (t) is forwarded when switch 160 is activated by the envelope signal S env (t) is activated. The phase signal S θ (t) is therefore a pulse width modulated signal with a variable duty cycle, which corresponds to the changes in the amplitude of the input signal S. i (t) corresponds. According to the invention, both the amplitude signal S env (t) as well as the phase signal S θ (t) on the amplifier 104 operating in switching mode ( Fig. 1a) are applied. However, in the illustrated embodiment, only S θ (t) applied as described below.

[0028] In the present invention, the quadrature modulator 150 performs a phase modulation with respect to the phase signal S θ(t) is carried out. The quadrature modulator 150 can include an in-phase mixer 136 and a quadrature mixer 138. The output of the in-phase mixer 136 and the output of the quadrature mixer 138 are summed by a summing node 140 to produce the phase signal S θ1 (t) to form, which is based on the phase information j (t) which is provided by the S i (t) is derived using the baseband processor 128. The quadrature modulator 150 can be configured according to the present invention in any manner known in the prior art. The signal generator 102 can also be configured according to the invention using other circuit techniques and methods known in the prior art. For example, the phase signal S θ1 (t) can be derived using PLL control loops or frequency synthesizers.

[0029] Fig. Figure 2a represents a circuit embodiment of a conventional switching amplifier 200. The switching amplifier 200 has a preamplifier 209, which has an input transistor TR1 charged by an RF coil 204. The output of the preamplifier 209 is coupled to the input of a power amplifier 211, which has a transistor TR3 charged by an RF coil 206 and driven by a voltage V. BIAS via a bias resistor R B is pre-tensioned. The matching network 202, which has an AC coupling capacitance C, ensures adjustment of the stages. BThe switching amplifier 200 is biased according to state-of-the-art biasing techniques. Since the switching amplifier 200 is used to amplify PWM signals, an input terminal Vin at the gate of transistor TR1 of the preamplifier is coupled to a PWM signal, and a high-order bandpass filter 212 is coupled to an antenna 214.

[0030] The transistor TR1 of the preamplifier is typically driven in such a way that the input of TR1 is below its threshold value V during time periods between individual pulses. T This effectively switches off transistor TR1 to save power between pulses (i.e., when no signal is present).

[0031] Fig. 2b represents a PWM-RF signal 220, which is connected to an input terminal Vin of the switching amplifier 200 ( Fig. 2a). The signal 220 can be an output signal S θ (t) from the signal generator 102 ( Fig. 1b). The signal 220 is modulated with respect to its phase and has an envelope 222, which is represented by the dashed lines in Fig. 2b is shown. A corresponding signal, which is received at an input node 210 of the power amplifier ( Fig. 2a) is present, is indicated by a signal 240 in Fig. Figure 2c shows that at the time the signal has been amplified by the preamplifier 209 and passes through the matching network 202, a demodulation 242 of the signal clearly occurs. This demodulation can be detected by the matching network 202 ( Fig. 2a) or caused by the interaction of matching network 202, preamplifier 209 and power amplifier 211. Since a signal is present during time periods in which no signal is desired, more power is consumed than necessary.

[0032] Fig. Figure 3a represents a switching amplifier 300 according to an embodiment of the invention. As with the conventional switching amplifier 200, the switching amplifier 300 comprises a preamplifier 313 with an input transistor TR1, an impedance matching network 302, and a power amplifier 311 with an output transistor TR3. However, in the switching amplifier 300, a switchable impedance element, implemented by an LDMOS transistor TR2, is coupled to an output of the preamplifier 313. The switchable impedance element is present to provide a first impedance value when a signal generated by pulse operation is active, and to provide a second impedance value, which differs from the first impedance value, when the signal generated by pulse operation is not active.Transistor TR2 can be coupled to the input of the matching network 302 and to the drain terminal of the input transistor TR1. AC coupling capacitors C1 and C2 are provided to isolate the DC voltage from each transistor. RF chokes and RF coils 304 and 306 provide bias current for the preamplifier 313 and the power amplifier 311, respectively, while a high upper headroom (e.g., allowing for the highest possible amplitude values ​​for passed signals) and a high impedance with respect to the passed signals are maintained.

[0033] Transistor TR2 is controlled by a signal ENV, which is linked to the envelope signal S. env (t) corresponds to the signal from signal generator 102 ( Fig. 1b) is output. The ENV signal can be a digital signal decoupled by an inverter 303. According to the invention, the ENV signal can also be provided by any other digital circuit known in the prior art, or the ENV signal can comprise an analog signal. Means for generating the envelope signal ENV can be present in the system to generate the ENV signal according to the signal generated by pulse operation.

[0034] According to the embodiment, the matching network 302 comprises an inductive element L1 in series. The values ​​for C1, C2, C3 and L1 can be selected according to the following equations. C1=C2=1πfcRS C3=12πRifcRSRi−RS L1=RS(1−Ri−RSRS)2πfc

[0035] R S R is the impedance exhibited by the drain terminal of transistor TR1 when transistor TR1 is switched on. iis the impedance exhibited by the gate terminal of transistor TR3 and f c is the carrier frequency of the system. It should be noted that the parasitic capacitance of the drain terminal of transistor TR1 contributes at least partially to the capacitance of C3. Alternatively, a matching network, as known from the prior art, can be used in the embodiment according to the invention.

[0036] For example, transistor TR2 can be designed to have a resistance between approximately 5 Ω and 20 Ω, for example 10 Ω, when transistor TR2 is switched on. The output impedance for transistor TR2 can be selected to be of a similar order of magnitude to the output impedance of transistor TR1 when transistor TR1 is switched on, according to the invention. In the illustrated embodiment, transistor TR2 does not include a bias network for a DC voltage (i.e., no DC bias); however, it is possible, according to the invention, for transistor TR2 to be biased by a resistor, for example an RF choke or a current source. If transistor TR2 is biased by a DC voltage, a compromise must be made in the design between the output resistance and the power consumption.In this case, the output impedance of transistor TR2 can be chosen to be higher than the output impedance of transistor TR1, as long as acceptable system performance is achieved. In alternative embodiments of the invention, the output impedance of transistor TR2 can even differ from the output impedance of transistor TR1 by orders of magnitude.

[0037] Fig. 3b - 3d represent intermediate signals in the signal path of the amplifier 300 operating in switching mode. Fig. 3b represents a representative signal 320, which is present at the gate of transistor TR1 and corresponds to the signal of the Fig. 2b is similar to the above. Signal 320 is a pulse-width modulated signal with an envelope 322. Signal 320 is applied to the gate of transistor TR1, so that the voltage at the gate of transistor TR1 remains below a threshold voltage V during time periods when the pulse is not active, for example, during time periods outside the envelope 322 (or when the envelope 322 drops to almost 0). T of transistor TR1.

[0038] Fig. 3c represents a time response of signal 330, which corresponds to the ENV signal applied to the gate of transistor TR2.

[0039] Fig. 3d represents a signal 340, which is connected to a node 310 at the input of the power amplifier 311 ( Fig. 3a) is present. In contrast to the one in Fig. In the signal shown in Figure 2c, signal 340 shows no demodulation at the input of the power amplifier 311. The lack of demodulation is due to transistor TR2, which is switched on when the pulse-width modulated signal is not active. According to the invention, transistor TR2 prevents demodulation by effectively providing an AC ground to the input of the matching network 302. Without transistor TR2, the series connection of C1, C2, C3, and L1 forms a resonant LC series circuit. Without transistor TR2 pulling capacitor C3 to ground while transistor TR1 is switched off, this resonant series circuit would cause demodulation of the PWM signal. Transistor TR2 therefore shunts any potentially demodulated signal to ground while transistor TR1 is switched off.

[0040] According to the invention, the transistor TR2 can also be replaced by a non-switching element, which establishes a connection to ground, such as a shunt resistor or a cascode bias stage.

[0041] According to the invention, additional power efficiency can also be achieved by designing the power amplifier 311 to operate stably only for impedance values ​​that are present when either transistor TR1 or transistor TR2 is switched on. Without transistor TR2, the power amplifier 311 would be exposed to a wider range of impedance values ​​(that is, a range that includes both the output impedance of transistor TR1 when it is conducting current and the output impedance of transistor TR1 when it is not conducting current). With transistor TR2, however, the impedance that the power amplifier 311 sees at its input does not change as much, since transistor TR2 is switched on when transistor TR1 is not conducting current or is switched off.The design of the power amplifier 311 can therefore be simplified, as it is no longer necessary for the power amplifier 311 to operate stably over a wide range of input impedances. Thus, the stability required by the prior art over a large impedance range can be effectively replaced by improved gain and power efficiency.

[0042] According to the invention, the switching amplifier 300 can be implemented with discrete 28 V CMOS LDMOS components. The source terminals of these LDMOS components can be coupled to ground and have a housing to allow direct heat conduction. Of course, other components can also be used according to the invention. The present invention can, for example, also be built in a fully integrated configuration, in which both amplifier stages share the same substrate. In this case, the components of the switching amplifier 300 can also be located on the same substrate as other circuits in the system, such as the signal generator 102 ( Fig. 1a and Fig. 1b) and other RF system components and baseband system components.

[0043] Another embodiment of the amplifier 400 operating in switching mode according to the invention is described in Fig. Figure 4 shows the switching amplifier 400, which represents a pseudo-differential version of the switching amplifier 300. Fig. 3a). At the in Fig. In the embodiment shown in Figure 4, the preamplifier 409, the matching network 402, and the power amplifier 411 are each configured such that the number of components has doubled in a pseudo-differential manner. For example, instead of transistor TR1 ( Fig. 3a) An input transistor TR1P and an input transistor TR1N, which are differentially driven by VINP and VINN during the periods in which the PWM signal is active. During the periods in which the PWM signal is not active, however, VINP and VINN are below the threshold voltage of transistors TR1P and TR1N. Instead of transistor TR2 ( Fig.3a) Transistors TR2P and TR2N exist, which are activated by the signals ENVa and ENVb respectively, as described below. For example, both transistors TR2P and TR2N are switched on when the input signal is inactive and when transistors TR1P and TR1N are switched off.

[0044] Since the amplifier 400, operating in switching mode, is differentially driven, the capacitors C3 can be combined into a single capacitor C3D, which is arranged between nodes 408n and 408p. The value of the combined capacitor C3D is half the value of the capacitor C3. The power amplifier 411 effectively consists of two non-differential power amplifiers, each coupled to a positive or negative phase of the preamplifier 409 and the matching network 402, respectively. According to the invention, the high-order bandpass filter 412 can be a non-differential filter, which is why a balun or symmetry transformer 416 is required to convert the differential signals VOUTP and VOUTN into a (non-differential) signal.

[0045] The pseudo-differential switching amplifier 400 offers the advantage of compensating for second-order interference caused by the active components TR1P, TR1N, TR3P, and TR3N, as is known in the prior art. Furthermore, for a given supply voltage, it allows for a doubling of the available voltage deflection, which is advantageous when high output power is required. In addition, the pseudo-differential switching amplifier 400 offers advantages over a fully differential implementation because the source terminals of components TR1P, TR1N, TR2P, TR2N, TR3P, and TR3N can be directly coupled to ground. Direct ground coupling improves the quality of the available AC ground and enables better heat dissipation and power efficiency.Of course, according to the invention, a fully differential version of the amplifier 400 operating in switching mode can also be used.

Claims

[1] System for amplifying an RF signal generated in pulse operation, wherein the system comprises (300; 400): a first amplifier (313; 409), wherein the first amplifier (313; 409) is configured to amplify the RF signal (320) which is applied to an input (Vin; VINP, VINN) of the first amplifier (313; 409), a switchable impedance element (TR2; TR2P, TR2N) which is coupled to an output of the first amplifier (313; 409), wherein the switchable impedance element (TR2; TR2P, TR2N) is designed such that it includes a first impedance, when the RF signal (320) is active, and that it includes a second impedance when the RF signal (320) is not active, and a second amplifier (311; 411) which is coupled to the output of the first amplifier (313; 409), wherein the switchable impedance element comprises an LDMOS transistor (TR2; TR2P, TR2N). [2] System according to claim 1, characterized by , that the switchable impedance element (TR2; TR2P, TR2N) is controlled by an envelope (322) of the RF signal (320). [3] System according to claim 1 or 2, characterized by , that the system (300; 400) also includes an matching network (302; 402) between the output of the first amplifier (313; 409) and an input of the second amplifier (311; 411). [4] System according to any one of the preceding claims, characterized by , that the second amplifier includes a power amplifier (311; 411). [5] System according to any one of the preceding claims, characterized by , that the system (300; 400) also includes a control to switch off the first amplifier (313; 409) when the RF signal (320) is not active. [6] System according to any one of the preceding claims, characterized by , that the first amplifier (409) and the second amplifier (411) each comprise a differential amplifier. [7] System according to any one of the preceding claims, characterized by , that the switchable impedance element comprises several switchable impedances (TR2P, TR2N). [8] System according to any one of the preceding claims, characterized by that the system also includes several drive amplifiers, several secondary amplifiers and several switchable impedance elements. [9] System according to claim 8, characterized by that the multiple drive amplifiers and the multiple second amplifiers are coupled in series. [10] Amplifier circuit for pulse operation, comprising: a first amplifier stage (313; 409) comprising an input (Vin; VINP, VINN) coupled to a switched RF signal source, a switchable impedance element (TR2; TR2P, TR2N) which is coupled to an output of the first amplifier stage (313; 409) and has a first impedance value when the switched RF signal source is active, and has a second impedance value when the switched RF signal source is not active, and an output stage (311; 411) which is coupled to the output of the first amplifier stage (313; 409), wherein the switchable impedance element comprises an LDMOS transistor (TR2; TR2P, TR2N). [11] Amplifier circuit according to claim 10, characterized by, that the first amplifier stage (313; 409) comprises a first transistor (TR1; TR1P, TR1N) which is coupled to the switched RF signal source, and that the switchable impedance element comprises the LDMOS transistor (TR2; TR2P, TR2N) as a second transistor (TR2; TR2P, TR2N), wherein the second transistor (TR2; TR2P, TR2N) is off when the switched RF signal source is active, and on when the switched RF signal source is not active. [12] Amplifier circuit according to claim 11, characterized by , that the amplifier circuit (300; 400) also includes an matching network (302; 402) which is coupled between the output of the first transistor (TR1; TR2P, TR2N) and an input (310) of the output stage (311; 411). [13] Amplifier circuit according to claim 12, characterized by, that the switched RF signal source is designed such that it provides a pulse-modulated signal (320), and that the second transistor (TR2; TR2P, TR2N) is designed such that it prevents the matching network (302; 402) from demodulating the pulse-modulated RF signal (320) provided by the switched RF signal source. [14] Amplifier circuit according to one of claims 11-13, characterized by , that the amplifier circuit also includes a control to switch off the first transistor when the switched RF signal source is not active. [15] Amplifier circuit according to one of claims 11-14, characterized by, that the impedance of the second transistor (TR2; TR2P, TR2N) when the second transistor (TR2; TR2P, TR2N) is switched on is essentially the same value or at least of the same order of magnitude as the impedance of the first transistor (TR1; TR1P, TR1N) when the first transistor (TR1; TR1P, TR1N) is switched on. [16] Amplifier circuit according to one of claims 11-15, characterized by , that the impedance of the second transistor (TR2; TR2P, TR2N), when the second transistor is switched on, has a value between approximately 5 ohms and approximately 20 ohms. [17] Amplifier circuit according to one of claims 10-16, characterized by , that the output stage includes a power amplifier (311; 411). [18] Method for amplifying a signal (320) which is generated in pulse operation, the method comprising: Applying the signal (320) to a first amplifier (313; 409), Coupling a switchable impedance element (TR2; TR2P, TR2N) with an output of the first amplifier (313; 409), controlling the switchable impedance element (TR2; TR2P, TR2N), wherein the controlling includes: Causing the switchable impedance element (TR2; TR2P, TR2N) to assume a first impedance value when the signal (320) is active, and Causing the switchable impedance element (TR2; TR2P, TR2N) to assume a second impedance value when the signal (320) is not active, and Coupling a second amplifier (311; 411) with the output of the first amplifier (313; 409), wherein the switchable impedance element comprises an LDMOS transistor (TR2; TR2P, TR2N). [19] Method according to claim 18, characterized by , that the procedure also includes switching off the first amplifier (313; 409) when the signal (320) is not active. [20] Method according to claim 18 or 19, characterized by, that the first impedance value is a high impedance value, and that the second impedance value is an impedance value which is on the order of magnitude of the output impedance of the first amplifier (313; 409) when the signal is active.

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