Device for low-interference and ISO26262 compliant operation of an inductive rotary angle sensor with m outputs

Abstract

Device for operating and evaluating an inductive rotary angle sensor head (303) a. with m, but at least three, inductive receiving loops (S1, S2, S3) and b. with a field-inducing excitation loop (2) and c. with a runner with a rotor loop distorting the field of the excitation loop (2), d. with at least one signal generator (320) which generates at least one excitation signal (322) and e. with at least one electrical current or voltage source (301) modulated by means of the excitation signal (322), which excites the excitation loop (2) of the inductive rotary angle sensor head (303), and f. wherein in particular the star voltage of the receiving loops (S1, S2, S3) of the inductive rotary angle sensor head (303) is zero volts in the ideal error-free case, characterized in that g. that the device has at least one sub-device, a signal transformation block (307), for performing a signal coordinate transformation, in particular an affine signal coordinate transformation, for an output signal vector (304) of the rotary angle sensor head (303), consisting of output signals (304) of the m receiving loops (S1, S2, S3), for forming n, but at least three and preferably m, transformed partial signals (308,i) of the transformed signals (308) and h. that for each transformed partial signal (308,i) of the transformed signals (308) it has at least one sub-device with two inputs (SIG1, SIG2) which, from the excitation signal (322) or a control signal correlated thereto for synchronous demodulation (321) as the first input signal (SIG1) of the sub-device and the respective transformed partial signal (308,i) as the second input signal (SIG2) of the sub-device, generates for each transformed partial signal (308,i) a level signal (310,j) of a total of k, but at least three - preferably n or m, level signals (310) associated with the sub-device and the respective transformed partial signal (308,i) by synchronous demodulation and / or scalar product formation, wherein this sub-device can in particular be a synchronous demodulator.

Description

Introduction

[0001] Various inductive angle sensors are known in the state of the art. One example is DE 197 38 839 A1. Such sensors are also used in safety-relevant applications, for example. Various standards exist for the requirements of such safety-relevant sensors. Examples include the ISO 26262 standard for the automotive sector or the ISO 62508 standard, which generally defines various safety integrity levels, or safety classes.

[0002] In order for a rotation angle sensor, in particular a rotation angle sensor based on an inductive measuring principle according to DE 197 38 836 A1, to meet the requirements of such standards, the evaluating electronics must meet certain requirements that arise from the difficulties of such rotation angle sensor systems.

[0003] For a better introduction to the technology used in DE 197 38 836 A1, the basic principle of such an inductive angle of rotation sensor is briefly described again at this point in order to work out the requirements for the evaluation electronics of such an inductive angle of rotation sensor and to illustrate the problems.

[0004] An inductive angle sensor according to DE 197 38 836 A1 is, in a first approximation, rotationally symmetric with an n-fold rotational symmetry (see Fig.1). It typically consists of a carrier, for example made of a conventional circuit board material, onto which typically three conduction loops, the receiver loops (S1, S2, S3), are applied in a meandering pattern. The meander structures of the receiver loops (S1, S2, S3) have an angular periodicity that is equal and constant with respect to the orbital angle. This can be referred to as an angular wavelength (Δφ) of the meander structures of the receiver loops (S1, S2, S3). The typically three receiver loops (S1, S2, S3) are rotated relative to each other by typically one-third of the said angular wavelength (Δφ). If m receiver loops (S1, S2,....Sm) are used instead of three receiver loops (S1, S2, S3), the rotation is typically only 1 / m of the angular wavelength (Δφ) of the meander structures. In the following, three receiver loops are discussed without limiting the number to these with regard to this disclosure.In addition, at least a fourth conduction loop, the excitation loop (2), is mounted rotationally symmetrically to the axis of symmetry (Sym), preferably without a meandering structure. This entire construction is typically used in a stationary position and is referred to below as the stator.

[0005] The rotation axis, which is to be measured in its angle of rotation relative to the stator, passes perpendicular to the image plane of the Fig. 1 the stator at its symmetry point, which is the intersection point of the system's symmetry axis (Sym). A rotor is mounted on this rotational axis, which itself serves as an electrically non-conductive support for a short-circuit loop, the rotor loop. This rotor loop preferably has the same structure and angular wavelength (Δφ) as the receiving loops (S1, S2, S3), which are preferably all identical to each other except for their relative rotation.

[0006] The excitation loop (2) now transmits a primarily magnetic signal that induces a current in the rotor. Since the excitation loop (2) is rotationally symmetrical with respect to the axis of symmetry (Sym) and thus rotationally symmetrical with respect to the axis of rotation, this induction into the rotor loop of the rotor is independent of its angular position and thus of the angle of rotation of the rotor relative to the stator, which is to be measured. The rotor, and thus the rotor loop, are typically firmly connected to this axis of rotation, which runs along the axis of symmetry (Sym). Therefore, the induction depends on the angle of rotation of the axis of rotation relative to the stator.

[0007] Likewise, the excitation loop (2) induces a current into each of the three receiver loops (S1, S2, S3). This current should be the same for all receiver loops (S1, S2, S3), since they should have an identical geometry and differ only by an angle of rotation (Δφ / m) around the rotation axis, i.e., the axis of symmetry (Sym). However, this angle is irrelevant due to the rotational symmetry of the excitation loop (2).

[0008] However, depending on the rotational position of the rotor with its rotor loop relative to the three receiver loops (S1, S2, S3), the rotor induces a different current into the respective receiver loops (S1, S2, S3). The back induction can be compared to the induction in an asynchronous motor, where the squirrel-cage rotor induces fields in the three stator coils (see, for example, column 5, lines 45 to 46 of DE 197 38 836 A1).

[0009] As always, such a system cannot be operated smoothly and error-free in reality. Since such inductive angle sensors are also intended for use in safety-relevant applications, particularly in automobiles, it is necessary to classify and correct these errors in electronic post-processing, unless they can be intercepted during the design of the sensor element.

[0010] For this purpose, DE 197 38 836 A1 (see, for example, column 5, lines 45 to 49 of DE 197 38 836 A1) proposes operating the three receiver loops (S1, S2, S3) in star connection and using the star point to correct interference, since this should have a value of 0V due to the design of the system, without proposing a specific method for this. The excitation loop of the inductive angle sensor head (303) is Fig.2 is excited with an alternating current by an excitation current source (301) using an excitation current that is an alternating current. As a result, if the inductive angle sensor head (303) functions correctly, its receiver loops (S1, S2, S3) each individually display an alternating voltage signal. The sum of these alternating voltage signals (304) is zero in the case of a star connection if the said inductive angle sensor head (303) functions correctly.

[0011] However, the interference compensation in this star connection is insufficient. In particular, the star connection is not capable of reliably suppressing common-mode signals that are coupled into the sensor system (300) via the excitation loop (2) in the event of EMC radiation. This limits the gain of the downstream amplifiers and thus reduces the robustness of the system.

[0012] For this reason, the method for correcting interference disclosed in DE 197 38 836 A1 is particularly unable to reliably detect certain common-mode faults. For example, a lack of excitation in the excitation loop (2) cannot be reliably detected and classified.

[0013] Furthermore, it was recognized within the scope of the invention that the star connection proposed in DE 197 38 836 A1 for interference suppression is also disadvantageous because it favors common-mode signals and thus leads to increased sensitivity, for example, to alternating magnetic fields that flow through the receiver loops (S1, S2, S3). This additional common-mode signal, which by definition should have the same amplitude in all three receiver loops (S1, S2, S3), leads to a loss of amplitude bandwidth in the subsequent three input amplifier stages for the output signals of the receiver loops of the inductive angle sensor head (303). If insufficient amplitude reserve is provided at their inputs, the input amplifiers can be overloaded.Since this overload typically does not occur simultaneously in all three input amplifiers, the faulty common-mode signal can ultimately be distorted into a differential-mode signal, thus leading to an incorrect angle measurement result. However, such an incorrect measurement resulting from a common-mode signal is considered a critical error in safety-critical applications and must therefore be reliably prevented or at least classified and detected. This is not possible with the current technology.

[0014] A similar device is known from DE 197 38 839 A1.

[0015] All of these aforementioned devices typically operate with a monofrequency signal. This makes them susceptible to narrowband interference.

[0016] DE 197 38 839 A1 discloses a device for operating and evaluating an inductive angle sensor head. It comprises more than two inductive receiving loops, a field-inducing excitation loop, a rotor with a rotor loop that distorts the field of the excitation loop, a signal generator that generates an excitation signal, and an electrical current or voltage source modulated by the excitation signal, which excites the excitation loop of the inductive angle sensor head. The star voltage of the receiving loops of the inductive angle sensor head is zero volts in the ideal, fault-free case.

[0017] Furthermore, the document presented here refers to the documents DE 197 38 836 A1 and DE 10 2010 031 142 A1.

[0018] Furthermore, all these devices are not capable of detecting, classifying and reliably distinguishing more complex disturbance modes as such. Object of the invention

[0019] The object of the invention is to provide a method in which the signals provided by the sensor can be sufficiently distinguished from the background noise and at the same time can be evaluated in such a way that the following errors can be detected, classified and differentiated: a) Common mode faults such as stray fields, breakage of the excitation loop, breakage of the rotor, etc. (hereinafter referred to as symmetrical fault conditions) b) Differential faults such as breakage of a receiver loop or connection to it, damage to the amplifier inputs, short circuits on the stator or rotor, EMC, etc. (hereinafter referred to as asymmetric fault conditions) c) Other errors that can be described by an error model.

[0020] To solve this problem, the significance of detection must be increased.

[0021] This object is achieved by a device according to claim 1.

[0022] In the state of the art, errors a to c are not reliably detected or classified in such sensor systems as described and are therefore not reliably distinguishable. Description of the invention

[0023] The sensor system (300) of the device according to the invention, which carries out the method according to the invention, is first Fig. 2. All figures below are simplified for clarity. The scope of the invention is determined solely by the claims.

[0024] The excitation loop (2) of the inductive angle sensor head (303) is again excited with an excitation current from the excitation current source (301), as previously described in DE 197 38 836 A1. Again, when the inductive angle sensor (303) is functioning correctly, its receiver loops (S1, S2, S3) each individually display an alternating voltage signal in the form of the output signals (304) of the inductive angle sensor head, with each of the output signals (304) of the inductive angle sensor head (303) being assigned to a receiver loop (S1, S2, S3). The sum of these output signals (304) of the inductive angle sensor head (303) should be zero in the case of a star connection if the said inductive angle sensor head (303) is functioning correctly. Typically, but not necessarily, the inductive angle sensor head (303) is already connected in a star configuration.In the following, the invention is described using the example of a star-connected angle sensor head, without limiting the invention thereto.

[0025] The output signals (304) of the rotation angle sensor head (303) are pre-filtered in an EMC filter block (305) to form filtered output signals (306) of the rotation angle sensor head (303). Typically, only those signal components are passed through the EMC filter block (305) that correspond to the excitation signal (322) used to control the excitation current source (301) that supplies the excitation current for the excitation loop (2) or that lie within the bandwidth of the excitation signal (322). Preferably, one filter is provided for each output signal (304) and / or each receiving loop (S1, S2, S3) of the inductive rotation angle sensor head (303).

[0026] According to the invention, however, it has now been recognized that it is more expedient to evaluate other, more significant signals after a signal coordinate transformation for the purpose of error detection instead of the exemplary star-point voltages, more precisely the m output signals (304) of the rotation angle sensor head (303). For this purpose, it was recognized that the signals of the m star-point voltages are m-dimensional vectors with m signals as vector elements, which have both a temporal and an m-dimensional extension. It has been recognized according to the invention that an affine mapping of this m-dimensional output signal vector (304) to a more significant signal vector allows the detection of more complex error states, in contrast to the prior art.According to the invention, it was further recognized that after this vector signal coordinate transformation, pattern recognition can be performed by forming a scalar product with prototypical error model signal vectors, which can be used to detect more complex faulty states. Only the combination of this signal coordinate transformation in the form of an affine mapping and the subsequent scalar product formation yields the desired result of increased significance and robustness, as well as the sensor system's ability to classify and differentiate between system states.

[0027] In the exemplary case of affine mapping, the exemplary output voltages at the outputs (304) of the inductive angle of rotation sensor (303) with, for example, m=3 receiving loops (S1, S2, S3) are now evaluated for simplified explanation. Since the induction of a common-mode interference should be the same in all m, here, for example, three receiver loops (S1, S2, S3), n, here, for example, three transformed signals of the m, here, for example, three star voltages, should no longer contain any common-mode signal components. Such a preferred transformation is described below without limiting this disclosure thereto. It is essential to recognize that the output signals (304) of the angle of rotation sensor head (303) must first be subjected to a transformation in order to be able to be adapted to the subsequent circuits.This transformation is to be carried out in a signal transformation block (307) which receives its input signals either directly from the rotation angle sensor head (303) as its output signals (304) or first pre-processes them, for example filtered as a filtered output signal (306) of the rotation angle sensor head (303) via a filter block (305).

[0028] Such an exemplary signal transformation, as an example of the affine signal vector transformation for significance enhancement, is carried out in the said signal transformation block (307), which is in Fig. 4 is shown in more detail. From the filtered output signals (306) of the rotation angle sensor head (303), for example, three differential amplifiers (401 to 403) each form a signal voltage of a transformed signal (308) by forming a respective difference.

[0029] This corresponds to the technical realization of the vector equation V→308=A↔307V→304+V→off

[0030] Here V means 304 the m-dimensional signal vector of the m signal voltages of the m output signals (304) of the angle sensor head (303). We have not always included the filtering of the output signals (304) of the angle sensor head (303) to form the filtered output signals (306) in this description, since it is obvious to those skilled in the art that filtering can also mean directly looping the signals, whereby the output signals (304) of the angle sensor head (303) can also be connected directly (304) or filtered (306) to the subsequent signal transformation block (307). 308 stands for the n-dimensional signal vector of the signal voltages of the n transformed signals (308). A 307represents a link matrix between the m-dimensional signal vector of the m signal voltages of the m partial signals (304,I) of the output signals (304) of the rotation angle sensor head (303) or the m filtered output signals (306) on the input side of the signal transformation block (307) and the n-dimensional signal vector of the n signal voltages of the n partial signals (308,i) of the transformed signals (308) of the signal transformation block (307) on the other side. V off is an n-dimensional offset vector which is converted by the signal transformation block (307) to the result of the rotation-stretch transformation by means of the link matrix (A 307 ) is added. In the case of an arbitrary transformation chosen only as an example for the representation of the invention, the link matrix (A 307 ) can be designed, for example, as follows: A↔307=(1−1001−1−101)

[0031] Such an exemplary link matrix (A 307 ), which is filled with ones, negative ones and zeros, can be realized by simple subtraction, for example in differential amplifiers. (see Fig. 4)

[0032] The transformed signals (308) according to the invention contain, when applying the above linking matrix (A 307 ) and thus preferably no longer contains a DC component after signal vector transformation in the signal transformation block (307). However, it is also possible, for example, to project the DC component out as a separate signal. The exemplary combination matrix (A 307 ) is then as m×n matrix no longer a 3×3 matrix in the exemplary case of m=3, but a 3×4 matrix, A↔307=(1−1001−1−101111) if the sensor system (300) is otherwise symmetrical, as intended in the fault-free case. Symmetrical here means that all receiver loops (S1, S2, S3), except for the aforementioned rotation around the axis of rotation, i.e., the axis of symmetry (Sym), have the same geometry with n-fold symmetry and are arranged exactly centrally, and that this also applies to the rotor loop. In this ideal case, the excitation loop (2) is also mounted completely symmetrically with respect to the axis of symmetry (Sym).

[0033] By means of the signal transformation, the voltage amplitude of the transformed signals (308) can also be increased with respect to their useful signal component by a value determined by the selected transformation (A 307 ) dependent factor, here for example 3, be greater than the amplitudes of the output signals (304) of the inductive rotation angle sensor head (303). This increases the level difference between the useful signal of the receiving loops (S1, S2, S3) after transformation and a potentially radiated interference signal, for example, an radiated common-mode interference signal, and, secondly, and this is essential, the control range of the subsequent amplifier stages is better utilized with regard to the useful signals.

[0034] Is the effect of a possible interference signal V stör known, this can be fed into the signal path in an exemplary linear relationship, for example by means of a weighting vector G, which can be measured or calculated from design data: V→308=A↔307V→304+V→off+G→Vsto¨r

[0035] In the case of an example common mode signal, G→Vsto¨r=(111)Vsto¨r

[0036] The interference voltage fed into the rotation angle sensor head (303) with three output signals (304).

[0037] In this exemplary case, the matrix multiplication in the signal transformation block (307) results in: A↔307G→Vsto¨r=(1−1001−1−101111)(111)Vsto¨r=(0 0 0 0 3)Vsto¨r

[0038] Here, we assumed G to be G=(1,1,1). In the first three channels, the common-mode signal is reliably suppressed, while the interference signal is amplified in the fourth channel. This makes the error pattern more significant and easier to detect, as it is separated from the useful signals in the first two channels.

[0039] This previous example is therefore an excellent example to illustrate the significance enhancement by an affine vector signal coordinate transformation in the signal transformation block (307).

[0040] Typically, the device according to the invention in a motor vehicle is supplied with a simple DC voltage that is not symmetrical about the zero point. Therefore, in a preferred version of the device according to the invention, a DC voltage vector signal (V off ), the offset, for example by differential amplifiers (401 to 403) of the Fig. 4 is added. In the example case, this offset (V off ) for each partial signal (308,i) preferably half of the operating voltage (V BD ) and / or is placed in the middle of the input amplifier's dynamic range. Typically, the offset voltages are the same for all channels. This state-of-the-art adding circuit is shown in Fig. 4 not shown additionally for simplification.

[0041] While the increase in significance described above by signal coordinate transformation in the exemplary form corresponds to a matrix multiplication of the output vector signal (304) of the angle measuring head (303), this addition of the offset now represents a shift of the resulting signal vector. Such a multiplication of a vector signal with a matrix, here a predetermined linking matrix (A 307 ) for a hard-wired signal coordinate transformation, in combination with an addition, here the addition of a fixed vector signal offset (V off) by adder circuits, i.e., a quasi vectorial adding device, represents an affine mapping of the output signal vector to a new signal vector, here the transformed signals (308). In addition to these affine mappings, more complex mappings are also possible. For example, a complete and / or partial feedback of the output signal vector of the signal transformation block (307) consisting of the transformed signals (308) as part of the input signals of the signal transformation block (307) into the signal transformation block (307) is conceivable, thereby enabling non-linear transformations, which are part of this disclosure (see also Fig. 12). This feedback can be weighted and mixed, for example, by a weighting matrix (G2).

[0042] Following the significance increase in the signal transformation block (307), the output signals of the signal transformation block (307), the transformed signals (308), are checked for the presence of various features. The most important feature is the presence of the measurement signal itself. The significance increase in the signal transformation block (307) is therefore preferably carried out in such a way that the measurement signal can be found on other sub-signals of the transformed signals (308) than the levels of the interference signals for error detection. The detection of measurement signals and error signals is now carried out by comparison with prototypical signals. This is done by forming a scalar product between the respective prototype signal of a prototypical signal vector, which can also be one-dimensional, and the respective output signal (308,i) of the signal transformation block (307), the respective transformed signal (308,i). The individual measurement signal level (V mess), which here is a measurement signal level for the respective prototype and can therefore also be an error signal level, can be determined, for example, by the formula ∑i=1nV308,i∗Vproto,i=∑i=1n(∑j=1mA307,ijV→304,j+V→off)∗Vproto,i=Vmess,i be described.

[0043] Preferably, the excitation signal (322) is used as a prototypical signal, typically one of several, e.g., u, prototypical signals. It is also useful to use an excitation signal (322) shifted by a predetermined phase (e.g., +90°) and / or a predetermined time ΔT as a further prototypical signal, for example, to be able to determine the angle directly as the arccos of the ratio of the resulting amplitudes.

[0044] The sum of the amplitudes of the signal voltages of the partial signals (308,i) of the transformed signals (308) should, for example, in the presented combination matrix (A 307) for the signal transformation to increase significance in the signal transformation block (307) must be zero volts. If this is not the case, a sum error, for example, may be detected.

[0045] According to the invention, it was recognized that this sum of the amplitudes of the transformed signals (308), i.e. not the sum of the star point voltages of the output signals (304) of the angle measuring head (303) as in the prior art itself, can be evaluated to assess the sensor state. According to the invention, it was recognized that in particular task-related error states can be detected, classified, and differentiated that affect the sensor and / or the evaluation in an asymmetric or symmetric manner. Common-mode errors influence the sum of the amplitudes of the signal voltages of those signals of the transformed signals (308) that represent the measured values, according to the significance increase with the above-mentioned linking matrices (A 307) in the signal transformation block (307) is no longer present. Thus, according to the invention, the detection of differential-mode errors and their differentiation from common-mode errors, as well as their differentiation from other errors, is possible. This is not known in the prior art. In the device according to the invention, the operating voltage is subtracted once from the voltage sum, or the aforementioned offset is subtracted twice, as a result of which the sum signal (326) is again in the middle of the control range of subsequent amplifiers.

[0046] This formation of the sum signal (326) by summing the level signals (310) that form the output signals of the synchronous demodulator block (307), in which the scalar product formation takes place, takes place, for example, in a summing device (324). The formation of the sum signal itself, incidentally, corresponds to a scalar product formation between the signal vector of the transformed signals (308) and a unit vector signal whose elements are occupied by constant one-signals.

[0047] The sum signal (326) thus formed is compared, for example, by means of a first comparator (341) with a first threshold value (327), for example in the form of a first reference voltage (V ref1 ) is evaluated by comparison, whereby the reference voltage (V ref1) is preferably above said offset, and for example by means of a second comparator (342) with a second threshold value (330), for example in the form of a second reference voltage (V ref2 ), which is preferably below the offset. This typically results in a first result signal (334) and a second result signal (333). If the voltage of the sum signal (326) lies outside the range defined by the first reference voltage (V ref1 ), i.e. the first threshold value (327), and the second reference voltage (V ref2 ), i.e., the second threshold value (330), a symmetry error, i.e., a differential-mode error, is present, which is to be detected according to the object of the invention. Furthermore, the device recognizes that this error is not a common-mode error, i.e., not a symmetrical error condition.

[0048] In the signal transformation block (307), the corresponding transformed signals (308) of the output signals (304) of the inductive angle sensor head (303) are transformed, as described above, using the described affine mapping. Depending on the selected link matrix (A 307 ) the DC interference is preferably removed, as described, and overloading of the subsequent amplifier inputs is thus less likely. The headroom in the subsequent signal path increases. Against this background, it is often useful to perform the signal transformation and scalar product formation several times, thus in multiple stages, one after the other, and in the first transformation, particular emphasis is placed on preventing overloading, for example, by a suitable choice of the combination matrix (A 307). For the sake of simplicity, only a single-stage transformation without feedback and filters is described here and below. The three transformed signals (308) thus formed are then scalar multiplied and filtered in a synchronous demodulator block (309) to transform the excitation signal (322) of the excitation current source (301) into a DC voltage signal or low-frequency signal corresponding to the respective reception level.

[0049] A clock generator (320) supplies, on the one hand, the excitation signal (322) for the excitation current source (301) and, on the other hand, simultaneously a control signal (321) for synchronous demodulation by scalar product formation. Strictly speaking, the transformed signals (308) are scalar-multiplied by the control signal (321) in the form of the excitation signal (322) of the excitation current source (301) and then low-pass filtered. This is a process familiar from synchronous demodulators. A scalar product is formed between the excitation signal (322) of the excitation current source and the respective transformed signal (308) in the synchronous demodulator block (309) in space multiplex by n, here, three separate synchronous demodulators, as an example. Time-division multiplexing is, of course, also possible with digital sampling. The low-pass filtering suppresses signal components with higher frequencies.In principle, instead of using scalar products, it is also possible to use any other linear forms, provided that the orthogonality of two different signals can be determined under determinable conditions with regard to the difference between these signals in the form of the disappearance of the result (V. mess,i ) the combination of these signals can be achieved using this linear form. Fig.Figure 3 shows a simple implementation of a unit for scalar product generation. A first signal (SIG1) is multiplied by a second signal (SIG2) in a multiplier (M1) and filtered in the downstream low-pass filter (F1) to produce the output signal (OUT). If one of the signals is a binary signal, for example, the first signal (SIG1), the multiplier can be implemented very easily. For example, the multiplier could be an impedance converter or similar device that can be restructured using switches activated by the first signal (SIG1) so that, depending on the first signal (SIG1), it has a gain of 1 at one time and a gain of -1 at another. Typically, the offset is subtracted beforehand and then added back up.If the second signal (Sig2) then also changes its sign synchronously with the first signal (Sig1) after offset subtraction, the resulting signal is always positive, the DC component of which is then separated by the subsequent low-pass filter. Other structures for scalar product formation are, of course, conceivable. The cutoff frequency of the low-pass filter (F1) is chosen so that it lies below the minimum frequency (ω). min ) of the excitation signal (322).

[0050] For the purposes of this disclosure, any two signals A and B are referred to as orthogonal with respect to a synchronous demodulator if they were fed into one of the synchronous demodulators of the synchronous demodulator block (309) instead of the first signal (SIG1) and the second signal (SIG2), and the associated output signal, i.e., the associated level signal (310), would then be 0. In reality, however, such a synchronous demodulator of the synchronous demodulator block (309) will exhibit system noise and a control error. This system noise and this control error can be unambiguously determined by feeding the excitation signal (322) into the respective synchronous demodulator instead of the first signal (SIG1) and simultaneously keeping the second signal (SIG2) of the respective synchronous demodulator constant. The output signal of the synchronous demodulator would then ideally be zero, but in reality, it typically exhibits the aforementioned noise and control error.

[0051] In the example of the synchronous demodulator block (309), for example, each of the transformed signals (308) is connected to a first signal input (SIG1) of a device for scalar product formation corresponding to Fig. 3. All second signals (SIG2) of these scalar product formation devices are typically connected to the synchronous demodulation control signal (321) from the signal generator (320). This synchronous demodulation control signal (321) typically correlates with the excitation signal (322). All three of these scalar product formation devices together form the synchronous demodulator block (309). Each of the three output signals (OUT) of the three scalar product formation devices is connected to one of the three level signals (310).

[0052] Preferably, in a special embodiment of the invention, the excitation signal (322) formed by the clock or signal generator (320) is a bandwidth-limited signal with a lower limit frequency ω min and a different upper limit frequency ω max , which should be different. The lower limit frequency ω min greater than half the frequency bandwidth Δω= ω max ,- ω min , The low-pass filter is now preferably designed so that it filters out signal components with a frequency greater than the lower cutoff frequency ω minand / or preferably greater than half the frequency bandwidth Δω / 2, and passes frequency components below half the frequency bandwidth Δω / 2 with as little attenuation as possible. All statements refer to the frequency magnitudes. The result of this scalar product formation are three level signals (310), the voltage value of which is typically a respective measure of the magnitude of the induction of the excitation signal (322) into the receiving loops (S1, S2, S3) of the inductive rotation angle sensor head (303).

[0053] According to the invention, instead of simple mono-frequency excitation signals (322), as in the prior art, spreading code-based excitation signals (322) are now generated by the signal generator (320). For this purpose, the respective spreading code is typically transmitted repeatedly. This results in a band-limited signal. The use of digital spreading codes consisting of v ordered bits is particularly preferred. In this case, it is particularly preferable if the v bits of the spreading code have an equal number of 1-bits and 0-bits. Only in this way can it be ensured that the generated excitation signal (322) has the angular frequency (ω=0s -1 ). Rather, it is desirable that the generated excitation signal (322) has a minimum frequency (ω min ) below which the amplitude of the excitation signal (322) disappears or is insignificant. This lower, minimum frequency (ω min) is determined by the inverse of the clock period T0, with which the bits of the spreading code are transmitted. The upper frequency (ω max ) is determined by the inverse of the transmission duration of a spreading code (T1=n*T0).

[0054] These spreading codes can be based on random signals and / or pseudorandom signals. The next spreading code to be transmitted can also be selected from among several spreading codes by a random signal at the end of the transmission of a previous spreading code.

[0055] The use of spreading codes is particularly useful in environments with particularly high levels of interference or when there are special reliability requirements.

[0056] During the development of the invention, it was recognized that it is advantageous to perform the synchronous demodulation or the scalar product formation after the significance increase in the signal transformation block (307) in the signal path, because then the symmetrical interference levels are already removed and an over- or under-modulation of the measurement signal path is less likely. It was thus recognized according to the invention that it is expedient to combine the signal coordinate system from the m, here three, output signals (304) of the inductive rotation angle sensor head (303) by the said matrix coordinate transformation with the linking matrix (A 307 ) in such a way that the significance of the resulting n-dimensional vector signal, here the transformed signals (308), is maximized. If the system were always affected in the same way by an asymmetric error, instead of the link matrix (A 307), as exemplified above, another equalizing matrix operation with an alternative combination matrix (A 307 ') by the signal transformation block (307). The above-specified link matrix (A 307 ) is just one possible example of such an operation. When reference is made to a significance increase in this disclosure, this typically means an affine mapping of the output signal vector from the output signals (304,1) of the output signal vector (304) of the inductive rotation angle sensor head (303) to an optimized transformed signal vector (308). In this disclosure, this is always referred to as "transformed signals (308)" for ease of understanding.

[0057] The invention thus consists of two processing stages: significance maximization in stage 1 by means of the signal transformation block (307) and correlation with excitation signals and error model signals that are as little correlated with interference as possible by means of the synchronous demodulator block (309) and the summer (324) in stage 2.

[0058] Such multi-stage, especially two-stage, processing is not known for inductive angle sensors from the state of the art.

[0059] However, the use of spreading codes increases the dead time of the sensor system. Therefore, in some cases, it is useful to dynamically switch the type of excitation signal (322) during operation and, for example, to switch the length of the spreading code depending on the permissible dead time and / or the required precision. A longer spreading code is used when greater precision or the suppression of significant interference is required; a shorter one, or even no spreading code at all, is used when a short dead time is required.

[0060] Up to this point in the signal processing sequence in the device according to the invention, processing has preferably been carried out in space division multiplexing. This means that up to this point in the signal path of the device according to the invention, there is preferably a signal path for each of the three receiver loops (S1, S2, S3), for example, or up to this point, three signal paths are preferably always run in parallel, for example. It has been shown that from this point onwards, it is advisable to carry out processing in the signal path preferably in time division multiplexing. For this purpose, for example, an analog multiplexer (311) switches cyclically between the k, here three, level signals (310,j) generated up to this point. The multiplexer output signal (312) is then preferably fed to a controllable amplifier (313).

[0061] The adjustable amplifier (313), with adjustable gain, amplifies the multiplexer output signal (312) to the amplifier output signal (314). Its gain is set once for all k, here three, level signals (310,j) by an amplifier control signal (318) to the same level for each measurement run. For the duration of the measurement run, in which all k, here three, level signals (310,j) of the level signals (310) are evaluated, the gain of the amplifier (313) remains the same and is only adjusted by a subsequent control logic (317) at the end of the measurement run after all k, here three, level signals (310,j) have been evaluated once by this control logic (317). This adjustment is carried out in such a way that at least one of the k, here three level signals (310,j) leads to a full modulation of the subsequent analog-to-digital converter (ADC, 315) without overloading it.For this purpose, the amplifier output signal (314) is fed to the analog-to-digital converter (ADC, 315), which forms the ADC output signal (316) therefrom. A control logic (317) downstream of the ADC (315) temporarily stores the conversion results of the ADC (315) in the form of the ADC output signal (316) during the measurement run and evaluates these conversion results of the ADC (315) in the form of the ADC output signal (316) at the end of the measurement run. The control logic (317) adjusts the gain of the amplifier (313) by means of the amplifier control signal (318) for one measurement run of all k, here three, level signals (310,j) up or down at the end of the measurement run, thereby closing the control loop for the gain of the amplifier (313).It should be noted that the information to be measured, namely the angle of rotation, is contained in the relative amplitudes of the level signals (310) and is precisely unchanged during a measurement run due to the uniform amplification for all level signals (310). The control logic (317) then controls the multiplexer (311) via a multiplexer control signal (319). The control logic (317) converts the k, here three, states of the ADC output signal (316) during the measurement of the k, here three different level output signals (310,j) into the desired measured value (345) of the angle of rotation.

[0062] According to the invention, it has now also been recognized that the value of the amplifier control signal (318) of the gain of the amplifier (313) is itself a parameter which is suitable for detecting a symmetrical error state, i.e. a common-mode error, of the rotation angle measuring system according to the invention.

[0063] If the value of the amplifier control signal (318) is above a third threshold value (335), the permissible maximum gain value, this may indicate, for example, a lack of excitation due to a lack of excitation current and / or an interrupted excitation loop or a missing rotor.

[0064] If the value of the amplifier control signal (318) is below a fourth threshold value (336), the permissible minimum gain value, this may indicate asymmetric radiation or an interrupted receiver loop (2).

[0065] For this purpose, the amplifier control signal (318) is compared with the said third threshold value (335), for example by a third comparator (343), and a third evaluation signal (337) is generated.

[0066] The amplifier control signal (318) is compared, for example, by a fourth comparator (344) with said fourth threshold value (336) and a fourth evaluation signal (338) is generated.

[0067] Furthermore, by comparing with a seventh threshold value and an eighth threshold value, it can be checked whether one level signal (310,j) of the level signals (310) is actually always in the upper control range, a second level signal (310,j) of the level signals (310) is in the middle control range, and a third level signal (310,j) of the level signals (310) is in the lower control range. This comparison is preferably performed by the control logic (317) at the end of a measurement run. Until then, the level values ​​of the individual level signals (310,j) are buffered by this control logic (317) after amplification in the adjustable amplifier (313) and digital conversion in the ADC (315), as described.

[0068] In its most general form, the invention thus relates to a device for operating and evaluating an inductive angle sensor head (303), which typically has three, more generally m, inductive receiving loops (S1, S2, S3). Furthermore, in the most general form, the angle sensor head (303) has one or more field-inducing excitation loops (2) and a rotor with one or more rotor loops that distort the field of the excitation loop (2). The device according to the invention has at least one signal generator (320) that generates at least one excitation signal (322). A modulated electrical current or voltage or energy source (301) excites the excitation loop (2) of the inductive angle sensor head (303) by means of said excitation signal (322).In contrast to the prior art, the device according to the invention has at least one sub-device for increasing the significance of the output signal vector of the rotation angle sensor head (303). In the example discussed above, this occurs in a signal transformation block (307). This sub-device carries out, for example, an affine mapping of the output signal vector (304) of the rotation angle sensor head (303) with m output signals to n new signal lines of a further transformed signal vector (308) of transformed signals (308, i). In the example above, these are the n, here three, transformed signals (308). Preferably, the dimension of this new signal vector is equal to the dimension of the output signal vector, i.e., m=n. However, this is not necessarily the case. Several mappings can also be connected in series. After this transformation mapping, the significance of the transformed signal vector (308) is increased.Incidentally, it is conceivable - although usually not opportune - that outputs of this block (307), which are part of its output signal vector (308), are again used as part of its input signal vector (304, 306), which leads to non-linear mappings.

[0069] After this signal coordinate transformation in the signal transformation block (307), a synchronous demodulator within a synchronous demodulator block (309) and a subsequent summation by means of a summer (324) forms a scalar product between a reference signal vector and the transformed signals (308) as the input signal vector of this synchronous demodulator block (309).

[0070] In the simplest case, the formation of the scalar product can be described as follows: P326=∑k=nF[(∑i=1mP304,i∗A307;ik+O307,k)P322]

[0071] The symbols have the following meaning: P 304,i: Level of the i-th output signal of m output signals of the inductive angle sensor head (303). These levels form the output signal vector of the angle sensor head. A 307,ik : Transformation matrix for the signal coordinate transformation of the output signal vector of the rotation angle sensor head (303) in the signal transformation block (307). This transformation matrix is ​​preferably hard-wired and leads to an increase in the significance of the resulting signal vector compared to the output signal vector of the rotation angle sensor head (303). O 307,ik : Offset signal vector for adapting the transformed output signal vector to the subsequent signal processing. P 322 : Level of the excitation signal (322). The level of this excitation signal (322) is multiplied by each vector signal component to form an intermediate signal vector. F[]: Low-pass filter function of the filter (F1). Without further differentiation, this is typically an integrating linear filter with low-pass characteristics. Typically, the upper cutoff frequency of this low-pass filter is chosen so that it lies below the minimum frequency (ω min ) of the excitation signal (322). The result of this filtering is the aforementioned level signals (310). The j-th level signal (310,j) gives, in particular with suitable, quadratic dimensioning of the linking matrix A i,j the amplitude of the component in the level of the j-th output signal (340,j) which correlates with the excitation signal (322). P 326: Level of the sum signal (326) of the level signals (310). Only through this summation is the total Euclidean distance between the excitation signal (322) and the output signal vector (304) of the angle sensor head (303) determined. This summation thus completes the scalar product formation. Extracting the square root to correctly determine the Euclidean distance is typically unnecessary in practice and would be too complex.

[0072] The device according to the invention thus has, for each transformed signal (308,i) of the transformed signals (308), at least one sub-device with two inputs (SIG1, SIG2), which generates, from the excitation signal (322) or a control signal correlating therewith for the synchronous demodulation (321) as the first input signal (SIG1) of the sub-device and the respective transformed signal (308) as the second input signal (SIG2) of the sub-device, a level signal of a total of k, but at least three, level signals (310) associated with the sub-device and the respective transformed signal (308,i) by synchronous demodulation and / or scalar product formation. This sub-device can, in particular, be a synchronous demodulator.

[0073] In a particular embodiment of the invention, this structure can now be used to detect other error cases and differentiate them from simple error modes such as common-mode and differential-mode errors. For this purpose, this embodiment has a fault model signal generator (1100) that generates a fault model signal (1101). This can be imagined, for example, as a distorted excitation signal. For example, a foreign body in the sensor can lead to a phase shift in the sensor's signal response in common mode and / or differential mode.

[0074] This error model signal vector (1101), which typically consists of several, for example u, error model signals (1101,u), is now linked to the output signals (304) of the rotation angle sensor head (303) in a similar manner.

[0075] In the simplest case, the scalar product formation for the error model vector signal can be described as follows: F1104=∑j=1nG1106,j∑k=1nF[(∑i=1mP304,i∗A307;ik+O→307,k)F1101,j]

[0076] The symbols have the following meaning: P 304,i Level of the i-th output signal of m output signals (304) of the inductive angle sensor head (303). These levels form the output signal vector of the angle sensor head. A 307,ik Transformation matrix for the signal coordinate transformation of the output signal vector (304) of the rotation angle sensor head (303) in the signal transformation block (307). This transformation matrix is ​​preferably hard-wired in the signal transformation block (307) and leads to a significance increase of the resulting signal vector of the transformed signals (308) compared to the output signal vector of the output signals (304) of the rotation angle sensor head (303). O 307,ikOffset signal vector for adapting the transformed output signal vector to the subsequent signal processing within the signal transformation block (307). F 1101,j Level of the j-th error model signal (1101). The level of this error model signal (1101) is multiplied by each vector signal component to form an intermediate signal vector. F[] Low-pass filter function of filter (F1). Without further elaboration, this is an integrating linear filter with low-pass characteristics. Typically, the upper cutoff frequency of this low-pass filter is chosen so that it lies below the minimum frequency of the excitation signal (322) and the error model signal (1101). The result of this filtering is the aforementioned error level signals (1110). These represent the amplitude of the component in the level of the i-th output signal (304) that correlates with the j-th error model signal (1101). G 1106Level of the weighting factor signal. This level can also be set to a constant level, i.e., hard-wired. This weighting factor signal is used to sum the scalar product results in the form of error level signals (1110) from the error synchronization demodulator block (1109) in the error summing device (1124). F 1104Level of the error sum signal (1104) of the error level signals (1110). Only through this weighted summation in the error summing device (1124) is the total Euclidean distance between the respective error model signal (1101) and the output signal vector (304) of the rotation angle sensor head (303) determined. This summation thus completes the scalar product formation. This summation typically takes place in the error summing device (1124) according to the level of the weighting factor signal (1106). Extracting the square root to correctly determine the Euclidean distance is typically unnecessary in practice and is too complex.

[0077] This embodiment of the invention thus has at least one sub-device with two inputs (SIG1, SIG2) for each transformed signal (308,i), which sub-device generates an error level signal (1102) associated with the sub-device and the respective transformed signal (308,i) from the error model signal (1101) as the first input signal (SIG1) of the sub-device and the respective transformed signal (308,i) as the second input signal (SIG2) of the sub-device for each transformed signal (308,i) by synchronous demodulation and / or scalar product formation, wherein this sub-device can in particular be a synchronous demodulator.

[0078] To complete the scalar product formation, the embodiment according to the invention has an error summing device (1124) which forms an error sum signal (1104) by summing a plurality of error level signals (1110) weighted with the weighting factor signal (1106).

[0079] Finally, it comprises an error detection unit (1142, 1141) that compares the error sum signal (1104) or a signal derived therefrom with one or more error thresholds (1127, 1130) to determine whether the error sum signal (1104) lies above or below one threshold or between two thresholds. It then outputs an error detection signal (1134, 1133) depending on the comparison result.

[0080] From the preceding text, the following list of thresholds results First threshold (327) for detecting a sum signal (526) of the level signals (310) that is too small Second threshold (330) for detecting an excessively large sum signal (326) of the level signals (310) Third threshold (335) for detecting an amplifier control signal that is too large (326) Fourth threshold (336) for detecting an amplifier control signal that is too small (326) Fifth threshold (527) for detecting a second sum signal (526) of the second level signals (510) that is too small (This threshold value will be explained later). Sixth threshold (530) for detecting an excessively large second sum signal (526) of the second level signals (510) (This threshold value will be explained later). Seventh threshold for detection by the control logic (317) of whether one of the level signals (310) is in the upper control range, the lower edge of which is defined by this seventh threshold value. Eighth threshold for detection by the control logic (317) as to whether one of the level signals (310) is in the lower control range, the upper edge of which is defined by this eighth threshold value. Ninth threshold for detection by the control logic (317) of whether at least one of the level signals (310) is in the upper control range but not yet in overload, the permissible upper edge of which is defined by this ninth threshold value. Further thresholds for evaluating the measurement result (345) (explained below)

[0081] Fig. 5 shows the exemplary sequence of an exemplary measurement run for measuring for exemplary three receiver loops (S1, S2, S3) in time division multiplex.

[0082] In a first step (600), a first of the k, here exemplary three, level signals (310) is selected by the multiplexer (311) and a first measured value (316) is determined, which is stored by the control logic (317).

[0083] In a second step (601), a second of the k, here exemplary three level signals (310) is selected by the multiplexer (311) and a second measured value (316) is determined, which is also stored by the control logic (317).

[0084] In a third step (602), a third of the k, here for example three level signals (310) is selected by the multiplexer (311) and a third measured value (316) is determined, which is typically also stored by the control logic (317), etc.

[0085] In the fourth step (603), the measured value (345) is determined from these k, here three, values ​​by the control logic (317), but is typically not yet output.

[0086] In the fifth step (604) the measured value (345) is typically evaluated.

[0087] If the measured value is evaluated as being OK by the control logic (317), in particular in comparison with other measured values ​​that were measured in parallel or previously, and / or with other threshold values, this is typically output in a sixth step (606) and the system is typically not reparameterized.

[0088] If the measured value is assessed as not correct by the control logic (317), in particular in comparison with other measured values ​​that were measured in parallel or previously, and / or with other threshold values, then in a fifth step (605), for example, the last measured value assessed as OK is output and the system is typically re-parameterized. In addition, an error message is typically sent to a higher-level system. However, it is also conceivable to record this error message in a counter and to have the control logic only issue an error message when a certain counter reading is reached. The behavior of this counter can be designed in such a way that a correct measurement immediately resets the counter and, if necessary, a corresponding all-clear message is sent to a higher-level system. However, it can also count down to zero at the same or a different speed as the counting up.

[0089] The aforementioned re-parameterization of the system typically affects a. the frequency of the excitation signal (322, 522) and / or b. a spreading code used to generate the excitation signal (322, 522) and / or c. a pseudo-random signal used to generate the excitation signal, in particular, for example, the feedback polynomial of a feedback shift register for generating a pseudo-random signal, which can be exchanged or modified.

[0090] After the sixth or fifth step, the measurement cycle in this example starts again with the first step.

[0091] In Fig. 6 shows an extension of the device according to the invention. In addition to the first excitation current source (301) of the Fig.2, a second excitation current source (501) is shown, which feeds a second excitation current into the excitation loop (2) of the inductive angle sensor head (303), superimposed on the first excitation current of the first excitation current source (301). This second excitation current source (501) is controlled by a second excitation signal (522). The signal path downstream of the inductive angle sensor head (303) is initially unchanged up to the synchronous demodulator block (309), in addition to which a typically identical second synchronous demodulator block (509) is now connected in parallel. Its second control signal for synchronous demodulation (521) is now based on the second excitation signal (322). It can, for example, also be identical to this.It is now preferred that the two excitation signals are selected in relation to one another such that the signal components of the output signals (304) of the inductive angle sensor head (303) that are attributable to the first excitation signal (322) do not generate any signals at the second level signals (510), i.e., the outputs of the second synchronous demodulator (509). Conversely, in this case, it is preferred that the signal components of the output signals (304) of the inductive angle sensor head (303) that are attributable to the second excitation signal (522) also do not generate any signals at the first level signals (310), i.e., the outputs of the first synchronous demodulator (309). In this case, the excitation signals are referred to as being orthogonal to one another.

[0092] The multiplexer (311) is now simply extended accordingly. Preferably, one measurement pass refers to the outputs of a synchronous demodulator block (309, 509).

[0093] It is now sensible to also check the correct function of the sensor system with the additional second level signals (510) which originate from the second synchronous demodulator block (509). The sum of the amplitudes of the transformed voltages of the transformed signals (308) after demodulation by the synchronous demodulator to form the second level signals (509) must again be zero volts. This sum of the amplitudes of the transformed voltages of the transformed signals (308) can therefore also be evaluated to assess the sensor state. This can also be used to detect error states (push-pull errors) which affect the sensor and / or the evaluation in an asymmetric manner. This can be done, for example, at a different frequency using a second excitation signal (522) which is different from the first excitation signal (322). In the device according to the invention, the operating voltage is therefore deducted from the voltage sum once orthe said offset is subtracted twice, whereby the second sum signal (526) is again in the middle of the control range of subsequent amplifiers.

[0094] This formation of the second sum signal (526) by summing the second level signals (510) takes place in a second summing device (524).

[0095] The second sum signal (526) thus formed is compared, for example, by means of a fifth comparator (541) with a fifth threshold value (527), for example also in the form of a first reference voltage (V ref1 ) which is above said offset, and, for example by means of a sixth comparator (542), with a sixth threshold value (530), for example also in the form of a second reference voltage (V ref2) which is below the offset. This typically results in a fifth result signal (534) and a sixth result signal (533). If the voltage of the second sum signal (526) lies outside the range defined by the first reference voltage (Vref1), i.e., the fifth threshold value (527), and the second reference voltage (V ref2 ), i.e. the sixth threshold value (530), defined, permissible total voltage range, then there is again a symmetry error to be detected in accordance with the object of the invention.

[0096] Of course, it is also conceivable to apply the measurement to all synchronous demodulator blocks (309, 509). However, this is disadvantageous in that the resolution of a block is then reduced.

[0097] If necessary, the thresholds specified above, i.e. the third and fourth, as well as the seventh to ninth thresholds, can be doubled accordingly.

[0098] It is particularly advantageous if at least one of the signal generators (320, 520) uses a random signal as the excitation signal (322, 522). The use of a pseudorandom signal is also possible.

[0099] A further advantageous embodiment is the use of a further spreading code for generating a further excitation signal (522) by the signal generator (320) or a further signal generator (520). However, with a very long further spreading code, the reaction time of the sensor system increases. Therefore, it is useful if, in this case, the other signal generator uses a shorter spreading code or no spreading code at all. The spreading codes or the base clock of the signal generators (520, 320) for generating the excitation signals (322, 522) should preferably be selected such that they result in mutually orthogonal excitation signals (322, 522).This can be ensured, for example, if the base clock (T0) of the first signal generator (322) is 2*w times the base clock (T1) of the second signal generator (522) and the spreading code length for the first spreading code of the first signal generator, assuming a digital spreading code, is 2*w clocks and the first spreading code contains w 1 values ​​and w 0 values. The second spreading code for signal generation by the second signal generator should also have a length of an even number of clocks and contain an equal number of 1 values ​​and 0 values. This ensures that the minimum frequency of the further excitation signal (522) lies above the maximum frequency of the first excitation signal (322). Their spectra therefore do not overlap. As a result, the two frequency channels do not interfere with each other.

[0100] The lower limit frequency of the synchronous demodulator block (309, 509) lies below the smallest minimum frequency of the excitation signals (322, 522). The output filters (F1) of the sub-devices of the synchronous demodulator block (309, 509), which perform the synchronous demodulation of the transformed signals (308) with the respective excitation signal (322, 522), filter out the frequency bands of the excitation signals (322, 522) from the output signal of the respective synchronous demodulator block (309, 509), the level signals (310, 510), due to their low-pass characteristics.

[0101] It is also conceivable for one signal generator (520, 320) or both signal generators (522, 322) to generate a monofrequency excitation signal (522, 322) that is not within the frequency range of the other signal generator. With two monofrequency signals, the frequency of one excitation signal (322) is preferably an integer multiple of the frequency of the other excitation signal (522), or vice versa. Excitation signals (322, 522) of different frequencies are, incidentally, the simplest examples of orthogonal excitation signals.

[0102] In this case, it is useful for the control logic (317) to output a second measured value (545) for the additional excitation signal (522).

[0103] It is also possible, of course, for one or more excitation signals (322, 522) to be changed after each measurement sequence, i.e. the measurement of all level signals (310, 510), as described. This can d. relate to the frequency of the excitation signal (322, 522), which corresponds to a frequency hopping method, and / or e. relate to a spreading code used to generate the excitation signal (322, 522) and / or f. a pseudo-random signal used to generate the excitation signal, in particular the feedback polynomial of a feedback shift register.

[0104] In this way, multiple measured values ​​(345, 545) can be obtained by spatial multiplexing and / or time multiplexing and / or frequency and / or spread code multiplexing, which can be compared with one another and with further threshold values, in particular by averaging and subtracting in an evaluation unit. This makes it possible, for example, to generate three measured values ​​from three parallel measured values ​​using three excitation current sources and three respectively assigned excitation currents controlled by three excitation signals, and thus to generate a majority vote for the probably correct measured value. For example, three level signals are generated by three synchronous demodulator blocks, which are then processed as already described. Each of the three branches is fed by one excitation signal, each originating from one of three signal generators.Of course, it is conceivable that a signal generator produces several of these different excitation signals simultaneously.

[0105] Depending on the evaluation result, the excitation signals can then be modified by the control logic or another control unit. This makes it possible to detect interference in the event of strong EMC radiation and minimize it by optimizing the excitation signals, as long as the interference does not affect all excitation signals equally, which is typically the case with narrowband interferers.

[0106] As in many other inductive sensor systems, it is also advisable in a sensor system according to DE 197 38 836 A1 to operate the excitation loop (2) within an LC resonant circuit, for example, together with a parallel resonant circuit capacitance (C). The advantage of the high resonance amplitude, however, results in the disadvantage of a very narrow-band resonance frequency. In parallel with the modulation of the AC power source, the modulation of the resonance frequency of the LC resonant circuit itself is therefore also possible and particularly preferred using a modulatable capacitance (701) and / or a modulatable inductance (801) as the excitation loop (2). This is achieved with the aid of the Fig. 7 to 10 explained.

[0107] Fig. Figure 7 shows an exemplary system with a modulatable capacitance (701).

[0108] In the case of a modulatable capacitance (701), this modulability can be achieved, for example, by connecting and disconnecting partial capacitances (C1, C2). Fig. Figure 8 shows an example of a very simple modulatable capacitance (701). The exemplary modulatable capacitance (701) consists of a first partial capacitance (C1), which is connected in parallel with the second partial capacitance (C1) via the switch (SW1) depending on the second excitation signal (722) or is separated from it. This allows the resonant frequency of the LC resonant circuit to be Fig.7 are modulated. Suitable switches (SW1) include mechanical switches such as relays, and in particular micromechanical relays, but also MOS and bipolar transistors and similar semiconductor switches that can fulfill the function of such a switch. These can also be more complex circuits that perform this function of a switch. In this case, the first excitation signal (322) of the first signal generator (320) is typically also switched synchronously so that the first excitation current source (301) receives an excitation signal (322) that matches the current resonant frequency of the LC resonant circuit comprising the excitation loop (2) and the modulatable capacitance (701) in accordance with the value of the second further excitation signal (722). A variant in which the second further excitation signal (722) has a period length that is a multiple of the period length of the first excitation signal (322) is particularly preferred.

[0109] The processing is carried out analogously to the description of the Fig. 6 in a second synchronous demodulator block (709), which forms second level signals (710) depending on a second control signal (721) for synchronous demodulation. The second control signal (721) for synchronous demodulation is typically identical to the second excitation signal (722). A second clock or signal generator (720) generates said second excitation signal (722). This typically occurs synchronously with the generation of the first excitation signal (322) in the signal generator (320).

[0110] It is obvious that the second additional level signals (710) should be symmetrical and have the same levels. Deviations can be detected as errors, as before.

[0111] Fig. 10 shows an exemplary system with a modulatable inductance (801) and a resonant circuit capacitance (C).

[0112] In the case of a modulatable inductance (801), this modulability can be achieved, for example, by switching partial inductances (L1, L2) on and off. Fig.Figure 10 shows an example of a very simple modulatable inductance (801). The exemplary modulatable inductance (801) consists of a first partial inductance (L1), which is short-circuited via the switch (SW1) as a function of the second further excitation signal (822) of the second partial inductance (L1), and is thus removed from the resonant circuit. This allows the resonance frequency of the LC resonant circuit consisting of the excitation loop (2) and the resonant circuit capacitance (C) to be modulated. The excitation loop itself can consist of several such switchable partial excitation loops, which has the same effect. Mechanical switches such as relays, and in particular micromechanical relays, can also be used as switches (SW1), but also MOS and bipolar transistors and similar semiconductor switches, etc., which can fulfill the function of such a switch.In this case, the first excitation signal (322) of the first signal generator (320) is typically also switched synchronously so that the first excitation current source (301) receives an excitation signal (322) that matches the current resonant frequency of the LC resonant circuit comprising the excitation loop (2) and the modulatable inductance (801) and the resonant circuit capacitance (C) according to the value of the second additional excitation signal (822). A particularly preferred variant is one in which the second additional excitation signal (822) has a period length that is a multiple of the period length of the first excitation signal (322).

[0113] The processing is again analogous to that described in the Fig.6 in a second, further synchronous demodulator block (809), which forms second, further level signals (810) depending on a second, further control signal (821) for the synchronous demodulation. The second, further control signal (821) for the synchronous demodulation is typically equal to the second, further excitation signal (822). A second, further clock or signal generator (820) generates said second, further excitation signal (822). This typically occurs again synchronously with the generation of the first excitation signal (322) in the signal generator (320).

[0114] It is obvious that the second additional level signals (810) should again be symmetrical and have the same levels. Deviations can be detected as errors, as before.

[0115] Of course, this modulation technique can also be used in combination by simultaneously using a modulatable capacitance and a modulatable inductance.

[0116] Fig. Figure 11 shows an embodiment of the invention in which the detection of a fault occurs according to a fault model signal (1101). This signal is generated in a fault model signal generator.

[0117] An error synchronous demodulator block (1109) generates, in the manner described above, the error level signals (1110) corresponding to the correlating components of the error model signals (1101) in the transformed signals (308).

[0118] An error summing device (1124) sums these error model signals (1101) weights with factors corresponding to the weighting factor signals (1106) to one or more error sum signals (1104).

[0119] One or more error detection devices (1142, 1141) compare the levels of these error sum signals (1104) with one or more error threshold values ​​(1130, 1127) and generate error detection signals (1133, 1134) depending on the result of this comparison.

[0120] Fig. 12 shows the system of Fig. 2, in which the transformed signals (308) are, for example, fed back as inputs of the signal transformation block (307). This results in a nonlinear transformation that is no longer an affine transformation. Advantages of the invention

[0121] In contrast to the prior art, the device and method according to the invention can thus detect the faults of the task with high EMC robustness. It is particularly noteworthy that the device is capable of classifying and distinguishing between common-mode and differential-mode faults and other faults, and, in particular, in the inventive combination of claims 1 and 2, can reliably detect all of these. List of reference symbols 2 excitation loop; 300 sensor system; 301 excitation current source; 303 inductive rotation angle sensor head; 304 output signals of the inductive angle sensor head (303) with m partial signals (304,I); 304,I I-th partial signal of m partial signals of the output signals (304) of the inductive rotation angle sensor head (303); 305 EMC filter block; 306 filtered output signals (306) of the rotation angle sensor head (303); 307 Signal transformation block; 308 transformed signals from n individual sub-signals (308,i); 308,i i-th sub-signal of n sub-signals of the transformed signals (308); 309 Synchronous demodulator block; 310 level signals from k partial signals (310,j); 310,j j-th partial signal of k partial signals of the level signals (310); 311 analog multiplexer; 312 multiplexer output signal; 313 adjustable amplifier; 314 amplifier output signal; 315 analog-to-digital converters (ADCs); 316 ADC output signal; 317 Control logic; 318 amplifier control signal; 319 Multiplexer control signal; 320 clock or signal generator; 321 Control signal for synchronous demodulation; 322 excitation signal; 324 summing device for forming the sum signal (326); 326 Sum signal of the level signals (310); 327 first threshold; 330 second threshold; 333 second result signal; 334 first result signal; 335 third threshold, the maximum permissible gain value; 336 fourth threshold, the permissible minimum gain value; 337 third result signal; 338 fourth result signal; 341 first comparator for forming the first result signal (334) by comparing the first threshold value (327) with the sum signal (326); 342 second comparator for forming the second result signal (333) by comparing the second threshold value (330) with the sum signal (326); 343 third comparator for forming the third result signal (337) by comparing the third threshold value (335) with the amplifier control signal (318); 344 fourth comparator for forming the fourth result signal (338) by comparing the fourth threshold value (336) with the amplifier control signal (318); 345 desired measured value (345) of the angle of rotation; 401 first differential amplifier for signal transformation of the filtered output signals (306) of the rotation angle sensor head (303) into a transformed signal (308); 402 second differential amplifier for signal transformation of the filtered output signals (306) of the rotation angle sensor head (303) into a transformed signal (308); 403 third differential amplifier for signal transformation of the filtered output signals (306) of the rotation angle sensor head (303) into a transformed signal (308); 501 second excitation current source; 509 second synchronous demodulator block; 510 second level signals; 520 second clock or signal generator; 521 second control signal for synchronous demodulation and / or scalar product formation; 522 second or further excitation signal; 524 second summing device for forming the second sum signal (526); 526 second sum signal of the second level signals (510); 527 fifth threshold; 530 sixth threshold; 533 sixth result signal; 534 fifth result signal; 541 fifth comparator for forming the fifth result signal (534) by comparing the fifth threshold value (527) with the second sum signal (526); 542 sixth comparator for forming the sixth result signal (533) by comparing the sixth threshold value (530) with the second sum signal (526); 545 second measured value of the angle of rotation; 600 first step of an exemplary measurement run (measurement of a first of the exemplary three level signals (310)); 601 second step of an exemplary measurement run (measurement of a second of the exemplary three level signals (310)); 602 third step of an exemplary measurement run (measurement of a third of the exemplary three level signals (310)); 603 fourth step of an exemplary measurement run (determination of the measurement result (345) before output by the control logic (317)); 604 fifth step of an exemplary measurement run (evaluation of the measurement result (345)); 605 sixth step of an exemplary measurement run (execution of the actions in case of a correct measurement result, e.g. output of the measurement result (345)); 606 seventh step of an exemplary measurement run (execution of actions in case of an incorrect measurement result, e.g. output of a previous measurement result (345)); 701 modulatable capacitance, which forms an LC resonant circuit with the excitation loop (2); 709 second additional synchronous demodulator block; 710 second additional level signals; 720 second additional clock or signal generator; 721 second additional control signal for synchronous demodulation; 722 second additional excitation signal; 801 modulatable inductance, which forms an LC resonant circuit with the excitation loop (2) and another capacitor; 809 second additional synchronous demodulator block; 810 second additional level signals; 820 second additional clock or signal generator; 821 second additional control signal for synchronous demodulation; 822 second additional excitation signal; 1100 error model signal generator; 1101 error model signals from u error model subsignals; 1101,u u-th error model sub-signal of the u error model sub-signals of the error model signals (1101); 1104 Error sum signal; 1105 Error detection signal; 1106 weighting factor signal; 1109 Error synchronization demodulator block; 1110 error level signal with q sub-signals; 1110,r r-th sub-signal of q sub-signals of the error level signals (110); 1110 error level signal; 1124 error summing means for forming the second error sum signal (1104); 1127 Error threshold; 1130 Error threshold; 1133 Error detection signal; 1134 Error detection signal; 1141 Error detection; 1142 Error detection; Δφ angular wavelength; Δω = ω max - ω min Bandwidth of the excitation signal (322) (magnitude); ω max upper limit frequency of the excitation signal (322) (magnitude); ω min lower limit frequency of the excitation signal (322) (magnitude); A 307 Link matrix for the signal transformation in the signal transformation block (307); A 307,ij Element of the i-th column and j-th row of the link matrix (A 307 ) for the signal transformation in the signal transformation block (307); C resonant circuit capacitance; C1 first partial capacity of the exemplary modulatable capacity (701); C2 second partial capacity of the exemplary modulatable capacity (701); F1 filter; G is an example weighting vector for an example error model. Here, the weighting vector specifies the factors with which a disturbance signal with the disturbance signal voltage V stör into the three exemplary receiver loops (S1, S2, S3); k Number of level signals (310, 510 etc.) L1 first partial inductance of the exemplary modulatable inductance (801); L2 second partial inductance of the exemplary modulatable inductance (801); m number of receiver loops; n number of transformed signals (308); Out Output signal of the device for scalar product formation. This output signal is typically connected to one of the outputs of a synchronous demodulator block (e.g., 309), e.g., a level signal (310,I) of the k level signals (310); S1 first receiver loop; S2 second receiver loop; S3 third receiver loop; SIG1 first input signal of the device for scalar product formation SIG2 second input signal of the device for scalar product formation Sm m-th receiver loop SW1 Switch of the exemplary modulatable capacitance (701) which can be actuated by the second excitation signal (522). Sym axis of symmetry V BD Operating voltage V 304 Signal voltage vector of the output signals (304) of the rotation angle sensor head V 304,i the i-th signal of the signal voltage vector of the output signals (304) of the rotation angle sensor head V 307 Signal voltage vector of the output signals (308) of the signal transformation block (307), the transformed signals (308) V 307,i the i-th signal of the signal voltage vector of the output signals (308) of the signal transformation block (307), the transformed signals (308) V messk-dimensional signal vector for encoding the errors and measured values. Typically, some sub-signals of this signal vector preferably contain only measured value signals, while other error level signals for error levels corresponding to one or more of the prototypical, m-dimensional signal voltage vectors (V proto ) are equivalent to. V mess,i i-th subsignal of the k-dimensional signal vector (V mess ) for the coding of the errors and measured values. Typically, some sub-signals of this signal vector preferably contain only measured value signals, while other error level signals for error levels corresponding to one or more of the prototypical, m-dimensional signal voltage vectors (V proto ) are equivalent to. V off exemplary offset voltage vector which is added to the transformed signal voltage vector of the output signals (304) of the rotation angle sensor head (303) in the signal transformation block (307). V protoprototypical, m-dimensional signal voltage vector for the prototype of an error or a measured value V proto the i-th signal of the prototypical, m-dimensional signal voltage vector (V proto ) for the prototype of an error or a measured value V stör,i Interference signal (here, for example, one-dimensional). This could be, for example, EMC radiation.

Claims

[1] Device for operating and evaluating an inductive angle sensor head (303) a. with m, but at least three, inductive receiving loops (S1, S2, S3) and b. with a field-inducing excitation loop (2) and c. with a rotor with a rotor loop distorting the field of the excitation loop (2), d. with at least one signal generator (320) which generates at least one excitation signal (322) and e. with at least one electrical current or voltage source (301) modulated by the excitation signal (322) which excites the excitation loop (2) of the inductive rotation angle sensor head (303), and f. wherein in particular the star voltage of the receiving loops (S1, S2, S3) of the inductive rotation angle sensor head (303) is zero volts in the ideal, error-free case, characterized in that g. that the device has at least one sub-device, a signal transformation block (307), for carrying out a signal coordinate transformation, in particular an affine signal coordinate transformation, for an output signal vector (304) of the rotation angle sensor head (303), consisting of output signals (304) of the m receiving loops (S1, S2, S3), for forming n, but at least three and preferably m, transformed sub-signals (308,i) of the transformed signals (308) and h. that for each transformed partial signal (308,i) of the transformed signals (308) it has at least one sub-device with two inputs (SIG1, SIG2), which sub-device generates from the excitation signal (322) or a control signal correlating therewith for the synchronous demodulation (321) as the first input signal (SIG1) of the sub-device and the respective transformed partial signal (308,i) as the second input signal (SIG2) of the sub-device for each transformed partial signal (308,i) a level signal (310,j) assigned to the sub-device and the respective transformed partial signal (308,i) from a total of k, but at least three - preferably n or m, level signals (310) by synchronous demodulation and / or scalar product formation, wherein this sub-device can in particular be a synchronous demodulator. [2] Device for operating and evaluating an inductive rotation angle sensor head (303) according to claim 1, characterized by , a. that it comprises an error model signal generator (1100) which generates an error model signal (1101), and b. that for each partial signal (308,i) of the transformed signals (308), it has at least one sub-device with two inputs (SIG1, SIG2), which sub-device generates a partial signal (1110,r) of the error level signals (1110) assigned to the partial device and the respective partial signal (308,i) by synchronous demodulation and / or scalar product formation from the error model signal (1101) as the first input signal (SIG1) of the sub-device and the respective partial signal (308,i) as the second input signal (SIG2) of the sub-device, wherein this sub-device can in particular be a synchronous demodulator. [3] Device for operating and evaluating an inductive rotation angle sensor head (303) according to claim 2, characterized by , a. that it comprises an error summing device (1124) which forms an error sum signal (1104) by weighted or unweighted summation of several partial signals (1110,r) of the error level signals (1110). [4] Device for operating and evaluating an inductive rotation angle sensor head (303) according to claim 3, characterized by , b. that it has an error detection means (1141, 1142) which compares the error sum signal (1104) or a signal derived therefrom with one or more error threshold values ​​(1127, 1130) to determine whether the error sum signal (1104) is above or below the respective error threshold value or between two error threshold values, and outputs an error detection signal (1133, 1134) depending on the comparison result. [5] Device for operating and evaluating an inductive rotation angle sensor head (303) according to one or more of claims 1 to 4 characterized by , a. that the signal generator (320) generates at least one excitation signal (322) based on a spreading code. [6] Device according to one or more of claims 1 to 5, characterized in that a. that the signal generator (320) generates an excitation signal (322) that is a band-limited signal with a non-zero upper limit frequency ω max and a non-zero lower limit frequency ω min with a frequency bandwidth Δω=ω max -ω min with respect to the magnitude of the frequency and b. that the frequency band thus defined does not include 0Hz and c. that the lower limit frequency ω min of the excitation signal (322) is greater than half the frequency bandwidth Δω / 2 of the excitation signal (322). [7] Device according to one or more of claims 5 to 6; characterized in that a. that the spreading code for generating the excitation signal (322) is a random signal or a pseudorandom signal [8] Device according to one or more of the preceding claims 5 to 7, characterized in that a. that the at least one signal generator (320) or a further signal generator (520) generates a further excitation signal (522) based on a further spreading code and b. that it has a further excitation current source (501) modulated by means of the associated further excitation signal (522), which excites the excitation loop (2) of the inductive rotation angle sensor head (303) with at least one additional superimposed further excitation current and c. that for each transformed partial signal (308,i) of the transformed signals (308), it has at least one sub-device with two inputs (SIG1, SIG2), which generates from the further excitation signal (522) or a control signal correlating therewith for the synchronous demodulation (521) and / or scalar product formation as the first input signal (SIG1) of the sub-device and the relevant partial signal (308,i) of the transformed signals (308) as the second input signal (SIG2) of the sub-device for each partial signal (308,i) an associated partial signal (310,j) of a total of k, but at least three, level signals (310) by synchronous demodulation and / or scalar product formation, wherein this sub-device can in particular be a synchronous demodulator and d. that the first and the further excitation signal (322, 522) are orthogonal with respect to the synchronous demodulation and / or the scalar product formation in the synchronous demodulation block (309, 509), where orthogonal means that if the two excitation signals (322, 522) in question were combined with one another by a synchronous demodulator in the synchronous demodulation block (309, 509) as its first input signal (SIG1) and its second input signal (SIG2), the output signal of this synchronous demodulator would then be zero except for control errors and system noise. [9] Device according to claim 8, characterized in that a. that the excitation signal (322) and the further excitation signal (522) are based on two different spreading codes, the spreading code and the further spreading code, which have a different spreading code length.

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