Electromagnetic pulse driver, circuit and method for charging an electropermanent magnet actuator

Abstract

Circuit comprising: a multi-ampere capacitor (506); a battery (502); a charging circuit (504) connected between the battery (502) and the capacitor (506); an actuator (514); an electropermanent magnet coil (604) configured to actuate the actuator (514); a switching circuit connected between the multi-ampere capacitor (506) and the electropermanent magnet coil (604), the switching circuit comprising an H-bridge circuit (512) of four transistors (910, 912, 914, 916); a diode (704) configured in parallel with the electropermanent magnet coil (604) to provide a current return path when the capacitor (506) is disconnected from the electropermanent magnet coil (604); a Measuring resistor connected to the electropermanent magnet coil (604);a comparator (508) having a first input connected to the measuring resistor and a second input connected to a reference voltage (606); and a digital controller (510) connected to an output of the comparator (508), the digital controller (510) having transistors performing a D flip-flop function, the digital controller (510) being programmed to provide a signal at an output of the digital controller (510) connected to the switching circuit to disconnect the capacitor (506) from the electropermanent magnet coil (604) when a current from the capacitor (506) to the electropermanent magnet coil (604) generates a voltage across the measuring resistor that reaches the reference voltage (606).

Description

Unless otherwise indicated herein, the materials described in this section are not prior art to the claims of this application and are not recognized as prior art by inclusion in this section.The present invention relates to drivers for electromagnets (EM) and electro permanent magnets (EPM) for actuating actuators, such as haptic devices. Examples are an EPM-based magnetic ratchet (for example for mouse scroll wheel), an EPM-based reluctance motor, EPM-controlled bistable mechanical locking systems and EPM-controlled ferromagnets. All of these devices are controlled by the magnitude of the instantaneous electrical current flowing through the coil winding.EM and EPM drivers can be used, for example, in real computer peripheral interface devices such as keyboards, mice, joysticks, wheels, etc., which can be real devices that a user manipulates to interface with a computing device. Real computer peripheral interface devices may include wheel input elements that a user can manipulate. Computer mice may include, for example, scroll wheels that may be used to pan a viewing window over an image or document displayed by a computing device in response to rotation of the scroll wheel about an axis. Interface wheels may operate over a variety of friction profiles. For example, a mouse scroll wheel may selectively operate between a free-wheeling mode and a ratchet mode. The ratchet may be provided using electromagnets, such as electro permanent magnets. By turning the magnetic field on and off, interactions with magnetic ribs in a scroll wheel can quickly change from a resistive force to no resistive force, which gives the wheel's turning a real ratchet feel. Similar effects may be provided to other haptic devices, such as Braille readers or simply vibration feedback inducing haptic devices.A pulse driver provides the current for the winding of the electromagnet. The pulse driver is often located in battery powered devices. The peak current of the battery may be several amperes, while the peak discharge current required by the haptic actuator may be dozen amperes. Therefore, a current transducer and a power reservoir are often used for actuation to provide the needed current surges. The energy reservoir may be a capacitor that can be charged sufficiently to deliver many tens of amperes. The capacitor forms a tank circuit with the coil of the electromagnetic windings when switched to connect the capacitor to the coil.The conventional approach consumes much power and electromagnetic interference (EMI).The document DE 297 00 618 U1 discloses an electromagnet with a power supply for lifting, holding and transporting magnetizable workpieces. In the event of a short circuit caused by a winding fault, the power supply is immediately switched off in order to avoid further damage to the electromagnet.US 2015 / 0 062 770 A1 discloses an energy-saving bistable permanent magnet drive system which can supply short current pulses to the control coil which generates a short secondary magnetic field in order to reverse a primary magnetic field from a permanent magnet in order that movable magnetic pole pieces can be alternately attracted into a bistable permanent magnet actuator. To solve the above problems and achieve the object, a circuit and method for charging an electropermanent magnetic actuator having the features of the appended independent claims are provided. Advantageous embodiments of the present invention are defined in the dependent claims, the description and the drawings of the present invention.In some embodiments, a driver circuit is provided to connect a charged capacitor through a switch to the inductive windings of a haptic feedback electromagnet or other application. A feedback circuit provides an indication of the current supplied. When the current reaches a desired level, the current is turned off rather than being controlled. This makes use of the realization that in a haptic feedback application and in similar applications the value of the current and thus the electromagnetic force at the tip is to be sensed and does not have to be controlled beyond the tip. By turning off the current, further discharge of the capacitor is avoided, thereby maintaining 80% or more of its charge for the next pulse in certain embodiments. The turn-off level may be slightly below the desired peak level to allow the desired peak level to be reached upon exceeding.In some embodiments, a diode is provided in parallel with an electro permanent magnet (EPM) winding when the current is turned off, thereby providing a feedback path that prevents voltage transfer across the EPM terminals and provides gradual current decay through the EPM or actuator winding. This causes a force drop when the EPM is part of a solenoid, a linear motor or similar electromotive device. The current cut-off is a pulsing method that consumes less energy altogether than the conventional continuous (linear or PWM) systems, since the cut-off occurs almost immediately once the target current through the coil winding is reached.In certain embodiments, the switching circuit is configured to allow for rapid switching, which allows for better control over the amount of overshoot. The amount of overshoot can be predicted because the magnetizing force of the electromagnet flattens the sinusoidal charging curve to linear in the region before the desired peak current. Thus, in a fast switching circuit, the overshoot can be predicted quite accurately. In particular, the high speed switching circuit is obtained with an H-bridge circuit transistor. The transistors in the H-bridge circuit are small (such as 10 times smaller) than required to provide a constant current at the desired current level. This makes use of the fact that such transistors can process a much higher current pulse than the constant rated current power. A smaller transistor has a smaller gate with a lower gate capacitance and thus a faster switching speed. Voltage-current conversion via a controlled impedance is used for the feedback path, which allows the use of a comparator to simplify the control circuit.In other embodiments, a fast digital control configuration is used for feedback control to further ensure the fast switching speed required to limit the overshoot. A flip-flop, which operates like a bidirectional thyristor or triac, is connected to the comparator output to enable the fast switching. The flip-flop transistors for the fast switching are configured as at least "silicon controlled rectifier" (SCR) or else as "thyristor", or at least two transistors are configured as SCR or flip-flop.In certain embodiments, a circuit is provided with a battery, a multi-amp charging capacitor, and a charging circuit connected between the battery and the capacitor. An actuator is actuated by an electromagnetic coil. A switching circuit is connected between the multi-ampere capacitor and the electromagnetic coil. The switching circuit includes an H-bridge circuit of four transistors. A diode is configured in parallel with the electromagnetic coil on one of the four transistors to treat the parasitic capacitance and provide a current feedback path when the charging capacitor is disconnected from the electromagnetic coil. A measuring resistor is connected to the electromagnetic coil. A comparator has a first input connected to the sense resistor and a second input connected to a reference voltage. A digital controller is connected to an output of the comparator. The digital controller is a D flip-flop and is programmed to provide a signal at an output of the digital controller connected to the switching circuit to disconnect the charging capacitor from the electromagnetic coil when a current from the charging capacitor to the electromagnetic coil generates a voltage at the sensing resistor that reaches the reference voltage.In another embodiment, a method of charging an electromagnetic actuator is shown. First, a multi-ampere capacitor is charged with a battery. The capacitor is switched from connection to the battery for charging to connection to the solenoid actuator to provide a current pulse for activating the solenoid actuator. The amount of current supplied to the solenoid actuator is detected. The capacitor is disconnected from the solenoid actuator when it is determined that the magnitude of the current has reached a threshold below a desired current peak. The multi-amp capacitor may then be recharged with the battery for the next cycle or pulse because it is only partially discharged by the capacitor being turned off at the threshold.This summary is not intended to identify key or essential features of the claimed subject matter, nor is it intended to be used in isolation to determine the scope of the claimed subject matter. The subject matter should be understood by reference to the corresponding portions of the entire specification of this disclosure, some or all of the drawings, and each claim.The foregoing, along with other features and examples, is described in more detail in the following specification, claims, and accompanying drawings.Aspects and features of the various embodiments will become more apparent from the description of examples and reference to the accompanying drawings, in which: FIG. 1 shows an example user input device suitable for use with the described embodiments and in the form of a wireless mouse. FIG. 2A shows an example electro-permanent magnet according to certain embodiments with dashed lines representing magnetic flux emitted from the electro-permanent magnet, showing how with both the first and second permanent magnets aligned in the same direction, magnetic flux is released from the electro-permanent magnet to generate well-defined north and south poles. FIG. 2B shows how, when the polarity of the first permanent magnet is opposite the polarity of the second permanent magnet, the magnetic flux generated by both permanent magnets remains substantially contained in and circulating through the ferromagnetic poles, the first permanent magnet, and the second permanent magnet, in accordance with certain embodiments. FIG. 3A shows a perspective view of an example implementation in accordance with certain embodiments, in which an electro-permanent magnet is configured to change a resistance profile of a rotary input controller compatible with the device illustrated in FIG. 1. FIG. 3B shows a side view of a rotary input controller according to certain embodiments, arranged through a support structure over a support surface. FIG. 3C shows an exploded view of the rotary input controller and support structure, in accordance with certain embodiments. FIG. 4A shows a cross-sectional view of a rotary input controller according to certain embodiments, in which a polarity of the magnetic fields emitted by permanent magnets is aligned in the same direction, and how a magnetic flux emitted by radially protruding teeth of the EPM assembly interacts with the ferromagnetic material of which the teeth on the wheel are comprised when offset (not aligned). FIG. 4B shows a cross-sectional view of a rotary input controller according to certain embodiments, in which a polarity of the magnetic fields emitted by the permanent magnet is aligned in the same direction when the radially protruding teeth are aligned with a corresponding one of the teeth on the wheel. FIG. 4C shows another cross-sectional view of a rotary input controller in which a polarity of a permanent magnet has been reversed, in accordance with certain embodiments. FIG. 5 is a system diagram of a force control architecture according to certain embodiments. FIG. 6 is a simplified circuit diagram for charging an electromagnet with a switch in a charging position, according to certain embodiments. FIG. 7 is a simplified circuit diagram for charging an electromagnet with a switch in a shut-down position, in accordance with certain embodiments. FIG. 8 is a simplified circuit diagram according to certain embodiments for the digital controller of FIGS. 6 and 7. FIG. 9A is a block diagram of an electromagnetic charging system having a switch in a charging position, in accordance with certain embodiments. FIG. 9B is a block diagram of FIG. 9A with further control details, in accordance with certain embodiments. FIG. 10 is a block diagram of an electromagnetic charging system having a switch in a shut-down and free-wheeling position, in accordance with certain embodiments. FIG. 11 is a circuit diagram of the comparator and bidirectional thyristor blocks and flip-flop blocks of FIGS. 9 and 10, respectively, in accordance with certain embodiments. FIG. 12 is a circuit diagram of the output stage block and H-bridge circuit block of FIGS. 5 and 10, respectively, in accordance with certain embodiments. FIG. 13A is a graph showing a charging and discharging cycle according to the related art with a curve of magnetization to saturation. FIG. 13B is a graph showing a charge and shut down cycle without saturation, in accordance with certain embodiments. FIG. 14 illustrates an embodiment of a method for charging an electromagnetic actuator. FIG. 15 illustrates a processing system for controlling certain features of the peripheral devices disclosed herein. FIG. 16 shows another diagram of a system for operating a peripheral device (e.g., a user input device, such as a computer mouse), in accordance with certain embodiments. FIG. 17 is a graph illustrating a typical EPM magnetization curve. FIG. 18 is a graph illustrating the moment versus the magnetizing current value. FIG. 19 illustrates a test station configuration for detecting the EPM magnetization curve according to embodiments. FIG. 20 illustrates a test station calibration setup using an absolute moment calibration method, according to embodiments.While certain embodiments are described, these embodiments are presented by way of example only and are not intended to limit the scope of protection. The devices and systems described herein may be embodied in a variety of other forms. Moreover, various omissions, substitutions, and changes in the form of the example methods and example systems described herein may be made without departing from the scope of protection.In some embodiments, a driver circuit is provided to connect a charged capacitor through a switch to the inductive windings of a haptic feedback electromagnet or other EM actuator application. A feedback circuit provides an indication of the current supplied. When the current reaches a desired level, conventional control systems continuously monitor and control the current. Embodiments of the present invention simply cut off the current rather than control it.The conventional linear control systems continuously monitor the error between the reference point and the output current setting. Indeed, under normal conditions, the above error will be minimized by the system either monotone (no decaying oscillations induced and no overshoot) or nonmonotone with the decaying oscillations. Due to the nature of the continuous control system and the constantly changing parameters of the system, the number of undesirable vibration cycles (normally decaying because the function corrects to the steady state) is not well controlled. That is, there would be several cycles of oscillation before the system converges to the steady state. The more reactive the system is (wider bandwidth), the more overshoot can be observed.Embodiments of the present invention do not exhibit any vibrations under any circumstances possible due to the design. This is because after arming, the flip-flop transitions from the ACTIVE state to the OFF state only once per discharge cycle before each cycle. In contrast, in the conventional linear control with the feedback loop, the step response function could have multiple decaying oscillations and large overshoot. The overshoot occurs due to the propagation delay in the error amplifier. In other words, a conventional linear controller with a randomly alternating actuator could tend to oscillate, while embodiments of the present invention do not tend to oscillate. Also, unwanted vibration will result in EMI emissions and non-precise EPM or transducer operation.The embodiments of the present invention take advantage of the realization that for a haptic feedback application and similar applications, the value of the current, and thus the electromagnetic force, is to be sensed at the tip and need not be controlled via the tip. By turning off the current, further discharge of the capacitor is avoided, with in certain embodiments 80-90% of its charge being retained for the next pulse. The turn-off value may be somewhat below the desired value to allow the desired level to be reached upon exceeding. A diode is provided in parallel with the electromagnetic winding when the current is cut off and provides a feedback path that prevents the current from being discharged, thereby losing less energy than conventional solutions. In one embodiment, the electromagnetic force is applied to a scroll wheel on a mouse, keyboard, or other computer peripheral. Hereinafter, such a scroll wheel application will be described in detail, and then the details of the novel driving circuit will be explained.Scroll wheel DetailA peripheral input device used as an interface between a user and a computing device may include a rotary input controller as a real element. The user may rotate the input controller to send a corresponding command to the computing device. An example of such input control is a scroll wheel that may be located between the left and right buttons on the top of a peripheral input device. Scroll wheels may be used to pan the field of view of a computer screen. For example, a scroll wheel may be used by a user to scroll through the view of a document displayed on the computer screen. Other possible controls are compatible with the described embodiments and may include, for example, a rotary wheel or rotary encoder. However, for simplicity, examples of a scroll wheel are used, which is not intended to limit the scope of the described embodiments under consideration.A scroll wheel may have various modes of operation. For example, an operating mode may be a free-wheeling mode in which the scroll wheel may be rotated about an axis with a relatively constant and low coefficient of friction (which may be referred to as a first friction profile). In such a mode, a user can quickly pan his view over a document with a single finger motion to rotate the wheel. Another mode may be a ratchet mode in which the scroll wheel has periodic segments of relatively high friction and lower friction segments therebetween (which may be referred to as a different friction profile from the first friction profile). Such a mode may allow a user greater control in panning through a document, as a single finger movement to rotate the wheel may result in a metered panning of a view.Some peripheral input devices allow a user to selectively activate a different friction profile for application to a scroll wheel to change the behavior of the scroll wheel according to a corresponding computer application, intended application, or user preference, for example. Various mechanisms are disclosed that may be used to change the friction profile applied to a wheel of a peripheral input device. Each of the mechanisms provides different characteristics with respect to power consumption, noise, user feel, and actuation time. In some embodiments, the friction profile may be changed in accordance with parameters provided by an active application. For example, the friction profile could increase sharply to signal a short pause / stop of scrolling to highlight a particular function. Additional force applied to overcome the increased friction profile may allow scrolling to continue and in certain cases trigger a change back to the original friction profile.These and other embodiments will be discussed below with reference to Figs. 2 to 9. However, those skilled in the art will readily appreciate that the detailed description provided herein with reference to these figures is for illustrative purposes only and is not to be construed as limiting.FIG. 1 shows an example user input device 100 suitable for use with the described embodiments and in the form of a wireless mouse. The wireless mouse 100 includes a housing 102 and input keys 104 and 106. Between the input keys 104 and 106, a rotatable input controller 150 in the form of a scroll wheel (also referred to as a "wheel") is disposed. The rotational input controller 150 may include a mechanism by which a ratchet-shaped friction profile for rotating the rotational input controller 150 may be realized. The mechanism may include an electro-permanent solenoid actuator ("EPM") for changing a friction profile associated with rotation of the rotary input controller 150, as further described in the following embodiments.FIGS. 2A and 2B show an example electro permanent magnet 200. In particular, the electro permanent magnet 200 comprises a first permanent magnet 202 and a second permanent magnet 204. The first permanent magnet 202 may have a higher self-coercive field strength than the second permanent magnet 204. In some embodiments, the first permanent magnet 202 may take the form of a rare earth magnet (e.g., neodymium-iron-boron or samarium-cobalt) and the second permanent magnet 204 may take the form of a ferromagnetic magnet (e.g., aluminum nickel cobalt alloy or ferrite). The lower inherent coercive force of the second permanent magnet 204 allows a magnetizing coil 206 to emit a magnetic field of sufficient strength to reverse a polarity of the magnetic field emitted by the second permanent magnet 204 without affecting the magnetization of the first permanent magnet 202. For example, in some embodiments, a self-coercive force of the first permanent magnet 202 may be more than ten times greater than a self-coercive force of the second permanent magnet 204. The lower intrinsic coercive force of the second permanent magnet 204 also reduces the amount of electrical energy used to reverse the polarity of the second permanent magnet 204, thereby enabling more efficient operation of the electropermanent magnet 200. The first permanent magnet 202 and the second permanent magnet 204 are each arranged between and in direct or at least close contact with the ferromagnetic poles 208. The ferromagnetic poles 208 may be formed of a ferritic material, such as structural steel, that has an even lower intrinsic coercive field strength than the second permanent magnet 204. The ferromagnetic poles 208 help direct the magnetic fields emanating from the first permanent magnet 202 and the second permanent magnet 204. In some embodiments, the size and shape of the ferromagnetic poles 208 may be adjusted to generate a magnetic field having a desired size and shape.FIG. 2A shows dashed lines 208 illustrating magnetic flux emitted from the electro-permanent magnet 200 that shows how with both the first and second permanent magnets 202 and 204 aligned in the same direction, magnetic flux is released from the electro-permanent magnet 200 to generate well-defined north and south poles. This magnetic field is symmetrical as shown when the strengths of the magnetic fields emitted by the two permanent magnets are approximately equal.FIG. 2B shows how, when the polarity of the first permanent magnet 202 is opposite the polarity of the second permanent magnet 204, the magnetic flux generated by both permanent magnets remains substantially contained within and circulates through the ferromagnetic poles 208 of the first permanent magnet 202 and the second permanent magnet 204. This results in the electropermanent magnet 200 emitting little to no magnetic field. These principles can be applied to the embodiments that follow in Figures 3A-7.FIG. 3A shows a perspective view of an example implementation in which an electro-permanent magnet is configured to change a resistance profile of a rotary input controller 150 compatible with the device illustrated in FIG. 1. The electro-permanent magnet 300 is disposed within a central opening defined by a ferromagnetic wheel 302. The ferromagnetic wheel 302 includes a plurality of teeth 304 that protrude into the central opening and toward the electropermanent magnet 300. The electro permanent magnet 300 comprises a first permanent magnet 306 and a second permanent magnet 308. The magnetizing coils 310 and 312 are configured to reverse a polarity of the magnetic field emitted from the second permanent magnet 308 to change a resistance profile of the rotary input controller 150. The ferromagnetic poles 314 each include radially projecting teeth 316 that are equidistant from the teeth 304 of the ferromagnetic wheel 302. The radially protruding teeth 316 concentrate the magnetic field emitted from the electro-permanent magnet 300 such that rotation of the ferromagnetic wheel 302 produces a resistance profile that provides the user with a varying amount of resistance, the resistance variation occurring with a wheel corresponding to a speed at which the ferromagnetic wheel 302 rotates. The variation in resistance is caused by the interactions between the magnetic field emitted by the electropermanent magnet 300 and the ferromagnetic materials in the teeth of the ferromagnetic wheel 302. It should be noted that throughout this disclosure, reference is made to "poles" that may include the two ends of a magnet or at the ends of a magnetically conductive structure coupled to the end of a magnetic pole, thereby effectively positioning the magnetic poles at the end of the magnetically conductive structure. For example, in Figure 4A, there are ferromagnetic poles 314 that couple magnetically conductive elements to the magnet itself. In operation, the lines of magnetic flux originate from the magnetically conductive elements, rather than from the end of the magnets (as they conduct the magnetic field), and magnetically interact with the teeth 304 of the wheel 302 as described below and as would be understood by one of ordinary skill in the art having the advantages of this disclosure.FIGS. 3B and 3C show a support structure for the rotary input controller 150. FIG. 3B shows a side view of the rotary input controller 150 disposed through a support structure 316 via a support surface 315. The central opening of the ferromagnetic wheel 302 is covered by a non-magnetic bearing assembly 318 that includes a self-lubricated axle 320 that can be configured to stabilize the ferromagnetic wheel 302 during use by engaging a bearing of the housing 102 (not shown). In some embodiments, the support surface 315 may take the form of a wall of an input device housing, such as the housing 102 shown in FIG. 1. In some embodiments, the support structure 316 may be integrated or incorporated into the wall of the input device housing.FIG. 3C shows an exploded view of the rotary input controller 150 and the support structure 316. In particular, the teeth 304 do not extend axially through the central opening defined by the ferromagnetic wheel 302, but instead leave space for a portion of the bearing assembly 318 to press fit into the ferromagnetic wheel 302. The interference fit provides a simple way for the bearing assembly 318 to be axially aligned with the ferromagnetic wheel 302. Alternatively, the ferromagnetic wheel 302 could also be bonded to one side of the ferromagnetic wheel 302. FIG. 3C also shows how the electro-permanent magnet 300 may be coupled to the support structure 316 and how a shaft 322 extends through a central portion of the electro-permanent magnet 300. In particular, the shaft 322 may extend between the first permanent magnet 306 and the second permanent magnet 308. The shaft 322 engages an aperture defined by the self-lubricated axle 320 to couple the ferromagnetic wheel 302 to the support structure 316. It should be noted that in some embodiments, the bearing mount 318 and the support structure 316 may both be made of polymeric material to avoid any undesired contact with the electro permanent magnet 300.FIGS. 4A and 4B show cross-sectional views of the rotary input controller 150 in which a polarity of the magnetic fields emitted from the permanent magnets 306 and 308 is oriented in the same direction. FIG. 4A shows how a magnetic flux emitted by the radially protruding teeth 316 interacts with the ferromagnetic material of which the teeth 304 consist. In the position shown, each of the teeth 304 is positioned between two adjacent radially protruding teeth, which results in the resistance to rotation of the ferromagnetic wheel 302 in both directions being low. That is, FIG. 4A shows protruding teeth 316 configured in their furthest position with respect to teeth 304, causing weaker magnetic coupling and thus providing lower resistance to rotation of the rotary input controller 150. However, when the radially protruding teeth 316 are aligned with a respective one of the teeth 304, as shown in FIG. 4B, rotation of the ferromagnetic wheel 302 becomes more difficult because rotation in both directions further removes the teeth 304 from a respective radially protruding teeth 316. That is, FIG. 4B shows protruding teeth 316 configured in their narrowest position with respect to teeth 304, which provides greater magnetic coupling and thus greater resistance to rotation of the rotary input controller 150. In this way, a resistance profile may provide ratchet feedback to a user without requiring movement of the electropermanent magnet 300. In some embodiments, the ferromagnetic wheel may include a tactile ribbed layer that improves the grip of a user's finger on the rotary input controller 150.FIG. 4C shows another cross-sectional view of the rotary input controller 150 in which a polarity of the permanent magnet 308 has been reversed. This results in the magnetic flux 402 being contained within the ferromagnetic poles 314 because the polarity of the permanent magnets allows the magnetic flux 402 to circulate within the magnetic circuit defined by the permanent magnets 306 / 308 and the ferromagnetic poles 314. This results in there being little to no interaction between the electro-permanent magnet 300 and the ferromagnetic wheel 302, whereby a user does not experience tactile feedback during rotation of the rotation controller 150. In practice, this may feel like a constant profile with less frictional resistance (e.g., a "free-wheeling" mode of operation).In some embodiments, multiple coils may be used to integrate more than two modes of operation, typically an "on" state in which there is magnetic induced rotational friction as shown in FIGS. 4A and 4B and an "off" state in which there is little to no magnetic induced rotational friction as shown in FIG. 4C. For example, two or more EPMs may each be individually "powered on" (fully energized; current flow) or "powered off" (fully de-energized; no current flow) in a series of different combinations to achieve different amounts of magnetic coupling between the protruding teeth 316 and the teeth 304. The two or more EPMs may be configured together, in series, parallel, in series-parallel, or in any other suitable configuration.EM DriverFIG. 5 is a system diagram of a force control architecture according to certain embodiments. A battery cell 502 provides the energy required to power the circuit. A power converter 504 transfers the charge from the battery cell and accumulates it in the main discharge capacitor, which is the energy reservoir 506. The energy reservoir 506 stores the energy that is directly supplied to an actuator 514 via an output stage 512.A fault comparator 508 continuously monitors the difference between the current actuator current and the current demanded by a microcontroller 518. A digital timing controller 510 starts an actuation cycle upon command of the microcontroller 518, but stops it upon reaching a reference value. The reaching of the reference value is signaled by an output of comparator 508.Output stage 512 is digital, with transistors acting only as switches. In one embodiment, the transistors form an H-bridge circuit. The actuator 514 is an electro permanent magnet (EPM) in one embodiment. The instantaneous current adjusts the magnetization of the EPM magnet. A current sense resistor 516 provides feedback to the fault comparator 508 and converts the current through the winding (coil) of the solenoid of the actuator 514 into a voltage that can be applied to the fault comparator. Microcontroller 518 performs several functions, specifically it provides the actuator peak reference current and the command to digital timing controller 510 to start the actuation cycle.FIG. 6 is a simplified circuit diagram for charging an electromagnet, in accordance with certain embodiments, with a switch in a charging position. FIG. 6 illustrates capacitor 506 connected to a solenoid 604 of actuator 514 via the H-bridge circuit of output stage 512. Also shown is the sense resistor 516 connected to an input of the comparator 508 which provides its output to the digital controller 510. Comparator 508 also receives a reference value from the microcontroller on line 606 and a voltage from current sense resistor 516 on line 610. The digital controller 510 controls the bridge 512 via the control line 608. An initial voltage 602 is applied to bridge 512, although in FIG. 6, the bridge has already switched to apply the voltage across capacitor 506 to the coil of electromagnet 604.In an embodiment where the electromagnet 604 is an inductive actuator (e.g., solenoid) the main parameters shown either in FIG. 6 or discussed with reference to the following equations include:L_actuator (or L) inductance of actuator 604;C_tank (or C) - is the tank capacity;V_initial (or V 0) - is the initial voltage of C_tank before discharge via the actuator;R_total (or R) - parasitic series resistance of bridge (switch) 512, current sense resistor 516 and the above-mentioned ESR (equivalent series resistance) of coil 604 and capacitor 506;- I_target - target peak current for the driver circuit;I_delta_error - current overshoot; andt ctrl- propagation delay in the error comparator (508 in FIG. 5 ), in the digital timing controller (510 in FIG. 5 ) and in the output stage (for example H-bridge circuit).FIG. 7 is the simplified circuit diagram for charging an electromagnet as in FIG. 6, with the switch in a shut-off position, in accordance with certain embodiments.In the initial state shown in FIG. 7, the capacitor C_tank 506 is charged by the initial voltage V_initial 602. The digital controller 510 controls the bridge 512 via the control line 608 to connect the capacitor 506 to the initial voltage 602 for charging the capacitor. Upon command of the microcontroller to the digital controller 510, the digital controller 510 allows the bridge 512 to switch to connect to the solenoid 604, as shown in FIG. 6, and the current begins to flow from C_tankof the capacitor 506 back to the capacitor (via the bridge 512, the actuator 604, and the current sensing resistor 516. The connection of the capacitance C of the capacitor 506 to the inductance L of the electromagnet 604 forms an LC tank circuit. The current may be expressed by the following equations (1) for under-attenuated oscillations in the improved LC tank: where: ω d= attenuated oscillation frequency, w 0= normal oscillation frequency of the LC system.Due to the propagation delay t ctrl through the comparator, digital controller and bridge, a systematic error occurs affecting the accuracy of the peak current setting. In other words, the system exceeds:Even if the control delay t ctrl can be assumed to be constant, the current systematic current error is not constant, but rather a function of the time (or of the reference current and thus of the time end point). The further the current proceeds from 0 to its peak, the less the systematic error.In this type of driver system, there is no problematic oscillation, since immediately after the moment the bridge is disconnected from the controller, a free wheeling diode 704 is activated, which closes the current path from the actuator coil, as shown in FIG. 7 with the current path 702.An RLC circuit is formed by the capacitor 506, the coil of the electromagnet 604, and the resistor R of the parasitic series resistance of the bridge (switch) 512, the current sense resistor 516 and the ESR of the coil 604 and the capacitor 506. Whether the RLC circuit is underdamped or critically attenuated depends on the inductance of the actuator, the capacitance of the tank capacitor, and the resistance comprised of the R sensor 516 and the overall series parasitic resistance of the circuit. The above-mentioned equation for the critically damped oscillation (in which no oscillations are possible even without the freewheeling diode) is as follows:The pulse driver of embodiments of the present invention takes advantage of advances in comparator technology along with the lower gate capacitance high current MOSFET transistors. The transistors in the H-bridge circuit are smaller in size than required to provide a constant current at the desired current level. This makes use of the fact that such transistors can process a much higher current pulse than the constant rated current capacity. A smaller transistor has a smaller gate with a lower gate capacitance and thus a faster switching speed. Voltage-current conversion via a controlled impedance is used for the feedback path, which allows the use of a comparator to simplify the control circuit. A fast comparator design for feedback control is used to achieve the fast switching speed required to limit the overshoot. A flip-flop, which operates like a silicon controlled rectifier (SCR) or also thyristor, is connected to the comparator output in order to enable the fast switching. The flip-flop transistors are configured at least as a silicon-controlled rectifier (SCR) or else as a thyristor or at least two transistors are configured as a silicon-controlled rectifier (SCR) or else thyristor or as a flip-flop.The faster the comparator is, the smaller the error between the desired current and the actual shut-down current value. In one embodiment, the comparator is also precise. In order to increase the efficiency of the circuit, it is important to reduce the galvanic (ohmic) losses. The sp resistance at which the current is measured is preferably minimized. This also reduces the voltage swing at the comparator input. Thus, the comparator is a precision class to detect a small voltage difference between the reference voltage (representing the desired current) and the shunt voltage drop (representing the immediate current in the H-bridge circuit). In one embodiment, the comparator has a propagation delay of less than 300 nano seconds and a voltage difference between the reference voltage and the shunt voltage drop of less than 10 mV, in one embodiment 8 mV. This is done at a shunt resistance of 0.1 ohm, + / -8 mV / 0.1 ohm= + / -80 mA. + / -80 mA / 20 A= + / -0.4%. With a voltage offset of 8 mV in the comparator, the static current adjustment can thus take place with an error of + / -0.4%. The greater the difference between a time period of the natural oscillation frequency of the LC resonator and the comparator delay, the better it is for the precision adjustments of the output current. Since the current cut-off condition is based on the result of the comparison of Vref (reference voltage) with the voltage across the shunt resistor, the lower the voltage offset in the comparator, the better the performance.The switching delay is the sum of several contributing factors:The major contribution to the overall control delay t ctrl is the delay within the bridge t bridge the delay of the control t controller is negligible.One of the design tradeoffs is the size of the H-bridge circuit transistors that are activated only for a very short time. The maximum pulse current is typically 10 times higher than the continuous maximum current for the same output transistors. Operation in the pulse mode allows selection of much smaller transistors with smaller gate capacitance, making the circuit even faster. The faster the circuit is, the less time is required to transition the output transistors from the high impedance state to a fully boosted state, and thus there are fewer losses.This dependency helps minimize the size of the transistors by knowing the maximum time required to achieve the peak current (RLC operates in the series resonant mode).In other words, by varying the design parameters such as L of the actuator 604 and C of the tank capacitor 506, the t peak- value for the driver may be adjusted. The driver operates between 0 and time t peak.In the embodiments, the invention provides a pulse resonance system in which the control method causes the maximum duration of the rising pulse to be less than or equal to π / 2, where π / 2 is the phase increment in the arc corresponding to 90°, π / 2 of the attenuated natural oscillation frequency ω d. The pulse then transitions into a monotone decay process. Accordingly, there is no tendency to swing due to design. Since the linear force profile of the EPM does not require that the full current must flow via the EPM windings, the pulsing method then offers a considerable energy saving, which makes it possible to send a series of fast discharge pulses from the same storage capacitor. In addition, it is possible to recharge the capacitor 10 times more quickly, since only 10% of the capacitor charge is consumed instead of 100% as in the previous resonance methods. Further, the systematic error (the constant error / offset due to the time delay) is calibrated and cancelled (in feedforward as offset or calibration constant) during a calibration period. This method works because the part of the current curve is linear with the force profile. Calibration is discussed in more detail below with reference to Figures 17-20.FIG. 8 is a simplified circuit diagram according to certain embodiments for the digital controller 510 of FIGS. 6 and 7. In one embodiment, flip-flop 802 is a D-type flip-flop. A D flip-flop is a clocked flip-flop that can be constructed from a clocked SR flip-flop with an inverter added between the S and R inputs to enable a single D input (data). Then, this single data input labeled "D" is used in place of the "set" signal and the inverter is used to generate the complementary "reset" input, thereby forming a level sensitive D flip-flop from a level sensitive SR flip-flop. Flip-flop 802 may be implemented with discrete semiconductors or an integrated circuit (IC).In Figure 8, data (D) input 806 is simply connected to a voltage (VDD) 801 to always apply a high level (digital one). A clock input 810 is connected via a line 808 to the microcontroller, which provides on this line a signal for starting the discharge of the capacitor. As shown, fault comparator 508 compares two inputs, a voltage corresponding to the sensed current from resistor 516 on line 610, and the target current reference value from the microcontroller on line 606. Flip-flop 802 is set to set input 810 in response to the microcontroller command on line 808 to start the actuation cycle. Flip-flop 802 is reset at reset input 812 by the feedback of comparator 508.FIG. 9A is a block diagram of an electromagnetic charging system, wherein a switch is in a charging position, according to certain embodiments. While FIGS. 6 and 7 were simplified diagrams showing a single switch, FIG. 9A shows the complete H-bridge circuit configuration with four transistor switches 910, 912, 914, and 916 controlled by four digital controllers 902, 904, 906, and 908. Transistors 910 and 916 are NMOS transistors, and transistors 912 and 914 are PMOS transistors. An integrator 918 converts a pulse width modulation (PWM) control signal from the microcontroller 518 to a voltage reference setting provided to the error comparator 508.An amplification stage 922 is connected to a simplified terminal 924 which is connected to the capacitor 506. The gain stage 922 and the terminal 924 implement the gain block 504 of FIG. 5. the microcontroller 518 controls the gain stage 922 and the terminal 924 to provide the charging current from the battery (not shown, battery 502 in FIG. 5 ) to the capacitor 506. The error comparator 508 provides a feedback signal to the D flip-flop 920. This signal controls the turn-on of transistor 910 in H-bridge circuit 512 (transistors 910, 912, 914, and 916) as described above. Although the control line is not shown in FIG. 9A, the D flip-flop 920 also turns on the H bridge circuit transistor 914, as described in more detail below with respect to FIG. 9B. When these two transistors are turned on, this allows current to flow from capacitor 506, as illustrated by arrows 926 and the other arrows that show the flow through transistor 914, through EPM coil 604, through transistor 910, and through sense resistor 518, which provides the feedback voltage to fault comparator 508.In the conventional approach discussed in the introduction, linear control of the transistor is used, while the embodiments of the present invention use digital control. The conventional linear control method requires the error amplifier to control the output current of the EPM coil by linear control of the output NMOS transistor. In this case, the excessive power (20-30%) needs to be dissipated in the output NMOS transistor. This immediately reduces the efficiency of the circuit and also increases the complexity, since the possible stability problems (driving the inductive load with the very high current) can lead to undesired oscillations, as well as current overshoot at the beginning of the discharge cycle. Moreover, since the NMOS transistor in the linear control circuit operates in the linear mode, its size must be adjusted to the amount of dissipated energy, making the device expensive, bulky and unreliable (thermal cycles).However, the circuit of the embodiment of Figures 9A and 9B does not have a fast current control circuit. It is digital. The NMOS output transistor is either turned off or amplified, thus energy dissipation is minimal. In addition, the thermal dissipation is tiny, which makes it possible to have a much smaller transistor pad, making the device cheaper, smaller and more reliable. Finally, an amount of energy is conserved for the next EPM cycle.FIG. 9B is the block diagram of FIG. 9A with multiple added control details, in accordance with certain embodiments. When microcontroller 518 is ready to start a new current pulse, the desired peak current value is applied in the form of a voltage to the input of comparator 508. The voltage corresponding to the desired peak current is converted from a digital output of the microcontroller to an analog voltage value at the comparator input using a digital-to-analog converter (DAC). The DAC includes a pulse width modulated (PWM) signal from the microcontroller averaged by a low pass filter (LPF) 919. The output of the LPF 919 is applied to the positive terminal of the comparator.Microcontroller (μC) 518 has two digital positive / negative discharge control lines across EPM 604 (defining the activation polarity of the H-bridge circuit). The first control line is the activated control line 930 that goes high to order the EPM pulse. This low-to-high transition on line 930 sets D flip-flop (DFF) 920 to an active state in which it can be reset by comparator 508 as discussed above. The output for the D flip-flop is one input to the AND gate 934, the other input being the activated control line 930. AND gate 934 triggers digital controller / driver 902 which turns on N-MOS transistor 910 of the H-bridge circuit. This is achieved because at the low-to-high transition in the clock of DFF 920, the logic one is latched into the DFF and enables AND gate 934, which in turn enables N-MOS driver 934. The corresponding P-MOS transistor 914 is enabled directly by the uC command on line 930 through the digital controller / driver 906.The current then begins to increase across the EPM coil 604, and feedback is accomplished by measuring the voltage drop across the resistor (Rsense) of the current sense resistor 516. The current setting is calculated using the following simple formula:Ipeak=Rbac / Rsense. For example, if Rsense is 0.1 ohm and the target current is 10 amps, the DAC voltage must be set to a 1V.The current increases via the EPM coil, i.e. the voltage drop across the current measuring resistor also increases.To drive the current in the opposite direction, inactive control line 932 goes high and turns on PMOS transistor 912 and NMOS transistor 908. Transistor 908 is turned on via AND gate 936, similar to AND gate 934.FIG. 10 is a block diagram of an electromagnetic charging system with the H-bridge circuit switch in a deactivated, free-wheeling position, in accordance with certain embodiments. FIG. 10 is identical to FIG. 9A, except for the addition of a diode 704 (see FIG. 7 ) in parallel with the transistor 912 to enable a free-wheeling mode. In this mode, charging via the H-bridge circuit 512 is stopped by the capacitor 506 by turning off the transistor 910. Transistor 914 remains on. Thus, as shown by the arrows, there is a circular current flow through transistor 914, EPM coil 604, diode 704, and back to transistor 914.The discharging operation of the capacitor is performed via the H bridge circuit. The digital controller is enabled by the microprocessor, but the DFF 920 is already enabled. This state is maintained under the timer function in the microprocessor for a fixed time period corresponding to the full amplitude EPM charge cycle. The duration of this cycle begins with the activation of the H-bridge circuit and the digital controller by the microprocessor. Roughly speaking, this cycle corresponds to π or 180° of the evolution of (natural oscillation frequency). After this period of time, the digital controller is turned off by the microprocessor, resulting in the turning off of transistor 914 (or 912, depending on which transistor pair or in which direction the current flows through EPM coil winding 604). Thereafter, the system is brought into the IDLE state.Reference is again made to FIG. 9B. At some point, the voltage across current sense resistor 516 will be slightly above the programmed DAC reference voltage representing the output current (1.0V as mentioned in the above example). In this case, the output of comparator 508 goes to ZERO, resetting DFF 920 and disabling N-MOS transistor driver 902 through AND gate 934, turning off NMOS transistor 910. While the PMOS transistor 914 remains in the ON state, the NMOS transistor 910 is turned off because the DFF 920 is reset. From this moment, the current through the EPM coil 604 ceases to increase and begins to flow via the freewheeling diode 940 of the opposing PMOS transistor 912. The diode 49 may be a body diode or an external diode.For a current pulse in the opposite direction, the mirror circuit is used. The NMOS transistor 916 is turned off and the current through the EPM inductor 604 ceases to rise (fall) and begins to flow across the free wheeling diode 942 of the opposing PMOS transistor 914.In one implementation, the final circuit is highly optimized with optimized circuits that replace the AND gates and the DFF using techniques known to those skilled in the art. For example, a simple silicon controlled rectifier (SCR) may also be used to function as a flip-flop and to block the action of the N-MOS transistors as soon as the output current reaches its destination until the end of the discharge cycle.FIG. 11 is a circuit diagram of the comparator and bidirectional thyristor, also triac blocks and flip-flop blocks of FIGS. 9 and 10, respectively, in accordance with certain embodiments. The circuit shown includes comparator 508 and flip-flop 802.FIG. 12 is a circuit diagram of the output stage block and H-bridge circuit block of FIGS. 5 and 10, respectively. The free wheeling diode 704 is also shown.FIG. 13A is a graph showing a charging and discharging cycle according to the related art with a curve of magnetization to saturation. Line 1302 shows the capacitor voltage starting at 29.4 V in one embodiment. Line 1304 shows the current from the capacitor to the EPM coil via the H-bridge circuit. As can be seen, the current peaks and then slowly drops to zero, completely dissipating the energy of the capacitor. The capacitor voltage starts at 29.4 V at point 1301 and decreases to 0 V at point 1303. Arrows 1305 indicate how much energy is lost in the capacitor during discharge. This is because the current value is continuously controlled to avoid oscillations as described above. As a result, the capacitor must be fully recharged before another pulse can be generated.FIG. 13B is a graph showing a charge and shut down cycle without saturation, in accordance with certain embodiments. Here, line 1306 represents the capacitor voltage and line 1308 represents the charging current. In contrast to FIG. 13A, the capacitor current is not constantly controlled, but rather is switched off shortly before its peak value is reached. As can be seen, the current falls rapidly, preventing complete discharge of the capacitor. In the example shown, capacitor voltage 1306 begins at 29.4 V and ends at 23.8 V. Thus, approximately 65% of the capacitor energy is still present after the cycle, as indicated by arrows 1310. This is actually a worst case representation and typically 80% of the capacitance energy will remain. Thus, the capacitor can be recharged much more quickly, so that a subsequent current pulse can be provided more quickly. Embodiments of the invention may shut down in 10 seconds of nanoseconds and the overshoot is one percent or less. In one embodiment, the discharge time is about 100 μs, and only about 50 μs full charge time is required, which makes a system about 500 times faster than the conventional design. As can be seen, the current falls immediately (the current flowing through the capacitor and resistor 516 prevents the capacitor 506 from fully discharging). However, after the capacitor is turned off, the current continues to flow through the EPM, decreasing rapidly.In summary, it can thus be concluded that in one embodiment a circuit is provided with a battery 502, a multi-ampere capacitor 506 and a charging circuit 504, which is connected between the battery and the capacitor. An actuator 514 is actuated by an electromagnetic coil 604. A switching circuit 512 is connected between the multi-ampere capacitor and the electromagnetic coil. The switching circuit includes an H-bridge circuit 512 of four transistors 910, 912, 914 and 916. A diode 704 is configured in parallel with the electromagnetic coil to provide a current feedback path when the capacitor is disconnected from the electromagnetic coil. A measuring resistor 516 is connected to the electromagnetic coil. A comparator 508 has a first input connected to the sense resistor and a second input connected to a reference voltage 606. The digital controller 510 is connected to an output of the comparator. The digital controller is a D flip-flop 802, and is programmed to provide a signal at an output of the digital controller that is connected to the switching circuit to disconnect the capacitor from the electromagnetic coil when a current from the capacitor to the electromagnetic coil generates a voltage at the sensing resistor that reaches the reference voltage.FIG. 14 shows an embodiment of a method for charging an electromagnetic actuator. Step 1402 is charging a multi-amp capacitor with a battery. Step 1404 is switching the capacitor from the connection to the battery for charging to the connection to the electromagnetic actuator for providing a current pulse for activating the electromagnetic actuator. Step 1406 is detecting the amount of current supplied to the solenoid actuator. Step 1408 is disconnecting the capacitor from the solenoid actuator when it is determined that the magnitude of the current reaches a threshold below a desired current peak. Step 1410 is recharging the multi-amp capacitor with the battery.FIG. 15 illustrates a system 1500 for operating a host computing device (e.g., host computing device for 1510) to control certain features of the peripheral devices disclosed herein, in accordance with certain embodiments. The system 1500 may be used to implement any of the host computing devices or peripheral interface devices discussed herein and the numerous embodiments defined herein or within the scope of this disclosure, but not necessarily explicitly described. The system 1500 may include one or more processors 1502 that may communicate with a number of peripheral devices (e.g., input devices) via a bus subsystem 1504. These peripheral devices may include a storage subsystem 1506 (including a storage subsystem 1508 and a file storage subsystem 1510), user interface input devices 1514, user interface output devices 1516, and a network interface subsystem 1512. The user input devices 1514 may be any of the types of input devices described herein (e.g., keyboard, computer mouse, remote control, etc.). User output devices 1516 may be a display of any type, including computer monitors, displays on handheld devices (e.g., smart phones, gaming systems), or the like, as understood by one of ordinary skill in the art. Alternatively or additionally, may include a virtual reality (VR) display, augmented reality display, holographic display, or the like, as will be understood by one of ordinary skill in the art.In some examples, internal bus subsystem 1504 may provide a mechanism to allow the various components and subsystems of computer system 1500 to communicate with each other as provided. Although internal bus subsystem 1504 is shown schematically as a single bus, alternative embodiments of the bus subsystem may use multiple buses. Additionally, network interface subsystem 1512 can serve as an interface for communication of data between computer system 1500 and other computer systems or networks. Embodiments of the network interface subsystem 1512 can include wired interfaces (e.g., Ethernet, CAN, RS232, RS485, etc.) or wireless interfaces (e.g., Bluetooth® BLE, ZigBee® Z-Wire® Wi-Fi, cellular protocols, etc.).In some cases, the input devices of the user interface 1514 may include a keyboard, a presenter, a pointing device (e.g., mouse, trackball, touchpad, etc.), a touch screen integrated into a display, audio input devices (e.g., voice recognition systems, microphones, etc.), human machine interfaces (HMI), and other types of input devices. In general, the use of the term "input device" is intended to encompass all possible types of devices and mechanisms for inputting information into the computer system 1500. Additionally, the output devices of the user interface 1516 may include a display subsystem, a printer, or non-visual displays such as audio output devices, etc. The display subsystem may be any known type of display device. In general, use of the term "output device" is intended to encompass all possible types of devices and mechanisms for outputting information from computer system 1500.Storage subsystem 1506 may include storage subsystem 1508 and file storage subsystem 1510. Storage subsystems 1508 and file storage subsystems 1510 represent non-transitory computer readable storage media that can store program code and / or data that provide the functionality of embodiments of the present disclosure. In some embodiments, a storage subsystem 1508 may include a number of memories, including main random access memory (RAM) 1518 for storing instructions and data during program execution and read only memory (ROM) 1520 in which fixed instructions may be stored. The file storage subsystem 1510 may provide persistent (i.e., non-transitory) storage for program and data files, and may include a magnetic or solid state hard disk drive, an optical drive along with associated removable media (e.g., CD-ROM, DVD, Blue-Ray, etc.), a removable flash memory-based drive or card, and / or other types of storage media known in the art.It should be understood that computer system 1500 is illustrative and is not intended to limit embodiments of the present disclosure. The various embodiments may further be implemented in a variety of operating environments, which in some cases may include one or more user computers, computing devices, or processing devices that may be used to operate multiple applications. User or client devices may include multiple general purpose personal computers, such as desktop or laptop computers running a standard or non-standard operating system, as well as cellular, wireless, and hand-held devices running mobile software, capable of supporting a variety of network and messaging protocols. Such a system may also include a number of workstations running a variety of commercially available operating systems and other known applications for purposes such as development and database management. These devices may also include other electronic devices, such as dummy terminals, thin clients, gaming systems, and other devices capable of communicating over a network.Most embodiments utilize at least one network known to those skilled in the art to support communication using a variety of commercially available protocols, such as TCP / IP, UDP, OSI, FTP, UPnP, NFS, CIFS, and the like. The network may be, for example, a local area network, a wide area network, a virtual private network, the Internet, an intranet, an extranet, a public telephone network, an infrared network, a wireless network, or any combination thereof.In embodiments using a network server, the network server may include a variety of server or mid-tier applications, including HTTP servers, FTP servers, CDI servers, data servers, Java servers, and business application servers. The server(s) may also be capable of executing programs or scripts in response to requests from user devices, for example, by executing one or more applications that may be implemented as one or more scripts or programs written in a programming language including, but not limited to, Java ®, C, C#, or C++, or a scripting language such as Perl, Python, or TCL, as well as combinations thereof. The server(s) may also comprise database servers, including, without limitation, those commercially available from Oracle ®, Microsoft ®, Sybase ® and IBM ®.Such devices may also include a read-only device for computer readable storage media, a communication device (e.g., a modem, a network card (wireless or wired), an infrared communication device, etc.), and memory as described above. The computer readable storage medium reader may be coupled to or configured to receive a non-transitory computer readable storage medium, which may be remote, local, fixed, and / or removable storage devices, as well as storage media for temporarily and / or permanently receiving, storing, transmitting, retrieving computer readable information. The system and the various devices will also typically include a number of software applications, modules, services, or other elements residing in at least one memory device, including an operating system and application programs such as a client application or browser. Custom hardware could also be used and / or certain elements may be implemented in hardware, software (including portable software such as applets), or both. Further, connections to other computing devices, such as network input / output devices, may be used.FIG. 16 illustrates a system 1600 for operating a peripheral device (e.g., user input device 100), in accordance with certain embodiments. The system 1600 includes process processor(s) 1610, memory array 1620, power management system 1630, communication system 1640, and input sensing module 1650. Each of the system blocks 1620-1650 may be in electrical communication with the processor(s) 1610 (e.g., via a bus system). System 1600 may further include additional systems not shown or discussed to prevent obscuring the novel features described herein. System blocks 1620-1650 may be implemented as separate modules, or alternatively, multiple system blocks may be implemented in a single module. It should be noted that system 1500 and system 1600 represent examples of some typical systems that may be used to implement the various embodiments described herein, as may be understood by one of ordinary skill in the art with the advantages of this disclosure.In certain embodiments, processor(s) 1610 includes(s) one or more microprocessors (μCs), and may be(can) configured to control the operation of system 1600. Alternatively, the processor(s) 1610(s) may include one or more microcontrollers (MCUS), digital signal processors (DSPS), or the like, with supporting hardware and / or firmware (e.g., memory, programmable input / output, etc.), as would be understood by one of ordinary skill in the art. The processor(s) 1610 may (may) control some or all aspects of the operation of the user input device 100 (e.g., system block 1620- 1650). Alternatively or additionally, some of the system blocks 1620-1650 may include an additional dedicated processor that may cooperate with the processor 1610.The storage array 1620 may be configured to store information related to one or more operating configurations of the user input device 100. As discussed further below, one or more operating configurations of the user input device 100 may include adjustment of the performance of the user input device 100, including, but not limited to, a computer mouse scroll speed, a sensitivity of the computer mouse motion sensor, mapping of keyboard hot keys, microphone volume, etc., and the like. The storage array 1620 may also store other configuration information used for communication with the user input device 100, as described further below.Additionally, the memory array 1620 may store one or more software programs executed by processors (e.g., in processor(s) 1610). It should be noted that "software" may refer to sequences of instructions that, when executed by processing unit(s) (e.g., processors, processing devices, etc.), cause the system 1600 to perform certain operations of software programs. The instructions may be stored as firmware residing in read-only memory (ROM) and / or as applications stored in a media memory that may be read into memory for processing by processing devices. Software may be implemented as a single program or as a collection of separate programs and may be stored in non-volatile memory and copied, in whole or in part, to volatile memory during program execution. With regard to the present disclosure, the memory array 1620 (along with the processor 1610) may include instructions (implemented via software, firmware, hardware, and a combination thereof) configured to implement the various control schemes described herein for the electropermanent magnet, such as determining the time for the capacitor current to shut down, recharging the capacitor, and introducing a second EM charge pulse.The power management system 1630 may be configured to manage power distribution, recharging, power efficiency, and the like for the user input device 100. In some embodiments, power management system 1630 may include a battery (not shown), a USB-based charging system for the battery (not shown), and power management devices (e.g., low-drop voltage regulators - not shown). In certain embodiments, the functions provided by the power management system 1630 may be incorporated into the processor(s) 1610. The power source may be a replaceable battery, a rechargeable energy storage (e.g., supercapacitor, lithium polymer battery, NiMH, NiMh), or a coded power supply. The charging system may be an additional cable (dedicated to the charging purpose) or it may use a USB connection to charge the battery.The communication system 1640 may be configured to provide wireless and / or wired communication between the processors 1610 and one or more of the user input devices 100, according to some examples. The communication system 1640 may be configured to provide radio frequency (RF), Bluetooth® BLE, WiFi, infrared (IR), ZigBee® Logitech, Unifying® or any suitable communication technology to communicate with other computing devices and / or peripherals. The communication system may also provide a hardwired connection to the processors 1610. The hard-wired connection may include a serial connection, such as a Universal Serial Bus (USB), FireWire® DisplayPort® etc.An example of communication system 1640 may be a dongle that can provide a combination of wireless and wired communication between processors 1610 and one or more of user input devices 100. For example, the dongle may include a wired connector interface (e.g., a USB connector) that can be plugged into a hardware interface port (e.g., a USB port). The hardware interface port, in turn, is connected to the processors 1610 via a hard-wired connection (e.g., USB buses). In addition, the dongle also included a wireless interface (e.g., a Bluetooth ®- wireless interface) to perform a wireless data transfer with the user input device 100. The dongle can receive sensor data from the user input device 100 via the wireless interface and transmit the sensor data to the processors 1610 via the hardwired connection.The input capture module 1650 may control the capture of user interaction with input items on the user input device 100. For example, the input acquisition module 1650 may acquire user input based on sensor data from the computer mouse 130. In some embodiments, the input acquisition module 1650 may operate in conjunction with the memory array 1620 to generate input data for the processors 1610 based on the sensor data received from the communication system 1640. For example, based on the scroll speed information stored in the memory array 1620 and the sensor data from the computer mouse 130, the input acquisition module 1650 may calculate a distance traveled by a mouse pointer on the display 120 and pass the calculation information to the processors 1610 (or a renderer) to present the movement of the mouse on the display 120.Although certain systems are not expressly discussed, they should be considered part of system 1600 as would be understood by one of ordinary skill in the art. For example, system 1600 may include a bus system to transfer power and / or data to and from the various systems therein.It should be noted that system 1600 is illustrative. Further, while the system 1600 will be described with reference to particular blocks, it will be understood that these blocks are defined for ease of description and are not intended to indicate a particular physical arrangement of components. Moreover, the blocks need not correspond to real different components. Blocks may be configured to perform various operations, for example, by programming a processor or providing suitable control circuitry, and various blocks may or may not be reconfigurable depending on how the initial configuration is obtained.CalibrationThe calibration method relies on instantaneous EPM coil current measurement during a single EPM magnetization cycle and curve distortion analysis, in which it is important to find the transition points of the magnetization curves (the correspondence between the magnetization current and EPM moment / force (the moment is referred to herein as UX)). The otherwise ideal half-sine current curve is severely distorted and the distortion range is the range in which the mechanical moment variation occurs. The typical EPM magnetization curve is shown in FIG. 17. Curve 1701 is the current flowing through the EPM coil winding. The corresponding tank capacitor voltage is represented by curve 1706. The range between the two bending points (marked by arrows 1702 and 1704) is the variable torque range that the calibration routine must find. The smaller, inserted diagram shows successive moment pulses 1708 and 1710 as a function of the magnetizing current. These curves lie between the two torque or mechanical force saturation ranges. The moment pulses are in a variable mechanical moment range in which the magnetizing current modulates the EPM moment.This EPM current curve distortion occurs due to the variable magnetic permeability under the magnetizing force throughout the EPM magnetization cycle, resulting in a dynamic variation of the EPM wheel coil inductance. Thus, the inductance varies along with the permeability value. In turn, this distorts the magnetization curve from the quarter sinusoidal shape to the shape with two bending points 1702 and 1704 on the slope (like a third order curve). By analyzing the current curve and finding the inflection points (calculating the second derivative of the current and searching for zero crossings on the rising current slope), it is possible to configure the magnetizing force curve solely by the realization of 4 parameters:(1) upper and (2) lower bending point stream;(3) Free wheel and (4) Full Ratchet Saturation Torque (value from the EPM Manufacturer's test bench - different for each EPM unit each time), resulting in an absolute calibration table with very high accuracy.As a result, there is a nearly exact correspondence between the magnetizing current bending points and the magnetizing force curve bending points (see the graph in FIG. 17 ). If the absolute torque value of the saturated EPM is not available from the manufacturer (the EPM manufacturer cannot provide its test data for each individual EPM and associate it with each individual EPM), the relative curve is still good enough for the variable relative torque UX.Even though the saturated EPM moment is unknown (not supplied by the manufacturer), it can be sized (only a single measurement) at the manufacturing line test station, thereby reducing the test time from nearly ten seconds to one second.Thus, the above method is the ultrafast alternative to the standard test method in which the magnetizing current is gradually increased while the EPM wheel rotates at a constant speed and the EPM moment is measured.By performing the function analysis and finding the two inflection points corresponding to the HIGH (Hi moment in FIG. 18 ) and LOW (Lo moment) moments, the other moment stages can be assigned by dividing the Hi-Lo curved band by the intervals. In the case of four torque stages + one free wheel, the band between the bending points has the following stage assignments:magnetizing current=0→un-running;25% the current band between the bending points->low moment;50% the current band between the bending points→mean moment;75% the current band between the bending points→high moment; andMagnetizing current = MAX -> saturated torque - (max. Ratchet).This method, without any mechanical calibration, allows well-defined torque bands with a perceptible increase in force between adjacent stages. However, the absolute force difference between different EPM units (EPM #1, EPM #2, etc.) is determined by the manufacturing tolerances, so the method applies only to the relative force. Since it is possible to distinguish the torque differences between adjacent stages, this satisfies the minimum requirement of the UX team and does not require the complex build-up of the roller force calibration.The EPM magnetization curve can be detected with a test station, as shown in FIG. 19. An example is shown where Unit Under Test (UUT) 1902 is a mouse with a scroll wheel 1904 with an EPM wrap 1906. The magnetization curve(s) is(are) obtained by supplying the full current to the EPM winding. In one embodiment, two curves are used for the two events R 2F (ratchet to free wheel) and F 2R (free wheel to ratchet). The test spots are placed in the area accessible by the test station's Cargo pins. Two contacts labeled "T" (one for GND and another for current measurement) are used to transmit the instantaneous voltage measurements at a current sense resistor 1908 to the measurement hardware, a digital storage oscilloscope (DSO) 1910 with an analog-to-digital converter (ADC) and sampler. In one embodiment, the bandwidth of the device is at least 1 MHz.An embodiment of calibration using an absolute torque calibration method is shown in FIG. 20. This method uses the same inflection point detection as the relative torque calibration method of FIG. 19 The difference is that after the corresponding current value is detected for 25%, 50%, 75% and 100% torque, these values are mechanically calibrated by rolling the EPM wheel and measuring the gear / ratchet friction.Four stages for one-way → ratchet and another four stages for one-way → freewheel can be measured with the tester shown in Fig. 20. There are 8 torque levels to be measured in total. Each step measurement is taken at least one full revolution of the EPM wheel 1904 at the test station. Assuming that the wheel is rotating at 300 rpm (5 rpm / second), the entire test is completed within 2 seconds. Then curve fitting (or a simpler polynomial or spline interpolation) is applied and the entire range of moments versus magnetizing current is calculated at the test station.The test assembly of Figure 20 includes a capstan 1912 biased against the EPM wheel 1904. The capstan is driven by a DC motor 1914 with an optical position transducer. The velocity from the optical position transducer is provided via output line 1916 to an analog velocity controller 1918. The speed controller 1918 receives a speed voltage on a line 1920 from a DC power source 1922. The DC power source 1922 is controlled by a test station 1926, which also receives a signal from the current sensor torque meter 1924. The haptic ratchet control signals for the EPM wheel 1904 are transmitted to the UUT 1902 via a dongle 1928 and an antenna 1930. The wireless signal is received by an antenna 1932, which is connected to a transceiver 1934, which provides the control signal to an EPM drive circuit 1936.Embodiments of the present invention may be implemented in devices including electronic devices implemented with a combination of circuits and software. Moreover, aspects and / or portions of system 1600 may be combined with or operated by other subsystems as required by the design. For example, the input sensing module 1650 and / or the memory array 1620 may operate within the processor(s) 1610 rather than functioning as a separate unit. Moreover, the inventive concepts described herein may also be applied to various peripheral devices and are not limited to computer mice, keyboards, or microphones. The system 1600 may be applied to any of the two embodiments described herein, regardless of whether it is explicitly, referencingly, or silently described (for example, if one of ordinary skill in the art would have known it to be applicable to a particular peripheral). The foregoing embodiments are not to be considered limiting, and those familiar with the advantages of this disclosure will understand the numerous applications and possibilities.The various embodiments illustrated and described are provided by way of example only to illustrate various features of the claims. However, the features shown and described with respect to a particular embodiment are not necessarily limited to the associated embodiment. Furthermore, the claims are not intended to be limited by a single embodiment.Although the present disclosure provides certain exemplary embodiments and applications, the scope of the present disclosure is intended to be defined only by reference to the appended claims.

Claims

A circuit comprising: a multi-amp capacitor (506); a battery (502); a charging circuit (504) connected between the battery (502) and the capacitor (506); an actuator (514); an electro-permanent magnet coil (604) configured to actuate the actuator (514); a switching circuit connected between the multi-amp capacitor (506) and the electro-permanent magnet coil (604), the switching circuit including an H-bridge circuit (512) of four transistors (910, 912, 914, 916); a diode (704) configured in parallel with the electro-permanent magnet coil (604) to provide a current feedback path when the capacitor (506) is disconnected from the electro-permanent magnet coil (604); a sensing resistor connected to the electro-permanent solenoid coil (604); a comparator (508) having a first input connected to the sensing resistor and a second input connected to a reference voltage (606); A digital controller (510) connected to an output of the comparator (508), the digital controller (510) comprising transistors performing a D-flip flop function, the digital controller (510) programmed to provide a signal at an output of the digital controller (510) connected to the switching circuit to disconnect the capacitor (506) from the electro-permanent solenoid coil (604) when a current from the capacitor (506) to the electro-permanent solenoid coil (604) generates a voltage at the sensing resistor that reaches the reference voltage (606).The circuit of claim 1, wherein the transistors performing a D-flip-flop function comprise at least two transistors configured as a thyristor or a flip-flop.The circuit of claim 1 or 2, wherein the four transistors (910, 912, 914, 916) of the H-bridge circuit (512) are configured for a constant current that is a multiple less than the current from the capacitor (506).The circuit of any of claims 1 to 3, wherein the reference voltage (606) is set below a desired peak charge voltage corresponding to a peak current from the capacitor (506).The circuit of any of claims 1 to 4, further comprising a haptic actuator coupled to the electro-permanent solenoid (604).A circuit comprising: a multi-amp capacitor (506); a battery (502); a charging circuit (504) connected between the battery (502) and the capacitor (506); an actuator (514); an electro-permanent solenoid coil (604) configured to actuate the actuator (514); a switching circuit connected between the multi-amp capacitor (506) and the electro-permanent solenoid coil (604); a sense resistor connected to the electro-permanent solenoid coil (604); a comparator (508) having a first input connected to the sense resistor and a second input connected to a reference voltage (606); A digital controller (510) connected to an output of the comparator (508), the digital controller (510) programmed to provide a signal at an output of the digital controller (510) connected to the switching circuit to disconnect the capacitor (506) from the electro-permanent solenoid coil (604) when a current from the capacitor (506) to the electro-permanent solenoid coil (604) generates a voltage across the sensing resistor that reaches the reference voltage (606).The circuit of claim 6, further comprising: a diode (704) configured in parallel with the electropermanent solenoid coil (604) to provide a current feedback path when the capacitor (506) is disconnected from the electropermanent solenoid coil (604).The circuit of claim 6 or 7, wherein the switching circuit comprises an H-bridge circuit (512) of four transistors.The circuit of claim 8, wherein the four transistors of the H-bridge circuit (512) are configured for a constant current that is a multiple of the current from the capacitor (506).The circuit of any of claims 6 to 9, wherein the digital controller (510) comprises transistors that perform a D-flip-flop function.The circuit of claim 10, wherein the transistors performing a D-flip-flop function comprise at least two transistors configured as a thyristor or a flip-flop.The circuit of any of claims 6 to 11, wherein the reference voltage (606) is set below a desired peak charge voltage corresponding to a peak current of the capacitor (506).The circuit of any of claims 6 to 12, further comprising a haptic actuator coupled to the electro-permanent solenoid (604).The circuit of claim 13, wherein the haptic actuator is a ratchet mechanism for a scroll wheel.A method of charging an electro-permanent magnet actuator, comprising: charging a multi-amp capacitor (506) with a battery (502); switching the capacitor (506) from connection with the battery (502) for charging to connection with the electro-permanent magnet actuator to provide a current pulse to activate the electro-permanent magnet actuator; detecting an amount of current supplied to the electro-permanent magnet actuator; disconnecting the capacitor (506) from the electro-permanent magnet actuator when it is determined that the amount of current reaches a threshold below a desired current peak; and recharging the multi-amp capacitor (506) with the battery (502).The method of claim 15, wherein a difference between the threshold and the desired current peak corresponds to the magnitude of the current provided by the capacitor (506) during a delay time between sensing the current pulse at a threshold and activating a circuit for switching the capacitor (506) to be disconnected from the electro-permanent solenoid actuator.The method of claim 15 or 16, wherein the threshold value and the desired current peak lie in a linear region of a charging curve of the current pulse.The method of any of claims 15 to 17, wherein the switching of the capacitor (506) is performed with four transistors (910, 912, 914, 916) in an H-bridge circuit (512), the four transistors (910, 912, 914, 916) being designed for a current that is many times smaller than the desired current peak to minimize their size and thus minimize their gate capacitance and switching time.The method of claim 18, wherein disconnecting the capacitor (506) from the electro-permanent solenoid actuator is performed by providing a control signal to the H-bridge circuit (512) using a D-flip-flop function with transistors operating as a thyristor or flip-flop.The method of any of claims 15 to 19, further comprising switching the capacitor (506) to connect to the electropermanent solenoid actuator in response to a ratchet signal for a scroll wheel in a computer peripheral.

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