RF amplifier system with hybrid power supply generator / modulator

Abstract

In some embodiments, a system comprises: a hybrid power supply generator / modulator comprising: an input for receiving an input voltage; an output for providing a modulated voltage; a multi-output control stage configured to provide two intermediate voltages with different voltage levels, wherein at least one of the two intermediate voltages is synthesized from the input voltage; and a multi-level converter configured to receive the two intermediate voltages and generate the modulated voltage with a voltage level corresponding to one of the two intermediate voltage levels or at least one synthesized voltage level that differs from both intermediate voltage levels;and a circuit arrangement coupled to the output of the hybrid power supply generator / modulator and to a power supply connection of at least one high-frequency (HF) amplifier, wherein the circuit arrangement is configured to modify the modulated voltage supplied to the at least one HF amplifier.

Description

[0001] The efficiency of high-frequency (HF) power amplifiers (PAs) can be improved through "supply modulation" (or "drain modulation" or "collector modulation"), in which the power supply voltage provided to the PA is dynamically adjusted ("modulated") over time depending on the synthesized HF signal. For the greatest efficiency improvements, a supply voltage can be adjusted discretely (between discrete levels) or continuously on a short timescale, tracking or dynamically responding to rapid variations in the HF signal amplitude (or envelope), such as can occur while data is being encoded in the HF signal, or when it is desired to change the HF signal amplitude with a high envelope bandwidth (e.g., as in envelope tracking, extended envelope tracking, polar modulation, "Class G" power amplification, multi-level backoff, multi-level LINC, asymmetric multi-level outphasing, etc.).The power supply voltage (or voltage level) provided to the PA can also be adjusted to accommodate longer-term changes in the desired RF envelope (e.g., "adaptive bias"), as is associated with adjusting the transmitter output strength to minimize data transfer errors, for RF "traffic" variations, etc.

[0002] “Continuous” supply modulation (e.g., “envelope tracking” or “adaptive biasing”) can advantageously be achieved by dynamically selecting an intermediate voltage from a set of discrete power supply voltages and then further controlling (reducing) this intermediate voltage to generate a continuously variable supply voltage provided to the PA, or by pulse width modulation between two or more levels and filtering the output to generate a continuously varying waveform.

[0003] Some RF amplifier systems utilize "discrete" supply modulation (or discrete "drain modulation"), in which the supply voltage is switched between a set of discrete voltage levels, possibly including additional filtering or modulation to shape the voltage transitions between levels. Systems of this type include "Class G" amplifiers, multi-level LINC (MLINC) power amplifiers, power amplifiers with asymmetric multi-level outphasing (AMO), multi-level backoff amplifiers (including "asymmetric multi-level backoff" amplifiers), and digitized polar transmitters, among others. Hybrid systems that utilize a combination of continuous and discrete supply modulation can also be implemented. SUMMARY

[0004] This document describes concepts, systems, circuits, devices, and techniques for use in and / or with power amplifier (PA) architectures that employ supply voltage modulation. These concepts, systems, circuits, devices, and techniques can provide both very rapid variations of the modulated power supply voltage (e.g., between multiple discrete levels) and the ability to slowly adjust the voltages of the discrete levels over a desired range. Such concepts, systems, circuits, devices, and techniques are used in a number of applications, including, but not limited to, PA architectures.

[0005] Using the described concepts, systems, circuits, devices, and techniques, it is possible to efficiently and compactly generate a set of m power supply voltages. Two of the m power supply voltages (e.g., V1 and V) mThe two independently controlled power supply voltages can be independently controlled. The other m-2 power supply voltages can be distributed in a prescribed relationship to the two independently controlled power supply voltages, such as being evenly spaced between and / or around them (e.g., with adjacent voltage levels separated by an approximate voltage ΔV). Thus, for example, the following m power supply voltages can be provided, where V1 and V2 m The independently controlled supply voltages are: Vk=V1+(k−1)⋅(Vm−V1) / (m−1)fu¨rk=1…m.

[0006] Such an arrangement is equivalent to enabling the independent specification or control of the following: (a) a minimal (V min ) and maximum (V max ) Voltage levels (where the distance between the voltage levels ΔV with respect to V) min and V maxand the total number of levels (m) is determined). (b) the minimum supply voltage level (V min ) and the inter-level voltage difference ΔV (where the maximum voltage level V max determined by ΔV and the total number of levels m). (c) the maximum supply voltage level (V max ) and the inter-level voltage difference ΔV (where the minimum voltage level V min determined by ΔV and the total number of levels m).

[0007] The concepts, systems, circuits, devices, and techniques described herein provide essentially all (or most) of the practical advantages available from supply modulation (e.g., in terms of PA efficiency), while avoiding the limitations associated with providing truly independent voltage level control or ratiometric levels. Thus, the concepts, systems, circuits, devices, and techniques described herein offer significant advantages in combinations of size, cost, efficiency, and performance compared to existing approaches.

[0008] Further advantages arise if only two regulated supply voltages need to be generated and it is possible to directly provide the power amplifier (PA) with one or more additional supply voltage levels, without the need for a separate supply generator element to generate these additional levels in cascaded form with a supply modulator to select between them. Combining the intermediate-level generation and supply modulation functions can reduce the number and size of required passive components (e.g., capacitors) as well as the number, area, and loss of semiconductor components (e.g., switches).

[0009] The object of the present invention is to provide a system and a method for modifying the frequency response of a circuit coupled to an output of a hybrid power supply generator / modulator and to a high-frequency (HF) amplifier, with improved characteristics.

[0010] This problem is solved by a system according to claim 1 and a method for changing a frequency response of a circuit coupled to an output of a hybrid power supply generator / modulator and to a high-frequency (HF) amplifier according to claim 21.

[0011] According to one aspect of the present disclosure, a system comprises: a hybrid supply generator / modulator comprising: an input for receiving an input voltage; an output for providing a modulated voltage; a multi-output control stage configured to provide two intermediate voltages with different voltage levels, wherein at least one of the two intermediate voltages is synthesized from the input voltage; and a multi-level converter configured to receive the two intermediate voltages and to generate the modulated voltage with a voltage level corresponding to one of the two intermediate voltage levels or at least one synthesized voltage level that differs from both intermediate voltage levels;and a circuit arrangement coupled to the output of the hybrid power supply generator / modulator and to a power supply connection of at least one high-frequency (HF) amplifier, wherein the circuit arrangement is configured to modify the modulated voltage supplied to the at least one HF amplifier.

[0012] In some embodiments, the circuit arrangement includes a disconnect switch configured to selectively supply a zero-voltage level to the at least one RF amplifier. In some embodiments, the circuit arrangement includes a disconnect switch configured to selectively isolate the output of the hybrid power supply generator / modulator from the at least one RF amplifier. In some embodiments, the disconnect switch is configured to be actuated in conjunction with the activation or deactivation of the RF amplifier.

[0013] In some embodiments, the circuit arrangement includes a pulse-shaping network (PSN) configured to filter the modulated voltage. In some embodiments, the PSN includes at least one passive element. In some embodiments, the at least one passive element is implemented as a discrete element. In some embodiments, the at least one passive element is implemented on an integrated circuit (IC) or module. In some embodiments, the at least one passive element results from parasitic resistance, parasitic inductance, or parasitic capacitance.In some embodiments, the PSN comprises: a first and second inductor connected in series between the output of the hybrid power supply generator / modulator and the power supply terminal of the at least one RF amplifier; a capacitor with a first terminal connected between the first and second inductors; and a third inductor connected between a second terminal of the capacitor and ground.

[0014] In some embodiments, the PSN comprises: a resistor having a first terminal connected to the output of the hybrid power supply generator / modulator; and a capacitor having a first terminal connected to a second terminal of the resistor and a second terminal connected to ground. In some embodiments, the PSN comprises: a capacitor having a first terminal connected to the output of the hybrid power supply generator / modulator and a second terminal connected to ground, and a resistor having a first terminal connected to the first terminal of the capacitor.

[0015] In some embodiments, the circuit arrangement further comprises a switching network. In some embodiments, the switching network is configured such that, in a first state, the switching network provides a first signal path with a first filter configuration between the output of the hybrid power supply generator / modulator and the power supply terminal of the at least one RF amplifier, and in a second state, the switching network provides a second signal path with a second, different filter configuration between the output of the hybrid power supply generator / modulator and the power supply terminal of the at least one RF amplifier.In some embodiments, the switching network has at least one passive element with a first terminal connected to the output of the hybrid power supply generator / modulator, wherein a switching network is configured such that in a first state the switching network connects a second terminal of the at least one passive element to ground and in a second state the switching network disconnects the second terminal of the at least one passive element from ground.

[0016] In some embodiments, the switching network is coupled to the hybrid power supply generator / modulator in a cascaded configuration. In some embodiments, the switching network is coupled via the PSN and is configured to selectively provide a signal path that bypasses the PSN. In some embodiments, the switching network is coupled in parallel with a passive element of the PSN and is configured to modify a transfer function of the PSN by selectively short-circuiting at least one passive element. In some embodiments, the switching network has a plurality of switches, and at least a first set of the plurality of switches is located on a first integrated circuit die / chip, and at least a second set of the plurality of switches is located on a second, different integrated circuit die / chip.

[0017] According to another aspect of the present disclosure, a system comprises: a hybrid supply generator / modulator comprising: an input for receiving an input voltage; an output for providing a modulated voltage; a multi-output control stage configured to provide two intermediate voltages with different voltage levels, wherein at least one of the two intermediate voltages is synthesized from the input voltage; and a multi-level converter configured to receive the two intermediate voltages and to generate the modulated voltage with a voltage level corresponding to one of the two intermediate voltage levels or at least one synthesized voltage level that differs from both intermediate voltage levels;wherein the output of the hybrid power supply generator / modulator is directly or indirectly coupled to a power supply input of at least one high-frequency (HF) amplifier.

[0018] According to another aspect of the present disclosure, a method for changing a frequency response of a circuit coupled to an output of a hybrid power supply generator / modulator and to a high-frequency (HF) amplifier is provided, wherein the method comprises: in a first state, configuring the circuit to provide a first signal path with a first filter configuration between the output of the hybrid power supply generator / modulator and the input of the HF amplifier; and in a second state, configuring the circuit to provide a second signal path with a second, different filter configuration between the output of the hybrid power supply generator / modulator and the input of the HF amplifier.

[0019] The manner and process of manufacturing and using the disclosed embodiments can be understood by referring to the figures in the accompanying drawings. It should be understood that the components and structures illustrated in the figures are not necessarily to scale, and the emphasis is instead placed on illustrating the principles of the concepts described herein. The same reference numerals denote corresponding parts throughout the different views. Furthermore, the embodiments illustrated in the figures are examples and not limiting.

[0020] Preferred embodiments of the present invention are explained in more detail below with reference to the accompanying drawings. These show: Fig. 1A a block diagram of an RF amplifier system that uses multiple supply levels capable of powering multiple PAs; Fig. 1B a block diagram of an RF amplifier system utilizing multiple supply levels and implemented as a single-inductor multi-output boost converter such as a supply generator, a parallel supply modulator and an LC filter; Fig. 2A a schematic diagram of an illustrative switching network provided as a series supply modulator; Fig. 2B a schematic diagram of an illustrative switching network provided as a parallel supply modulator; Fig. 3 a block diagram of a power supply modulator with a cascaded off switch; Fig. 4 a schematic diagram of an illustrative implementation of the series modulator of Fig. 2A; Fig.5 a block diagram of an RF amplifier system that uses a hybrid power supply generator / modulator to provide multiple power supply levels; Fig. 5A a block diagram of an RF amplifier system with a hybrid power supply generator / modulator and a filter network; Fig. 5B a block diagram of an RF amplifier system with a hybrid power supply generator / modulator and a PA off switch; Fig. 6 a block diagram illustrating an architecture of a hybrid power generator / power modulator suitable for use in an RF amplifier system; Fig. 6A a block diagram of another hybrid power generator / power modulator architecture where one of the intermediate voltages corresponds to the input voltage; Fig.7A-E schematic diagrams illustrating implementations of a multi-output control stage suitable for use in a hybrid power generator / power modulator; Fig. 8 a schematic diagram of a differential multi-level converter suitable for use in a hybrid power generator / power modulator, wherein the converter is able to provide three (3) effective output levels using four (4) switching states; Fig. 9 and Fig. 9A Diagrams of state machines used to control a switch state in the differential multi-level converter of Fig. 8 can be used; Fig. 10A-10C a series of plots illustrating voltage patterns and state transitions that can be described using the state machine of Fig. 9 are associated; Fig.11 a schematic diagram of a differential multi-level converter suitable for use in a hybrid power generator / power modulator, wherein the converter is able to provide four (4) effective output levels using eight (8) switching states; Fig. 12A and Fig. 12B Diagrams of state machines used to control a switch state in the differential multi-level converter of Fig. 11 can be used; Fig. 13A and Fig. 13B Schematic diagrams of differential multi-level converters suitable for use in a hybrid power generator / power modulator, wherein the converters have a circuit arrangement for controlling the charge on a flying capacitor; Fig.14 a schematic diagram of an RF power amplifier system comprising a hybrid supply generator / supply modulator with multiple outputs to supply multiple power amplifiers; Fig. 14A a schematic diagram of an RF power amplifier system comprising several differential multi-level converters magnetically coupled to supply a single power amplifier; and Fig. 15 a schematic diagram of a switching network comprising one or more pulse shaping networks (PSNs) coupled to one or more hybrid supply generators / supply modulators.

[0021] With reference to Fig.Figure 1A shows an overview of an illustrative RF power amplifier system architecture that utilizes supply modulation, wherein a supply modulator switches between several voltages generated by a separate supply generator (elements and aspects of signal processing and control for such a system are omitted for clarity).

[0022] The illustrative system 10 comprises a multi-output power supply generator subsystem (or, more simply, a "power supply generator") 12, which provides multiple power supply voltages V1-V m can synthesize from a single input source 11. In some examples, the supply generator 12 can provide one or more power supply voltages V1-V m regulate. The supply generator 12 provides one or more of the voltages V1-V mat inputs of one or more power supply modulator subsystems (or, more simply, "power supply modulators") 14a-14N of a power supply modulator system 14. The power supply modulators 14a-14N can switch (and ideally switch quickly) between the different power supply voltages provided by the power supply generator 12 in order to thus modulate supply voltages V supply#1 -V supply#Nto provide at one of its outputs. Switches can be modulated quickly enough to provide a power supply voltage to the PA so that the PA can provide the required RF output envelope while maintaining high efficiency, according to techniques known in the art as discrete drain modulation, extended envelope tracking (ETA), discrete envelope tracking, and digital envelope tracking (digital ET). Such techniques are described, for example, in one or more of US patents 8,829,993; 9,160,287; 9,166,536; 9,172,336; 9,209,758; 9,755,672. The supply voltage inputs can be coupled to supply terminals of one or more PAs 18a-18N. In some examples, the PAs 18a-18N can be provided as RF power amplifiers. In some examples, the supply generator 12 can supply the same or different voltages to the supply modulators 14.In some examples, a different number of voltages can be coupled between the supply generator 12 and the supply modulators 14.

[0023] In some examples, some or all of the supply voltages can be coupled to the PA supply terminals via optional filter or voltage regulation stages 16a-16N. The filter / regulation stages 16a-16N can include filter networks, including passive filters and / or active filters, and / or additional means for regulating the voltage (e.g., including low-dropout regulators, LDOs) to the PA, V supply , from the modulated voltage, V mod .

[0024] In some examples, one, some, or all of the supply modulator subsystems 14a-14N may include one or more switches to couple one or more voltages provided by the supply generator 12 to PA supply terminals 19a-19N. A variety of different switching circuits (i.e., switches with a wide variety of switch configurations or switch topologies) can be used to implement supply modulator subsystems 14. For example, a supply modulator subsystem may include a variety of series-coupled switches configured to provide a "series" modulator. Alternatively, a supply modulator subsystem may include a variety of parallel-coupled switches configured to provide a "parallel" modulator.As another alternative, a power supply modulator subsystem can have one or more serially coupled switches and one or more parallel coupled switches.

[0025] It is understood that the manner in which the voltages are synthesized by the supply generator affects the required switch ratings in the one or more supply modulators 14a-14N. This can be an important consideration, as the required modulator switch ratings can affect (and in some cases strongly affect) the switching speed (and achievable modulation rate) and modulator efficiency, both of which are significant system factors. Regardless of the modulator switch topology used, if there are m supply levels, ordered in increasing voltage V1,...,V m (i.e. V1 < V2 <... < V m), the plurality (or chain) of switches that connect the j-th supply voltage V j and the output V mod are coupled, ideally dimensioned to at least a negative voltage with magnitude (V) m - V j ) and a positive voltage, which is either (V j - V1) or V j The need to block the power supply depends on whether the modulator draws a minimum voltage V1, or ideally should be able to supply zero volts directly to the PA. In some examples with designs of the latter type, if the power supply provided to the PA needs to be "cut off" (discharged to a zero-volt power supply), a separate low-frequency "off" or "disconnect" switch can advantageously be placed in series with the output of a supply modulator capable of modulator output voltages V1,...,V mto obtain. Such a shutdown switch can meet the requirements for the voltage blocking of the modulator switch chain of V j on (V j - V1); this can be advantageous for the modulator design.

[0026] In some examples, an RF power amplifier system, such as System 10, may incorporate a series modulator in a form suitable for integrated circuit (IC) fabrication and use with ratiometric supply voltages (e.g., V2 = 2.1V, V3 = 3.1V, V4 = 4.1V). Such a design illustrates the influence of supply levels on the required nominal voltage of individual modulator devices. By correctly selecting the level voltages, optimal use of integrated CMOS processes can be achieved using both core devices and extended voltage devices to obtain the required voltage blocking characteristics of the modulator switching chains. Furthermore, such a circuit demonstrates the use of the generated levels for gate drive of the devices. This type of drive approach enables high efficiency and switching speed.To control the device gates between adjacent voltage levels (e.g., between V. j and V j-1 To take advantage of this, however, level voltages for this design should be maintained with sufficient clearance; otherwise, more sophisticated gate drive designs may be required, which could limit the achievable switching performance.

[0027] Fig. 1B shows an implementation of the in Fig. Figure 1A illustrates an architecture that may be suitable, for example, for discrete supply modulation.

[0028] As shown, a system 30 can include a multi-output power supply generator 34, a power supply modulator 36, and optionally a filter 38 and a PA 40 with an RF input 40a, an RF output 40b, and a power supply connection 40c. In this example of Fig.In 1B, the supply generator 34 is provided as a single-inductance, three-output boost converter, having an inductor L with a first terminal coupled to a voltage supply 32 and a second terminal coupled to a node 34a. One or more switches (here three switches S1, S2, S3) have a first terminal coupled to node 34a and a second terminal coupled to at least one voltage node produced via a capacitor stack C1, C2, C3, C4 (e.g., a plurality of capacitors C1, C2, C3, C4 coupled in series between a first voltage node and ground to produce a plurality of voltage nodes V1–V3). A fourth switch S0 has a first terminal coupled to node 34a and a second terminal coupled to ground. In the example of Fig.1A is the second connection of the switches S1, S2, S3 coupled to their respective voltage nodes V1, V2, V3.

[0029] Furthermore, the example of Fig. 1B the power supply modulator a multitude of switches S m1 -S m3 on, with a first connection of each switch S m1 -S m3 is coupled with a corresponding voltage connection V1 - V3 and a second connection of each switch S m1 -S m3 is coupled to node 36a.

[0030] In the example of Fig.In 1B, node 36a is coupled to the supply terminal 40c of PA 40 via filter 38. In some examples, filter 38 may be implemented as an LC filter comprising an inductor Lf, a resistor Rf, and capacitors C4 and C5. In other examples, node 36a may be coupled to the supply terminal 40c of PA 40 via a different circuit arrangement (i.e., a different circuit arrangement than the filter circuit arrangement). In still other examples, node 36a may be directly coupled to the supply terminal 40c of PA 40.

[0031] The in Fig. 1A, Fig.1B The illustrated systems comprise two separate subsystems: (a) a supply generator that can synthesize multiple power supply voltages from a single input source and may regulate one or more of these power supply voltages, and (b) one or more supply modulators that can each rapidly switch between the power supply voltages provided by the supply generator to provide a modulated supply voltage to a PA.

[0032] According to the concepts described herein, the inventors have recognized that the best way to implement (or "realize") these two subsystems may depend on the power level, voltage level, and application space of the RF amplifier system. The inventors have also recognized that for many mobile applications (e.g., mobile phones, smartphones, personal devices, and the like), it may be desirable to monolithically integrate electronic elements of both the power generator and the power modulator on a single semiconductor die / chip (e.g., in a CMOS or BCD process). The inventors have further recognized that in some cases, it may be desirable to integrate electronics for the power generator, the one or more power modulators, and power amplifiers on a single die.In other cases (especially at high power), it may be desirable to implement the subsystems with discrete components connected on one or more printed circuit boards.

[0033] A variety of different switching circuits can be used to implement / realize a power supply modulator subsystem. Two illustrative networks are shown in Fig. 2A, Fig. 2B shown. Fig. 2A illustrates a series modulator 200, the switches S1-S4, S 34 and S 234 exhibits which are connected as shown. Fig. Figure 2B illustrates a parallel modulator 240, which has switches S1-S4 connected as shown. Additionally, filter networks can be used, including passive and / or active filters, and / or additional means for regulating the voltage (e.g., including low-dropout regulators, LDOs) to the PA, V. supply , from the modulated voltage, Vmod , as in Fig. 1 illustrates.

[0034] Now with reference to Fig. 3 In some embodiments, the modulated power supply provided to the PA (e.g. V) must be supply ), possibly “cut off” (discharged to a level of zero volts). For example, this can be used to allow the reduction of modulator switch rated voltages in cases where a zero output must be provided to a power amplifier. In such cases, a circuit 300 may include a separate low-frequency cut-off switch 302 (or “disconnect switch”) connected in series between an output of a supply modulator 304 (which is capable of modulator output voltages V1,...,V) m to obtain) and is coupled to a PA 306.

[0035] Fig. Figure 4 shows an implementation of the series modulator of Fig.2A in a form suitable for the fabrication of integrated circuits and for use with ratiometric supply voltages (e.g., V2 = 2.1V, V3 = 3.1V, V4 = 4.1V). An illustrative circuit 400 includes switches S1, S2, and S3, implemented as NMOS transistors, and switches S4, S 34 and S 234 , which are implemented as PMOS transistors. Circuit 400 also includes CMOS gate drivers that are supplied with different levels of power.

[0036] Circuit 400 illustrates the influence of supply levels on the required nominal voltage of individual modulator devices. By correctly selecting the level voltages, optimal use of integrated CMOS processes can be achieved using both core devices and extended voltage devices to obtain the required voltage blocking characteristics of the modulator switching chains. Furthermore, Circuit 400 demonstrates the use of the generated levels for gate driving of the devices (e.g., transistors). This type of drive approach can enable high efficiency and switching speed.

[0037] To control the device gates between adjacent voltage levels (e.g., between V j and V j-1To fully exploit the potential of this design, voltage levels with sufficient spacing must be maintained. Otherwise, more sophisticated gate drive designs may be required to limit the achievable switching performance. The devices, circuits, and techniques described herein enable the maintenance of voltage levels suitable for achieving high-performance integrated circuit-based modulators and gate drives by maintaining desired voltage relationships between the levels.

[0038] Power supply generators can be implemented using a variety of methods. For example, power supply generators can be implemented using multiple separate converters, multi-output magnetic converters, multi-output switching capacitor converters, and hybrid magnetic / switching capacitor converters that provide a ratiometric set of output voltages. Another approach is to implement a multi-output power supply generator that produces two independently controllable DC voltages (e.g., with a magnetic conversion stage) and further uses a differential capacitive energy transfer stage to realize one or more additional DC supply voltages that are ratiometrically distributed between or around the two independently controllable voltages.Each of these approaches has limitations (which some may consider significant limitations) regarding the achievable size, cost, efficiency and / or performance (e.g., modulation bandwidth) of supply-modulated RF amplifier systems.

[0039] Using separate power converters to generate the multiple supply voltages provides a flexible solution, allowing each output voltage to be independently regulated to desired values, regardless of input voltage variations, and enabling continuous adjustment of the output voltages over time (e.g., to provide adaptive biasing of the power amplifier). Unfortunately, this solution is inherently large and expensive due to the large number of physically large power supply components (e.g., magnetic components) required.

[0040] Single-inductor multi-output converters (sometimes called "SIMO" converters) allow multiple output voltages to be regulated independently while requiring only a single magnetic component, somewhat mitigating the size challenge of multiple power converters. However, because SIMO designs inherently rely on temporal sharing of the inductor to power the multiple outputs, performance and efficiency can degrade, and control complexity can increase with a growing number of outputs. This characteristic can limit the effectiveness of this approach in multi-level supply modulator systems, which typically utilize between three and seven supply levels to achieve high performance (with even more levels potentially desirable in some cases).

[0041] Some types of converters, such as conventional multi-output magnetic converters (e.g., multi-output flyback converters), multi-output switching capacitor converters, and hybrid magnetic / switching capacitor converters, produce multiple ratiometrically related output voltages while reducing the number of magnetic components required compared to using multiple independent power converters. Traditional multi-output magnetic converters typically use transformers with scaled turns ratios to generate multiple (ideally) ratiometrically scaled output voltages. These designs can only regulate a single output, with the ratiometric relationships of the other outputs being approximately maintained by the transformer turns ratios (unless additional "post-regulation" is provided to the other outputs, such as through the use of added linear regulators).The use of transformers tends to reduce the achievable efficiency in these designs (often to unacceptable levels), and such designs can suffer from significant cross-regulation between the outputs in practice (i.e., one output voltage varies depending on the load at another output). This leads to undesirable performance in the RF amplifier systems unless post-regulation is used, which can further degrade performance.

[0042] Multi-output switching capacitor converter circuits can generate several ratiometrically related output voltages while achieving very high efficiency and small size, with the rational (ideal) ratios between output voltages determined by the circuit topology and / or switching pattern. However, with this type of circuit, the output voltages are all scaled versions of the input voltage, which provides no means of continuously regulating the output voltages independently of variations in the input voltage; this is a significant disadvantage in many systems.

[0043] Some limitations of these previous approaches to multi-output supply generation can be addressed via hybrid magnetic / switching capacitor circuits with ratiometrically scaled outputs. In these designs, a magnetic control stage independently regulates a single output voltage (independent of the system input voltage), with additional ratiometrically related output voltages synthesized and enforced by the action of a switching capacitor voltage balancing stage. For example, in an m-output supply generator, the magnetic stage can regulate an input voltage V X take and a single output voltage V Y regulate, whereby the switching capacitor voltage action (ideally) voltages k1·V Y , k2·V Y ,...,k m-1· V Y synthesized, where the constants k1,..,k m-1rational numbers are determined by the circuit topology and / or the circuit pattern.

[0044] Advantages of this approach include relatively high efficiency and small size requirements for synthesizing multiple related output voltages and relative simplicity of control.

[0045] Despite the advantages of the design approaches mentioned above, all designs that result in ratiometric supply generator voltage outputs have limitations (which some may consider significant) for PA systems that utilize multi-level supply modulation.

[0046] One limitation of ratiometric outputs relates to the usable supply voltage ranges for available PAs. Some PAs can function well with wide supply voltage ranges of up to 4:1 or even higher (e.g., well over a power supply voltage range with a maximum voltage of V). maxdown to a minimum voltage equal to or less than V min = V max / 4 function). Many other PAs—including those typically used in applications such as WiFi, mobile handsets, and MIMO transmitters for LTE and 5G applications—can only operate over much narrower supply voltage ranges (e.g., 3:1 or even less). With ratiometric supply voltages, if the maximum generated voltage is reduced (e.g., for conditions of reduced average PA output power), then the synthesized ratiometric voltages are all reduced proportionally. This often means that one or more of the lowest synthesized voltages will become unusable for supply modulation under such conditions, as they fall below the allowable minimum PA power supply voltage. This, in turn, reduces the achievable PA efficiency improvement that can be provided via supply modulation under these conditions.In many applications, it would be desirable if the power supply voltages were not maintained as a fixed set of ratios, so that all (or almost all) of the synthesized supply voltage levels in reduced-power operation would remain above the allowable minimum voltage for the PA.

[0047] Another limitation of ratiometric outputs relates to how the spacing between voltages varies as the largest synthesized supply voltage is reduced. In a supply generator with a ratiometric output, two adjacent voltages can be represented as k j ·V Y and k j-1· V Y expressed where k is a scaling value, j is an integer index, V is a voltage, and Y is an index corresponding to the number of voltage levels, V Y the Y-th voltage level, where the value of V YThe voltage can be scaled up or down while the average transmit power of the PA is adjusted. The difference between voltage levels can thus be expressed as (k j - k j-1 )·V Y can be expressed as being proportional to V Y scaled up and down. As above in conjunction with Fig.As described in section 4, this can be problematic for driving integrated modulator switches, especially if the gate drive voltages are derived from intermediate-level voltages (voltage differences between levels). This can lead to increased gate drive complexity in an integrated modulator and may limit the achievable switching performance of the modulator. In many applications, it would be desirable for the power supply voltages not to be maintained as a fixed set of ratios, so that the spacing between adjacent levels can be controlled independently of the maximum synthesized supply voltage. A hybrid power supply generator / power supply modulator

[0048] For PA architectures that use supply modulation, it may be desirable to provide a system that can offer both very fast variations of the modulated power supply voltage (e.g., between several discrete levels) while also providing the ability to slowly adjust the voltages of the discrete levels over a desired range.

[0049] In particular, and as discussed previously, it would be useful to be able to efficiently and compactly generate a set of m discrete levels for supply to a PA, where two of the m voltage levels are independently controllable and the other m-2 voltage levels are distributed in a prescribed relationship to the two independently controlled levels.

[0050] Although not quite as flexible as truly independent control of all voltages, one would gain most of the practical benefits available from supply modulation (e.g., in terms of PA efficiency) while avoiding the limitations described above that are associated with providing truly independent voltage level control or ratiometric levels. Such a design would offer significant advantages in combinations of size, cost, efficiency, and performance compared to existing approaches.

[0051] Furthermore, it would be advantageous if only two regulated supply voltages needed to be generated and if it were possible to directly provide one or more additional supply voltage levels to the PA without the need for a separate supply generator element to generate these additional levels in cascaded form with a supply modulator to select between them. Combining the functions of intermediate level generation and supply modulation can reduce the number and size of required passive components (e.g., capacitors) as well as the number, required area, and loss of semiconductor elements (e.g., switches).

[0052] Now with reference to Fig. Figure 5 shows a general system architecture of an RF amplifier system (or “RF module”) 500 which includes a hybrid supply generator / modulator 502 configured to provide a modulated supply voltage V smto generate and supply these to a PA 504. The hybrid supply generator / modulator 502 can have a hybrid magnetic / switching capacitor power converter that can synthesize three or more discrete levels (which can be related discrete voltage levels) at its output (i.e., the voltage V). SM (can correspond to one of three or more discrete voltage levels at any given time). As described above, the voltage V can SM based on a pair of two independently controlled DC supply voltages, which he derives from an input voltage V IN synthesized. In some implementations, one of the two DC supply voltages can be directly equal to the voltage V. IN The second supply voltage can be controlled independently.

[0053] It should be noted that "hybrid magnetic / switching capacitor converters" are a broad class of converters. Disclosed systems described as "hybrid power generators / modulators" can be considered a specific subtype of this broad class. It should also be noted that in the term "hybrid magnetic / switching capacitor converter," the word "hybrid" refers to the hybridization of a magnetic converter and a switching capacitor converter together. In contrast, in the term "hybrid power generator / modulator," the word "hybrid" refers to the hybridization of the power generator and the power modulator.

[0054] As shown in the figure, the system 500 can optionally include an additional circuit arrangement 506 coupled between an output of the hybrid supply generator / modulator 502 and the PA 504. The additional circuit arrangement 506 can, for example, include a disconnect switch, one or more filter networks including passive and / or active filters, low-dropout regulators (LDOs), and / or additional means for regulating or controlling the PA supply voltage, V. supply , from the modulated voltage, V sm In some embodiments, the additional circuit arrangement 506 may have more than two terminals. For example, the additional circuit arrangement 506 may optionally be connected to ground, as shown.

[0055] The system 500 may also include one or more controllers 550 to operate switches and / or other devices of the hybrid supply generator / modulator 502 using control techniques described in detail below. For example, the one or more controllers 550 may operate one or more state machines described herein for modulating V smimplement, execute, or otherwise use. In some embodiments, the hybrid power generator / modulator 502 may include a multi-output control stage (or "first stage") and a differential multi-level converter (or "second stage"), and the one or more controllers 550 may include at least one controller for controlling the switching state of the multi-output control stage and at least one controller for controlling the switching state of the differential multi-level converter. These may be the same controller or different controllers. In some embodiments, at least one of the one or more controllers 550 may be configured to issue commands for selecting / controlling a voltage level of the modulated voltage, V sm , to receive.

[0056] In some embodiments, the one or more controllers 550 can include a digital controller configured to provide one or more control signals to one or more hybrid power generators / modulators. In response, the hybrid power generators / modulators can provide variable supply voltages to one or more RF amplifiers. That is, the control signals can be used to select discrete supply voltages for the amplifiers from several independent supply voltages generated by the hybrid power generators / modulators. More specifically, the control signals can be used to operate one or more switches within the hybrid power generator / modulator, directly or indirectly (e.g., switches within a multi-output control stage, as in Fig.7A-E illustrates, and / or switches within the differential multi-level converter, as in Fig. 8 or Fig. 11 illustrated).

[0057] Fig. Figure 5A shows an RF amplifier system 520 similar to that of Fig. 5, wherein the additional circuit arrangement 506 is implemented as a filter network 522. The illustrative filter network 522 comprises a first and second inductor 524a, 524b, which is connected in series between an output of the hybrid power supply generator / modulator 502 and an input of the PA 504. The filter network 522 further comprises a capacitor 526 with one terminal connected between the first and second inductors 524a, 524b, and a third inductor 524c, which is connected between the other terminal of the capacitor 526 and ground. The filter network 522 may sometimes be referred to as a pulse-shaping network (PSN).

[0058] The 522 filter network of Fig. 5 is merely an example, and other types of filter networks can be used. For example, a reconfigurable filter can be used, where one or more switches are configured to change the filter parameters (or characteristics) of the filter network.

[0059] Fig. Figure 5B shows an RF amplifier system 540 similar to that of Fig. 5, wherein the additional circuit arrangement 506 is implemented as a PA switch-off switch 542 (or "disconnect" switch). As discussed previously, in some cases the modulated power supply provided to the PA (e.g., V) must be supply), possibly “cut off” (discharged to a level of zero volts). For example, this can be used to allow the reduction of modulator switch ratings in cases where a zero output must be provided to a power amplifier. In such cases, a low-frequency cutoff switch 542 can be connected in series between an output of the hybrid power supply generator / modulator 502 and an input of the power amplifier 504, as shown. The cutoff switch 542 can be used to selectively isolate the output of the multi-output control stage from the RF amplifier.

[0060] In some embodiments, the RF amplifier system 540 can be part of a transceiver system that includes a modem, among other components, external to the RF amplifier system. The modem or one of the other components can directly or indirectly generate control signals to cause the off switch 542 to be actuated in conjunction with the activation / deactivation of the RF amplifier.

[0061] Fig. Figure 6 shows an illustrative architecture of a hybrid power generator / power modulator 600 according to some embodiments. The illustrative hybrid power generator / power modulator 600 can be integrated within the system 500 of Fig. 5 can be provided. For example, the Modulator 600 can be used by Fig. 6 completely or partially to the modulator 502 of Fig. 5 correspond.

[0062] The illustrative hybrid power supply generator / power supply modulator 600 comprises a multi-output control stage 602 and a differentially coupled multi-level converter 604.

[0063] The multi-output control stage 602 is configured to provide an input voltage V IN to receive, couple or otherwise accept, and synthesizes two intermediate voltages V A and V B In some embodiments, the intermediate voltages V A and V B be independently controllable. In some embodiments, the multi-output control stage 602 can be provided as a magnetic or hybrid magnetic / switching capacitor control stage.

[0064] The differentially coupled multi-level converter 604 has an input that differentially switches between the intermediate voltages V A and V Bis connected or otherwise coupled, and is configured to provide three or more instantaneous output voltage levels V SM to synthesize from it. In some embodiments, the differentially coupled multi-level converter 604 can use switching capacitors to synthesize instantaneous levels while allowing desired voltage / charge balancing across the capacitors. The differentially coupled multi-level converter 604 can be based on a differentially coupled multi-level converter with a flying capacitor (FCML) or on another capacitor-based energy transfer topology, including those in which the charge transfer requirements prevent one or more of the synthesizable levels from being continuously provided without switching the differentially coupled multi-level converter.

[0065] The multi-output control stage 602 and the differentially coupled multi-level converter 604 can include one or more devices (e.g., switches) configured to be controlled according to one or more control techniques described in detail below.

[0066] The multi-output control stage 602 and / or the differential multi-level converter 604 can be controlled by one or more controllers, such as the one or more controllers 550 from Fig. 5.

[0067] Fig. Figure 6A shows a different architecture of a hybrid supply generator / supply modulator 620, where one of the intermediate voltages directly corresponds to the input voltage. As shown, the input voltage V can be in provided as an input to a control stage 622, and is also connected to supply an intermediate voltage V to the differential multi-level converter 604. Bto provide. The other intermediate voltage V A The signal provided to the differential multi-level converter 604 can be generated by the control stage 622. In other words, it can be said that V B bypasses the control stage 622. The control stage 622 may resemble any of the multi-output control stage implementations or sections thereof described herein, but may be adapted to have only one signal output and an associated circuit arrangement. Although Fig. 6A shows an example where V B V in If this corresponds, the same general approach can be used in other examples so that V A V in corresponds.

[0068] The multi-output control stage 602 and / or the differential multi-level converter 604 can be controlled by one or more controllers, such as the one or more controllers 550 from Fig. 5.

[0069] Now with reference to Fig. Figures 7A-7E show implementations of a multi-output control stage, which is used within the hybrid power supply generator / power supply modulator 600. Fig. 6 according to the embodiments described in this disclosure. For example, the implementations of the multiple output control stage of Fig. 7A-7E completely or partially of the multiple output control stage 602 of Fig. 6 correspond. Likewise, the implementations of Fig. 7A-7E within the 500 system of Fig. 5 can be used.

[0070] Fig.Figure 7A illustrates a multi-output control stage 700, which features a single-input multi-output (SIMO) converter with two outputs based on a 4-switch buck-boost converter, but with an additional output and switch. The multi-output control stage 700 has two outputs 702a and 702b (e.g., terminals) to which each of the synthesized voltages V can be applied. A and V B can be provided. A first switch S1 is connected between a voltage source, V IN , and an inductive element L. A second switch S2 is connected at one end between S1 and L and at the other end to ground (or more generally, a reference voltage). A third switch S3 is connected between L and an output 702b (the V B(provides) connected. A fourth switch S4 is connected at one end between L and S3 and at the other end to ground. A fifth switch S5 is connected at one end between S4, L and S3 and at the other end to another output 702a (the V A (provides) connected. Independent decoupling capacitors C A and C B They can be connected to outputs 702a and 702b respectively, as shown.

[0071] As shown, additional optional switches S opt,A and S opt,B It must be included to allow direct energy transfer from output 702a to output 702b or vice versa. As also shown, a differential decoupling capacitor C can optionally be included. D enclosed and connected via the two outputs 702a, 702b.

[0072] Fig.Figure 7B illustrates a multi-output control stage 720, which has two outputs 722a and 722b (e.g. terminals) to which each of the voltages V A and V B can be provided. The voltages V A and V B can be synthesized using separate 4-switch buck-boost converters 724a and 724b, respectively. As shown, each buck-boost converter 724a and 724b has four switches S1-S4 and an inductive element (L). A or L B ) on. Optional output decoupling capacitors C A , C B and / or C D They can also be provided as shown.

[0073] Although the exemplary embodiment of Fig.7B comprises two buck-boost converters; depending on the voltage range requirements, two different types of converters can be used. For example, in other embodiments, one converter can be implemented as a 4-switch buck-boost, while the other converter can be either a simple buck converter (e.g., to supply the smaller of the two voltages V1). A and V B ) or as a simple boost converter (to supply the larger of the two voltages V) A and V B ) will be implemented.

[0074] Fig. Figure 7C illustrates a multi-output control stage 740, which includes a 4-switch buck-boost converter to provide a first output voltage V. B to synthesize at an output 742b, and includes a buck converter that is driven by the voltage V B is supplied with power to provide a second independently regulated voltage V Ato synthesize at another output 742a, where V A less than or equal to V B More specifically, the buck-boost converter has switches S1-S4 and an inductive element L. B on, whereas the buck converter switches S5, S6 and an inductive element L A Features optional output decoupling capacitors C A , C B and / or C D They can also be provided as shown.

[0075] It is understood that other similar conversion approaches can be used to generate two independently controllable voltages V. A and V B to synthesize. For example, the structure of Fig. 7C can be adjusted so that the initial buck-boost stage is omitted and the input (V in ) directly with capacitor C B is connected so that the voltage V B is equal to the input voltage and the voltage V AIt is independently controllable. Identical structures can be created in such a way that the voltage V A is equal to the input voltage and the voltage V B independently controllable (e.g. from the voltage V) A (derived with a boost converter).

[0076] Fig. Figure 7D illustrates a multi-output control stage 760, where V B directly the input voltage V in corresponds and V A from V in generated using a buck converter, so that V A < V B The buck converter can, for example, have an input capacitor C. in , comprising a pair of switches S1, S1 and an inductor L, connected as shown. The output decoupling capacitors C A , C B and / or C D They may be provided optionally.

[0077] Fig. Figure 7E illustrates a multiple output control stage 780, where V Adirectly the input voltage V in corresponds and V B from V in generated using a boost converter, so that V A < V B The boost converter can, for example, consist of a pair of switches S1, S1, an inductor L, and at least one of the capacitors C. B or C D include components connected as shown. The decoupling capacitors C A and / or C IN They may be provided optionally.

[0078] It should be noted that, whereas the hybrid power generator / power modulator 620 from Fig. 6A provides the input voltage as one of the intermediate voltages by bypassing the control stage 622; the same result can be achieved without surrounding the control stage, as for example by using the control stage implementation of Fig. 7D or Fig. 7E.

[0079] The various switches that are in Fig.7A-E, illustrated, can be controlled by one or more controllers, such as the one or more controllers 550 of Fig. 5. For example, one or more controllers can generate control signals that are coupled to selectively turn individual switches on or off.

[0080] Fig. Figure 8 shows the structure of an illustrative differential multi-level converter 800, which may be the same as or similar to the differential multi-level converter used in conjunction with Fig. Section 6 is described. The differential multi-level converter is operated with two voltages V. A and V B operated, which are received at respective inputs 802a and 802b (e.g., terminals). The two voltages V A , V Bcan be synthesized, for example, by a multiple-output control stage, which may be the same as or similar to one of the multiple-output control stages described above in conjunction with Fig. 7A-7E are described. In some embodiments, the voltage V B so that they are greater than the voltage V A is. In some embodiments, V B or V A an input voltage V in are equivalent to.

[0081] The differential multi-level converter 800 includes four switches S A , S B , S A' , S B', the connection between the two inputs 802a, 802b and a floating capacitor C f are connected in series, with one end between S A and S B and at the other end between S A' and S B' is connected. An output 804 (e.g., a terminal) can be connected between S B and S B'be connected as shown to produce a modulated output voltage V sm to provide a ground reference, for example. The 800 converter can optionally include a differential decoupling capacitor C. D include (not shown) the connection between inputs 802a and 802b.

[0082] The structure of the 800 converter is partly similar to that of a flying capacitor multi-level converter (FCML), with the differences that: (1) the output 804 is taken at discrete levels instead of being filtered to provide a continuously variable output; and (2) the DC input 802a, 802b is differentially switched between V A and V B is taken instead of an input source and shared. The ground-referenced output voltage V SM can therefore assume discrete values ​​that lie between voltage values ​​V A and V B are distributed, with energy from and between VA and V B and is transferred to output 804.

[0083] The switches S A , S B , S A' , S B' They can be implemented in a variety of ways, with some or all of the switches exhibiting unidirectional blocking capability. For example, one or all of the switches can be configured to block less than or equal to the maximum difference between V. A and V B , but block more than the maximum difference between them.

[0084] In a CMOS or BCD process, one or more of the switches S A , S B , S A' , S B' They can be implemented solely as NMOS devices, a combination of NMOS and PMOS devices, or solely as PMOS devices. In some implementations, it is possible to select the switches S. A and S A' as a CMOS pair and the switches S B and SB , to be implemented as a CMOS pair. The selection of a specific implementation can be based on the size / performance of the switches and / or the ease of gate control of the switches, with the gate drive levels optionally derived from V A and V B can be derived.

[0085] In some embodiments, the differential multi-level converter can be Fig. 8 can be operated according to four (4) different switch states, which control the output voltage V SM provide the controller specified in Table 1. That is, a controller (e.g., the one or more 550 controllers from Fig. 5) can be connected and configured to control individual switches S A , S B , S A' , S B' to switch on or off in order to achieve a given state as shown in Table 1. TABLE 1: Condition Switches turned on V SM V SM = level V cf 1 S A , S B V B V B L3 Unchanged 2 S A , S B' V B - V cf (V A + V B ) / 2 L2 Increasingly (under load) 3 S B , S A' V A + V cf (V A + V B ) / 2 L2 Decreasing (under load) 4 S B', S A' V A V A L1 Unchanged

[0086] Table 1 shows the behavior of the differential multi-level converter of Fig. 8 for different switch states. In the table, V denotes cf the voltage across the flying capacitor C f State one (1) conducts V B an V SM over, while state four (4) V A an V SM transitions. State two (2) leads to V SM = V B - V cf , while state (3) is V SM = V A + V cf leads. It is understood here that, with suitable switching control of the converter (i.e., suitable selections between states two (2) and three (3)), V cf near (V B - V A ) / 2 is maintained, resulting in both states two (2) and three (3) having an output voltage V SM = (V A + V B ) / 2, as shown in Table 1.

[0087] Controlled using the switch states from Table 1, the converter 800 allows three effective levels (i.e., approximate level values) to be synthesized at its output 804: V A , V B and (V A + V B ) / 2. The 800 converter can therefore be used to provide similar functionality to a 3-level supply modulator, which, for example, sets level L1 = V A , L3 = V B and L2 = (V A + V B ) / 2 generated.

[0088] One advantage of the design of Fig. 8 consists in combining the differential capacitive energy transfer supply function with the supply modulation function, reducing device area and component size compared to using a separate differential supply generator and supply modulator.

[0089] In some embodiments, V cf(the voltage across the flying capacitor C) f ) near (V B - V A ) / 2 can be maintained by selectively choosing between states two (2) and three (3) of Table 1 when attempting to maintain an output voltage level close to (V A + V B ) / 2 to synthesize. Under a positive load current (e.g., derived from a PA), V cf over time it will increase in state two (2) and will decrease over time in state three (3).

[0090] In some embodiments, when selecting which of the states two (2) or three (3) is used to supply a voltage close to (V) A + V B ) / 2 is to be used at the modulator output (i.e., synthesizing L2), state two (2) can be selected if V cf too far below (V B - V A ) / 2 falls, and state three (3) can be selected if V cf too far beyond (V B - VA ) / 2 increases, where "too far" can be defined using one or more threshold values, as described below.

[0091] In some embodiments, switching between states two (2) and three (3) can be controlled hysterically and / or with clocked switching transitions. For example, such switching can occur immediately when the voltage V cf a permissible voltage deviation ΔV above or below (V B - V A ) / 2 exceeds, or with clocked transitions, such as with a discrete supply modulation clock rate. In some embodiments, the permissible value of the voltage deviation ΔV can be hard-coded in the controller, can be a programmable value of the controller, or can be transmitted via an external signal.

[0092] It is understood that the circuit arrangement can be provided to regulate the voltage V cfabout the flying capacitor C f to measure / detect. For example, a differential sensor (not shown) can be provided, with each of its two inputs connected to a differential input of C. f is connected and its output is connected to the controller that controls the switching states.

[0093] Now with reference to Fig. Figure 9 shows an illustrative state machine 900 that can be used to control the switching state of the differential multi-level converter 800 to maintain L2 (i.e., V). SM = (V A + V B) / 2), by switching back and forth (e.g., alternating) between states two (2) and three (3). For example, when L2 is selected (e.g., when the hybrid supply generator / supply modulator is instructed to output the voltage level L2), the controller can use the state machine 900 to determine whether state two (2) or three (3) is used to synthesize L2.

[0094] In the example of Fig. 9 are transitions between state two 902 and state three 903 a function of the capacitor voltage V cf More specifically, starting from state two 902, the state machine 900 can transition to state three 903 if V cf greater than (V B - V A ) / 2 + ΔV. The state machine 900 can transition back to state two 902 if V cf smaller than (V B - V A ) / 2 - ΔV is.

[0095] Such state transitions can be performed on either an immediate or clocked basis. The permissible deviation ΔV from V cf of (V B - V A ) / 2 determines the maximum deviation of the L2 output voltage from (V A + V B ) / 2 and can be used as a fixed value or a programmable value. Reducing ΔV results in a more precise supply modulator output voltage for L2 (smaller difference between the two states that synthesize L2), but requires more frequent switching between states.

[0096] In some embodiments, the state machine 900 can be reinitialized each time L2 is selected (i.e., transitions from L1 or L3), with the initialization being determined by the voltage V cf depends. For example, it may be desirable to initialize the state machine to state two 902 if V cf < (V B - VA ) / 2 is, and to initialize the state machine from state three 903 if V cf > (V B - V A ) / 2. Such a selection - or a similar one with different threshold(s) - can reduce the number of transitions required between states two (2) and three (3) to synthesize L2, and therefore improve efficiency.

[0097] Now with reference to Fig. 9A can, according to some embodiments, be the illustrative state machine of Fig. 9 to maintain the L2 voltage into a larger state machine 920 to control all three levels L1, L2 and L3.

[0098] In addition to states 902 and 903, which correspond to L2, the illustrative state machine 920 also includes a state 901, which corresponds to L3, and a state 904, which corresponds to L1.

[0099] In some cases, the state machine 920 can be initialized to one of the states 901 - 904 based on a standard voltage level (e.g., during startup / boot-up), which is programmed or hard-coded into the controller, for example.

[0100] The state machine 920 can transition between states 901–904 in response to level selection commands, such as commands received from an external circuit or control unit. For example, as shown, the state machine 920 can transition to state 904 if L1 is selected and to state 901 if L3 is selected. If L2 is selected, the state machine 920 can transition to either state 902 or state 903, depending on V. cf , the voltage across the flying capacitor.

[0101] One or more system controllers (e.g., controller(s) 550 of Fig. 5) can the state machine of Fig.9 and / or Fig. 9A implement, execute or otherwise use to control a differential multi-level converter.

[0102] Fig. 10A-10C illustrate voltage patterns ( Fig. 10A, Fig. 10B) and state transitions ( Fig. 10C), which is connected to the state machine of Fig. 9 are associated when used on an unclocked basis.

[0103] Fig. Figure 10A shows a plot 1000 with a vertical axis representing voltage and a horizontal axis representing time t. A curve 1002 represents V cf Over time. A horizontal line 1004 gives (V A - V B ) / 2, a horizontal line 1006a gives (V B - V A ) / 2 + ΔV and a horizontal line 1006b gives (V B - V A ) / 2 - ΔV on.

[0104] Fig.Figure 10B shows a plot 1020 with a vertical axis representing voltage and a horizontal axis representing time t. A curve 1022 represents V sm Over time. A horizontal line 1024 gives (V A + V B ) / 2 (i.e., the effective L2 value), a horizontal line 1026a indicates (V A + V B ) / 2 + ΔV and a horizontal line 1026b gives (V A + V B ) / 2 - ΔV on.

[0105] Fig. Figure 10C shows a plot 1040 with a vertical axis corresponding to the L2 switch state (i.e., state two (2) or state three (3)) and a horizontal axis corresponding to time t. A curve 1042 indicates the L2 state, which is a function of V. cf is ( Fig.10A). For example, at time t1, curve 1042 changes from state (2) to state (3) when curve 1002 crosses (or approaches) the horizontal line 1006a. Subsequently, at time t2, curve 1042 changes from state (3) to state (2) when curve 1002 crosses (or approaches) the horizontal line 1006b.

[0106] Out of Fig. 10A-10C can be seen to be the approach of transitions between switch states as a function of V cf within a maximum voltage deviation ΔV from a central voltage (V B - V A ) / 2 maintains and the supply modulator voltage v sm within a maximum voltage deviation ΔV from a central voltage (V B + V A ) / 2 maintains.

[0107] L2 state selection can additionally or alternatively be performed on a clocked basis. One way to do this, for example, is to use a state machine like the one in Fig. 9 is used to select how L2 is synthesized, updating the state of a state machine on a clock edge. The clock can be the same clock as the one used to make level selections, or it can be different.

[0108] In some embodiments, when transitioning from L2 states on a clocked basis, the way in which L2 is synthesized at each clock cycle can be updated by choosing state two (2) if V cf < (V B - V A ) / 2, or state three (3), if V cf > (V B - V A) / 2, optionally using a small degree of hysteresis in the comparator on which the decision is made. For a fully clocked selection, this approach can lead to a minimal deviation in V. sm of (V B + V A ) / 2 for L2, at the cost of a potentially high number of transitions for it.

[0109] Now with reference to Fig. 11. The general concepts, structures and techniques described above can be extended to provide a hybrid supply generator / supply modulator with more than three (3) output levels.

[0110] Fig. Figure 11 shows a differential multi-level converter 1100 for a hybrid power supply generator / power supply modulator. The converter 1100 operates with two voltages V. A and V B , which are received at respective inputs 1102a and 1102b, and synthesized V smat output 1104. In some embodiments, the voltage V B so that they are greater than the voltage V A is. In some embodiments, V B or V A an input voltage V in are equivalent to.

[0111] The converter 1100 comprises two (2) flying capacitors C f1 , C f2 and three (3) complementary switch pairs S1 / S 1' , S2 / S 2' and S3 / S 3' The switch pairs can be operated (e.g. by a controller) according to eight (8) different switch states to produce four (4) effective output levels (V). sm to provide.

[0112] A voltage V cf1 about first flying capacitors C f1 can be controlled so that they are close to (V B - V A ) / 3 is, and a voltage V cf2 via second flying capacitors C f2can be controlled so that it is close to 2(V) B - V A ) / 3 is what allows four effective output levels to be synthesized: L1=VA; L2=VB−2(VB−VA) / 3≈VA+(VB−VA) / 3; L3=VB−(VB−VA) / 3≈VA+2(VB−VA) / 3; and L4=VB.

[0113] There is therefore an evenly spaced distribution of effective output levels between V A and V B .

[0114] In some embodiments, switches S1, S 1' , S2, S 2' , S3, S 3' selected, a nominal voltage of more than 1 / 3 of the maximum difference between V B and V A to demonstrate, but does not have to be for more than the maximum difference between V B and V A be dimensioned. Gate control sources for the switches can again be derived from input voltages V. A , V B and / or voltages V cf1 , V cf2flying capacitors are derived.

[0115] Table 2 shows the behavior of the differential multi-level converter 1100 from Fig. 11 for eight (8) different switch states. TABLE 2: Condition Switches turned on V SM V SM = level V cf1 V cf2 1 S 1' , S 2' , S 3' V A V A L1 Unchanged Unchanged Condition Switches turned on V SM V SM = level V cf1 V cf2 2 S 1' , S 2' , S3 V B - V cf2 V B - 2(V B - V A ) / 3 L2 Unchanged Increasingly 3 S 1' , S2, S 3' V A + V cf2 - V cf1 V B - 2(V B -V A ) / 3 L2 Increasingly Decreasing 4 S 1' S2, S3 V B - V cf1 V B - (V B - V A ) / 3 L3 Increasingly Unchanged 5 S1, S 2' , S 3' V A + V cf1 V B - 2(V B -V A ) / 3 L2 Decreasing Unchanged 6 S1, S 2' , S3 V B -V cf2 +V cf1 V B - (V B - V A ) / 3 L3 Decreasing Increasingly 7 S1, S2, S3, V A + V cf2 V B - (V B - V A ) / 3 L3 Unchanged Decreasing 8 S1, S2, S3 V B V B L4 Unchanged Unchanged

[0116] As can be seen from Table 2, there is one way to synthesize L2 (i.e., state one (1)), one way to synthesize L4 (i.e., state eight (8)), and three ways to synthesize each of the two intermediate effective output levels L2 and L3. In particular, L2 can be synthesized using states two (2), three (3), and five (5), while L3 can be synthesized using states four (4), six (6), and seven (7).

[0117] The states that synthesize each of the two intermediate effective levels each exhibit different effects on the charging and discharging of C. f1 and C f2 on. (i.e. V cf1 or V cf2(increase or decrease under load). By dynamically selecting which state is used to synthesize a desired intermediate output level, the capacitor voltages V can be cf1 and V cf2 each close to their target voltages of (V B - V A ) / 3 and 2(V B - V A ) / 3 are maintained. These control choices can be made through a variety of means, including the use of one or more appropriate state machines. For example, a control scheme / state machine can be selected such that V cf1 within a voltage ΔV1 of the target voltage (V B - V A ) / 3 is maintained and V cf2 within a voltage ΔV2 of the target voltage 2(V) B - V A) / 3 is maintained. The values ​​ΔV1 and ΔV2 can be hard-coded in the controller, can be programmable values ​​of the controller, or can be transmitted via one or more external signals.

[0118] It is understood that a circuit arrangement can be provided to control the voltage V cf1 about the flying capacitor C f1 and the voltage V cf2 about the flying capacitor C f2 to measure / detect. For example, two differential sensors (not shown) can be provided, one of which receives its inputs via C. f1 has connected and the other of these has its inputs via C f2 The outputs of both differential sensors can be connected to the controller that controls the switching states.

[0119] Fig. 12A and Fig.Figure 12B shows state machines used to control the switch state of the differential multi-level converter 1100. Fig. 11 can be used.

[0120] Fig. Figure 12A shows a state machine 1200 that can be used to control the switch state of the converter 1100 to switch L2 between state two 1202, state three 1203 and state five 1205 as a function of V cf1 or V cf2 to maintain. For example, when L2 is selected (e.g., when the hybrid supply generator / supply modulator is instructed to output the voltage level L2), the controller can use the state machine 1200 to determine whether state two (2), three (3), or five (5) is used to synthesize L2.

[0121] Fig.Figure 12B shows a state machine 1220 that can be used to control the switch state of the converter 1100 to switch L3 between state four 1204, state six 1206 and state seven 1207 as a function of V cf1 or V cf2 to maintain. For example, if L3 is selected, the controller can use the state machine 1220 to determine whether state four (4), six (6) or seven (7) is used to synthesize L3.

[0122] The illustrative state machines of Fig. 12A, Fig.Section 12B will select which states to use for synthesizing the intermediate output levels within the desired tolerances. Again, the state machines can be implemented on a clocked or unclocked basis. It is understood that there is adequate control authority in the proposed system for selecting states, both to synthesize the desired output levels and to keep the voltages of the flying capacitors within an acceptable range; many other means can be implemented equally well.

[0123] The illustrative state machines of Fig. 12A, Fig. 12B can be integrated into a larger state machine with eight different states, as shown in Table 2, to control all four levels L1-L4, with transitions occurring in response to level selection commands (similar to the one described above in the context of Fig.9A for a 3-level converter state machine).

[0124] After reading the disclosure provided herein, average professionals in the field will understand that hybrid supply generator / supply modulator designs incorporate more flying capacitors and achieve more effective output levels by directly extending the 3-level design of Fig. 8 and the 4-level design of Fig. 11 can be synthesized.

[0125] In general, designs that provide N effective output levels can utilize N-2 flying capacitors and N-1 complementary switch pairs (which can provide up to 2 (N-1) (resulting in switch states that can be used for level synthesis).

[0126] It is understood that an additional circuit arrangement can be used to connect the flying capacitors (e.g., C). f in Fig. 8 or C f1 and C f2 in Fig.11) to preload to the desired levels before starting the modulation.

[0127] In some embodiments, an additional circuit arrangement may also be provided to control the voltages of flying capacitors during periods when the synthesized output levels do not provide charge / discharge control of the capacitor voltage(s), when the system is not operating (e.g., in standby mode or during startup), and / or when the voltage levels V A , V B or their references can be adjusted to maintain them within a desired range (e.g., close to their respective target voltages). Such a circuit arrangement could include linear circuits (e.g., current sources, linear regulators, etc.) implemented in the integrated circuit.

[0128] Fig. Figure 13A shows an example of a differential multi-level converter 1300 with a flying capacitor C. fand a circuit arrangement for regulating or controlling a charge on it, according to some embodiments.

[0129] The 1300 converter operates with two voltages V A and V B operated, which are received at inputs 1302a and 1302b respectively. The two voltages can be synthesized, for example, by a multi-output control stage, which can be the same as or similar to one of the multi-output control stages described above in conjunction with Fig. 7A-7E are described. In some embodiments, the voltage V B so that they are greater than the voltage V A Output 1304 provides a modulated output voltage V. sm ready, which can be mass-referenced.

[0130] As shown, the differential multi-level converter circuit comprises a set of four switches S A , S B , S A' , S B',which are connected in series between inputs 1302a and 1302b, with a floating capacitor C f1 a first connection between the switches S A and S B is connected, and a second connection that runs between the switches S A' and S B' is connected, exhibits.

[0131] To charge at capacitor C f1 To regulate the charge, the 1300 converter also includes a charge control circuit arrangement (sometimes referred to as a "pre-charge" or "charge-hold" circuit arrangement). In the example of Fig. 13A The charge control circuit arrangement includes a controller 1306 (e.g. a linear controller) which provides a reference voltage V P receives, a fifth switch S E , which is between an output of the controller 1306 and a first terminal of the flying capacitor C f1 is connected, and a sixth switch S F, which is between the output of the regulator 1306 and a second terminal of the flying capacitor C f1 is connected. The linear controller can differentially operate between V A and V B It is supplied with power and is designed to draw current and / or sink current to cause its output voltage to be close to the reference voltage V. P lies. The reference voltage V P can be based on the levels V A and V B can be selected. To control the charge on the flying capacitor in a three-level system, the reference voltage V can be used. P as half between V A and V B selected so that the linear controller attempts to move vcf1 towards half of (V B - V A) to charge. Alternatively, it can be dynamically programmable (e.g., by an external circuit arrangement). In a four-level system, one can have a charge control circuit for each of the flying capacitors, where reference voltages are selected to drive the capacitor voltages in the direction of (V). B - V A ) / 3 or 2(V B - V A ) / 3 to load.

[0132] The 1300 converter provides a ground-referenced output voltage V SM ready to go between the switches S B and S B' is taken as shown. The output voltage V SM can assume discrete values ​​that lie between voltage values ​​V A and V B are distributed, with energy from and between V A and V B and is transferred to the output.

[0133] The switches of Fig. Article 13A can be implemented in a variety of ways, including those mentioned above. Fig. 8 different switch implementations described.

[0134] The controller 1306 and the switches S E , S F can be used to measure the voltage V cf1 of the flying capacitor during periods when the synthesized output levels do not provide charge / discharge control of the capacitor voltage(s), when the system is not operating (e.g., in standby mode or during startup) and / or when the voltage levels V A , V B or their references can be adjusted within a desired range (e.g., close to the reference voltage V). p to maintain

[0135] In a state where switch S A If the switch is kept switched on, the S F be switched on, and the controller 1306 can supply current through C f1 sink or draw in, so that the voltage at C F near a desired reference voltage V Pis maintained. Likewise, in a state where the switch S A' If the switch is kept switched on, the S E be switched on, and the controller 1306 can supply current through C f1 sink or draw in, so that the voltage at C f1 near the desired reference voltage V P is maintained.

[0136] The reference voltage V P can be used with the intermediate voltages V A , V B be related. For example, V P at the midpoint between V A and V B lie, that is, V P = (V B +V A ) / 2. One could optionally use the reference voltage V P through a voltage divider between V A and V B generate. Alternatively, the reference voltage V can be used. P be provided externally in either analog or digital form.

[0137] The 1306 controller can be configured to regulate the voltage via C f1 to control them so that they are connected to the reference voltage V P matches, or can be configured to act to increase the voltage across C f1 in the direction of V P to target only if the difference between V P and V cf1 a threshold value ΔV p > exceeds 0. In some embodiments, the threshold ΔV can be p so that it is greater than the voltage ΔV used to select the switch state in order to keep the output voltage level close to (V B +V A ) / 2 to synthesize.

[0138] In some embodiments, the controller 1306 (e.g., a linear controller) can be configured to draw current or sink if V cf1 by more than an amount ΔV p by V pis removed. This can include a hysteresis comparison. In some examples, a comparator with hysteresis can be used. In other examples, two comparators can be used: a first comparator to detect when V cf greater than V p + ΔV p is, and a second comparator to detect when V cf smaller than V p - ΔV p If the first comparator is activated, the controller 1306 can be activated to adjust V cf to control downwards. If the second comparator is activated, controller 1306 can be activated to control V cf to drive upwards. If the voltage is within V p - ΔV p < V cf < V p + ΔV P The regulator may not work if the situation is such that the controller is inactive.

[0139] In some embodiments, the charge control circuit arrangement (e.g., controller 1306 and switch S) can be E , S F) based on an operating mode, a modulator state, the duration of a specific modulator state, etc. For example, the charge control circuit can be selectively activated and deactivated during modes such as start and standby, or during specific modulator states (such as when switch S is open). A or S A' is active) or when specific states of the modulator are held for more than a certain duration, or when the voltages V A and V B They can be adjusted and selectively activated.

[0140] Fig. Figure 13B shows another example of a differential multi-level converter 1340 with a flying capacitor and a circuit arrangement for regulating or controlling a charge on it, according to some embodiments. Similar to the converters of Fig. 8 and Fig. The converter 1340 includes 13A. Fig. 13B a ​​set of four switches S A , S B , SA' , S B', which are connected in series between inputs 1302a and 1302b, with a floating capacitor C f1 a first connection between the switches S A and S B is connected, and a second connection that runs between the switches S A' and S B , is connected, exhibits.

[0141] Additionally, the converter includes 1340 of Fig. Figure 13B comprises a differential sensor 1342, a controller 1344, and a charge control circuit 1346, which can be collectively referred to as the "charge control circuit arrangement". The differential sensor 1342 has inputs that are connected to opposite terminals of C. f1 are connected to increase the voltage V cf1 via C f1to detect, and an output that is connected as an input to the charge control circuit 1346. The controller 1344 can include controllable source elements 1348a-1348c (e.g., controllable current sources) that control the capacitor C f1 They can charge and discharge. The charge control circuit 1346 receives a reference input V. p (e.g. a reference voltage) and the output of the differential sensor 1342 and is configured to control the controller and in particular to control individual source elements 1348a-1348c of the controller 1344.

[0142] In some embodiments, the charge control circuit 1346 can be configured to C cf1 to charge or discharge, if its voltage V cf1 by more than an amount ΔV p by V p removed. In some examples, this can be done using two comparators: a first comparator to detect when V cf1 more than V p+ ΔV p is, and a second comparator to detect when V cf smaller than V p - ΔV p If the first comparator is activated, then the middle power source 1348b can be activated / switched on to provide V cf1 to drive downwards. If the second comparator is activated, current sources 1348a and 1348c can be activated to drive V cf1 to drive upwards. If the voltage is within V p - ΔV p < V cf < V p + ΔV p If the power source is switched off, all 1348a-c power sources can be turned off.

[0143] In some embodiments, the reference input V p with the intermediate voltages V A , V B are related, as above for Fig. 13A discussed.

[0144] In some embodiments, the power sources 1348a-c can be Fig.13B can be controlled to adjust the voltage at each terminal of the capacitor in response to a V A -Target voltage, a V B -To regulate target voltage, a system operating mode and / or other conditions.

[0145] Similar to the one above with Fig. The charge control circuit arrangement discussed in 13A can be used to describe the charge control circuit arrangement of Fig. 13B the voltage across C f1 control so that they are connected to the reference input V p matches, or can be configured to act to increase the voltage across C f1 in the direction of V P to target only if the difference between V P and V cf1 a threshold value ΔV p exceeds. In some embodiments, the threshold ΔV can p so that it is greater than the voltage ΔV used to select the switch state in order to keep the output voltage level close to (V B +VA ) / 2 to synthesize.

[0146] As also mentioned above with Fig. As discussed in section 13A, the charge control circuit arrangement of Fig. 13B can be selectively activated and deactivated based on the operating mode, the modulator state, the duration of a specific modulator state, etc.

[0147] Fig. 13A and Fig. Figure 13B illustrates charge control used with a differential multi-level converter 1340, which has a single flying capacitor, and how it can be used to provide three effective output levels. For converters with larger numbers of flying capacitors (e.g., the converter of Fig. 11, which is configured to provide four effective output levels) can demonstrate the general structures and techniques associated with Fig. 13A and Fig.The methods shown and described in Figure 13B can be extended to control the charge on each of the flying capacitors. For example, using the approach of Fig. 13A a separate linear regulator 1306 and a pair of switches S E , S F provided for each flying capacitor. As another example, using the approach of Fig. 13B a ​​separate charge control circuit 1346 and a set of controllable source elements 1348a-1348c are provided for each flying capacitor.

[0148] Now with reference to Fig. 14 According to embodiments of the present disclosure, a hybrid power supply generator / power supply modulator can have multiple outputs to supply one or more power amplifiers (e.g. with different voltages).

[0149] The illustrative hybrid power supply generator / power supply modulator 1400 comprises a single multi-output control stage 1402 and a plurality of differentially coupled multi-level converters 1404a, 1404b, ..., 1404n. The multi-output control stage 1402 receives an input voltage V IN and synthesizes two intermediate voltages V A and V B , which can be controlled independently. Each of the differentially coupled multi-level converters 1404a-n can handle the intermediate voltages V A and V B receive and be configured to synthesize one of three or more instantaneous output voltage levels, the voltage levels of which can be used to power a corresponding number of PAs 1406a-n.

[0150] In some embodiments, the differentially coupled multi-level converters 1404a-n can supply voltages V supply1...Ndirectly to the PAs 1406a-n, as shown. In other embodiments, the outputs of the differentially coupled multi-level converters 1404a-n can be filtered / regulated using an additional circuit arrangement to accommodate the multitude of PA supply voltages V. supply1...N to provide. Examples of such an additional circuit arrangement are given above in the context of Fig. 5, Fig. 5A and Fig. 5B described. In any case, it is understood that different PA supply voltages V supply1...N using the system architecture of Fig. 14 can be controlled independently.

[0151] The multi-output control stage 1402 can be the same as, or similar to, any of the embodiments of multi-output control stages disclosed herein. Likewise, any of the differentially coupled multi-level converters 1404a-n can be the same as, or similar to, any of the embodiments of differentially coupled multi-level converters disclosed herein.

[0152] In some embodiments, the multi-output control stage 1402 can be implemented on a single IC. In some embodiments, the differential multi-level converters 1404-n can be implemented on one or more ICs. In some embodiments, the IC of the multi-output control stage 1402 can differ from the IC(s) of the differential multi-level converters 1404-n. In some embodiments, the multi-output control stage 1402 and one or more differential multi-level converters 1404-n can be implemented on the same IC.

[0153] Now with reference to Fig.In some embodiments, the outputs of two or more differential multi-level converters can be combined (e.g., using coupled magnetics) to supply a single PA. The illustrative system 1440 comprises differential multi-level converters 1442a and 1442b, both configured to provide the same intermediate voltages V. A and V B to receive and a respective modulated voltage V SMA and V SMB to provide. The system 1440 further includes a combiner circuit 1444, which is configured to modulate the voltages V SMA , V SMB into a combined modulated voltage V SM to combine, which is used to supply a PA 1446. For example, V SM can be coupled directly or indirectly to a bias port of the PA 1446.

[0154] The system 1440 can optionally include a filter circuit 1448 or other additional circuitry coupled between the combiner circuit 1444 and the PA 1446, as shown. For example, the filter circuit 1448 can be configured to filter and smooth the transitions of the supply voltage provided to the PA 1446.

[0155] A controller 1450 can be coupled to control the switching states of the differential multi-level converters 1442a and 1442b (e.g., to control them independently). In some embodiments, the controller 1450 can be provided as part of a hybrid power supply generator / modulator that also includes the differential multi-level converters 1442a and 1442b. For example, the controller 1450 can be implemented on the same IC as the converters 1442a and 1442b.

[0156] It should also be acknowledged that the combiner circuit 1444 can be provided as any means capable of combining two or more modulated power supply output signals provided therein to provide a combined modulated signal suitable for use in an application of interest. Examples of such combining means are provided below. After reading the disclosure provided herein, an average person skilled in the art will understand how to select a means of combining appropriately for use in a particular application.

[0157] In the illustrative embodiment of Fig.In 14A, the combiner circuit 1444 is provided as a coupled magnetic structure exhibiting high coupling (e.g., a coupling coefficient in the range of approximately 0.5 or greater, or in some applications it may be desirable or necessary to utilize a coupling coefficient in the range of approximately 0.9 or greater) between windings 1452a and 1452b (e.g., an interphase transformer combiner). In this embodiment, the output, V SM , the combiner circuit 1444 a linear combination of the inputs from the individual converters 1442a, 1442b.

[0158] The 1444 combiner circuit is not limited to interphase transformer combiners. It can, for example, include uncoupled magnetics (e.g., separate inductors or uncoupled windings on a single core structure), a transmission line transformer combiner, a lumped or distributed GF combiner, a hybrid circuit, or another lumped or distributed passive three-port network.

[0159] Furthermore, while in the example of Fig. Figure 14A shows two converters 1442a, 1442b and a two-way combiner; a system generally comprises M modulators and an M-way combiner. This can enable increasingly higher performance (e.g., reduced content of unwanted frequencies for a given desired frequency content) with increasing M.

[0160] Two or more multi-level converters can be used together to create a single output V SMto provide. One such technique, which can be used with the illustrative system 1440, is so-called "split pulse transitions" (SPT). With SPT, the controller 1450 can receive a control input signal specifying a voltage level desired at the combiner output and generate outputs that, in turn, serve as inputs to the differential multi-level converters. The controller 1450 can set the output voltage levels V SMA and V SMB of two multi-level converters 1442 and thus V SM Based on the desired voltage level specified by the control input signal, they can be controlled independently. A person skilled in the field will understand that more than two differential multi-level converters can be combined in a similar way to implement SPT (Single Step-Through) technology.

[0161] In some embodiments, the controller 1450 can be configured to delay the output signal for the converter 1442b or the converter 1442a, so that at V SMA The observed transition occurs after the same transition at V SMB This delay can be referred to as a "split pulse delay" or "SPT delay". In this case, the voltages V SMA and V SMB The SPT delay must be unequal for a period corresponding to the SPT delay. The controller 1450 can apply the SPT delay to either the converter 1442b or the converter 1442a according to a state machine implemented as part of the controller. Which input of the multi-level converter 1442b or 1442a receives the delay can change as frequently as each transition of the desired voltage level specified by the controller input signal.

[0162] In some embodiments, when the SPT delay is configured to a value greater than zero, the overall transition of V SM The transition from a first level to a second level consists of three voltage levels spanning three sections. During the first section of the overall transition at V SM show both V SMA as well as V SMB the same output voltage, so that V SM equal V SMA and V SMB is. During the second period, the controller commands the desired transition for the multi-level converter 1442a, so that V SMA transitions to the second voltage level, but the output V SMB The output of 1442b remains at its original level. During this section, current flows between the outputs of 1442a and 1442b through combiner 1444. SM takes a value between V SMA and V SMBproportional to the ratio of the number of windings 1452a to the number of windings 1452b. Assuming an equal number of windings, the voltage V SM during this second period (V SMA -V SMB ) / 2 + V SMA The controller can delay the command for 1442a or 1442b according to an internal state machine. During the final period, the controller commands the desired transition for the multi-level converter 1442b, so that the output of 1442a and 1442b is once again the same, and V SM will be equal to V SMA and V SMB be.

[0163] SPT has the effect of changing the transition waveform while maintaining the high efficiency of the hybrid supply modulator / generator. Without SPT, the voltage V changes. SM quickly from the first voltage level to the second voltage level. The above example SPT implementation reduces dv / dt (voltage rise rate) of V.SM during the transition: V SM It starts at the first voltage level, moves to a voltage level between the first and second voltage levels during the SPT delay time, and then moves to the second voltage level after the SPT delay time. The SPT delay time between 1442a and 1442b changes the frequency content of the output V. SM and can be configured to determine the time domain shape and frequency domain content of V SM to optimize. A transceiver can command the controller 1450 to adjust the SPT delay time to optimize RF performance for challenging operating conditions (e.g., high or low bandwidth transmission, strict linearity or emission requirements).

[0164] Although Fig.Figure 14A shows an example of two differential multi-level converters 1442b, 1442a, and the preceding discussion explains how one of the converters can be delayed relative to the other. The general concepts and techniques described can be applied to other numbers of converters. For example, a hybrid power supply generator / modulator can have at least three differential multi-level converters, and the control system can apply two or more different delays to the at least three differential multi-level converters.

[0165] In some embodiments, a system disclosed herein (e.g., a system of Fig. 14A) utilize one or more control techniques described in U.S. Patent No. 11,909,358, issued on February 20, 2024, entitled “Multilevel Amplifier Systems and Related Techniques”, the patent being incorporated by reference in its entirety.

[0166] Now with reference to Fig. 15. As discussed previously, an RF power amplifier system may include a controller (e.g., one of the 550 controllers from Fig.5) configured to provide one or more control signals to one or more hybrid power generators / modulators. In some embodiments, the variable supply voltages (or "power biases") can be provided in the form of pulses, each pulse having a discrete number of voltage levels. That is, the hybrid power generators / modulators can each provide one of a variety of discrete supply voltages to the power / bias terminal of a corresponding RF amplifier (or, in some cases, multiple RF amplifiers). Such discrete voltage supply levels can be predetermined or adjusted over time based on required average transmit power levels or other factors.

[0167] Transitions between pulses with different voltage levels (i.e., transitions from one voltage level to another) can introduce unwanted frequency components into the varying supply voltage signals. In some embodiments, such variable supply voltages can be provided to the amplifier's supply / bias terminal via one or more pulse-shaping networks (PSNs). A PSN functions to filter out or otherwise remove unwanted frequency components in a supply voltage signal (i.e., the PSN filters or shapes the trajectory of the supply voltage signal). Thus, a filtered supply voltage signal is provided to the RF amplifier's supply terminal. In the context of Fig.For example, PSNs can correspond to or form part of the additional circuit arrangement 506. In some cases, the power supply generator, power supply modulator, and PSN can be provided as an integrated power management circuit (PMIC).

[0168] Fig.Figure 15 shows an example of a switching network that has one or more PSNs and is coupled to one or more hybrid power generators / modulators. The illustrative switching network 1500 has one or more inputs, each of which can be coupled to one or more hybrid power generators / modulators, with two hybrid power generators / modulators 1511a and 1511b shown in this example. Alternatively or additionally, instead of outputs, separate hybrid power generators / modulators 1511a and 1511b, signals 1501a and 1501b (and so on), or multiple outputs of a single hybrid power generator / modulator (e.g., the differential multi-level converter outputs 1404a and 1404b of the hybrid power generator / modulator 1400 in Figure 1500) can be used. Fig.14) or represent combinations of outputs of a single hybrid multi-output power generator / power modulator and one or more other hybrid power generator(s) / power modulator(s).

[0169] In some implementations, the switching network 1500 can be coupled in a cascaded configuration with the hybrid power supply generators / modulators 1511a and 1511b. In this exemplary embodiment, the switching network 1500 has two inputs 1501a and 1501b, which are coupled to outputs of the respective hybrid power supply generators / modulators 1511a and 1511b. The hybrid power supply generators / modulators (A and B) each provide modulated voltage signals VSMA and VSMB as inputs to the switching network 1500.

[0170] The switching network 1500 has a first plurality of outputs 1502a - 1502N, each connected to one or more of a second plurality of RF amplifiers (in Fig. (15 not shown) can be coupled. In exemplary embodiments, the number of switching network outputs can be the same as the number of RF amplifiers, so that there is a one-to-one correspondence between the number of switching network outputs and RF amplifiers. In this case, each RF amplifier can be coupled to one of the switching network outputs 1502a-1502N.

[0171] In the exemplary embodiment of Fig. Figure 15 illustrates the switching network 1500 as providing N outputs 1502a-1502N (where N is any integer greater than 1), at which each of the voltages V O1 -V ON The switching network outputs can be provided. The switching network outputs can be coupled to one or more bias connections (e.g., a supply connection) of one or more RF power amplifiers, and thus the output voltages V can be O1 -V ONIt may be coupled to a bias connection of one or more RF power amplifiers. In some embodiments, N can be equal to 2 (thus output voltages V). O1 , V O2 provide). In some embodiments, N can be equal to 4 (thus output voltages V). O1 -V O4 provide). In some embodiments, N can be equal to 6 (thus output voltages V). O1 -V O6 provide). In some embodiments, N can be equal to 8 (thus voltages V). O1 -V O8 provide). In some embodiments, N can be equal to 10 (thus voltages V). O1 -V O10 provide).

[0172] For example, in one embodiment, two or more output ports can be coupled to provide output signals; in some cases, all N output signals can be V O1 -V ONcoupled to a bias connection (e.g., a supply connection) of a single RF power amplifier. In other embodiments, one or more or each output signal V can be coupled. O1 -V ON each output amplifier may be coupled to its own respective RF power amplifier (i.e., a bias terminal of the respective RF amplifier). In yet other embodiments, one or more output amplifiers may be coupled to a single output terminal of the switching network 1500, while other RF output amplifiers may be coupled to two or more output signals of the switching network 1500.

[0173] The switches S1-S 11 The switching network 1500 can be coupled with a controller (not shown) that controls the switches S1-S 11 can open and close to control the output signals V O1 -V ONto control. In this way, the switching network 1500 can be coupled via one or more PSNs (and in some cases configured to provide a signal path that bypasses one or more of the PSNs). For example, when switch S1 is closed, the modulated voltage signal VSMA is coupled to output 1502a, where the voltage VO1 is provided. When switch S1 is open and switches S2 and S3 are closed, the modulated voltage signal VSMA is coupled to output 1502a via PSN 1503 (also known as filter network 1503). And when switches S1 and S3 are open, the output signal V O1Output 1502a is not connected to the voltage signal VSMA and can be floating or connected to any other potential, such as ground, by a circuit arrangement not shown. Thus, switches S1, S2, and S3 can be used to adaptively (or dynamically) filter the output signal V in real time (this filtering is performed by a filter network 1503). O1 to activate or deactivate. It should be acknowledged that the filter circuit arrangement can be provided in a variety of different circuit configurations to provide filter characteristics that are selected to meet the needs of a specific application. Assuming a filter network (PSN) 1503, which is illustrative for filter networks 1504-1510, the filter network 1503 has one or more electronic elements. In the example of Fig. 15. Filter network 1503 comprises four electronic elements, which are passive circuit elements (sometimes referred to herein as a "passive component"). Filter networks 1503-1510 can comprise various passive or active circuit elements, selected to meet the needs of a specific application. After reading the disclosure provided herein, a person skilled in the art will understand how to design one or more filter networks to meet the needs of a specific application.

[0174] In some embodiments, a PSN may comprise one or more passive elements implemented as discrete components. In some embodiments, a PSN may comprise one or more passive elements implemented on an IC or module. In some embodiments, a PSN may comprise one or more passive elements resulting from parasitic resistance, parasitic inductance, or parasitic capacitance.

[0175] In some embodiments, a PSN may comprise a resistor having a first terminal connected to the output of the hybrid power supply generator / modulator, and a capacitor having a first terminal connected to the second terminal of the resistor and a second terminal connected to ground.

[0176] Whether a given PSN (e.g., one or more of the filter networks 1503, 1508, 1510, etc.) is connected between a hybrid power generator / modulator (e.g., one or more of the hybrid power generators / modulators 1511a, 1511b) and an output (e.g., one of the outputs 1502a-1502N) depends on a variety of factors, including, but not limited to: the RF frequency band in which the RF signal supplied to the RF input of the power amplifier is located; the bandwidth of the RF signal supplied to the RF input of the power amplifier; a peak-to-average ratio of the RF signal supplied to the RF input of the power amplifier; and the power level of the RF signal supplied to the RF input of the power amplifier. other aspects or characteristics of the RF signal to be supplied to the PA's RF input; the mode of supply modulation used (e.g.digital envelope tracking versus average power tracking versus fixed supply); and / or by the characteristics of an operating or application scenario (e.g., observed noise or amplifier behavior) among other factors.

[0177] When switch S4 is closed, the modulated voltage signal VSMA is coupled through the filter network 1504 to output 1502b, where the voltage V O2 When switch S5 is closed, the modulated voltage signal VSMA is coupled to output 1502c via filter network 1506. Similarly, when switch S6 is closed, the modulated voltage signal VSMB is coupled to output 1502c via filter network 1506. Therefore, switches S5 and S6 can be used to select which modulated voltage signal, VSMA or VSMB (or both in parallel), is coupled to provide an output signal at output 1502c.

[0178] When switch S7 is closed, the modulated voltage signal VSMB is coupled through filter network 1508 to terminal 1502d and made available. It should be noted that one, some, or all of filters 1503-1510 can be provided as reconfigurable filters. For example, filter 1508 has a switch S8. Switch S8 is configured to modify the filter parameters (or properties) of filter network 1508. In this case, closing switch S8 creates a short-circuit signal path through inductor L8, effectively removing inductor L8 from filter 1508, which will affect the transfer function of filter network 1508. In this way, filter properties of filter 1508 can be modified (e.g., adaptively, on-the-fly, or in real time) to meet the needs of a specific application or operating scenario.For example, it might be desirable to dynamically adjust the filter characteristics depending on the RF band in which the signal is transmitted by the power amplifier, the bandwidth, peak-to-average ratio, power level, or other aspects of the signal being transmitted, or the mode of supply modulation used (e.g., digital envelope tracking versus adaptive power tracking versus fixed supply), or the characteristics of an operating or application scenario (e.g., observed noise or amplifier behavior), among other factors.

[0179] The switches S9 can be switched between open and closed states to selectively couple the modulated voltage signal VSMB through the filter 1510 to node 1512. The switches S 10 , S 11They can be switched between open and closed states to couple node 1512 to one or both of the outputs 1502N-1 and 1502N, where the respective voltages VN-1 and VN are provided. When switches S9 and S 10 When closed, the modulated voltage signal VSMB is coupled through the filter network 1510 to the output 1502N-1, at which the voltage V N-1 is provided. If switches S9 and S are equally available... 11 Once closed, the modulated voltage signal VSMB is coupled through the filter network 1510 to node 1512, where the voltage V N can be provided. If all three switches S9, S 10 and S 11 When the circuits are closed, the modulated voltage signal VSMB is coupled through the filter network 1510 to both output terminals 1502N-1 and 1502N.

[0180] The switches S1-S 11 in Fig.Fifteen generated signal paths are provided as examples. A person skilled in the field will recognize that other configurations of signal paths, filter parameters, and properties are possible by changing the number, arrangement, and control of the switches in the 1500 switching network. For example, switches S1-S 11They can be operated or controlled (i.e., placed in an open or closed state) so that either of the two modulated voltage signals VSMA, VSMB can be coupled to any of the 1502a-1502N terminals. In general, switches in the 1500 switching network can be configured to allow one or more on-die / chip supply modulator outputs to be routed to one or more power amplifier terminals; to adjust the filtering of a provided modulator output (e.g., to...to provide a reconfigurable pulse-shaping network); to reconfigure how different (possibly spatially separated) filter stages are used when connecting one or more modulator outputs to one or more RF amplifiers via one or more filter stages; and to turn off one or more switches to allow a supply modulator output to be disconnected from a power amplifier and / or filter, or to perform other tasks that modify and / or control the output signals that supply power to RF power amplifiers.

[0181] The specific way in which the 1500 switching network is implemented can depend on the power level, voltage level, and application of the system in which the switching network is used (e.g., an RF amplifier system). For some mobile device applications (e.g., a mobile phone, a smartphone, a tablet PC with cellular communication capabilities), it may be desirable to monolithically integrate electronic elements (e.g., circuit components) of both the power supply generator and the power supply modulator, as well as switching elements and sections of the auxiliary circuitry, onto a single semiconductor die / chip (e.g., in a CMOS or BCD process) or IC. In some cases, it may be desirable to integrate electronics such as the modulator(s) and the 1500 switching network together with power amplifiers onto a single die.Furthermore, in some cases it may be advantageous to encapsulate the modulator, switches, and some of the filter components in a single module, while locating other filter components and the RF amplifier in a physically separate location. In still other applications, it may be advantageous to encapsulate the modulator and some switches on a first die and place further switches on at least one additional die located some distance from the first die. This second die may also contain one or more power amplifiers or be physically close to power amplifier(s), for example, in a module or colocated on a printed circuit board.In these latter cases, the switches on the first die can be used for some of the functions described above and can be placed close to one or more first filter stages, while the second die can also implement some of the functions described above and can be placed closer to one or more second filter stages. Communication lines can also be provided between the first die and the second die (or between a controller and the second die) to allow the configuration to be changed.

[0182] Although reference is sometimes made herein to specific materials, devices and / or components, it is understood that other materials, devices and / or components having similar functional and / or structural properties may be appropriately substituted, and after reading the description provided herein, a person skilled in the art would understand how to select such materials, devices and / or components and incorporate them into the embodiments of the concepts, techniques and structures set forth herein without departing from the scope of protection of these teachings.

[0183] Various embodiments of the concepts, systems, devices, structures, and techniques to be protected are described herein with reference to the accompanying drawings. Alternative embodiments may be developed without deviating from the scope of protection of the concepts, systems, devices, structures, and techniques described herein. It is noted that various connections and positional relationships (e.g., above, below, adjacent, etc.) between elements are set forth in the following description and in the drawings. These connections and / or positional relationships may be direct or indirect unless otherwise specified, and the described concepts, systems, devices, structures, and techniques are not intended to be limited in this respect.Accordingly, a coupling of entities can refer to either a direct or an indirect coupling, and a positional relationship between entities can be a direct or indirect positional relationship.

[0184] As an example of an indirect positional relationship, references in the present description to the formation of layer "A" via layer "B" encompass situations in which one or more intermediate layers (e.g., layer "C") lie between layer "A" and layer "B," provided that the relevant characteristics and functionalities of layer "A" and layer "B" are not substantially altered by the intermediate layer(s). The following definitions and abbreviations are to be used for the interpretation of the claims and the patent specification. As used herein, the terms "has," "incorporating," "comprises," "including," "has," "with," "contains," or "containing," or any other variation thereof, are to cover a non-exclusive inclusion.For example, a composition, mixture, process, procedure, object or facility which includes a list of elements is not necessarily limited to those elements, but may include other elements which are not expressly listed or inherent in such composition, mixture, process, procedure, object or facility.

[0185] Additionally, the term "exemplary" is used herein to mean "serving as an example, instance, or illustration." Any embodiment or design described herein as "exemplary" is not necessarily to be construed as preferable or advantageous over other embodiments or designs. The terms "one or more" and "one or more" are to be understood as including any integer greater than or equal to one, i.e., one, two, three, four, etc. The terms "a multitude" are to be understood as including any integer greater than or equal to two, i.e., two, three, four, five, etc. The term "connection" may include an indirect "connection" and a direct "connection."

[0186] References in the patent specification to an "embodiment" or "embodiments" indicate that the one or more described embodiments may comprise a specific feature, structure, or property, but that each embodiment may comprise the specific feature, structure, or characteristic. Furthermore, such expressions do not necessarily refer to the same embodiment. Moreover, if a specific feature, structure, or property is described in connection with an embodiment, it is presented as being within the knowledge of a person skilled in the art to understand that such a feature, structure, or property may influence other embodiments, whether explicitly described or not.

[0187] For the purposes of the following description, the terms "upper," "lower," "right," "left," "vertical," "horizontal," "top," "bottom," and derivatives thereof shall refer to the described structures and processes as oriented in the drawings. The terms "overlying," "above," "on," "positioned on," or "positioned above" mean that a first element, such as a first structure, is located on top of a second element, such as a second structure, and that intermediate elements, such as an interface structure, may be present between the first and second elements. The term "direct contact" means that a first element, such as a first structure, and a second element, such as a second structure, are connected without any intermediate elements.

[0188] The use of ordinal terms, such as "first", "second", "third", etc., in the claims to modify a claim element does not in itself imply any priority, precedence, or order of one claim element over another, nor does it imply any temporal order in which actions of a process are carried out. Rather, they are merely used as identifiers to distinguish one claim element with a certain name from another element with the same name (with the exception of the use of the ordinal term) in order to differentiate the claim elements.

[0189] The terms "approximately" and "about" can be used to mean within ±20% of a target value in some embodiments, within ±10% of a target value in some embodiments, within ±5% of a target value in some embodiments, and still within ±2% of a target value in some embodiments. The terms "approximately" and "about" can include the target value. The term "essentially the same" can be used to refer to values ​​that are within ±20% of each other in some embodiments, within ±10% of each other in some embodiments, within ±5% of each other in some embodiments, and still within ±2% of each other in some embodiments.

[0190] The term "substantially" can be used to refer to values ​​that are within ± 20% of a comparative measure in some embodiments, within ± 10% in some embodiments, within ± 5% in some embodiments, and still within ± 2% in some embodiments. For example, a first direction that is "substantially" perpendicular to a second direction can refer to a first direction that is within ± 20% of a 90° angle with the second direction in some embodiments, within ± 10% of a 90° angle with the second direction in some embodiments, within ± 5% of a 90° angle with the second direction in some embodiments, and still within ± 2% of a 90° angle with the second direction in some embodiments.

[0191] As used herein, the terms "processor" and "controller" are used to describe an electronic circuit arrangement that performs a function, an operation, or a sequence of operations. The function, operation, or sequence of operations may be hard-coded in the electronic circuit or soft-coded by means of instructions held in a memory device. The function, operation, or sequence of operations may be performed using digital values ​​or using analog signals. In some embodiments, the processor or controller may be embodied in an application-specific integrated circuit (ASIC), which may be an analog ASIC or a digital ASIC, in a microprocessor with associated program memory, in a digital signal processor (DSP), and / or in a discrete electronic circuit, which may be analog or digital.A processor or controller may comprise internal processors or modules that perform sections of the function, operation, or sequence of operations. Likewise, a module may comprise internal processors or internal modules that perform sections of the function, operation, or sequence of operations of the module. A single processor or other unit may perform the functions of several means mentioned in the claims.

[0192] As used herein, the term "predetermined" is used to refer to a value or signal that is set or fixed in the factory at the time of manufacture or subsequently by external means, such as programming. As used herein, the term "determined" is used to refer to a value or signal that is identified by a circuit during operation, after manufacture.

[0193] It should be understood that the disclosed subject matter is not limited in its application to the details of the construction and arrangement of the components set forth in the following description or illustrated in the drawings. The disclosed subject matter is capable of other embodiments and various implementations and executions. Furthermore, it should be understood that the phraseology and terminology used herein serve the purpose of description and should not be considered limiting. As such, those skilled in the art will understand that the concept on which this disclosure is based can simply be used as a basis for designing other structures, processes, and systems for carrying out the various purposes of the disclosed subject matter.Therefore, the claims should be considered to include such equivalent constructions, insofar as they do not deviate from the concept and scope of protection of the disclosed subject matter.

[0194] Although the disclosed subject matter has been described and illustrated in the foregoing exemplary embodiments, it is understood that the present disclosure has been made only by way of example, and that numerous changes can be made to the details of the implementation of the disclosed subject matter without deviating from the idea and scope of protection of the disclosed subject matter.

[0195] Other variations of the disclosed embodiments can be understood and effected by those skilled in the field when practicing the claimed invention, from a study of the drawings, the disclosure and the attached claims.

[0196] The mere fact that certain measures are cited in mutually differing dependent claims does not indicate that a combination of these measures can be used to obtain an advantage.

[0197] Any reference numerals in the claims should not be interpreted as limiting the scope of protection.

[0198] All publications and references cited herein are expressly incorporated in their entirety by reference. QUOTES INCLUDED IN THE DESCRIPTION

[0000] This list of documents cited by the applicant was automatically generated and is included solely for the reader's convenience. The list is not part of the German patent or utility model application. The DPMA accepts no liability for any errors or omissions. Cited patent literature

[0000] US 8,829,993

[0022] US 9,160,287

[0022] US 9,166,536

[0022] US 9,172,336

[0022] US 9,209,758

[0022] US 9,755,672

[0022] US 11,909,358

[0165]

Claims

[1] System (500) which has the following features: a hybrid power supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) which features: an input for receiving an input voltage (V in ); an output to provide a modulated voltage (V) SM ); a multi-output control stage (602, 700, 720, 740, 760, 780, 1402) configured to provide two intermediate voltages (V A , V B ) with different voltage levels, wherein at least one of the two intermediate voltages (V A , V B ) from the input voltage (V in ) is synthesized; and a multi-level converter (604, 800, 1100, 1300, 1340, 1442a, 1442b) configured to convert the two intermediate voltages (V A , V B ) to receive and the modulated voltage (V SM) to generate a voltage level corresponding to one of the two intermediate voltage levels or at least one synthesized voltage level that differs from both intermediate voltage levels; and a circuit arrangement (506) coupled to the output of the hybrid power supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) and to a power supply connection of at least one high-frequency (HF) amplifier (504, 1406a-1406n, 1446), wherein the circuit arrangement (506) is configured to modulate the voltage (V) supplied to the at least one HF amplifier (504, 1406a-1406n, 1446). SM ) to modify. [2] System (500) according to claim 1, wherein the circuit arrangement (506) comprises a disconnect switch (542) configured to selectively supply a zero voltage level to the at least one RF amplifier (504, 1406a-1406n, 1446). [3] System (500) according to claim 1, wherein the circuit arrangement (506) comprises a disconnect switch (542) configured to selectively isolate the output of the hybrid power supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) from the at least one RF amplifier (504, 1406a-1406n, 1446). [4] System (500) according to claim 3, wherein the disconnect switch (542) is configured to be actuated in conjunction with the activation or deactivation of the RF amplifier (504, 1406a-1406n, 1446). [5] System (500) according to claim 1, wherein the circuit arrangement (506) comprises a pulse shaping network, PSN, (522, 1503-1510) configured to modulate the voltage (V SM ) to filter. [6] System (500) according to claim 5, wherein the PSN (522, 1503-1510) comprises at least one passive element. [7] System (500) according to claim 6, wherein the at least one passive element is realized as a discrete element. [8] System (500) according to claim 6, wherein the at least one passive element is implemented on an integrated circuit (IC) or a module. [9] System (500) according to claim 6, wherein the at least one passive element results from parasitic resistance, parasitic inductance or parasitic capacitance. [10] System (500) according to claim 5, wherein the PSN (522, 1503-1510) comprises: a first and second inductor (524a, 524b) connected in series between the output of the hybrid supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) and the power supply terminal of the at least one RF amplifier (504, 1406a-1406n, 1446); a capacitor (526) with a first terminal connected between the first and second inductors (524a, 524b); and a third inductor (524c) which is connected between a second terminal of the capacitor (526) and ground. [11] System (500) according to claim 5, wherein the PSN (522, 1503-1510) comprises: a resistor with a first terminal connected to the output of the hybrid power supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b); and a capacitor with a first terminal connected to a second terminal of the resistor, and a second terminal connected to ground. [12] System (500) according to claim 5, wherein the PSN (522, 1503-1510) comprises: a capacitor with a first terminal connected to the output of the hybrid power supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) and a second terminal connected to ground; and a resistor with a first terminal that is connected to the first terminal of the capacitor. [13] System (500) according to claim 5, wherein the circuit arrangement (506) further comprises a switching network (1500). [14] System (500) according to claim 13, wherein the switching network (1500) is configured such that in a first state the switching network (1500) provides a first signal path with a first filter configuration between the output of the hybrid power supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) and the power supply terminal of the at least one RF amplifier (504, 1406a-1406n, 1446) and in a second state the switching network (1500) provides a second signal path with a second, different filter configuration between the output of the hybrid power supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) and the power supply terminal of the at least one RF amplifier (504, 1406a-1406n, 1446). [15] System (500) according to claim 13, wherein the switching network (1500) includes at least one passive element (C f , C f1 , C f2) having a first terminal connected to the output of the hybrid supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b), wherein a switching network (1500) is configured such that in a first state the switching network (1500) has a second terminal of the at least one passive element (C f , C f1 , C f2 ) connects to ground and in a second state the switching network (1500) the second terminal of the at least one passive element (Cr, C) f1 , C f2 ) separates from mass. [16] System (500) according to claim 13, wherein the switching network (1500) is coupled in a cascaded configuration with the hybrid supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b). [17] System (500) according to claim 13, wherein the switching network (1500) is coupled via the PSN (522, 1503-1510) and is configured to selectively provide a signal path that bypasses the PSN (522, 1503-1510). [18] System (500) according to claim 13, wherein the switching network (1500) is connected in parallel with a passive element (C f , C f1 , C f2 ) of the PSN (522, 1503-1510) and is configured to perform a transfer function of the PSN (522, 1503-1510) by selectively short-circuiting at least one passive element (C f , C f1 , C f2 ) to change. [19] System (500) according to claim 13, wherein the switching network (1500) comprises a plurality of switches (S A , S B , S A' , S B' , S1, S2, S3, S 1' , S 2' , S 3') exhibits and at least a first set of the plurality of switches is located on a first integrated circuit chip and at least a second set of the plurality of switches is located on a second, different integrated circuit chip. [20] System (500) which has the following features: a hybrid power supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) which features: an input for receiving an input voltage (V in ); an output to provide a modulated voltage (V) SM ); a multi-output control stage (602, 700, 720, 740, 760, 780, 1402) configured to provide two intermediate voltages (V A , V B ) with different voltage levels, wherein at least one of the two intermediate voltages (V A , V B ) from the input voltage (V in ) is synthesized; and a multi-level converter (604, 800, 1100, 1300, 1340, 1442a, 1442b) configured to convert the two intermediate voltages (V A , V B ) to receive and the modulated voltage (V SM ) to generate a voltage level corresponding to one of the two intermediate voltage levels or at least one synthesized voltage level that differs from both intermediate voltage levels, wherein the output of the hybrid power supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) is coupled directly or indirectly to a power supply input of at least one high frequency (HF) amplifier (504, 1406a-1406n, 1446). [21] Method for modifying a frequency response of a circuit coupled to an output of a hybrid power supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) and to a high-frequency (HF) amplifier (504, 1406a-1406n, 1446), the method comprising: in a first state, configure the circuit to provide a first signal path with a first filter configuration between the output of the hybrid supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) and the input of the RF amplifier (504, 1406a-1406n, 1446); and In a second state, configure the circuit to provide a second signal path with a second, different filter configuration between the output of the hybrid supply generator / modulator (502, 600, 620, 1400, 1511a, 1511b) and the input of the RF amplifier (504, 1406a-1406n, 1446).

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