Amplifier circuit for broadband and low-noise amplification of a capacitive current source and a sensor system

Abstract

The invention relates to an amplifier circuit (1) for broadband and low-noise amplification of a capacitive current source, preferably a pyroelectric sensor, comprising a signal input (3) that can be connected to the capacitive current source at a node K1, an input stage (A1), wherein the input stage (A1) is connected to the node K1 at an input (7) of the input stage (A1), and has a node K2 at the output (9) of the input stage (A1), wherein the input stage (A1) is designed to amplify an input voltage at least by a factor of 3, wherein the input stage (A1) is designed to provide a high-value input resistance at the input of the input stage (A1), wherein the input stage (A1) is designed to provide a stable and load-independent voltage at the output of the input stage (A1), an amplifier cascade, wherein the amplifier cascade (11) has at least one first and one second amplifier (A2, A3) with a respective input (13, 17) and output (15, 19), wherein the output (19) of the first amplifier (A1) is connected to the input (17) of the second amplifier (A3) at a node K3, wherein the input (13) of the first amplifier (A2) is connected to the node K2, wherein the output (19) of the second amplifier (A3) is connected to a node K4, wherein the amplifier cascade (11) is designed to generate a high signal intensity with low phase rotation over a wide frequency range, a feedback network (F1), wherein the feedback network (F1) is connected to the input (7) of the input stage (A1) at the node K1 and to the output (19) of the second amplifier (A3) at the node K4, wherein the feedback network (F1) is designed to provide a high-value feedback resistance with a parasitic capacitance of less than 0.5 pF, wherein the feedback network (F1) is designed to provide a negative feedback to a structure comprising the input stage (A1) and the amplifier cascade (11), and a signal output (21) that is connected to a node K5, wherein the node K5 is connected to the node K4 or corresponds to the node K4. The invention also relates to a sensor system.

Description

[0001] Amplifier circuit for broadband and low-noise amplification of a capacitive current source and a sensor system

[0002] The invention relates to an amplifier circuit for broadband and low-noise amplification of a capacitive current source, preferably a pyroelectric sensor, as well as a sensor system. The subject matter of the invention is defined in the appended claims.

[0003] The highly sensitive detection of electromagnetic radiation in the infrared and terahertz range is relevant for a wide variety of applications. For example, heat sensors are used in motion and fire detectors, as well as for gas analysis and in spectrometers for chemical material analysis. For demanding measurements at room temperature, capacitive current sources such as pyroelectric sensors are very widely used due to their simple design and associated low cost. The core of a pyroelectric sensor is a crystal made of a pyroelectric material. In these materials, the charge centers of positive and negative ions do not coincide, which is why an electrical polarization forms that can be aligned along a crystal axis. Even the smallest changes in the crystal's temperature, e.g., due to the impact of thermal radiation, lead to a change in this polarization.An increase in temperature directly reduces spontaneous polarization, but it also indirectly mediates a change in dipole orientation through the expansion of the material. As a result, surface charges form at the crystal's interfaces, perpendicular to its polar axis, proportional to the temperature change. The charges can flow away via electrodes attached to these surfaces and are thus measurable as a current.

[0004] Since the currents generated by a pyroelectric crystal are typically on the order of a few picoamperes, low-noise measurement electronics with a high-gain amplifier circuit are required for evaluation. Achieving the widest possible frequency bandwidth simultaneously places high demands on the amplifier circuit used. This circuit is largely responsible for the overall performance of the sensor system for a given sensor element.

[0005] Circuits for measuring such small currents are known in the state of the art.

[0006] A disadvantage of prior art amplifier circuits is their narrow bandwidth at high signal amplification levels from capacitive current sources, such as those of pyroelectric sensors. Furthermore, the amplified signals often exhibit significant noise. Furthermore, prior art amplifier circuits often exhibit a strong dependence on their input capacitance.

[0007] The object of the present invention is therefore to provide a transimpedance amplifier (TIA) or an amplifier circuit which enables low-noise amplification of a signal from a capacitive current source, in particular a pyroelectric sensor, with a high bandwidth and thereby eliminates the disadvantages of the prior art.

[0008] The object is achieved in a first aspect of the invention by the amplifier circuit according to the invention according to claim 1. The object is also achieved in a second aspect of the invention by the sensor system according to the invention according to claim 20. Preferred embodiments of the invention emerge from the subclaims and the following explanations.

[0009] The object is achieved in the first aspect of the invention by the amplifier circuit for broadband and low-noise amplification of a capacitive current source, preferably a pyroelectric sensor, according to the features of claim 1. The amplifier circuit according to the invention for broadband and low-noise amplification of a capacitive current source, preferably a pyroelectric sensor, according to claim 1 comprises a signal input that can be connected to the capacitive current source at a node Ki, an input stage, wherein the input stage is connected to the node Ki at an input of the input stage and has a node K2 at the output of the input stage, wherein the input stage is configured to amplify an input voltage at least 3 times, wherein the input stage is configured to provide a high-ohm input resistor at the input of the input stage, wherein the input stage is configuredto provide a stable and load-independent voltage at the output of the input stage, an amplifier cascade, wherein the amplifier cascade has at least a first and a second amplifier, each with an input and an output, wherein the output of the first amplifier is connected to the input of the second amplifier at a node K3, wherein the input of the first amplifier is connected to the node K2, wherein the output of the second amplifier is connected to a node K4, wherein the amplifier cascade is configured to generate a high signal amplification with low phase shift over a wide frequency range, a feedback network, wherein the feedback network is connected to the input of the input stage at the node K1 and the output of the second amplifier at the node K4, wherein the feedback network is configured to provide a high-ohm feedback resistor with a parasitic capacitance of less than 0.5 pF,wherein the feedback network is configured to provide negative feedback to a structure comprising the input stage and the amplifier cascade, and a signal output connected to a node K5, wherein the node K5 is connected to the node K4 or corresponds to the node K4.,

[0010] The amplifier circuit according to the invention is a transimpedance amplifier (TIA) circuit. In the following, the terms amplifier circuit and transimpedance amplifier are used synonymously.

[0011] A capacitive current source, within the meaning of the invention, is a current source with an output impedance that can be described to a good approximation by an electrical capacitance. This output impedance corresponds to the source impedance at the downstream TIA input. Examples of capacitive current sources can be photodiodes, CCD pixels, tunneling current sensors, pressure and touch sensors, (Geiger-Müller) counter tubes, photomultipliers (e.g., microchannel plates), acceleration sensors, or, preferably, a pyroelectric sensor. Within the scope of this invention, a pyroelectric sensor is a component in which, due to its pyroelectric properties, a temperature difference causes a change in the component's electrical voltage.

[0012] The pyroelectric sensor can be represented by an equivalent circuit consisting of a parallel connection of current source l py , crystal capacity C py and loss resistance R pyThe resistivity of pyroelectric materials can be very high, typically in the order of several 10 10 Q cm. The electrical capacitance of a sensor element can vary between 100 pF and 1 nF, depending on the crystal thickness, for typical crystal sizes of a few mm in diameter.

[0013] Within the scope of the invention, a pyroelectric sensor comprises a crystal with a pyroelectric material or is a crystal that comprises or consists of a pyroelectric material. The electrical capacitance C py The performance of a pyroelectric sensor is primarily determined by the thickness and area of ​​the pyroelectric crystal. Broadband amplification of different sensor elements therefore requires a transimpedance amplifier that is particularly insensitive to changes in the source impedance.

[0014] The amplifier circuit according to the invention, i.e., the transimpedance amplifier, converts an incoming current signal into a proportional output voltage; it can thus be considered a current-controlled voltage source. This behavior is particularly advantageous for measuring and amplifying small current signals. In particular, this behavior is suitable for measuring small currents in a pyroelectric sensor element. The current generated at the interfaces of the pyroelectric material, also called a pyroelectric, is thus converted into an easily measurable voltage. This voltage can then be read out, for example, by an analog-to-digital converter. The capacitive current source preferably has a current on the order of magnitude of a pyroelectric sensor.

[0015] Within the scope of the invention, a signal input describes a hardware interface of the amplifier circuit to which the current signal from the capacitive current source can be applied. For this purpose, the signal input can be connected to the capacitive current source. The signal input can, for example, have one or more of the following connection devices for connecting to the capacitive current source: clamp connectors, crimp connectors, plug connectors, screw connectors, solder pads, solder joints, and high-frequency connectors.

[0016] It is also conceivable for the signal input of the amplifier circuit according to the invention to be connected directly to the capacitive current source. In this case, the amplifier circuit according to the invention can form a single unit with the capacitive current source. For example, the amplifier circuit according to the invention can be integrated into the capacitive current source.

[0017] A connection with a node K x, where x is a natural number, describes in the context of the invention a direct or indirect electrical connection with the node K x , preferably a direct electrical connection. A node, in the sense of the invention, describes a connection point of at least two conductor paths of the amplifier circuit. A direct connection to the node K x is an electrically conductive connection with node K x , which does not include any additional components or diversions via other nodes. An indirect connection to node K x is an electrically conductive connection with node K x, which comprises at least one electrical and / or electronic component or one or more component groups. A component group, within the meaning of the invention, is a group of components comprising at least two or more electrical and / or electronic components. Additionally or alternatively, the indirect connection can also include a rerouting via one or more additional nodes.

[0018] An input stage in the sense of the invention is a component or a group of components that is arranged downstream of the signal input and upstream of a subsequent stage, for example the amplifier cascade. The input stage is designed and configured such that an input signal, i.e. a current signal from the capacitive sensor that is present at the signal input or node Ki, is amplified. The amplification is at least 3-fold, preferably at least 5-fold, more preferably at least 10-fold. Additionally or alternatively, the amplification can be a maximum of 25-fold, preferably a maximum of 20-fold. Amplification in the range of 5-fold to 10-fold is particularly preferred. In this range, good amplification with optimally low noise is typically achieved. The signal output by the input stage at the output of the input stage can be inverted or non-inverted.Amplification is preferably achieved by linear control, preferably using a linear amplifier. Alternatively, amplification can also be achieved by utilizing a favorable, nonlinear characteristic of a corresponding amplifying component.

[0019] The input stage is configured to provide a high-impedance input resistor at the input of the input stage and, at the same time, to provide a stable and load-independent voltage at the output of the input stage. Preferably, the input stage is very low-noise. A low-noise input stage, in the sense of the invention, is an input stage with a spectral noise density of less than 5 nV. A low-noise input stage results from minimizing the voltage and current noise U associated with the input of the input stage. n or I n .

[0020] To achieve this, the input stage is preferably constructed discretely, i.e., consisting of at least two semiconductor components configured to control electrical voltages and / or currents. For example, the input stage can be constructed of transistors. The transistors can be either bipolar transistors or field-effect transistors, or a combination of at least one bipolar transistor and at least one field-effect transistor.

[0021] Preferably, the discrete design of the input stage comprises a component or group of components that provides amplification of the input signal and a component or group of components that lowers or adjusts the impedance of the input stage's output. Thus, the input stage provides a voltage at its output whose value is independent of the load from the downstream circuit.

[0022] Preferably, the discrete structure comprises at least one low-pass filter, also called a low-pass filter, for noise suppression of the supply voltage of the input stage.

[0023] An amplifier cascade, in the sense of the invention, is a series connection or chain of at least two amplifiers, i.e., at least a first amplifier with a first gain factor GQ3 and a second amplifier with a second gain factor GQ4, which amplify an input signal. The amplifier cascade is arranged downstream of the input stage.

[0024] The amplifier cascade may comprise two, three, four, or more amplifiers. The amplifier cascade preferably comprises two amplifiers.

[0025] A high signal amplification by the amplifier cascade describes in the context of the invention an intrinsic amplification of an input signal by at least a factor of 10 4. A low phase shift of the entire amplifier cascade, including the input stage, describes a phase shift of a maximum of -160° in the sense of the invention. A wide frequency range describes a frequency bandwidth of at least 500 Hz, preferably at least 1 kHz, more preferably at least 10 kHz, particularly preferably at least 100 kHz. A high signal amplification by a factor of 10 8 at a frequency bandwidth of 10 kHz is advantageous. Signal amplification by a factor of 10 is particularly advantageous. 9 at a frequency bandwidth of 10 kHz or a signal amplification by a factor of 10 8 with a frequency bandwidth of 100 kHz.

[0026] The amplification can be in open-loop gain or closed-loop gain. Amplification is preferably in open-loop gain. Preferably, the first amplifier is a linear controller that controls the function of the amplifier circuit via the feedback network. Preferably, the second amplifier has a low gain G Q4. The gain can be fixed or adjustable. In this case, the second amplifier can amplify the output signal of the first amplifier with a larger bandwidth and reduced phase shift. In addition, with a given negative feedback, it can provide a larger frequency range with a small input impedance. The first and second amplifiers and any further downstream amplifiers can be inverting, non-inverting, or a combination of inverting and non-inverting amplifiers. Preferably, the first and second amplifiers are non-inverting.

[0027] Preferably, the first gain factor G Q3 is much larger than the second gain factor GQ4: GQ3 » GQ4. For example, the ratio G Q3 / GQ4 may be at least 10 3 , preferably 10 4 , more preferably 10 5 be.

[0028] A feedback network, in the sense of the invention, describes a component group comprising at least one high-ohm feedback resistor Rfb for providing negative feedback to the configuration consisting of the input stage and amplifier cascade. This resistor defines the transimpedance gain as the ratio of the output voltage to its input current. A high-ohm feedback resistor with a transimpedance gain of 10 GV / A in the sense of the invention has an ohmic resistance of 10 GΩ.

[0029] To amplify the very low currents of capacitive current sources, preferably pyroelectric sensors, a very high signal amplification is required. The amplification factor should preferably be a factor of 10 6 Particularly advantageous is a signal amplification by a factor of 10 9 at a frequency bandwidth of 10 kHz or a signal amplification by a factor of 10 8 at a frequency bandwidth of 100 kHz. In other words, the amplifier circuit according to the invention can have a TIA gain bandwidth product of more than 10 TV / AHz.

[0030] For the noise characteristics of the amplifier circuit according to the invention, only the true transimpedance part of the amplifier circuit according to the invention is relevant. Downstream voltage amplification cannot improve the noise, even if it is a voltage filter circuit. With a downstream voltage amplification or voltage filter circuit, only the noise bandwidth can be changed, but not the noise density, which is important for the quality of the amplified signal.

[0031] In order to achieve an optimal signal-to-noise ratio of the amplified signal, it is important that the input current noise of the amplifier circuit is as low as possible for a given high amplification by the amplifier circuit according to the invention. The input current, including the input current noise, is converted into a voltage by the amplifier circuit according to the invention. The input current noise is significantly influenced by the selected feedback, i.e., by the Johnson-Nyquist noise of the feedback resistor. Preferably, the input current noise at room temperature with an amplification of 500 MV / A is fA fA approximately 10 -=, more preferably 5.8 -= . Further preferably, the input current noise at room temperature with an amplification of 5 GV / A is approximately 4-==, particularly preferably Hz

[0032] . " fA at about 1.8-== .

[0033] VHz

[0034] The amplifier circuit according to the invention is preferably a linear amplifier circuit. This is achieved by the negative feedback of the input stage and the amplifier cascade with the linear feedback network. For this reason, the feedback network is connected to the nodes K1 and K4.

[0035] The feedback network preferably comprises a compensation circuit configured to compensate for any potentially present undesired parasitic capacitance of the high-ohm feedback resistor. In other words, the compensation circuit is configured to minimize, preferably eliminate, a frequency dependence of the feedback network. For example, the compensation circuit may comprise at least one capacitor that advantageously interacts with the high-ohm feedback resistor. Additionally or alternatively, the compensation circuit may comprise a bandpass filter, for example, a high-pass and / or low-pass filter.

[0036] Preferably, the compensation circuit comprises electronic or electrical components whose ohmic resistance or capacitance can be variably adjusted. This allows the compensation circuit to be precisely tuned to the parasitic capacitance of the ohmic resistor in such a way that the parasitic capacitance is balanced and the frequency dependence of the feedback network (almost) disappears.

[0037] Within the scope of the invention, a signal output describes a hardware interface of the amplifier circuit at which the broadband, low-noise amplified output signal can be output. For further processing and use of the output signal, the signal output can be connected to external devices. For example, the signal output can be connected to an analog-to-digital converter and a downstream measuring and analysis device, such as a measurement computer. For connection to external devices, the signal output can, for example, have one or more of the following connection devices: clamp connectors, crimp connectors, plug-in connectors, screw connectors, solder pads, solder joints, or high-frequency connectors.

[0038] The amplifier circuit according to the invention has the surprising advantage of enabling extremely low-noise and simultaneously very high amplification with a wide frequency bandwidth. As a result, the amplifier circuit according to the invention provides an optimal signal-to-noise ratio. For example, gains G of, for example, G « 10 9 V / A and bandwidths greater than 10 kHz, especially up to 100 kHz, are possible, which is significantly faster than conventional commercially available amplifier circuits at this amplification. The amplifier circuit is robust against increases in input capacitance at the signal input. This allows the use of thinner and therefore more sensitive pyroelectric sensor elements even at high amplifications.

[0039] In a preferred embodiment according to the first aspect of the invention, the input stage comprises a junction field effect transistor and a bipolar transistor, wherein a drain terminal of the junction field effect transistor is connected to a base terminal of the bipolar transistor, wherein the junction field effect transistor is connected as a source circuit, wherein the bipolar transistor is connected as an emitter follower, wherein the first and second amplifiers of the amplifier cascade are operational amplifiers, wherein the first operational amplifier has a voltage gain of more than 10 4 wherein the second operational amplifier has a voltage gain of maximum 10 3 wherein the feedback network comprises a high-ohm feedback resistor with a parallel capacitance, wherein the feedback network comprises a series-connected low-pass filter.

[0040] Junction field-effect transistors are known in the art. Junction field-effect transistors typically have three terminals: source, gate, and drain. Bipolar transistors are known in the art. Bipolar transistors typically have three terminals: collector, base, and emitter.

[0041] Preferably, the gate terminal of the junction field-effect transistor is connected to node K1. Preferably, the emitter terminal of the bipolar transistor is connected to node K2.

[0042] The source circuit of junction field-effect transistors and the emitter-follower circuit of bipolar transistors are well known in the art.

[0043] The source circuit is configured to invertingly amplify the input signal at the junction field-effect transistor by at least a factor of 3, preferably by a factor of 5 to 20. However, it is also conceivable that the input signal is amplified non-invertingly.

[0044] The emitter follower is configured to reduce the output impedance of the input stage without additional voltage amplification. This allows for stable voltage transfer at node K2, the value of which is independent of the load from the downstream circuit. The emitter follower compensates for an offset potential at the drain output of the field-effect transistor with a fixed voltage drop between the base and emitter of the bipolar transistor. This offset correction reduces the need for the two operational amplifiers in the amplifier cascade following the input stage to counteract their input signal to zero. In a real operational amplifier, the common-mode input voltage should be close to zero. Without compensation, the signal level shift caused by the field-effect transistor would saturate the downstream high-gain amplifier cascade or limit the dynamic range of the gain.Preferably, the drain terminal of the junction field-effect transistor and / or the collector terminal and emitter terminal of the bipolar transistor each have a resistor or a series circuit of at least two resistors. The optimal operating point of the input stage can be adjusted using suitable resistors or series circuits of resistors at the drain terminal of the junction field-effect transistor and / or at the collector terminal and emitter terminal of the bipolar transistor. An optimal operating point of the input stage is achieved with maximum gain, high bandwidth, and low output signal noise.

[0045] Preferably, at least the series connection of resistors has a parallel low-pass filter at the drain terminal of the junction field-effect transistor and / or at the collector terminal and emitter terminal of the bipolar transistor. The low-pass filter can advantageously suppress noise from the supply voltage. Further preferably, each series connection of resistors has a parallel low-pass filter at the drain terminal of the junction field-effect transistor and at the collector terminal and emitter terminal of the bipolar transistor.

[0046] Preferably, the first operational amplifier has a voltage gain GQ3 of more than 10 6 , even more preferably more than 10 7 on.

[0047] Preferably, the second operational amplifier has a voltage gain GQ4 of maximum 1000, more preferably maximum 400, even more preferably maximum 250.

[0048] The capacitance Cs connected in parallel with the high-impedance feedback resistor and the low-pass filter connected in series with it, for example, consisting of a resistance Rs and a capacitance Ce, form a possible embodiment of the aforementioned compensation circuit. The compensation circuit is designed to reduce an unavoidable parasitic capacitance of the high-impedance feedback resistor to zero or close to zero. Close to zero in this case describes a parasitic capacitance of less than 0.5 pF, preferably less than 0.1 pF.

[0049] Preferably, the low-pass filter is connected to node K4. Preferably, the parallel circuit comprising high-impedance feedback resistor R7 and capacitor C5 is connected at least to node Ki.

[0050] The parasitic capacitance of the high-impedance feedback resistor R7 creates a pole in the control loop that can be compensated by the low-pass filter consisting of Rs and Ce. This can be demonstrated by decomposing the compensation network into its two components: The transfer function of an inverting transimpedance amplifier with the feedback IF consisting of R7 and C5 alone is: V Kf = - lin

[0051] V Kf is the voltage at node Kf. The directly upstream low-pass filter consisting of Rs and Ce alone has the following transfer function: with the potential V K4 at node K4. By substituting, the total transfer function of the compensation circuit is:

[0052] VK4lin

[0053] This clearly shows that the transfer function loses its frequency dependence when R8C6 = R7C5. A suitable setting of the low-pass filter can therefore compensate for the parasitic capacitance of the high-impedance feedback resistor Rs.

[0054] Preferably, the high-impedance feedback resistor Rs is a combination of fixed resistor and potentiometer.

[0055] The amplifier circuit according to the invention has the advantage that it is constructed from easily available, inexpensive and reliable electronic components.

[0056] In addition, the special arrangement of the various component groups, in particular the nesting of the input stage, amplifier cascade and feedback network, achieves a performance not previously achieved in the state of the art, consisting of broadband and low-noise high amplification of the signal of the capacitive current source, preferably of the pyroelectric sensor.

[0057] Junction field-effect transistors have very low input current noise. This keeps the signal noise at the signal output of the amplifier circuit according to the invention low and improves the signal-to-noise ratio.

[0058] This embodiment therefore offers in particular a very cost-effective amplifier circuit with the advantages of a broadband and low-noise high amplification already mentioned above.

[0059] In a preferred embodiment according to the first aspect of the invention, the amplifier circuit additionally comprises an output stage, wherein the output stage is connected to the node K4 at an input of the output stage and is connected to the signal output in the node K5 at an output of the output stage, wherein the output stage is configured to amplify a voltage at the input of the output stage by up to 20 times, wherein the output stage is configured to filter DC voltage interference at the input of the output stage and to adapt a signal level at the output of the output stage to Ks.

[0060] An output stage in the sense of the invention is a component group downstream of the amplifier cascade, which is designed to amplify a voltage at the input of the output stage, to filter DC voltage interference at the input of the output stage and to adapt a signal level of the voltage at the output of the output stage.

[0061] Preferably, the voltage amplification at the input of the output stage is a maximum of 15 times, more preferably a maximum of 10 times. The amplification can be inverting or non-inverting. The amplification can be achieved, for example, using one or more of the following amplifying components: field-effect transistor, unipolar transistor, bipolar transistor, operational amplifier.

[0062] Capacitive current sources, preferably pyroelectric sensors, typically measure temperature changes. The voltage signal derived (and amplified) from the current signal of the capacitive current source is therefore also a time-varying signal. DC components in the voltage signal, e.g., from suboptimal offset correction in the input stage, therefore provide no information about the measurement and are therefore undesirable. Filtering DC noise at the input of the output stage thus significantly improves signal quality.

[0063] DC interference can be filtered, for example, using a frequency-dependent circuit, e.g., a bandpass filter, or at least one frequency-dependent component, e.g., a capacitor. The frequency-dependent circuit or the at least one frequency-dependent component is preferably connected downstream of the input of the output stage and preferably upstream of the amplifying component.

[0064] The signal level can be adjusted using state-of-the-art solutions, for example, using an amplifier circuit based on an operational amplifier.

[0065] The output stage and any associated amplification have the advantage that the signal level can be optimally adapted to the dynamic range of the typically subsequent evaluation electronics (e.g., an analog-to-digital converter). Furthermore, the additional boost in the signal level makes any interference less significant relative to the signal.

[0066] This results in an optimal signal-to-noise ratio in the downstream measurement electronics. In a preferred embodiment according to the first aspect of the invention, the output stage comprises an inverting bandpass amplifier, with one input of the bandpass amplifier being AC-coupled.

[0067] Preferably, the input of the output stage amplifier is connected to node K4 via a series-connected capacitor. Preferably, the output of the output stage amplifier is connected to node K5 and the signal output. Alternatively, the bandpass amplifier can also be designed as non-inverting.

[0068] The output stage can be constructed from readily available, inexpensive, and reliable electronic components. This design thus offers a very cost-effective amplifier circuit with the aforementioned advantages of broadband, low-noise, high gain.

[0069] In a preferred embodiment according to the first aspect of the invention, the amplifier circuit additionally comprises an amplitude limiter, wherein the amplitude limiter is connected to the node K4 or K5 at an input of the amplitude limiter and to the node K3 at an output of the amplitude limiter, wherein the amplitude limiter is configured to limit the amplitude of the output signal at the signal output when a threshold value at the node K4 or node K5 is exceeded.

[0070] In the sense of the invention, an amplitude limiter is a group of components that is designed to limit the amplitude of the output signal at the signal output when a threshold value is exceeded.

[0071] Amplitude limitation is achieved through a nonlinear transfer function. Nonlinear amplitude limitation can be achieved, for example, using Zener diodes (Z-diodes). Alternatively, other components with a nonlinear transfer function can also be used.

[0072] This nonlinear feedback increases the stability of the circuit. Especially when amplifying the signal of a pyroelectric sensor, too much light can easily be applied to the sensor, which would saturate the amplifier stage. This prevents overloading at the signal output of the amplifier circuit according to the invention.

[0073] The following synergies arise, particularly in the interaction with the output stage: The nonlinear negative feedback is implemented at node K3 instead of node K1 in order to reduce or prevent an increase in the noise of the output signal at the signal output at node K5 due to a possible capacitance of at least one component in the amplitude limiter. To ensure controlled negative feedback at node K3, the polarity of the gain along a closed feedback loop must be inverting (negative) overall. For this purpose, the output stage is preferably designed to be inverting, and the amplitude limiter is arranged along the connection between nodes K5 and K3.

[0074] Another reason for placing the amplitude limiter after the first amplifier in the amplifier cascade is that the amplifiers in the amplifier cascade are very sensitive to additional capacitance and reduced input impedance. Nonlinear components such as diodes, in particular, would add significant noise.

[0075] Finally, oscillation of the amplifier cascade or overdriving of the amplifier cascade can be reduced or completely prevented by the interaction of the amplitude limitation with the output stage.

[0076] In a preferred embodiment according to the first aspect of the invention, the amplitude limitation comprises two Z-diodes connected in series with opposite polarity.

[0077] Zener diodes are known in the art. Zener diodes have a cathode and an anode. When two Zener diodes are connected in series with opposite polarity, either the cathode (or the anode) of each Zener diode is preferably connected to node K3 and the signal output at node K5. Both anodes (or both cathodes) of the two Zener diodes are connected to each other. Negative feedback occurs when the voltage at K5 exceeds a threshold. This threshold depends on the breakdown voltage of the Zener diode connected to K5. The threshold can be adjusted by selecting the appropriate Zener diodes.

[0078] Preferably, the anodes of the Zener diodes are connected to ground via a series circuit consisting of an ohmic resistor and a capacitor.

[0079] Zener diodes are simple, reliable, and inexpensive components. They are therefore particularly well-suited for amplitude limiting. In addition, the same advantages for amplitude limiting that were already mentioned in the previous paragraphs apply in this context. To avoid unnecessary redundancy, reference is made to the above statements and they will not be repeated again.

[0080] In a preferred embodiment according to the first aspect of the invention, the amplifier circuit has a frequency response compensation, wherein the frequency response compensation is connected to node K3 and node Ki, wherein the frequency response compensation is configured to reduce a tendency of the amplifier cascade to oscillate. Within the meaning of the invention, a frequency response compensation is a component or group of components that reduces an undesired tendency of the entire amplifier circuit to oscillate. Within the scope of the invention, an oscillation tendency of the amplifier circuit describes a tendency of the amplifier circuit to oscillate undesirably.

[0081] For example, the frequency response compensation may comprise frequency-dependent components or frequency-dependent component groups. The frequency response compensation may, for example, comprise at least one or more of the following frequency-dependent components or frequency-dependent component groups: a series-connected capacitor, a parallel-connected capacitor, a bandpass filter, for example, a high-pass filter and / or a low-pass filter.

[0082] The frequency-dependent components or frequency-dependent component groups serve to stabilize the amplifier cascade through internal frequency compensation. If the phase of the signal rotates by 180° when passing through a feedback loop of the amplifier circuit according to the invention (loop gain close to -1), the negative feedback can become positive feedback, and the amplifier cascade can tend to oscillate. This behavior can be compensated for by the frequency-dependent components or frequency-dependent component groups. The frequency-dependent components or frequency-dependent component groups are preferably dimensioned such that the circuit stability of the amplifier cascade and the entire amplifier circuit according to the invention is ensured, but at the same time the bandwidth is not excessively restricted.

[0083] Frequency response compensation thus makes it possible to increase the stability of the amplifier circuit according to the invention at high frequencies. Oscillation of the amplifier circuit is thus reduced or avoided, and the performance of the amplifier circuit is improved. In particular, frequency response compensation enables a flat transfer, i.e., a linear transfer function, to be achieved at high frequencies and high gains. The ripple of the transfer function is reduced, preferably to less than 5%, particularly preferably to less than 3% of the maximum amplitude. This is particularly necessary for making quantitative statements about the signal amplitude in broadband measurements.

[0084] In a preferred embodiment according to the first aspect of the invention, the frequency response compensation comprises negative feedback via a capacitor.

[0085] The negative feedback capacitor is preferably connected to the nodes Ki and K3.

[0086] The negative feedback capacitor serves to stabilize the entire amplifier circuit through internal frequency compensation. If the phase of the signal rotates by 180° when passing through a feedback loop of the amplifier circuit according to the invention (loop gain close to -1), the negative feedback can become positive feedback, and the amplifier circuit can tend to oscillate. This behavior can be compensated for by the negative feedback capacitor. The value of the capacitor's capacitance is preferably dimensioned such that the circuit stability of the amplifier cascade and the entire amplifier circuit according to the invention is ensured, but at the same time the bandwidth is not excessively restricted.

[0087] Capacitors are simple, reliable, and inexpensive components. Implementing frequency response compensation with a negative feedback capacitor is therefore a simple and cost-effective solution.

[0088] In addition, the same advantages of frequency response compensation apply in this context as already mentioned in the previous paragraphs. To avoid unnecessary redundancy, reference is made to the above statements and they will not be repeated again.

[0089] In a preferred embodiment according to the first aspect of the invention, the input stage comprises a component with a negative capacitance.

[0090] For the purposes of the invention, a component with a negative capacitance is one in which a reduction in the applied voltage results in an increase in the component's charge. For this purpose, the component can comprise a material with a negative capacitance. Initial experimental evidence for such materials is provided, for example, by M. Hoffmann, S. Slesazeck, and T. Mikolajick, "Progress and future prospects of negative capacitance electronics: A materials perspective," APL Mater. 9, 020902 (2021) and AK Yadav, KX Nguyen, Z. Hong, et al. "Spatially resolved steady-state negative capacitance," Nature 565, 468 (2019).

[0091] Examples of materials in which a negative capacitance can develop are ferroelectric materials such as HfCh or lead zirconate titanate or heterostructures made of these ferroelectric and dielectric materials such as strontium titanate.

[0092] Preferably, the component with negative capacitance is combined with the signal input or input stage in such a way that the negative capacitance of the component balances the normal, positive capacitance of the capacitive current source, preferably the pyroelectric sensor. Alternatively or additionally, the component with negative capacitance can also be included in the feedback network, where it compensates for the parasitic capacitance of the high-ohm feedback resistor.

[0093] The use of such a component significantly reduces noise and thus improves the signal-to-noise ratio when measuring the signal from the capacitive current source, preferably the pyroelectric sensor. In a preferred embodiment according to the first aspect of the invention, the amplifier circuit or at least a portion of the amplifier circuit is mounted on a circuit board, with at least one electronic component of the amplifier circuit being soldered to the circuit board using solder pads, with a region of the circuit board below the at least one electronic component being removed outside the solder pads.

[0094] For the purposes of the invention, a printed circuit board is any substrate suitable for the permanent mechanical attachment and electrical connection of the amplifier circuit according to the invention or parts of the amplifier circuit according to the invention. Examples of printed circuit boards are flexible or rigid printed circuit boards. Printed circuit boards can, for example, comprise fiber-reinforced plastic or laminated paper, as well as metallic coatings and / or conductor tracks.

[0095] In the sense of the invention, solder pads describe areas on a printed circuit board that are typically provided with metallic coatings and are suitable for a solder connection using solder, for example for connecting an electronic component.

[0096] In the context of the invention, removing a region of the circuit board describes a region of the circuit board within which the circuit board material has been partially or completely removed. In other words, there is no circuit board material (complete removal) or less circuit board material (partial removal) present in this removed region. The removal of the circuit board material can be performed, for example, by machining processes, e.g., milling, drilling, or sawing. Alternatively, corresponding recesses without circuit board material can also be provided during the manufacture of the circuit board material.

[0097] Every component, e.g., a resistor, has a parasitic capacitance that forms between the component's terminals and between the component and its environment. Especially with a resistor, the parasitic capacitance increases with component dimensions. In particular, the high-ohm feedback resistor and the transistors of the input stage are sensitive to changes in environmental conditions. Even the slightest deviations in the surrounding material of the circuit board, a protective coating applied to the components and the circuit board, or flux residues from the soldering process can generate leakage currents on the order of the current in the negative feedback circuits of the amplifier circuit according to the invention, which can significantly adversely affect the performance of the amplifier electronics according to the invention.In particular, these effects cause undesirably large quality fluctuations within a production batch during manufacturing, which require rework or generate scrap, thus increasing costs. Preferably, the circuit board is removed centrally below a component, leaving only the area of ​​the solder pads. A component connected to the solder pads can, once the component's connections are firmly soldered, span the gap in the circuit board created by the removed area, like a bridge. In this case, any contamination beneath the component is no longer possible.

[0098] This significantly increases the reproducibility of the amplifier circuit's performance during production. Furthermore, costs for any subsequent improvements to the amplifier circuit's performance after production can be reduced or eliminated.

[0099] In a preferred embodiment according to the first aspect of the invention, the at least one electronic component is one or more electronic components of the feedback network or the entire feedback network.

[0100] The feedback network is particularly important for the function of the amplifier circuit according to the invention and, at the same time, is very sensitive to parasitic capacitances. Reducing parasitic capacitances by removing the circuit board material thus leads to a significant improvement in the performance of the amplifier circuit according to the invention.

[0101] In a preferred embodiment according to the first aspect of the invention, the at least one electronic component is one or more electronic components of the input stage, preferably the field effect transistor and / or the bipolar transistor.

[0102] The input stage is particularly important for the function of the amplifier circuit according to the invention and, at the same time, is very sensitive to parasitic capacitances. Reducing parasitic capacitances by removing the circuit board material thus leads to a significant improvement in the performance of the amplifier circuit according to the invention.

[0103] The amplifier circuit according to the invention according to one of claims 1 to 19 can preferably be used to measure a current signal of a capacitive current source, preferably a pyroelectric sensor.

[0104] The amplifier circuit according to the invention according to one of claims 1 to 19 can preferably be used to measure a current signal of a capacitive current source, preferably a pyroelectric sensor, in an infrared spectrometer, preferably an FTIR spectrometer (Fourier transform infrared spectrometer).

[0105] The FTIR spectrometer (Fourier Transform Infrared Spectrometer) can comprise the following components: an infrared radiation source, an interferometer with at least one arm variable in length, a reference laser, a measuring cell with a sample interface, preferably an ATR crystal.Attenuated Total Reflection (ATR) that can be brought into contact with a sample, an infrared detector, a control system that is configured to change the length of at least one arm of the interferometer, and a mirror arrangement outside the interferometer with at least two mirrors, each with a reflective surface and a base body that comprises the reflective surface, wherein the mirror arrangement is at least configured to direct a light beam from the interferometer to the sample interface and to direct the light beam from the sample interface to the infrared detector, wherein the base body of at least one mirror or all of the mirrors of the mirror arrangement is / are made of a plastic material and / or 3D printed metal, or the base body of at least one mirror or all of the mirrors comprises / comprises plastic material and / or 3D printed metal.

[0106] With the help of the ATR crystal, an evanescent wave can couple into the sample material or sample in contact with the ATR crystal. This effect is also called the optical tunneling effect. The remaining light carries information about the interaction with the sample, is guided back out of the ATR crystal by total internal reflection, and can then be guided, for example, by reflection to an infrared detector.

[0107] The combination of all the aforementioned preferred embodiments is in turn a preferred embodiment and describes a four-stage amplifier circuit, ie an amplifier circuit comprising four amplifier stages, with three feedback loops.

[0108] The first amplifier stage is formed by the input stage. The second and third amplifier stages are formed by the first and second amplifiers of the amplifier cascade. The fourth amplifier stage is formed by the output stage.

[0109] The first feedback loop is formed by the feedback network. The second feedback loop is formed by the amplitude limiter. The third feedback loop is formed by the internal frequency compensation.

[0110] Preferably, the input stage and the output stage are each an inverting amplifier stage. Preferably, the first and second amplifiers of the amplifier cascade are each or form a non-inverting amplifier stage. The interaction of inverting and non-inverting amplifier stages enables separate feedback loops, each including only the first or last inverting stage and otherwise only non-inverting stages. The condition for negative feedback (an odd number of inverting amplifier stages) is thus met for all feedback loops. The first feedback loop contains a linear component network for high gains with capacitance compensation. The second feedback loop generates the nonlinear amplitude limiting. The third feedback loop serves to compensate the frequency response of the entire circuit.In a preferred embodiment according to the first aspect of the invention, the amplifier circuit additionally comprises a decoupler, wherein the decoupler is connected to the node K4 and the input of the output stage, wherein the decoupler is configured to decouple an output of the amplifier cascade from the output stage.

[0111] In a preferred embodiment according to the first aspect of the invention, the decoupler comprises an impedance converter, preferably a non-inverting impedance converter or a combination of two inverting impedance converters, or wherein the decoupler is an impedance converter, preferably a non-inverting impedance converter or a combination of two inverting impedance converters. Particularly preferably, the decoupler is a non-inverting impedance converter or comprises a non-inverting impedance converter.

[0112] Impedance converters are technically simple, cost-effective, and reliable components or component groups that can achieve the desired decoupling. Non-inverting impedance converters have the advantage of typically having a very high-impedance input, which is particularly advantageous for this circuit.

[0113] In a preferred embodiment according to the first aspect of the invention, the impedance converter comprises at least one or more of the following components or is one of the following components: MOSFET, bipolar transistor, operational amplifier, operational amplifier with high input resistance and low inherent noise, operational amplifier with high input resistance and low inherent noise, wherein the output of the operational amplifier is fed back directly or indirectly to its inverting input.

[0114] In a particularly preferred embodiment according to the first aspect of the invention, the impedance converter comprises at least one operational amplifier with high input resistance and low inherent noise, wherein the output of the operational amplifier is directly fed back to the inverting input.

[0115] Within the scope of the invention, decoupling of the output of the amplifier cascade from the input of the output stage by the decoupler describes in particular a decoupling of the frequency dependence of the output stage from the upstream amplifier cascade. This frequency decoupling can be complete or only partial. With complete frequency decoupling, the ideal frequency dependence of the output stage is not distorted and the frequency dependence of the amplifier cascade also remains unaffected. With partial frequency decoupling, the output stage loads the upstream amplifier cascade and distorts the frequency dependence of the output stage due to the frequency-dependent output impedance of the amplifier cascade. This distortion becomes more pronounced the greater the feedback resistance (i.e. the lower the feedback strength) of the amplifier cascade. Preferably, the decoupling is complete or almost complete.

[0116] MOSFETs, bipolar transistors, and operational amplifiers are inexpensive, reliable, and technically simple components suitable for use as or in decouplers. Operational amplifiers with high input resistance and low inherent noise are preferred components for use in decouplers because they can provide particularly strong decoupling, up to or almost complete decoupling. At the same time, due to their low inherent noise, they contribute little or no corruption to the signal amplified in the amplifier circuit according to the invention. Operational amplifiers with high input resistance and low inherent noise, where the output of the operational amplifier is directly fed back to the inverting input, are particularly preferred because they have the same advantageous properties as operational amplifiers with high input resistance and low inherent noise and additionally improve decoupling.

[0117] The decoupler is preferably connected to node K4 and the input of the output stage. In particular, the decoupler can be connected to node K4 and the input of the output stage directly or indirectly. A direct connection in this context describes the connection "K4 - decoupler - output stage input." An indirect connection can, for example, involve additional electronic components between K4 and the decoupler and / or the decoupler and the input of the output stage.

[0118] The provision of the decoupler or non-inverting impedance converter provides frequency decoupling between the amplifier cascade and the input of the output stage. This results in a significantly flatter transfer function between the amplifier cascade and the output stage. As a result, the frequency dependence of the output of the amplifier circuit according to the invention is reduced or improved.

[0119] At the same time, the decoupler improves the gain of the amplifier circuit according to the invention. In particular, the input signal of the amplifier circuit according to the invention can be amplified with a flat transfer function, for example, at a gain of 50 MV / A up to 360 kHz.

[0120] In a preferred embodiment according to the first aspect of the invention, the amplifier circuit additionally comprises an intermediate load, wherein the intermediate load is connected to the output of the first amplifier and the input of the second amplifier of the amplifier cascade in the node K3, wherein the intermediate load is configured to minimize natural oscillations of the amplifier circuit.

[0121] For the purposes of the invention, the connection of the intermediate load to the output of the first amplifier of the amplifier cascade at node K3 means that the intermediate load can be connected directly or indirectly, e.g., via one or more interposed electronic components, to the output of the first amplifier at node K3. A preferred electronic component for an indirect connection can be an ohmic resistor.

[0122] For the purposes of the invention, the connection of the intermediate load to the input of the second amplifier of the amplifier cascade at node K3 means that the intermediate load can be connected directly or indirectly, e.g., via one or more interposed electronic components, to the input of the second amplifier at node K3. A preferred electronic component for an indirect connection can be an ohmic resistor.

[0123] In a preferred embodiment according to the first aspect of the invention, the intermediate load is configured to minimize natural oscillations by a phase correction.

[0124] In a preferred embodiment according to the first aspect of the invention, the output of the first amplifier is connected to node K3 via a first ohmic resistor (R17), with node K3 being connected to ground via a second ohmic resistor (Ris). The first resistor (R17) and the second ohmic resistor (Ris) preferably form a voltage divider, from which a partial voltage can be tapped at node K3. The voltage divider advantageously limits excessive loading, particularly on the upstream amplifier stage or the upstream operational amplifier.

[0125] In a preferred embodiment according to the first aspect of the invention, the intermediate load has or consists of at least one or more of the following component combinations: an inductance and / or capacitance connected in series to ground and at least one ohmic resistor, preferably the second ohmic resistor; an inductance and / or capacitance connected in parallel to ground and at least one ohmic resistor. Particularly preferably, the intermediate load has or consists of an inductance or capacitance connected in series to ground and at least the second ohmic resistor. Further preferably, the intermediate load has or consists of an inductance connected in series to ground and at least the second ohmic resistor, or consists of an inductance connected in series to ground and at least the second ohmic resistor R15.Within the scope of the invention, an intermediate load describes any component or combination of components that is or are suitable for minimizing the natural oscillation of the amplifier circuit according to the invention. The intermediate load is preferably configured to introduce a phase correction that changes the frequency response such that the amplifier circuit according to the invention is more robust against natural oscillations. Within the scope of the invention, robustness against natural oscillations means that the feedback circuit, in particular the feedback network and / or the frequency response compensation, meets the Nyquist stability criterion. This means that the phase shift of the feedback is so small that the desired negative feedback does not unintentionally become positive feedback.

[0126] The aforementioned component combinations are simple in design and robust in use. Despite their simplicity, they also have the surprising effect of significantly improving the advantageous properties of the amplifier circuit according to the invention.

[0127] Particularly preferably, the inductance of the intermediate load in the component combination of inductance and at least one ohmic resistor, preferably the first and second ohmic resistors, has a value in the range from 1 pH to 20 pH, preferably in the range from 1 pH to 5 pH. Particularly preferably, the first and second ohmic resistors in the component combination of the intermediate load comprising inductance and ohmic resistors have a value in the range from 10 ohms to 100 ohms, preferably in the range from 10 ohms to 20 ohms. Preferably, the first and second resistors are of equal size. These aforementioned values ​​surprisingly show particularly good results in minimizing unwanted natural oscillations.

[0128] The provision of the intermediate load advantageously stabilizes the amplifier circuit according to the invention in the event of significant changes in the input capacitance of the capacitive current source. In particular, this enables the use of different capacitive current sources and / or pyroelectric sensors. This makes the circuit according to the invention more universal and usable without the need for costly or technically complex adaptations to the circuit.

[0129] The object of the invention is achieved in the second aspect of the invention by the sensor system having the features of claim 20. The sensor system according to the invention according to claim 20 comprises a capacitive current source and an amplifier circuit according to one of claims 1 to 19.

[0130] The statements made in connection with the first aspect of the invention also apply in connection with the sensor system according to the second aspect of the invention. To avoid unnecessary redundancies, reference is made here to the statements made above and a repeat of them will be omitted.

[0131] The capacitive current source may preferably comprise a component with a negative capacitance.

[0132] For the purposes of the invention, a component with a negative capacitance is one in which a reduction in the applied voltage results in an increase in the component's charge. For this purpose, the component can comprise a material with a negative capacitance. Initial experimental evidence for such materials is provided, for example, by M. Hoffmann, S. Slesazeck, and T. Mikolajick, "Progress and future prospects of negative capacitance electronics: A materials perspective," APL Mater. 9, 020902 (2021) and AK Yadav, KX Nguyen, Z. Hong, et al. "Spatially resolved steady-state negative capacitance," Nature 565, 468 (2019).

[0133] Examples of materials with a negative capacitance can be ferroelectric materials such as HfCh or lead zirconate titanate or heterostructures of these ferroelectric and dielectric materials such as strontium titanate.

[0134] Preferably, the component with negative capacitance is connected to the capacitive current source, for example, at an output of the capacitive current source. Alternatively, the component with negative capacitance can be part of the capacitive current source, preferably the pyroelectric sensor. In both cases, the negative capacitance of the component can balance the normal, positive capacitance of the capacitive current source, preferably the pyroelectric sensor.

[0135] The use of such a component significantly reduces noise and thus improves the signal-to-noise ratio when measuring the signal of the capacitive current source, preferably the pyroelectric sensor.

[0136] In a preferred embodiment according to the second aspect of the invention, the capacitive current source is a pyroelectric sensor.

[0137] The pyroelectric sensor can contain, for example, lithium tantalate (LiTaOä) or triglycine sulfate (TGS).

[0138] Due to the special properties of the amplifier circuit according to the invention in the sensor system according to the invention, a particularly broadband and low-noise amplification of the signal of a pyroelectric sensor is possible.

[0139] In a preferred embodiment according to the second aspect of the invention, the pyroelectric sensor is plate-like and has a maximum thickness of 40 pm, preferably a maximum of 10 pm.

[0140] A pyroelectric sensor typically has a crystal made of a pyroelectric material.

[0141] In the context of this invention, the thickness of the pyroelectric sensor describes the thickness of the pyroelectric material, ie the thickness or average thickness of the pyroelectric crystal.

[0142] The temperature increase of the pyroelectric sensor, or the crystal, is directly proportional to the absorption A th of the radiation in the crystal and inversely proportional to its heat capacity c t h (optical-thermal conversion). The largest possible temperature change is made possible by a suitable, broadband absorbing coating of the crystal. In addition, a low heat capacity can be achieved by a small crystal volume or, for a given sensor area, by a small crystal thickness. The heating process of the sensor element is counteracted by cooling, e.g., through thermal conduction of the crystal holder. The poorest possible thermal conduction ensures the highest possible temperature difference ΔT between the irradiated and unirradiated crystal and therefore leads to a correspondingly high current signal. With increasing modulation frequency of the incident light Δc, ΔT decreases and ΔT oc applies.

[0143] Therefore, a low heat capacity (a thin crystal) is required for the fastest possible c th wThe thermal response of the sensor is advantageous. A pyroelectric sensor with a small thickness thus ensures the fastest possible thermal response and advantageous behavior of the pyroelectric sensor, provided that the increased capacitance of the sensor element due to the small thickness does not exceed the reasonable limits of the input capacitance of the subsequent amplifier circuit. Preferably, the thickness or average thickness of the crystal in the context of this invention is between 2 pm and 40 pm.

[0144] Another factor is the thermal-electric conversion: The temperature change of the pyroelectric crystal generates a surface charge Q proportional to the area A of the crystal and to the pyroelectric coefficient p, which describes the specific strength of the pyroelectric effect in a material: Q = p A AT

[0145] A temporal change in temperature thus generates a time-varying charge, i.e., an electric current I. This means that a pyroelectric sensor can only react to changes in radiation. In the stationary case, no current flows. At high frequencies, the pyroelectric current is independent of the frequency of the light excitation.

[0146] 1 constant. The following applies to the current: I oc — , so a low heat capacity (a thin crystal) Cth ​​is desirable at high frequencies, even for the largest possible current signal.

[0147] Furthermore, the reduced thickness of the pyroelectric sensor results in an increase in bandwidth, up to the electronic bandwidth of the amplifier circuit according to the invention. In this case, both the optical-thermal conversion and the thermal-electrical conversion are optimized due to the reduced heat capacity of the pyroelectric sensor.

[0148] The sensor system according to the invention according to one of claims 20 to 22 can preferably be used in and / or with an FTIR spectrometer (Fourier transform infrared spectrometer). In particular, the sensor system according to the invention can be part of or used in an FTIR spectrometer as described above.

[0149] It should be clarified that one or more of the preferred embodiments described above, as long as they are consistent, can be combined with one another and also represent preferred embodiments. In particular, one or more of the preferred embodiments of the various aspects of the invention described above, as long as they are consistent, can also be combined with one another and also represent preferred embodiments of the invention.

[0150] Preferred embodiments of the invention are explained and described in more detail below with reference to the accompanying drawings.

[0151] Fig. 1 is a schematic representation of a circuit of a first embodiment of the amplifier circuit according to the invention,

[0152] Fig. 2 is a circuit diagram of an embodiment of an input stage,

[0153] Fig. 3 is a circuit diagram of an embodiment of an amplifier cascade with feedback network,

[0154] Fig. 4 is a schematic representation of a circuit of a second embodiment of the amplifier circuit according to the invention,

[0155] Fig. 5 is a circuit diagram of an embodiment of an output stage,

[0156] Fig. 6 is a schematic representation of a circuit of a third embodiment of the amplifier circuit according to the invention,

[0157] Fig. 7 is a circuit diagram of an embodiment of an amplitude limitation,

[0158] Fig. 8 is a schematic diagram of a circuit of a fourth embodiment of the amplifier circuit according to the invention,

[0159] Fig. 9 is a circuit diagram of an embodiment of an amplifier cascade with feedback network and frequency compensation,

[0160] Fig. 10 is a circuit diagram of a fifth embodiment of the amplifier circuit according to the invention,

[0161] Fig. 11 a,b a representation of a component on a circuit board with a removed area,

[0162] Fig. 12 shows an embodiment of a sensor system according to the invention,

[0163] Fig. 13 a,b Representation of measurement results of the performance of the amplifier circuit according to the invention,

[0164] Fig. 14 a,b,c Representation of measurement results of the performance of the sensor system according to the invention,

[0165] Fig. 15 Representation of measurement results of the transmission behavior at different amplifications,

[0166] Fig. 16 is a schematic representation of a circuit of a sixth

[0167] Embodiment of the amplifier circuit according to the invention,

[0168] Fig. 17 is a schematic diagram of a circuit of a seventh embodiment of the amplifier circuit according to the invention,

[0169] Fig. 18 is a circuit diagram of an embodiment of an intermediate load,

[0170] Fig. 19 is a schematic diagram of a circuit of an eighth embodiment of the amplifier circuit according to the invention, Fig. 20 is a circuit diagram of a ninth embodiment of the amplifier circuit according to the invention, and

[0171] Fig. 21 shows a second embodiment of a sensor system according to the invention.

[0172] Figure 1 shows a schematic representation of a circuit of a first embodiment of the amplifier circuit 1a according to the invention. The amplifier circuit 1a is an amplifier circuit for broadband and low-noise amplification of a capacitive current source, preferably a pyroelectric sensor. The amplifier circuit 1a comprises a signal input 3, which is connectable to the capacitive current source at a node Ki, an input stage Al, wherein the input stage Al is connected to the node Ki at an input 7 of the input stage Al and has a node K2 at the output 9 of the input stage Al, wherein the input stage Al is configured to amplify an input voltage at least three times, wherein the input stage Al is configured to provide a high-impedance input resistor at the input 7 of the input stage Al, wherein the input stage Al is configured to provide a stable and load-independent voltage at the output 9 of the input stage Al.an amplifier cascade 11, wherein the amplifier cascade 11 has at least a first amplifier A2 and a second amplifier A3, each with an input 13 or 17 and an output 15 or 19, wherein the output 15 of the first amplifier A2 is connected to the input 17 of the second amplifier A3 at a node K3, wherein the input 13 of the first amplifier A2 is connected to the node K2, wherein the output 19 of the second amplifier A3 is connected to a node K4, wherein the amplifier cascade 11 is configured to generate a high signal amplification with low phase shift over a wide frequency range, a feedback network Fl, wherein the feedback network Fl is connected to the input 7 of the input stage A1 at the node K1 and to the output 19 of the second amplifier A3 at the node K4, wherein the feedback network Fl is configured to provide a high-ohmic feedback resistor with a parasitic capacitance of less than 0.5 pF,wherein the feedback network Fl is configured to provide a negative feedback to a structure comprising the input stage Al and the amplifier cascade 11, and a signal output 21 connected to a node K5, wherein the node K5 is connected to the node K4 or corresponds to the node K4.,

[0173] Figure 2 shows an example of an input stage A1. The input stage A1 comprises a junction field-effect transistor Qi and a bipolar transistor 2. The junction field-effect transistor Qi has three terminals: source S, gate G, and drain D. The bipolar transistor 2 has three terminals: collector C, base B, and emitter E. The drain terminal of the junction field-effect transistor Qi is connected to the base terminal B of the bipolar transistor 2. The junction field-effect transistor Qi is connected as a common-source circuit. The bipolar transistor 2 is connected as an emitter follower.

[0174] The input stage A1 may additionally or alternatively comprise a component with a negative capacitance. Figure 3 shows an example of an amplifier cascade 11 with an example of a feedback network F1. The first and second amplifiers A2, A3 of the amplifier cascade 11 are each an operational amplifier Q3, Q4. The first operational amplifier Q3 may have a voltage gain of more than 10 4 The second operational amplifier Q4 can provide a voltage gain of up to 10 3 have.

[0175] The feedback network Fl comprises a high-ohm feedback resistor R7 with a parallel capacitance C5. In addition, the feedback network Fl has a series-connected low-pass filter. The low-pass filter has a resistance Rs and a capacitance Ce to ground. The low-pass filter is connected to the feedback resistor R7 and the parallel capacitance C5 at node Kf.

[0176] Figure 4 shows a schematic representation of a circuit of a second embodiment of the amplifier circuit 1b according to the invention. The circuit shown in Figure 4 is an extension of the circuit of Figure 1 and thus includes all the elements of Figure 1 and their functions. Figure 4 additionally shows an output stage A4. The output stage A4 is connected to node K4 at an input 23 of the output stage A4 and to the signal output 21 at node K5 at an output 25 of the output stage A4.

[0177] Output stage A4 is configured to amplify a voltage at the output stage input by up to 20 times. Output stage A4 is also configured to filter DC noise at input 23 of output stage A4 and to adjust a signal level at output 25 of output stage A4 to K5.

[0178] Figure 5 shows a circuit diagram of an example of an output stage A4. The output stage A4 includes an inverting bandpass amplifier Q5. An inverting input 27 of the bandpass amplifier A4 is AC-coupled by means of a series-connected capacitor C7.

[0179] Figure 6 shows a schematic representation of a circuit of a third embodiment of the amplifier circuit 1c according to the invention. The circuit shown in Fig. 6 is an extension of the circuit of Figs. 1 and 4 and thus includes all elements of Figs. 1 and 4 as well as their functions.

[0180] Figure 6 additionally shows an amplitude limiter F2 included in the amplifier circuit 1c. The amplitude limiter F2 is connected to node K4 or K5 at an input 31 of the amplitude limiter F2 and to node K3 at an output 29 of the amplitude limiter F2. The amplitude limiter F2 is configured to limit the amplitude of the output signal at signal output 21 when a threshold value at node K4 or node K5 is exceeded.

[0181] Figure 7 shows a circuit diagram of an example of an amplitude limiter F2. The amplitude limiter F2 comprises two Zener diodes Qe and Q7 connected in series with opposite polarity.

[0182] Figure 8 shows a schematic representation of a circuit of a fourth embodiment of the amplifier circuit 1d according to the invention. The circuit shown in Fig. 8 is an extension of the circuit from Figs. 1, 4 and 6 and thus has all the elements from Figs. 1, 4 and 6 as well as their functions.

[0183] Figure 8 additionally shows a frequency response compensation F3 included in the amplifier circuit 1d. The frequency response compensation F3 is connected to the node K3 and the node Ki and is configured to reduce the oscillation tendency of the amplifier cascade 11.

[0184] Figure 16 shows a schematic representation of a circuit of a sixth embodiment of the amplifier circuit 1f according to the invention. The amplifier circuit 1f shown in Fig. 16 is an extension of the circuit from Fig. 8. Figure 16 shows the amplifier circuit 1d shown in Fig. 8 with an additional decoupler A5. The decoupler A5 is connected to the node K4 and the input 23 of the output stage A4. The decoupler is configured to decouple an output of the amplifier cascade 11 from the output stage A4.

[0185] Preferably, the decoupler A5 comprises or is a non-inverting impedance converter Ch. Alternatively, the decoupler A5 may also be or comprise a combination of two inverting impedance converters.

[0186] Figure 17 shows a schematic representation of a circuit of a seventh embodiment of the amplifier circuit 1g according to the invention. The amplifier circuit 1g shown in Fig. 17 is an extension of the circuit from Fig. 8. Figure 17 shows the amplifier circuit 1d shown in Fig. 8 with an additional intermediate load F4. The intermediate load F4 is connected to the output 15 of the first amplifier A2 and the input 17 of the second amplifier A3 of the amplifier cascade 11 at node K3. The intermediate load F4 is configured to minimize natural oscillations of the amplifier circuit 1g. Preferably, the intermediate load F4 is configured to minimize the natural oscillations of the amplifier circuit 1g by means of a phase correction.

[0187] Figure 18 shows an example of an implementation of an intermediate load F4. The intermediate load F4 is illustrated by way of example as an inductance Li connected in series to ground and a nonreactive resistor R15. However, alternative embodiments of the intermediate load F4 are also conceivable. Figure 19 shows a schematic representation of a circuit of an eighth embodiment of the amplifier circuit 1h according to the invention. The amplifier circuit 1h shown in Fig. 19 is an extension of the circuit from Fig. 16 with an additional intermediate load F4.

[0188] Figure 9 shows the example of an amplifier cascade 11 shown in Figure 3 with an additional example of a frequency response compensation F3. The frequency response compensation F3 has a series-connected capacitor, for example, a capacitor C4. The capacitor C4 is connected to the nodes K1 and K3. The frequency response compensation F3 is configured to reduce the oscillation tendency of the amplifier cascade 11.

[0189] Figure 10 shows a circuit diagram of a fifth embodiment of the amplifier circuit 1e according to the invention. The amplifier circuit combines the component groups shown in Figs. 2, 3, 5, 7 and 9 to form an entire amplifier circuit 1e according to the invention. In addition, Fig. 10 shows a capacitive current source 41, which serves as a pyroelectric sensor D py which is connectable and connected to the amplifier circuit le at the node Ki. The amplifier circuit le is particularly preferred.

[0190] Figure 20 shows a circuit diagram of a ninth embodiment of the amplifier circuit lk according to the invention. The amplifier circuit lk combines the amplifier circuit shown in Fig. 10 with the intermediate load shown in Fig. 18 and the decoupler A5 shown in Fig. 16 in the form of a further operational amplifier Qs. The output of the operational amplifier Qs is directly fed back to its inverting input.

[0191] The amplifier circuit lk is particularly preferred.

[0192] The output of the first operational amplifier Q3 is connected to the node K3 via a first resistor R17, wherein the node K3 is connected to the non-inverting input of the second operational amplifier Q4 via an ohmic resistor R9.

[0193] The intermediate load F4 is illustrated by way of example as an inductor Li connected in series to ground and the second ohmic resistor R15. However, alternative embodiments of the intermediate load F4 are also conceivable. The ohmic resistor R17 and the ohmic resistor R15 form a voltage divider. The voltage divider advantageously limits excessive loading of the upstream amplifier stage or the upstream operational amplifier.

[0194] Preferably, resistors R15 and R17 are equal. These values, surprisingly, show particularly good results in minimizing unwanted self-oscillations.

[0195] The first to fifth embodiments of the amplifier circuits 1a, 1b, 1c, 1d, 1e can preferably be used to measure a current signal from a capacitive current source, preferably a pyroelectric sensor. Each embodiment, in particular also the sixth to ninth embodiments of the amplifier circuits 1a, 1b, 1c, 1d, 1e, 1f, 1g, 1h, 1k according to the invention (simplified: amplifier circuit 1), can preferably be used to measure a current signal from a capacitive current source, preferably a pyroelectric sensor.

[0196] The amplifier circuit la, lb, lc, ld, le, lf, lg, lh, lk according to the invention can preferably be used to measure a current signal of a capacitive current source, preferably a pyroelectric sensor, in an infrared spectrometer, preferably an FTIR spectrometer (Fourier transform infrared spectrometer).

[0197] Figure 11 shows a section of a printed circuit board 33, which can also be called a circuit card. The amplifier circuit 1a, 1b, lc, ld, le, lf, lg, lh, lk (hereinafter referred to simply as amplifier circuit 1 for the sake of better readability) or at least a part of the amplifier circuit 1 is applied to a printed circuit board 33. By way of example, Fig. 11 shows an electronic component 35 of the amplifier circuit 1 on the printed circuit board 33. The electronic component 35 is soldered to the printed circuit board 33 using solder pads 37. A region 39 of the printed circuit board 33 below the at least one electronic component 35 outside the solder pads 37 has been removed.

[0198] The region 39 of the circuit board 33 can preferably be located below an electronic component 35 or several components of the feedback network Fl or below the entire feedback network Fl.

[0199] Alternatively or additionally, the region 39 of the circuit board 33 below one or more electronic components of the input stage A1 can preferably be removed. Particularly preferably, the region 39 below the field-effect transistor Qi and / or the region 39 below the bipolar transistor Cb is removed.

[0200] Figure 12 shows an embodiment of a sensor system 43 according to the invention comprising a capacitive current source 41 and an amplifier circuit 1. Preferably, the capacitive current source of the sensor system 43 is a pyroelectric sensor D py .

[0201] Figure 21 shows a further embodiment of a sensor system 43 according to the invention comprising a capacitive current source 41 and an amplifier circuit lk. Preferably, the capacitive current source of the sensor system 43 is a pyroelectric sensor D py .

[0202] The pyroelectric sensor D pycan preferably be plate-like and have a maximum thickness d of 40 pm, preferably a maximum of 10 pm.

[0203] The sensor system 43 according to the invention can preferably be used in and / or with an FTIR spectrometer (Fourier Transform Infrared Spectrometer). Figures 13a, b show a representation of measurement results of the performance of the amplifier circuit according to the invention. Figures 14a, b, and c each show a representation of measurement results of the performance of the sensor system according to the invention, and Figure 15 shows a representation of measurement results of the transmission behavior at different amplifications.

[0204] Figure 13 shows measured data for the transmission of the four-stage amplifier circuit 1 according to the invention and demonstrates the performance and advantages of the invention. Despite a 40-fold higher gain of 400 MV / A compared to a conventional TIA, where the gain is typically only 10 MV / A, the amplifier circuit 1 according to the invention shows a flat transmission up to a 3 dB cutoff frequency of 90 kHz. The amplifier circuit 1 thus achieves the same bandwidth as known TIAs in the prior art with a lower gain of 10 MV / A. In other words, the amplifier circuit 1 achieves a significantly greater gain with a comparable bandwidth compared to the prior art. Furthermore, the presented circuit is robust against changes in the input capacitance, in particular due to the capacitive current source 41, for example, a pyroelectric sensor D pyThe transfer functions from 0 pF to 270 pF are almost identical. Even with the large input capacitance of 2.2 nF, there is no exaggeration in the frequency response as is the case with state-of-the-art TIAs.

[0205] The noise voltage of the amplifier circuit according to the invention is shown in Fig. 13b and is only slightly higher at low frequencies than the minimum possible Johnson-Nyquist noise of a 400 MΩ feedback resistor of . 4k B ■ 300 K ■ 400 Mfl = 2.6 ■ 10" 6 - . Hz

[0206] At higher frequencies from 10 4Hz, the noise depends on the input capacitance: the higher this is, the sooner and more pronounced the noise increases. This is due, on the one hand, to a reduction in the input impedance of the capacitive current source 41, which leads to a greater amplification of the input voltage noise. On the other hand, the dielectric loss of the sensor, which increases with frequency, effectively reduces its resistance R py This creates a further contribution to the noise.

[0207] Preferably, the capacity of the pyroelectric sensor D py be compensated by a passive component with a negative capacitance (see above). A combination of a component with negative capacitance with a capacitive current source 41, preferably a pyroelectric sensor D pywith conventional positive capacitance, promises a further significant reduction in noise and thus a significantly improved signal-to-noise ratio, as shown in Fig. 13b. The performance of the amplifier circuit 1 according to the invention was tested in conjunction with a pyroelectric sensor D py , ie as a sensor system according to the invention, was tested in an optical setup (see Fig. 14a to c). For this purpose, the sensitivity was measured with a pulsed diode laser with a power of 145 pW at an amplification of 5 GV / A over a frequency range up to 100 kHz and with a pyroelectric sensor D py with the same gain compared with a prior art single-stage TIA. With the same thickness of the pyroelectric sensor D pyof 30 pm, the bandwidth of the four-stage amplifier circuit 1 according to the invention (see Fig. 13a to c) is 5 kHz, which is significantly larger than the 200 Hz bandwidth of the single-stage TIA (see Fig. 13a to c). The cutoff frequency of 5 kHz corresponds in this case to the thermal time constant of the pyroelectric sensor D py with a thickness of 30 pm. The amplifier circuit 1 according to the invention has a very high electronic cutoff frequency of 8 kHz at this advantageous high amplification.

[0208] Significantly more signal and a further increase in the bandwidth up to the electronic bandwidth of the amplifier circuit 1 according to the invention is obtained by reducing the thickness of the pyroelectric sensor D pyIn this case, both the optical-thermal conversion and the thermal-electrical conversion are optimized due to the reduced heat capacity of the pyroelectric sensor. The corresponding curves in Fig. 14a to c demonstrate this effect using a 7 pm thick pyroelectric sensor D py . The signal is compared to the thicker pyroelectric sensor D py with the same amplifier circuit according to the invention, the bandwidth is more than doubled and the bandwidth increases up to the electronic bandwidth of 8 kHz.

[0209] As in Fig. 13b, the noise component in Fig. 14a to c at low frequencies is primarily Johnson-Nyquist noise of the feedback resistor. As expected, the noise of the sensor system with the four-stage amplifier 1 according to the invention increases with increasing frequency. The absolute value of the noise depends significantly on the capacitance C. py of the pyroelectric sensor D pyand its loss factor tan(ö). Both quantities are smaller for the thinner 7 pm pyroelectric sensor (C py = 250 pF) is larger than that of the 30 pm thick pyroelectric sensor (C py ~ 120 pF). This effect is overcompensated for the selected thicknesses of the pyroelectric sensor by the signal increase of the thinner pyroelectric sensor, so that it still has a significantly lower noise equivalent power (NEP) or a better signal-to-noise ratio (see Fig. 14c).

[0210] Overall, the performance of both sensor systems with the four-stage amplifier circuit 1 according to the invention is significantly better than prior art sensor systems across the entire frequency range, but especially at frequencies higher than 200 Hz. The bandwidth is larger with the same gain, and measurements above 1 kHz are easily feasible. Thanks to the insensitivity of the four-stage amplifier circuit 1 according to the invention to larger input capacitances, thinner pyroelectric crystals can be used in the pyroelectric sensor with higher capacitance and a larger signal without reducing the bandwidth or causing excessive transmission.

[0211] The bandwidth of the 7 pm thick pyroelectric sensor can be increased well beyond 8 kHz with the inventive four-stage amplifier circuit 1 with a flat transfer function by reducing the gain (see Fig. 15). With a gain of 400 MV / A, the detector's cutoff frequency of 70 kHz is no longer limited by the electronic bandwidth (90 kHz).

Claims

Patent claims 1. Amplifier circuit (1) for broadband and low-noise amplification of a capacitive current source, preferably a pyroelectric sensor, comprising a signal input (3) which can be connected to the capacitive current source at a node Ki, an input stage (Al), wherein the input stage (Al) is connected to the node Ki at an input (7) of the input stage (Al) and has a node K2 at the output (9) of the input stage (Al), wherein the input stage (Al) is configured to amplify an input voltage at least three times, wherein the input stage (Al) is configured to provide a high-ohmic input resistor at the input of the input stage (Al), wherein the input stage (Al) is configured to provide a stable and load-independent voltage at the output of the input stage (Al), an amplifier cascade, wherein the amplifier cascade (11) has at least a first and a second amplifier (A2, A3), each having an input (13, 17) and an output (15,19), wherein the output (15) of the first amplifier (A2) is connected at a node K3 to the input (17) of the second amplifier (A3), wherein the input (13) of the first amplifier (A2) is connected to the node K2, wherein the output (19) of the second amplifier (A3) is connected to a node K4, wherein the amplifier cascade (11) is configured to generate a high signal amplification with low phase shift over a wide frequency range, a feedback network (Fl), wherein the feedback network (Fl) is connected to the input (7) of the input stage (A1) at the node K1 and to the output (19) of the second amplifier (A3) at the node K4, wherein the feedback network (Fl) is configured to provide a high-ohmic feedback resistor with a parasitic capacitance of less than 0.5 pF, wherein the feedback network (Fl) is configured to provide negative feedback to a structure comprising the input stage (A1) and the amplifier cascade (11) provideand a signal output (21) connected to a node K5, wherein the node K5 is connected to the node K4 or corresponds to the node K4., 2. Amplifier circuit (1) according to claim 1, wherein the input stage (Al) comprises a junction field effect transistor (Qi) and a bipolar transistor (2), wherein a drain terminal of the junction field effect transistor (Qi) is connected to a base terminal of the bipolar transistor, wherein the junction field effect transistor (Qi) is connected as a source circuit, wherein the bipolar transistor (2) is connected as an emitter follower, wherein the first and second amplifiers (A2, A3) of the amplifier cascade (11) are operational amplifiers (Q3, Q4), wherein the first operational amplifier (Q3) has a voltage gain of more than 10 4 wherein the second operational amplifier (Q4) has a voltage gain of maximum 10 3wherein the feedback network (Fl) has a high-ohmic feedback resistor with a parallel capacitance, wherein the feedback network (Fl) has a series-connected low-pass filter.

3. Amplifier circuit (1) according to claim 1 or 2, wherein the amplifier circuit additionally has an output stage (A4), wherein the output stage (A4) is connected to the node K4 at an input of the output stage (A4) and is connected to the signal output in the node K5 at an output of the output stage (A4), wherein the output stage (A4) is configured to amplify a voltage at the input of the output stage by up to 20 times, wherein the output stage (A4) is configured to filter DC voltage interference at the input of the output stage and to adapt a signal level at the output of the output stage (A4) to K5.

4. Amplifier circuit (1) according to claim 3, wherein the output stage (A4) comprises an inverting bandpass amplifier, wherein an input of the bandpass amplifier is AC-coupled.

5. Amplifier circuit (1) according to one of claims 1 to 4, wherein the amplifier circuit additionally comprises an amplitude limiter (F2), wherein the amplitude limiter (F2) is connected to the node K4 or K5 at an input of the amplitude limiter (F2) and to the node K3 at an output of the amplitude limiter (F2), wherein the amplitude limiter (F2) is configured to limit the amplitude of the output signal at the signal output when a threshold value at the node K4 or node Ks is exceeded.

6. Amplifier circuit (1) according to claim 5, wherein the amplitude limiter (F2) comprises two Z-diodes (Q.&, Q7) connected in series with opposite polarity.

7. Amplifier circuit (1) according to one of claims 1 to 6, wherein the amplifier circuit (1) has a frequency response compensation (F3), wherein the frequency response compensation (F3) is connected to the node K3 and the node Ki, wherein the frequency response compensation (F3) is arranged to reduce a tendency of the amplifier cascade (11) to oscillate.

8. Amplifier circuit (1) according to claim 7, wherein the frequency response compensation (F3) comprises a negative feedback via a capacitor.

9. Amplifier circuit (1) according to one of claims 1 to 8, wherein the input stage (Al) comprises a component with a negative capacitance.

10. Amplifier circuit (1) according to one of claims 1 to 9, wherein the amplifier circuit (1) or at least a part of the amplifier circuit (1) is applied to a printed circuit board (33), wherein at least one electronic component of the amplifier circuit (1) is soldered on the printed circuit board (33) to solder pads (37) on the printed circuit board (33), wherein a region (39) of the printed circuit board (33) below the at least one electronic component is removed outside the solder pads.

11. Amplifier circuit (1) according to claim 10, wherein the at least one electronic component is or are an electronic component or several components of the feedback network (Fl) or the entire feedback network (Fl).

12. Amplifier circuit (1) according to claim 10 or 11, wherein the at least one electronic component is or are one or more electronic components of the input stage (Al), preferably the junction field effect transistor (Qi) and / or the bipolar transistor (Ch).

13. Amplifier circuit (1) according to one of claims 3 to 12, wherein the amplifier circuit (1) additionally comprises a decoupler (A5), wherein the decoupler (A5) is connected to the node K4 and the input (23) of the output stage (A4), wherein the decoupler (A5) is arranged to decouple an output of the amplifier cascade (11) from the output stage (A4).

14. Amplifier circuit (1) according to claim 13, wherein the decoupler (A5) comprises an impedance converter, preferably a non-inverting impedance converter or a combination of two inverting impedance converters, or wherein the decoupler (A5) is an impedance converter, preferably a non-inverting impedance converter or a combination of two inverting impedance converters.

15. Amplifier circuit (1) according to claim 14, wherein the impedance converter comprises at least one or more of the following components or is one of the following components: MOSFET, bipolar transistor, operational amplifier (Qs), operational amplifier (Qs) with high input resistance and low inherent noise, operational amplifier (Qs) with high input resistance and low inherent noise, wherein the output of the operational amplifier (Qs) is directly or indirectly fed back to its inverting input.

16. Amplifier circuit (1) according to one of claims 1 to 15, wherein the amplifier circuit (1) additionally comprises an intermediate load (F4), wherein the intermediate load (F4) is connected to the output (15) of the first amplifier (A2) and the input (17) of the second amplifier (A3) of the amplifier cascade (11) in the node K3, wherein the intermediate load (F4) is configured to minimize natural oscillations of the amplifier circuit (1).

17. Amplifier circuit (1) according to claim 16, wherein the intermediate load (F4) is arranged to minimize natural oscillations by a phase correction.

18. Amplifier circuit (1) according to one of claims 16 or 17, wherein the output (15) of the first amplifier (A2) is connected to the node K3 via a first ohmic resistor (R17), wherein the node K3 is connected to ground via a second ohmic resistor (Ris).

19. Amplifier circuit (1) according to claim 18, wherein the intermediate load comprises or consists of at least one or more of the following component combinations: an inductance (Li) and / or capacitance connected in series to ground and at least one ohmic resistor, preferably the second ohmic resistor (R15); an inductance and / or capacitance connected in parallel to ground and at least one ohmic resistor (Ris).

20. Sensor system (43) comprising a capacitive current source and an amplifier circuit (1) according to one of claims 1 to 19.

21. Sensor system (43) according to claim 20, wherein the capacitive current source is a pyroelectric sensor (D py ) is.

22. Sensor system (43) according to claim 20 or 21, wherein the pyroelectric sensor (D py ) is plate-like and has a maximum thickness of 40 pm, preferably a maximum of 10 pm.

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