LLC DC / DC converter

Abstract

To provide a LLC type DC / DC converter capable of suppressing increase in a boost current when a step-up ratio is large.SOLUTION: A LLC DC / DC converter 10 includes a transformer circuit Tr, a primary side switching circuit 1, a secondary side switching circuit 3, a resonant circuit 2 including a resonant inductance Lr and a resonant capacitance Cr, and a control unit 4. The control unit 4 performs boost control to cause the main circuit unit to perform a boost operation. When power is transferred from the primary side switching circuit 1 to the secondary side switching circuit 3, the control unit 4 during boost control reduces a drive frequency of the primary side switching elements Q1 to Q4 and turns on one of the secondary side switching elements Q5 to Q8 to form a short circuit in the secondary side switching circuit 3.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] The present invention relates to an LLC type DC / DC converter. [Background technology]

[0002] In recent years, in order to achieve carbon neutrality, development has been progressing on V2H (Vehicle to Home) systems that supply battery power from electric vehicles to household loads, and distributed power supply systems that combine solar power generation equipment and storage batteries, and there is a demand for smaller, less costly, and more efficient power supplies in these systems.For this reason, CLLC DC / DC converters, which are current resonant DC / DC converters with a small number of parts and capable of high efficiency, are being adopted as bidirectional DC / DC converters used in power supplies.

[0003] On the other hand, the CHAdeMO standard, which is the charging standard for electric vehicles, requires support for a wide range of voltages from 150V to 450V. For this reason, the power supply included in the V2H system needs to not only step down but also step up the direct current voltage output by the DC / DC converter to match the battery voltage of the electric vehicle. However, CLLC DC / DC converters generally control the output voltage by performing frequency modulation control, which has the disadvantage that the range of voltages that can be output is narrow.

[0004] Fig. 16 shows the main circuit section of a conventional CLLC-type DC / DC converter 10C. The main circuit section of the CLLC-type DC / DC converter 10C has a symmetrical circuit configuration capable of bidirectional operation. That is, the main circuit section includes a primary-side switching circuit 1 and a primary-side resonant circuit 2c on the primary side of a transformer Tr, and a secondary-side resonant circuit 5c and a secondary-side switching circuit 3 on the secondary side of the transformer Tr. The primary-side resonant circuit 2c includes a resonant coil Lr11 and a resonant capacitor Cr11, and the secondary-side resonant circuit 5c includes a resonant coil Lr21 and a resonant capacitor Cr21.

[0005] Since the CLLC DC / DC converter has a bidirectional circuit configuration of the LLC DC / DC converter, it has a resonant inductance (resonant coil) and a resonant capacitance (resonant capacitor) on both the drive circuit side during forward operation and the drive circuit side during reverse operation to obtain the required input / output ratio during forward operation and reverse operation. For example, the CLLC DC / DC converter 10C in FIG. 16 has a primary side resonant circuit 2c and a secondary side resonant circuit 5c. The resonant coils constituting these resonant circuits 2c and 5c are usually composed of a core and a winding, and are large in size and weight, which hinders miniaturization and cost reduction. In addition, film capacitors and multilayer ceramic capacitors are usually used for the resonant capacitors constituting the resonant circuits 2c and 5c, but the size is large in the case of a film capacitor, and the mounting area needs to be large in the case of a multilayer ceramic capacitor, which hinders miniaturization in either case. For this reason, compared to LLC type DC / DC converters in which the resonant circuit is only on the drive circuit side, CLLC type DC / DC converters have resonant circuits on both the primary and secondary sides (drive circuit side and rectifier circuit side), which means an increased number of components, resulting in larger and more expensive main circuit sections, and increased losses (reduced power conversion efficiency) due to the resonant circuit on the rectifier circuit side.

[0006] Patent Document 1 discloses an LLC DC / DC converter capable of bidirectional operation. In the LLC DC / DC converter described in Patent Document 1, the switch transistor on the input side is driven in a mode with a fixed drive frequency and a duty of 50%, and the switch transistors of the two half bridges on the rectification side are driven in a mode of phase shift control, thereby performing a boost operation. The LLC DC / DC converter described in Patent Document 1 adjusts the amount of phase shift to adjust the boost ratio in the boost operation, thereby making it possible to make the gain greater than 1.

[0007] The LLC DC / DC converter described in Patent Document 1 has the advantage of reducing the number of parts and making the main circuit smaller and less expensive because it has only one resonant circuit, but it has the problem that the inductance value of the main circuit decreases and a large short-circuit current (boost current) flows in a short time during boost operation, making it difficult to control the output. Furthermore, when the input / output voltage ratio (step-up ratio) of the main circuit increases, the boost ratio also increases, causing a problem that a larger boost current flows.

[0008] Furthermore, with LLC DC / DC converters, even if the length of the short-circuit period (boost period) is the same, the magnitude of the boost current differs if the drive frequency or input voltage differs due to input / output characteristics that depend on the LLC type drive frequency, so there is a problem in that it is difficult to determine the optimal operating parameters (for example, at what drive frequency should the boost operation be performed).

[0009] Patent Document 2 also discloses an LLC DC / DC converter capable of bidirectional operation. In the LLC DC / DC converter described in Patent Document 2, a secondary side switching element (short circuit switch) is turned on in synchronization with a primary side switching element to perform boost operation. The LLC DC / DC converter described in Patent Document 2 can adjust the boost ratio achieved by boost operation by controlling the on-duty of the short circuit switch. However, the LLC DC / DC converter described in Patent Document 2 has the same problems as the LLC DC / DC converter described in Patent Document 1.

[0010] Non-Patent Document 1 discloses a method for controlling the boost operation in an LLC DC / DC converter. The control method described in Non-Patent Document 1 gradually increases the boost angle and drive frequency as the input voltage decreases in a low input voltage region. However, if the drive frequency is increased as the input voltage decreases, the boost ratio increases as the step-up ratio increases, causing a larger boost current to flow, which makes it difficult to control the output. [Prior art documents] [Patent documents]

[0011] [Patent Document 1] Chinese Patent No. 111030464 [Patent Document 2] JP 2005-224012 A [Non-patent literature]

[0012] [Non-Patent Document 1] Chunyang Zhao, et al., "Design and Control of a High-Power Wide-Gain-Range LLC Resonant Converter", 20-24 March 2022, 2022 IEEE Applied Power Electronics Conference and Exposition (APEC), pp.40-44 Summary of the Invention [Problem to be solved by the invention]

[0013] The present invention has been made in consideration of the above circumstances, and an object of the present invention is to provide an LLC type DC / DC converter that can suppress an increase in boost current when the step-up ratio is large. [Means for solving the problem]

[0014] In order to solve the above problems, the LLC type DC / DC converter according to the present invention comprises: a main circuit section including a transformer circuit, a primary side switching circuit including a primary side switching element provided on a primary side of the transformer circuit, a secondary side switching circuit including a secondary side switching element provided on a secondary side of the transformer circuit, and a resonant circuit including a resonant inductance and a resonant capacitance; a control unit that generates a control signal for controlling on / off of the primary side switching element and the secondary side switching element; An LLC type DC / DC converter comprising: The control unit performs boost control to cause the main circuit unit to perform a boost operation, The control unit during the boost control is When the main circuit unit transmits power from the primary side switching circuit to the secondary side switching circuit, The control signal reduces a drive frequency of the primary side switching element and turns on the secondary side switching element to form a short circuit in the secondary side switching circuit.

[0015] In this configuration, the control unit reduces the drive frequency of the primary switching element to turn on the secondary switching element for a time period corresponding to the boost ratio during boost control, so that the output can be increased by the LLC output characteristics associated with the reduction in drive frequency. Therefore, with this configuration, the boost period can be shortened, and an increase in boost current can be suppressed when the boost ratio is large.

[0016] In the LLC type DC / DC converter, The control unit during the boost control is When the main circuit unit transmits power from the secondary side switching circuit to the primary side switching circuit, The control signal can be configured to reduce the drive frequency of the secondary side switching element and turn on the primary side switching element to form a short circuit in the primary side switching circuit.

[0017] In the LLC type DC / DC converter, The control unit is The drive frequency may be controlled to a value equal to or higher than the resonant frequency of the resonant circuit, and the boost control may be performed when the drive frequency is lower than a predetermined boost start frequency.

[0018] In the LLC type DC / DC converter, The control unit is When the drive frequency is equal to or higher than the boost start frequency, a frequency modulation control is performed to modulate the drive frequency; When the boost control is performed, the drive frequency is used to calculate a boost pulse width of the control signal, and the short circuit can be formed during the period of the boost pulse width.

[0019] In the LLC type DC / DC converter, The control unit is The boost start frequency can be varied so that the smaller the boost ratio at which the boost operation is performed, the lower the boost start frequency, and the higher the boost ratio, the higher the boost start frequency.

[0020] In the LLC type DC / DC converter, The control unit is When the boost pulse width of the control signal is Tb, the drive frequency is fs, and the gain of the boost ratio for performing the boost operation is G, Formula 1 (coefficient a / driving frequency fs)-constant b Formula 2 [(coefficient a / driving frequency fs)-constant b] x gain G The third formula is (-coefficient a × driving frequency fs) + constant b Formula 4 [(-coefficient a × drive frequency fs) + constant b] × gain G The boost pulse width can be calculated from any one of the following equations, and the short circuit can be formed during the period of the boost pulse width.

[0021] In the LLC type DC / DC converter, The gain G can be configured to have a larger value as the step-up ratio increases, and to have a smaller value as the step-up ratio decreases.

[0022] In the LLC type DC / DC converter, The primary side switching circuit and the secondary side switching circuit include It can be a half-bridge circuit, a full-bridge circuit, or a multi-phase bridge circuit including three phases. The full bridge circuit and the multi-phase bridge circuit of the secondary side switching circuit can be configured to include a bridgeless circuit in which the secondary side switching element in an upper arm is replaced with a diode. Effect of the Invention

[0023] According to the present invention, it is possible to provide an LLC DC / DC converter capable of suppressing an increase in boost current when the step-up ratio is large. [Brief description of the drawings]

[0024] [Figure 1] 1 is an example of a circuit diagram of an LLC DC / DC converter according to a first embodiment. [Diagram 2] 5 is a diagram showing an example of control timing of each switching element during forward operation and boost control according to the first embodiment. FIG. [Diagram 3] 2. FIG. 4 is a current path diagram in the first embodiment, where (A) is a diagram of a period of mode 1B in FIG. 2, (B) is a diagram of a period of mode 1 in FIG. 2, (C) is a diagram of a period of mode 2B in FIG. 2, and (D) is a diagram of a period of mode 2 in FIG. 2. [Figure 4] 5 is a diagram showing an example of control timing of each switching element during reverse operation and boost control according to the first embodiment. FIG. [Diagram 5] 5A is a current path diagram in the first embodiment, where (A) is a diagram of the period of mode 1B' in FIG. 4, (B) is a diagram of the period of mode 1' in FIG. 4, (C) is a diagram of the period of mode 2B' in FIG. 4, and (D) is a diagram of the period of mode 2' in FIG. 4. [Figure 6] FIG. 5 is a diagram showing an example of the relationship between the drive frequency fs and the boost pulse width Tb when the inverter is operated in the reverse direction with the control timing of FIG. 4 (when V2=300 V). [Figure 7]3A and 3B are diagrams showing examples of the relationship between the drive frequency fs and the boost pulse width Tb when the device is operated in the forward direction with the control timing of FIG. 2, where (A) is a diagram for V2=350 V and (B) is a diagram for V2=450 V. [Figure 8] FIG. 4 is an example of a current control flow diagram during reverse operation of the LLC DC / DC converter according to the first embodiment. [Figure 9] FIG. 4 is an example of a current control flow diagram during forward operation of the LLC DC / DC converter according to the first embodiment. [Figure 10] 1 is an example of a circuit diagram of an LLC DC / DC converter according to a first modified example. [Figure 11] FIG. 13 is a diagram showing an example of control timing of each switching element according to a first modified example during forward operation and boost control. [Figure 12] 11A and 11B are current path diagrams in the first variant, where (A) is a diagram of the period of mode 1B in FIG. 11, (B) is a diagram of the period of mode 1 in FIG. 11, (C) is a diagram of the period of mode 2B in FIG. 11, and (D) is a diagram of the period of mode 2 in FIG. 11. [Figure 13] 13 is an example of a circuit diagram of an LLC DC / DC converter according to a second modified example. [Figure 14] FIG. 13 is a diagram showing an example of control timing of each switching element according to a second modified example during forward operation and boost control. [Figure 15] 15A and 15B are current path diagrams in a second modified example, in which (A) is a diagram of a period of mode 1B in FIG. 14, and (B) is a diagram of a period of mode 1 in FIG. [Figure 16] 1 is an example of a main circuit diagram of a conventional CLLC type DC / DC converter. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0025] DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, an embodiment of an LLC type DC / DC converter according to the present invention will be described with reference to the accompanying drawings.

[0026] Fig. 1 shows an LLC DC / DC converter 10 according to one embodiment of the present invention. The LLC DC / DC converter 10 is a current-resonant LLC DC / DC converter capable of bidirectional operation, and includes a pair of terminals T1, T2, a capacitor C11, a primary-side switching circuit 1, a resonant circuit 2, a transformer Tr, a secondary-side switching circuit 3, a capacitor C21, a pair of terminals T3, T4, and a control unit 4. The above configuration excluding the control unit 4 corresponds to the "main circuit unit" of the present invention.

[0027] For example, the DC side terminal of a bidirectional inverter is connected to the terminals T1 and T2, and for example, a battery (for example, a battery of an electric vehicle) is connected to the terminals T3 and T4. If the power transmission from the terminals T1 and T2 (primary side) to the terminals T3 and T4 (secondary side) is called forward power transmission, the operation during the forward power transmission (hereinafter, forward operation) is a charging operation to the battery connected between the terminals T3 and T4. On the other hand, if the power transmission from the terminals T3 and T4 (secondary side) to the terminals T1 and T2 (primary side) is called reverse power transmission, the operation during the reverse power transmission (hereinafter, reverse operation) is a discharging operation from the battery connected between the terminals T3 and T4. During the forward operation, the primary side switching circuit 1 becomes a drive circuit, and the secondary side switching circuit 3 becomes a rectifier circuit. On the other hand, during the reverse operation, the secondary side switching circuit 3 becomes a drive circuit, and the primary side switching circuit 1 becomes a rectifier circuit.

[0028] Capacitor C11 is connected between terminals T1 and T2 and functions as a capacitor for reducing output ripple during reverse operation. Capacitor C21 is connected between terminals T3 and T4 and functions as a capacitor for reducing output ripple during forward operation.

[0029] The primary side switching circuit 1 includes a first leg and a second leg connected in parallel, and each leg includes an upper arm and a lower arm connected in series to form a full bridge circuit. A connection point between the upper arm of the first leg and the upper arm of the second leg is connected to a terminal T1. A connection point between the lower arm of the first leg and the lower arm of the second leg is connected to a terminal T2. A connection point X1 between the upper arm of the first leg and the lower arm of the first leg and a connection point X2 between the upper arm of the second leg and the lower arm of the second leg are connected to a resonant circuit 2.

[0030] The upper arm of the first leg includes a switching element Q1, a diode D1, and a capacitor C1. Similarly, the lower arm of the first leg includes a switching element Q2, a diode D2, and a capacitor C2, the upper arm of the second leg includes a switching element Q3, a diode D3, and a capacitor C3, and the lower arm of the second leg includes a switching element Q4, a diode D4, and a capacitor C4. The switching elements Q1 to Q4 correspond to the "primary side switching elements" of the present invention. The diodes D1 to D4 function as freewheeling diodes during forward operation, and function as rectifying or reverse current prevention diodes during reverse operation. The capacitors C1 to C4 function as partial resonance capacitors during forward operation.

[0031] The switching element Q1 may be a power semiconductor switching element capable of switching at high frequency, such as an IGBT (insulated gate bipolar transistor), a MOSFET (metal oxide semiconductor field effect transistor) using Si (silicon), SiC (silicon carbide) or GaN (gallium nitride), etc. The same applies to the switching elements Q2 to Q4 and the switching elements Q5 to Q8 described later.

[0032] The diode D1 is connected in parallel in the reverse direction to the current path of the switching element Q1. The diode D1 may be a parasitic diode of the switching element Q1, or an external diode independent of the switching element Q1, or both. The same applies to the diodes D2 to D4 and the diodes D5 to D8 described later.

[0033] Capacitor C1 is connected in parallel to the current path of switching element Q1 and diode D1. Capacitor C1 may be a parasitic capacitor of switching element Q1, or an external capacitor independent of switching element Q1, or both. The same applies to capacitors C2 to C4 and capacitors C5 to C8 described later.

[0034] The resonant circuit 2 includes a resonant inductance Lr by a resonant coil (inductor) connected in series to the primary winding of the transformer Tr, and a resonant capacitance Cr by a resonant capacitor. The resonant inductance Lr and the resonant capacitance Cr form a series resonant circuit together with the excitation inductance (excitation coil) of the transformer Tr. Since the problem in operation is the resonant frequency determined by the resonant inductance Lr and the resonant capacitance Cr (resonant components), the circuit constants of these resonant components (inductance Lr and resonant capacitance Cr) are called the resonance constant. The excitation inductance may be formed by a coil separate from the transformer Tr, but is omitted from the illustration as being included in the transformer Tr. The resonant inductance Lr may be the leakage inductance of the transformer Tr, a coil having an individual core, or both. The resonant capacitance Cr may be formed by an individual capacitor, a parasitic capacitance of the switching elements Q1 to Q4, or both. Note that the arrangement of the resonant inductance Lr and the resonant capacitance Cr can be appropriately changed as long as they are connected in series to the primary winding of the transformer Tr to form a series resonant circuit.

[0035] The resonant inductance Lr and the resonant capacitance Cr of the resonant circuit 2 form a series resonant circuit through the transformer Tr from the secondary side as an equivalent circuit during reverse operation. That is, the resonant circuit 2 can perform LLC resonant operation even during reverse operation. However, if the resonant constant of the resonant circuit 2 is a value optimized for forward operation, it may not be the optimal value during reverse operation.

[0036] The transformer Tr corresponds to the "transformer circuit" of the present invention, and is composed of one or more high-frequency isolated transformers. The primary winding of the transformer Tr is connected to a primary-side switching circuit 1 via a resonant circuit 2. The secondary winding of the transformer Tr is connected to a secondary-side switching circuit 3. The turns ratio of the primary and secondary windings of the transformer Tr is related to the step-up ratio, which is the input / output voltage ratio. For example, if a value is optimized for forward operation, it may not be the optimal value for reverse operation.

[0037] The secondary-side switching circuit 3 includes a third leg and a fourth leg connected in parallel, and each leg includes an upper arm and a lower arm connected in series to form a full-bridge circuit. A connection point between the upper arm of the third leg and the upper arm of the fourth leg is connected to a terminal T3. A connection point between the lower arm of the third leg and the lower arm of the fourth leg is connected to a terminal T4. A connection point X3 between the upper arm of the third leg and the lower arm of the third leg is connected to one end of the secondary winding of the transformer Tr, and a connection point X4 between the upper arm of the fourth leg and the lower arm of the fourth leg is connected to the other end of the secondary winding of the transformer Tr.

[0038] The upper arm of the third leg includes a switching element Q5, a diode D5, and a capacitor C5. Similarly, the lower arm of the third leg includes a switching element Q6, a diode D6, and a capacitor C6, the upper arm of the fourth leg includes a switching element Q7, a diode D7, and a capacitor C7, and the lower arm of the fourth leg includes a switching element Q8, a diode D8, and a capacitor C8. The switching elements Q5 to Q8 correspond to the "secondary side switching elements" of the present invention. The diodes D5 to D8 function as freewheeling diodes during reverse operation, and function as rectifying or reverse current prevention diodes during forward operation. The capacitors C5 to C8 function as partial resonance capacitors during reverse operation. In this way, the secondary side switching circuit 3 has the same configuration as the primary side switching circuit 1.

[0039] The control unit 4 includes a processing unit that generates a control signal for controlling the on / off of the switching elements Q1 to Q8, and a driving unit that turns the switching elements Q1 to Q8 on and off based on the control signal. The control unit 4 may be configured with a digital circuit such as a microprocessor or a digital signal processor, an analog circuit, or a circuit that combines a digital circuit and an analog circuit. The control unit 4 may further include a detection unit. The detection unit includes, for example, a current sensor and / or a voltage sensor that detects a current value and / or a voltage value required for the control of the control unit 4, and peripheral circuits thereof.

[0040] During forward operation, the control unit 4 monitors the input (input current, input voltage, or input power) to the terminals T1, T2 (primary side) and / or the output (output current, output voltage, or output power) from the terminals T3, T4 (secondary side), and controls the on / off of the switching elements Q1 to Q8 so that the output becomes a desired value. Similarly, during reverse operation, the control unit 4 monitors the input (input current, input voltage, or input power) to the terminals T3, T4 (secondary side) and / or the output (output current, output voltage, or output power) from the terminals T1, T2 (primary side), and controls the on / off of the switching elements Q1 to Q8 so that the output becomes a desired value.

[0041] The control unit 4 performs frequency modulation control to modulate the drive frequency of the switching element of the drive circuit as on / off control, and performs synchronous rectification control by controlling the switching element of the rectifier circuit in synchronization with the on timing of the switching element of the drive circuit. If the drive frequency during frequency modulation control is lower than a preset boost start frequency, the control unit 4 performs boost control and synchronous rectification control to short-circuit the rectifier circuit and cause the main circuit unit to perform a boost operation.

[0042] Although details will be described later, during boost control, the control unit 4 reduces the drive frequency of the switching element of the drive circuit and turns on the switching element of the rectifier circuit (the short-circuit switching element for short-circuiting to pass the boost current). For example, the desired output can be obtained by increasing the on-time (boost period) of the short-circuit switching element in response to the reduction in the drive frequency.

[0043] Fig. 2 shows an example of the control timing of the switching elements Q1 to Q8 during forward operation and boost control. The switching elements Q1 to Q8 are turned on when the control signal shown in Fig. 2 is at a high level, and turned off when the control signal is at a low level (the same applies to the following control timing diagrams).

[0044] The control unit 4 turns on and off the upper arm (switching element Q1) of the first leg and the lower arm (switching element Q4) of the second leg at the same timing with the same drive frequency and the same on-duty (for example, 50%) for the switching elements Q1 to Q4 of the primary side switching circuit 1, which is a drive circuit, and turns on and off the lower arm (switching element Q2) of the first leg and the upper arm (switching element Q3) of the second leg at the same timing. The switching elements Q1 and Q2 have a phase difference of 180° excluding dead time, and the switching elements Q3 and Q4 also have a phase difference of 180° excluding dead time.

[0045] The control unit 4 controls the switching elements Q5 to Q8 of the secondary switching circuit 3, which is a rectifier circuit, to turn on and off the upper arm (switching element Q5) of the third leg and the lower arm (switching element Q8) of the fourth leg at the same timing with the same drive frequency and on-duty as the switching elements Q1 to Q4, and to turn on and off the lower arm (switching element Q6) of the third leg and the upper arm (switching element Q7) of the fourth leg at the same timing. The switching elements Q5 and Q6 have a phase difference of 180° excluding dead time, and the switching elements Q7 and Q8 also have a phase difference of 180° excluding dead time.

[0046] Furthermore, the control unit 4 phase-shifts the on / off timings of the switching elements Q5 to Q8 by a phase difference θ [°] (phase shift amount θ) relative to the on / off timings of the switching elements Q1 to Q4. Specifically, the control unit 4 performs phase shift control to delay the turn-off timings of the switching elements Q6 and Q7 by the phase difference θ from the turn-on timings of the switching elements Q1 and Q4, and delay the turn-off timings of the switching elements Q5 and Q8 by the phase difference θ from the turn-on timings of the switching elements Q2 and Q3. The phase difference θ ranges from 0° to 180° (excluding dead time).

[0047] For the sake of simplicity, the dead time is omitted in FIG. 2 (this also applies to the control timing diagrams below). That is, in actual control, when the upper and lower arms of the same leg in the drive circuit are turned from off to on (for example, when the switching element Q2 is turned off and the switching element Q1 is turned on, or when the switching element Q1 is turned off and the switching element Q2 is turned on), there is a time for performing soft switching and a time for preventing through-current (for example, the time for preventing through-current of the switching elements Q1 and Q2). Similarly, when the upper and lower arms of the same leg in the rectifier circuit are turned from off to on (for example, when the switching element Q6 is turned off and the switching element Q5 is turned on, or when the switching element Q5 is turned off and the switching element Q6 is turned on), there is a time for performing soft switching and a time for preventing through-current (for example, the time for preventing through-current of the switching elements Q5 and Q6). The on-duty of the switching elements Q1 to Q8 is reduced from 50% by the dead time, for example, or the off-duty of the switching elements Q1 to Q8 is increased from 50% by the dead time, for example.

[0048] Figure 3 shows the current paths of each mode in the control timing diagram of Figure 2. In Figure 3, (A) is a diagram of the period of mode 1B in Figure 2, (B) is a diagram of the period of mode 1 in Figure 2, (C) is a diagram of the period of mode 2B in Figure 2, and (D) is a diagram of the period of mode 2 in Figure 2.

[0049] In mode 1B shown in FIG. 3(A), at the start of the mode, switching elements Q2 and Q3 are turned off and switching elements Q1 and Q4 are turned on, so that a resonant current flows through resonant circuit 2 and a current flows through the secondary side of transformer Tr. On the secondary side, switching elements Q6 and Q7 are turned on, so that a short circuit is formed by switching element Q6 and diode D8, and a short circuit is also formed by switching element Q7 and diode D5. Since the current on the secondary side flows through these short circuits, a large short-circuit current (boost current) flows through the secondary side in a short time, and on the primary side, a large resonant current flows through resonant circuit 2, and a large amount of energy is stored in resonant inductance Lr. The period of mode 1B is the boost period.

[0050] In mode 1 shown in Fig. 3(B), at the start of the mode, the switching elements Q6 and Q7 on the secondary side are turned off and the switching elements Q5 and Q8 are turned on, so the boost period ends, synchronous rectification is performed by the switching elements Q5 and Q8, and the large energy stored in the resonant inductance Lr flows to the terminals T3 and T4 as a load current. The period of mode 1 is the synchronous rectification period.

[0051] In mode 2B shown in FIG. 3(C), at the start of the mode, switching elements Q1 and Q4 are turned off and switching elements Q2 and Q3 are turned on, so that a resonant current flows in the opposite direction in resonant circuit 2, and a current flows in the opposite direction in the secondary side of transformer Tr. On the secondary side, switching elements Q5 and Q8 are turned on, so that a short circuit is formed by switching element Q5 and diode D7, and a short circuit is also formed by switching element Q8 and diode D6. Since the current on the secondary side flows through these short circuits, a large short-circuit current (boost current) flows in the secondary side in a short time, and a large resonant current flows in resonant circuit 2 on the primary side, and a large amount of energy is stored in resonant inductance Lr. The period of mode 2B is the boost period.

[0052] In mode 2 shown in Fig. 3(D), at the start of the mode, the switching elements Q5 and Q8 on the secondary side are turned off and the switching elements Q6 and Q7 are turned on, so the boost period ends, synchronous rectification is performed by the switching elements Q6 and Q7, and the large energy stored in the resonant inductance Lr flows to the terminals T3 and T4 as a load current. The period of mode 2 is the synchronous rectification period.

[0053] In this way, during the boost periods of modes 1B and 2B, a short circuit is formed on the secondary side, causing a large short circuit current (boost current) to flow, and on the primary side, a large resonant current flows in resonant circuit 2, storing a large amount of energy in resonant inductance Lr. During the subsequent synchronous rectification periods of modes 1 and 2, synchronous rectification is performed on the secondary side, causing a large load current to flow to terminals T3 and T4.

[0054] In the LLC DC / DC converter 10, the control unit 4 performs phase shift control during boost control to shift the on / off timing of the switching elements Q5 to Q8 by a phase difference θ (phase shift amount θ) relative to the on / off timing of the switching elements Q1 to Q4, thereby alternately generating boost periods and synchronous rectification periods. In the LLC DC / DC converter 10, the boost operation that forms a short circuit during the boost period and the synchronous rectification operation that outputs a load current during the synchronous rectification period are alternately performed, thereby performing the boost operation during boost control.

[0055] The phase shift amount θ [°] corresponding to the boost period of mode 1B and mode 2B may be expressed as a phase period [time] or as a phase ratio [%] (proportion to one drive cycle). Since the phase shift amount θ corresponds to the boost period [time] of the "boost pulse width Tb" described later, it may be expressed as a boost rate [%] or boost ratio as a ratio of the boost period to one drive cycle.

[0056] In this embodiment, the control timing of the switching elements Q5 to Q8 is simply phase-shifted from the control timing of the switching elements Q1 to Q4, and has the same drive frequency and on-duty as the switching elements Q1 to Q4, so it is relatively easy to generate control signals related to the control timing of the switching elements Q5 to Q8.

[0057] In this embodiment, synchronous rectification is performed after the boost operation, but if synchronous rectification is not performed without boost operation, the rectification after the boost operation will be diode bridge rectification by the diodes of the rectifier circuit. Generally, the on-resistance of a switching element is lower than the on-resistance of a diode, so performing synchronous rectification after the boost operation as in this embodiment results in lower loss and higher efficiency during rectification than performing diode bridge rectification after the boost operation.

[0058] FIG. 4 shows an example of the control timing of the switching elements Q1 to Q8 during reverse operation and boost control.

[0059] During reverse operation, the control timing is the same as during forward operation, except that the secondary side switching circuit 3 serves as a drive circuit and the primary side switching circuit 1 serves as a rectifier circuit. That is, the control unit 4 shifts the on / off timing of the switching elements Q1 to Q4 by a phase difference θ with respect to the on / off timing of the switching elements Q5 to Q8. As a result, the control unit 4 alternates between boost periods (periods of modes 1B' and 2B') and synchronous rectification periods (periods of modes 1' and 2'), causing the main circuit unit to perform a boost operation.

[0060] Figure 5 shows the current paths of each mode in the control timing diagram of Figure 4. In Figure 5, (A) is a diagram of the period of mode 1B' in Figure 4, (B) is a diagram of the period of mode 1' in Figure 4, (C) is a diagram of the period of mode 2B' in Figure 4, and (D) is a diagram of the period of mode 2' in Figure 4.

[0061] In mode 1B' shown in FIG. 5(A), at the start of the mode, switching elements Q6 and Q7 are turned off and switching elements Q5 and Q8 are turned on, so that a drive current flows to the secondary side and a resonant current flows to resonant circuit 2 on the primary side via transformer Tr. On the primary side, switching elements Q2 and Q3 are turned on, so that a short circuit is formed by switching element Q2 and diode D4, and a short circuit is also formed by switching element Q3 and diode D1. As the current on the primary side flows through these short circuits, a large resonant current flows in a short period of time and a large amount of energy is stored in resonant inductance Lr. The period of mode 1B' is the boost period.

[0062] In mode 1' shown in Fig. 5(B), at the start of the mode, the switching elements Q2 and Q3 are turned off on the primary side and the switching elements Q1 and Q4 are turned on, so the boost period ends, synchronous rectification is performed by the switching elements Q1 and Q4, and the large energy stored in the resonant inductance Lr flows to the terminals T1 and T2 as a load current. The period of mode 1' is the synchronous rectification period.

[0063] In mode 2B' shown in FIG. 5(C), at the start of the mode, switching elements Q5 and Q8 are turned off and switching elements Q6 and Q7 are turned on, so that a drive current flows in the opposite direction on the secondary side, and a resonant current flows in the opposite direction in resonant circuit 2 on the primary side via transformer Tr. On the primary side, switching elements Q1 and Q4 are turned on, so that a short circuit is formed by switching element Q1 and diode D3, and a short circuit is also formed by switching element Q4 and diode D2. The current on the primary side flows through these short circuits, so that a large resonant current flows in a short time, and a large amount of energy is stored in resonant inductance Lr. The period of mode 2B' is the boost period.

[0064] In mode 2' shown in Fig. 5(D), at the start of the mode, the switching elements Q1 and Q4 are turned off on the primary side, and the switching elements Q2 and Q3 are turned on, so that the boost period ends, synchronous rectification is performed by the switching elements Q2 and Q3, and the large energy stored in the resonant inductance Lr flows to the terminals T1 and T2 as a load current. The period of mode 2' is the synchronous rectification period.

[0065] In this way, during the boost periods of Mode 1B' and Mode 2B', a short circuit is formed on the primary side, a large resonant current (boost current) flows, and a large amount of energy is stored in the resonant inductance Lr. During the subsequent synchronous rectification periods of Mode 1' and Mode 2', synchronous rectification is performed on the primary side, and a large load current flows through terminals T1 and T2. That is, in the LLC DC / DC converter 10, a boost operation that forms a short circuit during the boost period and a synchronous rectification operation that outputs a load current during the synchronous rectification period are alternately performed, thereby performing a boost operation during boost control.

[0066] Figure 6 shows an example of the relationship between the drive frequency fs [kHz] and the boost pulse width Tb [μs] of the switching elements Q5 to Q8 when they are operated in the reverse direction with the control timing of Figure 4. The boost pulse width Tb corresponds to the phase shift amount θ in Figure 4. The current value [A] shown in the figure is an example of the current value of the load current (output current) output from terminals T1 and T2 when the input voltage V2 input to terminals T3 and T4 is V2=300 [V].

[0067] In the LLC DC / DC converter 10 during reverse operation, the input voltage V2 is controlled within the range of 150[V] to 450[V] under the control of the control unit 4, and the output voltage V1[V] output from the terminals T1 and T2 is controlled to a constant 350[V]. The resonant frequency of the resonant circuit 2 is set to 80[kHz].

[0068] When the drive frequency fs is 200 [kHz] or more, the control unit 4 performs frequency modulation control to control the output, while when the drive frequency fs is less than 200 [kHz], the control unit 4 performs boost control to control the output. When the boost control is performed, the control unit 4 performs control to increase the boost pulse width Tb as the drive frequency fs decreases. In other words, even when the drive frequency fs is less than 200 [kHz], the control unit is the drive frequency fs, and by lowering the drive frequency fs, the boost pulse width Tb is increased to perform the boost operation, thereby performing the boost control.

[0069] In the example of FIG. 6, the control unit 4 determines the drive frequency fs so that the current value of the output current output from the terminals T1 and T2 approaches a predetermined target value, and controls the boost pulse width Tb based on the control formula shown in the following formula (1).

number

[0070] The drive frequency fs is determined in a range equal to or higher than the resonant frequency and in which the value of the control formula is zero or a positive value. In the example of FIG. 6, the control unit 4 determines the drive frequency fs in a range equal to or higher than the resonant frequency of 80 [kHz] and equal to or lower than 250 [kHz]. The drive frequency fs is set to equal to or higher than the resonant frequency of 80 [kHz] because, if the drive frequency fs is equal to or higher than the resonant frequency, the input / output gain of the main circuit unit in the LLC type DC / DC converter 10 becomes 1 or less, and the boost operation becomes only the boost operation (the necessary boost ratio can be obtained only by the boost operation). As a result, the change in the input / output gain with respect to the frequency change, i.e., the boost ratio, becomes gentle and linear (linear function), making the control easier.

[0071] When the drive frequency fs is less than the resonant frequency, the input / output gain exceeds 1 due to the output characteristics of the LLC method, and the rate of change in the input / output gain relative to the frequency change is much steeper than in the region where the drive frequency fs is greater than the resonant frequency. In other words, as the drive frequency fs decreases, the change in the input / output gain due to the output characteristics of the LLC method increases, and the input / output gain value also increases. As a result, the change in the input / output gain relative to the frequency change, i.e., the boost ratio, is no longer linear (a linear function), making control difficult.

[0072] The upper limit of the drive frequency fs is set to 250 kHz because if the drive frequency fs is made too high (exceeding 250 kHz), losses will increase in the switching elements Q1 to Q8 as well as in the transformer Tr and resonant inductance Lr, resulting in increased heat generation.

[0073] In the LLC DC / DC converter 10, the required boost ratio can be obtained when the drive frequency fs is within a certain range (a range of 80 [kHz] or more and 250 [kHz] or less). In the example of Fig. 6, when V2 = 300 [V], the boost pulse width Tb increases with a constant gain as the drive frequency fs decreases, with Tb = 0.28 [μs] at fs = 180 [kHz], Tb = 0.63 [μs] at fs = 160 [kHz], and Tb = 1.07 [μs] at fs = 140 [kHz]. As a result, the load current increases to 3.0 [A], 11.6 [A], and 20.9 [A].

[0074] When the drive frequency fs is less than 200 [kHz] and boosting is required, the control unit 4 starts boost control, reduces the drive frequency fs, and increases the boost pulse width Tb. Since the control unit 4 can perform output control with the drive frequency fs as the control unit during boost control as well as during frequency modulation control, there is no need to switch the control unit, and control is simplified compared to a conventional LLC type DC / DC converter in which the drive frequency is fixed and output is controlled only by the boost pulse width during boost control.

[0075] Furthermore, because the output increases due to the output characteristics of the LLC system accompanying a decrease in drive frequency fs, the LLC DC / DC converter 10 can suppress the boost current compared to a conventional LLC DC / DC converter that fixes the drive frequency during boost control and controls the output only by the boost pulse width. In other words, the LLC DC / DC converter 10 has the advantage of being able to increase (boost) the output with a smaller boost pulse width Tb and a smaller boost current.

[0076] FIG. 7 shows an example of the relationship between the drive frequency fs [kHz] and the boost pulse width Tb [μs] of the switching elements Q1 to Q4 when they are operated in the forward direction with the control timing of FIG. 2. The boost pulse width Tb corresponds to the phase shift amount θ in FIG. 2. The current value [A] shown in FIG. 7(A) is the current value of the load current (output current) output from terminals T3 and T4 when the output voltage V2 output from terminals T3 and T4 is V2=350 [V]. The current value [A] shown in FIG. 7(B) is the current value of the load current when V2=450 [V].

[0077] In the LLC DC / DC converter 10 during forward operation, the input voltage V1 input to the terminals T1 and T2 is controlled to a constant 350 [V], and the output voltage V2 is controlled within the range of 150 [V] to 450 [V] under the control of the control unit 4. The resonant frequency of the resonant circuit 2 is 80 [kHz].

[0078] When the drive frequency fs is 200 kHz or more, the control unit 4 controls the output by performing frequency modulation control, while when the drive frequency fs is less than 200 kHz, the control unit 4 controls the output by performing boost control. During boost control, the control unit 4 performs control to increase the boost pulse width Tb as the drive frequency fs decreases.

[0079] In the example of FIG. 7, the control unit 4 determines the drive frequency fs so that the current value of the output current output from the terminals T3 and T4 approaches a predetermined target value, and controls the boost pulse width Tb based on the control formula shown in the following formula (2).

number

[0080] The drive frequency fs is determined in the range of 80 [kHz] or more of the resonant frequency to less than 200 [kHz] where the value of the above control formula becomes positive, and the output voltage V2 is controlled in the range of more than 300 [V] to 450 [V] (corresponding to the upper limit voltage of the battery). Note that when the output voltage V2 is 150 [V] or less than 300 [V], the step-down operation is performed and there is no need to perform boost control, which is a step-up operation, so frequency modulation control is performed at a drive frequency of 200 [kHz] or more.

[0081] As shown in Figure 7(A), when V2=350[V], the boost pulse width Tb increases with a constant gain as the drive frequency fs decreases: Tb=0.11[μs] at fs=180[kHz], Tb=0.18[μs] at fs=170[kHz], Tb=0.25[μs] at fs=160[kHz], Tb=0.33[μs] at fs=150[kHz], Tb=0.43[μs] at fs=140[kHz], and Tb=0.54[μs] at fs=130[kHz]. As a result, the load current increases from 1.1[A] to 1.6[A], 2.9[A], 5.2[A], 7.9[A], and 11.9[A].

[0082] As shown in FIG. 7B, in the case of V2=450[V], the boost pulse width Tb increases with a constant gain as the drive frequency fs decreases, i.e., Tb=0.33[μs] at fs=180[kHz], Tb=0.53[μs] at fs=170[kHz], Tb=0.75[μs] at fs=160[kHz], and Tb=1.0[μs] at fs=150[kHz]. As a result, the load current increases significantly to 1.3[A], 3.1[A], 6.2[A], and 11.0[A], compared to the case of V2=350[V]. That is, in the LLC DC / DC converter 10, the boost pulse width Tb increases for the same drive frequency fs as the output voltage V2 (input / output voltage ratio or boost ratio) increases. The input / output voltage ratio (boost ratio) can be expressed, for example, as the output voltage / input voltage.

[0083] In addition to the advantages of reverse operation and boost control, the LLC DC / DC converter 10 during forward operation and boost control has the following advantage: That is, in the LLC DC / DC converter 10, when the output voltage V2 increases during forward operation and boost control, the boost pulse width Tb increases and the gain of the boost control changes depending on the value of the output voltage V2 (input / output voltage ratio or boost ratio), so that even when the output voltage V2 is high and a larger input / output voltage ratio (boost ratio) is required, a load current commensurate with the input / output voltage ratio (boost ratio) can be obtained with a boost pulse width Tb suitable for control.

[0084] In the LLC DC / DC converter 10, whether it outputs a low output voltage V2 or a high output voltage V2, it is possible to obtain an appropriate boost pulse width Tb within a certain driving frequency range according to the input / output voltage ratio. That is, compared to the case where the boost control gain is not changed according to the input / output voltage ratio (boost ratio), it is possible to obtain an appropriate boost pulse width Tb within a narrower driving frequency range. Therefore, in the LLC DC / DC converter 10, it is not necessary to search for an optimal operating point for the input / output voltage ratio and the boost pulse width Tb, and it is possible to handle a wider range of input / output voltage ratios.

[0085] 8 shows an example of a control flow during reverse operation of the LLC DC / DC converter 10. Reverse operation corresponds to the operation during discharging of the battery connected to terminals T3 and T4.

[0086] The control unit 4, which has started the control, acquires output information on the output current output from the terminals T1 and T2 and target information on the target value (target current) of the output current (S101). The target value of the target current may be, for example, a current value preset by the control unit 4, a current value calculated by the control unit 4 from the input / output conditions of the main circuit unit, or a current value input to the control unit 4 from an external device. The current value of the output current is, for example, a current value detected by a detection unit provided at the terminal T1 or the terminal T2. Note that the target information may be information on a target voltage and / or a target power instead of or in addition to the target current. The output information may be information on an output voltage and / or an output power instead of or in addition to the output current.

[0087] The control unit 4 compares the target current with the output current, and determines the drive frequency fs of the switching elements Q5 to Q8 so that the output current approaches the target current, thereby performing frequency modulation control (S102). When the output current is smaller than the target current, the control unit 4 during frequency modulation control decreases the drive frequency fs to increase the output current, whereas when the output current is larger than the target current, it increases the drive frequency fs to decrease the output current. Therefore, when the load is heavy and the output current value is large, the drive frequency fs is low, and when the load is light and the output current value is small, the drive frequency fs is high.

[0088] The control unit 4 determines whether the current drive frequency fs (the drive frequency fs determined in step S102) is less than 200 [kHz] (S103). 200 [kHz] is a boost start frequency preset in the control unit 4. If the drive frequency fs is less than 200 [kHz] (Yes in S103), the control unit 4 calculates the boost pulse width Tb based on the drive frequency fs and the control formula (1) and performs boost control (S104).

[0089] Next, the control unit 4 judges whether or not to continue the control (S105). The control unit 4 performs the judgment of step S105 based on, for example, whether or not a control end command has been received from an external device. If the control unit 4 has not received the control end command, it judges to continue the control (Yes in S105), returns to step S101, and repeats the processes from step S101 onwards. As a result, when boost control is performed, the boost pulse width Tb is controlled together with the drive frequency fs so that the output current coincides with the target current. Having received the control end command, the control unit 4 judges not to continue the control (No in S105), and ends the control.

[0090] 9 shows an example of a control flow during forward operation of the LLC DC / DC converter 10. The forward operation corresponds to the operation during charging of the battery connected to terminals T3 and T4.

[0091] After starting the control, the control unit 4 acquires output information regarding the output current output from the terminals T3 and T4 and the output voltage V2, and target information regarding the target value (target current) of the output current (S201).

[0092] The control unit 4, which has acquired the target information and output information, compares the target current with the output current, and determines the drive frequency fs of the switching elements Q1 to Q4 so that the output current approaches the target current, thereby performing frequency modulation control (S202). When the output current is smaller than the target current, the control unit 4 during frequency modulation control decreases the drive frequency fs to increase the output current, whereas when the output current is larger than the target current, it increases the drive frequency fs to decrease the output current. Therefore, when the load is heavy and the output current value is large, the drive frequency fs is low, and when the load is light and the output current value is small, the drive frequency fs is high.

[0093] Next, the control unit 4 judges whether the current output voltage V2 (the output voltage V2 acquired in step S201) exceeds a preset threshold value (300 [V] in this embodiment) (S203). If the output voltage V2 exceeds 300 [V] (Yes in S203), the control unit 4 judges whether the current drive frequency fs (the drive frequency fs determined in step S202) is less than 200 [kHz] (S204). If the drive frequency fs is less than 200 [kHz] (Yes in S204), the control unit 4 calculates the boost pulse width Tb based on the drive frequency fs and the control formula of Equation (2) to perform boost control (S205).

[0094] If the output voltage V2 is 300 [V] or less in step S203 (No in S203), if the drive frequency fs is 200 [kHz] or more in step S204 (No in S204), and if the boost control is performed in step S205, the control unit 4 determines whether or not to continue the control (S206), similar to step S105 during reverse direction operation. If the control unit 4 determines to continue the control (Yes in S206), it repeats the processing from step S201 onward, and if it determines not to continue the control (No in S206), it ends the control. As a result, when the boost control is performed, the boost pulse width Tb is controlled together with the drive frequency fs so that the output current coincides with the target current.

[0095] In the LLC DC / DC converter 10, the drive frequency fs is reduced as the output increases even during boost control, so that the output rises due to the output characteristics of the LLC system associated with the reduction in the drive frequency fs. Therefore, compared to a conventional LLC DC / DC converter that fixes the drive frequency during boost control and controls the output only by the boost pulse width, the LLC DC / DC converter 10 can suppress the boost current. That is, the LLC DC / DC converter 10 can increase (boost) the output with a smaller boost pulse width Tb and a smaller boost current. Furthermore, the LLC DC / DC converter 10 can obtain an appropriate boost pulse width Tb within a certain drive frequency range according to the input / output voltage ratio (boost ratio) whether it outputs a low output voltage V2 or a high output voltage V2. Therefore, the LLC DC / DC converter 10 does not need to search for an optimal operating point of the input / output voltage ratio and the boost pulse width Tb, and can accommodate a wider range of input / output voltage ratios.

[0096] In this embodiment, the resonant circuit 2 is provided on the primary side, but the resonant circuit 2 may be provided on the secondary side instead of the primary side. In addition, either or both of the resonant inductance Lr and the resonant capacitance Cr may be divided and arranged on the primary side and the secondary side. For example, the resonant inductance Lr of the resonant circuit 2 may be divided and arranged on the primary side and the secondary side, or the divided resonant inductance Lr on the primary side and the secondary side may be transformer-coupled to the transformer Tr. Here, when comparing the inductance value when the resonant inductance Lr of the resonant circuit 2 is divided and arranged on the primary side and the secondary side with the inductance value in the case of the CLLC method having the resonant circuits 2c and 5c on the primary side and the secondary side as shown in FIG. 16, the inductance value can be made smaller in the former case. Dividing the resonant inductance Lr increases the mounting area, but the inductance value is reduced, so that a smaller coil is required than when it is realized with a single coil, which has the advantages of easiness of procurement, cost, size, heat dissipation, etc. Therefore, even if the resonant inductance Lr of the resonant circuit 2 is divided and arranged on the primary side and the secondary side, the problems of the increase in size and cost of the main circuit due to the increase in the number of parts, and the increase in loss (reduction in power conversion efficiency) due to the resonant circuit on the rectifier circuit side are solved. Furthermore, if the resonant capacitance Cr is divided into the primary side and the secondary side, the mounting area increases, but since the DC component can be cut, it can be used in common with the capacitor for preventing bias magnetism of the transformer Tr. In the case of the LLC type DC / DC converter 10A according to the first modified example described later, the inductance value can also be made smaller than that of the CLLC type.

[0097] In this embodiment, the control unit 4 shifts the on / off timing of the switching elements Q5 to Q8 by the phase difference θ with respect to the on / off timing of the switching elements Q1 to Q4 to short-circuit the secondary switching circuit 3 during the period of the phase difference θ (boost period) to allow the boost current to flow, but the on / off timing of the switching elements Q5 to Q8 that creates the boost period can be changed as appropriate. That is, the control unit 4 may control the switching elements of the rectifier circuit so that a short circuit is formed in the rectifier circuit only during the boost period.

[0098] For example, in mode 1B shown in FIG. 3A, the switching element Q6 may be turned off to form a short circuit only with the switching element Q7 and diode D5 of the upper arm, or the switching element Q7 may be turned off to form a short circuit only with the switching element Q6 and diode D8 of the lower arm. Alternatively, the switching element Q6 may be turned off and the switching elements Q5 and Q7 may be turned on to form a short circuit only with the switching elements Q5 and Q7 of the upper arm to reduce the conduction loss of the short circuit, or the switching element Q7 may be turned off and the switching elements Q6 and Q8 may be turned on to form a short circuit only with the switching elements Q6 and Q8 of the lower arm to reduce the conduction loss of the short circuit. In this way, both switching elements constituting the short circuit are turned on, thereby reducing the conduction loss of the boost current flowing through the short circuit and improving efficiency.

[0099] 3B, for example, the current flowing through the third leg or the fourth leg may be detected, and the switching elements Q5, Q8 may be turned on only during a period when the synchronous rectification current flowing through the switching elements Q5, Q8 is positive, and when the synchronous rectification current becomes equal to or less than a certain value or becomes 0, the switching elements Q5, Q8 may be turned off. Note that the period when the synchronous rectification current is positive may be detected by other methods, such as detecting the voltage of the switching elements Q5, Q8, in addition to the method of detecting the current.

[0100] In addition, although switching element Q5 is turned on in mode 1, rectification may be performed by turning off switching element Q5 and making diode D5 conductive. If only forward operation is required, the upper arms of the third leg and the upper arms of the fourth leg may be replaced with diodes only, rather than switching elements Q5 and Q7.

[0101] The same applies to mode 2B shown in FIG. 3(C) and mode 2 shown in FIG. 3(D) as described above for mode 1B and mode 1.

[0102] In this embodiment, the control unit 4 during reverse direction operation controls the boost pulse width Tb based on the control formula shown in formula (1), but this is not limited thereto, and the boost pulse width Tb may be controlled based on the control formula shown in the following formula (3) (corresponding to the "first formula" of the present invention). When the control formula shown in formula (3) is used, the drive frequency fs is determined within a range that is equal to or higher than the resonant frequency of the resonant circuit 2 and in which the value of the control formula is zero or a positive value.

number

[0103] The coefficient a and constant b are preset based on the constant of the resonant circuit 2 and the input / output voltage ratio (boost ratio) of the main circuit. The coefficient a corresponds to the inverse proportional ratio to the drive frequency fs, that is, the ratio (gain) of the increase in the boost pulse width Tb relative to the decrease in the drive frequency fs. The constant b is obtained from the boost start frequency [kHz], which is the drive frequency fs at which boost control starts, and the coefficient a, using the following formula (4).

number

[0104] In this embodiment, the control unit 4 during forward operation controls the boost pulse width Tb based on the control formula shown in formula (2), but this is not limited thereto, and the boost pulse width Tb may be controlled based on the control formula shown in the following formula (5) (corresponding to the "second formula" of the present invention). When the control formula shown in formula (5) is used, the drive frequency fs is determined within a range that is equal to or higher than the resonant frequency of the resonant circuit 2 and in which the value of the control formula is zero or a positive value.

number

[0105] The coefficient a and the constant b are preset according to the constant of the resonant circuit 2 and the input / output voltage ratio (step-up ratio) of the main circuit, as in the formula (4). G is a gain related to the input / output voltage ratio (step-up ratio), and is a function that increases the boost pulse width Tb in proportion to the output voltage V2 of 300 [V] or more at which the boost control starts, as shown in the formula (2), for example. In the case of the formula (2), the output voltage V2 is increased as a percentage of the minimum voltage of 150 [V].

[0106] In addition, the control equations of formulas (1) and (2) are expressed as equations inversely proportional to the drive frequency fs, but they may be expressed as equations negatively proportional to the drive frequency fs. The same is true for the control equations of formulas (3) and (5). That is, the control equation of formula (6) below (corresponding to the "third equation" of the present invention) may be used instead of the control equation of formula (3), and the control equation of formula (7) below (corresponding to the "fourth equation" of the present invention) may be used instead of the control equation of formula (5).

number

number

[0107] The LLC type DC / DC converter 10 can increase (boost) the output with a smaller boost pulse width Tb than conventional LLC type DC / DC converters, so no upper limit is set for the boost pulse width Tb. However, an upper limit may be set for the boost pulse width Tb, and control may be performed so that the boost pulse width Tb does not exceed the upper limit even if the drive frequency fs drops, thereby limiting the boost current.

[0108] In the example of FIG. 6, the output voltage V1 during reverse operation is constant at 350 [V], and in the example of FIG. 7, the input voltage V1 during forward operation is constant at 350 [V], but this is not limited thereto. The control unit 4 may control the input / output voltage ratio (step-up ratio) between the voltages V1 and V2 so as to vary the voltage V1 in accordance with the variation of the voltage V2. In this case, the control unit 4 performs boost control so as to vary the boost pulse width Tb according to the input / output voltage ratio. That is, for example, when the input voltage V2 is 300 [V] during reverse operation, if the output voltage V1 is changed from 350 [V] to 320 [V], the input / output voltage ratio (step-up ratio) decreases compared to when the input voltage is 350 [V], so that the boost pulse width Tb can be made smaller than that in FIG. 6. Furthermore, when the output voltage V2 during forward operation is 450[V], if the input voltage V1 is increased from 350[V] to 380[V], the input / output voltage ratio (boost ratio) will decrease compared to when the input voltage is 350[V], and the boost pulse width Tb can be made smaller than that in Figure 7(B).

[0109] In the examples of Figures 6 and 7, boost control is performed at a drive frequency fs of less than 200 [kHz] during both reverse and forward operation, but boost control may be started at a different drive frequency fs. Furthermore, in the example of Figure 7, boost control is performed at a drive frequency fs of less than 200 [kHz] when the output voltage V2 (battery voltage) is 350 [V] or 450 [V], but the drive frequency fs at which boost control is started may be changed depending on the output voltage V2 (input / output voltage ratio or boost ratio).

[0110] For example, when the output voltage V2 is lower than a predetermined value (e.g., 350 [V]), boost control may be started when the drive frequency fs is less than 170 [kHz]. When the input / output voltage ratio (boost ratio) is low, even if the boost start frequency is lowered to widen the frequency range in which frequency modulation control is performed (even if the frequency range in which boost control is performed is narrowed), control can be performed without significantly changing the increase rate of the boost pulse width Tb relative to the decrease in the drive frequency fs. In other words, even if the boost start frequency is lowered, control can be performed without significantly changing the gain during boost control.

[0111] In the examples of Figures 6 and 7, frequency modulation control is performed when the drive frequency fs is 200 kHz or higher, but this is not limited to frequency modulation control, and other controls such as phase shift control, PWM control, burst (intermittent) control, etc. may be performed, or a combination of frequency modulation control and other controls may be performed.

[0112] In the example of Fig. 9, the boost pulse width Tb is calculated using the drive frequency fs and the output voltage V2 (battery voltage) based on the control formula (2), but the present invention is not limited to this. For example, since the change in the battery voltage during charging is small, a gain related to the input / output voltage ratio may be determined in advance for each predetermined range of the output voltage V2 (battery voltage), and the gain may be stored in advance in the control unit 4. The control unit 4 can calculate the boost pulse width Tb based on the stored gain.

[0113] Although the embodiments of the LLC type DC / DC converter according to the present invention have been described above, the present invention is not limited to the above-mentioned embodiments.

[0114] [First Modification] Fig. 10 shows an LLC DC / DC converter 10A having a half-bridge circuit configuration according to a first modified example. The LLC DC / DC converter 10A is a current-resonant LLC DC / DC converter capable of bidirectional operation, and includes a pair of terminals T1, T2, a capacitor C11, a primary-side switching circuit 1a, a resonant circuit 2a, a transformer Tr, a secondary-side switching circuit 3a, a capacitor C21, a pair of terminals T3, T4, and a control unit 4a. The above configuration except for the control unit 4a corresponds to the "main circuit unit" of the present invention. The terminals T1 to T4, the capacitors C11, C21, and the transformer Tr are the same as those in the above embodiment.

[0115] The primary side switching circuit 1a and the secondary side switching circuit 3a are the same as those in the above embodiment, except that they are configured as half-bridge circuits.

[0116] The resonant circuit 2a includes a resonant inductance Lr and resonant capacitances Cr1 and Cr2 on the primary side, and resonant capacitances Cr3 and Cr4 on the secondary side. Note that the resonant circuit 2a also includes an exciting inductance of a transformer Tr (not shown).

[0117] The resonant inductance Lr and the resonant capacitance Cr1 form a series resonant circuit between the primary winding of the transformer Tr and the terminal T1, and the resonant inductance Lr and the resonant capacitance Cr2 form a series resonant circuit between the primary winding of the transformer Tr and the terminal T2. The resonant capacitances Cr1 and Cr2 divide the voltage V1.

[0118] The resonant capacitance Cr3 forms a series resonant circuit together with an inductance obtained by transforming the resonant inductance Lr to the secondary side at the turns ratio of the transformer Tr. Similarly, the resonant capacitance Cr4 forms a series resonant circuit together with an inductance obtained by transforming the resonant inductance Lr to the secondary side at the turns ratio of the transformer Tr. The resonant capacitances Cr3 and Cr4 divide the voltage V2.

[0119] With the above configuration, the resonant circuit 2a can perform power transmission by the LLC method (LLC resonant operation) not only during forward operation from the primary side to the secondary side, but also during reverse operation from the secondary side to the primary side.

[0120] The control unit 4a includes a processing unit that generates a control signal for controlling the on / off of the switching elements Q1 to Q4, and a driving unit that turns the switching elements Q1 to Q4 on and off based on the control signal. The control unit 4a may further include a detection unit. The configurations of the processing unit, the driving unit, and the detection unit are the same as those of the above embodiment.

[0121] The control unit 4a performs frequency modulation control to modulate the drive frequency fs of the switching elements Q1 to Q4 as on / off control, as in the above embodiment. When the drive frequency fs during frequency modulation control is less than a preset boost start frequency, the control unit 4a performs boost control to short-circuit the rectifier circuit and cause the main circuit unit to perform a boost operation, as in the above embodiment.

[0122] FIG. 11 shows an example of the control timing of the switching elements Q1 to Q4 during forward operation and boost control.

[0123] The control unit 4a turns on and off the switching elements Q1 and Q2 of the primary switching circuit 1a on the drive circuit side with the same drive frequency, the same on-duty (for example, 50%), and a phase difference of 180°. The control unit 4a also turns on and off the switching elements Q3 and Q4 of the secondary switching circuit 3a on the rectifier circuit side with the same drive frequency, the same on-duty, and a phase difference of 180° as the switching elements Q1 and Q2. Furthermore, the control unit 4a phase-shifts the on / off timing of the switching elements Q3 and Q4 by the phase difference θ with respect to the on / off timing of the switching elements Q1 and Q2. Note that the dead time is omitted to simplify the explanation, as in the above embodiment.

[0124] Figure 12 shows the current path of each mode in the control timing diagram of Figure 11. In Figure 12, (A) is a diagram of the period of mode 1B in Figure 11, (B) is a diagram of the period of mode 1 in Figure 11, (C) is a diagram of the period of mode 2B in Figure 11, and (D) is a diagram of the period of mode 2 in Figure 11.

[0125] In mode 1B shown in Fig. 12(A), the switching element Q1 is turned on at the start of the mode, so that a resonant current flows through the resonant inductance Lr and the resonant capacitance Cr2, and a current flows on the secondary side of the transformer Tr. On the secondary side, the switching element Q4 is turned on, so that a short circuit is formed by the switching element Q4 and the resonant capacitance Cr4. As a result, a large short-circuit current (boost current) flows through the short circuit, and a large amount of energy is stored in the resonant inductance Lr on the primary side.

[0126] In mode 1 shown in FIG. 12(B), at the start of the mode, switching element Q4 on the secondary side is turned off and switching element Q3 is turned on, so that synchronous rectification is performed by switching element Q3 and the large energy stored in resonant inductance Lr flows to terminals T3 and T4 as a load current.

[0127] In mode 2B shown in Fig. 12(C), switching element Q2 is turned on at the start of the mode, so that a resonant current flows in the opposite direction through resonant inductance Lr and resonant capacitance Cr1, and a reverse current flows in the secondary side of the transformer Tr. On the secondary side, switching element Q3 is turned on, so that a short circuit is formed by switching element Q3 and resonant capacitance Cr3. As a result, a large short-circuit current (boost current) flows in the short circuit, and a large amount of energy is stored in resonant inductance Lr on the primary side.

[0128] In mode 2 shown in FIG. 12(D), at the start of the mode, switching element Q3 is turned off and switching element Q4 is turned on on the secondary side, so that synchronous rectification is performed by switching element Q4 and the large energy stored in resonant inductance Lr flows to terminals T3 and T4 as a load current.

[0129] In this way, during the boost periods of modes 1B and 2B, a short circuit is formed on the secondary side, causing a large short circuit current (boost current) to flow, and a large amount of energy is stored in the resonant inductance Lr on the primary side. During the subsequent synchronous rectification periods of modes 1 and 2, synchronous rectification is performed on the secondary side, causing a large current to flow through terminals T3 and T4.

[0130] In the LLC DC / DC converter 10A, the control unit 4a performs phase shift control during boost control to shift the on / off timing of the switching elements Q3 and Q4 by a phase difference θ (phase shift amount θ) relative to the on / off timing of the switching elements Q1 and Q2, thereby generating boost periods and synchronous rectification periods alternately. In the LLC DC / DC converter 10A, the boost operation that forms a short circuit during the boost period and the synchronous rectification operation that outputs a load current during the synchronous rectification period are performed alternately, thereby performing the boost operation during boost control.

[0131] Since the LLC DC / DC converter 10A performs boost control in the same manner as in the above embodiment, the control flows shown in Figs. 8 and 9 can also be applied to the LLC DC / DC converter 10A.

[0132] In the LLC DC / DC converter 10A, even during boost control, the boost pulse width Tb is changed to increase in accordance with the reduction in the drive frequency fs, so that control can be performed only with the drive frequency fs. Also, even during boost control, the drive frequency fs is reduced as the output increases, so the output rises due to the output characteristics of the LLC system associated with the reduction in the drive frequency fs. Therefore, compared to a conventional LLC DC / DC converter that fixes the drive frequency during boost control and controls the output only with the boost pulse width, the LLC DC / DC converter 10A can suppress the boost current. Note that, even during reverse operation from the secondary side to the primary side, power transmission by the LLC system (LLC resonant operation) can be performed, so that a short circuit is formed on the primary side, a large short circuit current (boost current) flows, and a large energy is stored in the resonant inductance Lr converted to the secondary side. During the subsequent synchronous rectification period, synchronous rectification is performed on the primary side, and a large current similarly flows through the terminals T1 and T2.

[0133] [Second modified example] Fig. 13 shows an LLC DC / DC converter 10B having a three-phase bridge circuit configuration according to a second modified example. The LLC DC / DC converter 10B is a current-resonant LLC DC / DC converter capable of bidirectional operation, and includes a pair of terminals T1, T2, a capacitor C11, a primary-side switching circuit 1b, a resonant circuit 2b, a transformer circuit (transformers Tr1 to Tr3), a secondary-side switching circuit 3b, a capacitor C21, a pair of terminals T3, T4, and a control unit 4b. The above configuration except for the control unit 4b corresponds to the "main circuit unit" of the present invention. The terminals T1 to T4 and the capacitors C11, C21 are the same as those in the above embodiment.

[0134] The primary side switching circuit 1b includes first, second, and third phase legs connected in parallel, and the secondary side switching circuit 3b also includes first, second, and third phase legs connected in parallel. The primary side switching circuit 1b and the secondary side switching circuit 3b are the same as those in the above embodiment, except that they are configured as a three-phase bridge circuit.

[0135] The resonant circuit 2b includes a first-phase resonant circuit including a resonant inductance Lr1 and a resonant capacitance Cr1, a second-phase resonant circuit including a resonant inductance Lr2 and a resonant capacitance Cr2, and a third-phase resonant circuit including a resonant inductance Lr3 and a resonant capacitance Cr3. One end of the resonant circuit of each phase is connected to a connection point of the upper and lower arms constituting each leg of the primary-side switching circuit 1b, and the other end of the resonant circuit of each phase is connected to each other (balanced point connection). In FIG. 13, the resonant circuits are connected in a Y-connection, but other balanced point connections such as a Δ-connection may also be used. That is, the resonant circuit 2b is connected between the primary-side switching circuit 1b and a transformer circuit (transformers Tr1 to Tr3), and the primary-side switching circuit 1b is provided on the primary side of the transformer circuit.

[0136] The transformer circuit includes a first-phase transformer Tr1, a second-phase transformer Tr2, and a third-phase transformer Tr3. The exciting inductances of the transformers Tr1 to Tr3 are omitted from the drawing since they are included in the transformers Tr1 to Tr3. The primary windings of the transformers Tr1 to Tr3 are connected to the primary-side switching circuit 1b via the resonant circuit 2b. One end of the secondary windings of the transformers Tr1 to Tr3 is connected to a connection point of the upper and lower arms constituting each leg of the secondary-side switching circuit 3b, and the other ends are connected to each other (balanced point connection). In FIG. 13, the transformers are connected in a Y-connection, but other balanced point connections such as a Δ-connection may be used.

[0137] The control unit 4b includes a processing unit that generates a control signal for controlling the on / off of the switching elements Q1 to Q12, and a driving unit that turns the switching elements Q1 to Q12 on and off based on the control signal. The control unit 4b may further include a detection unit. The configurations of the processing unit, the driving unit, and the detection unit are the same as those of the above embodiment.

[0138] The control unit 4b performs frequency modulation control to modulate the drive frequency fs of the drive circuit of the switching elements Q1 to Q12 as the on / off control, as in the above embodiment. When the drive frequency fs during frequency modulation control is less than a preset boost start frequency, the control unit 4b performs boost control to short-circuit the rectifier circuit and cause the main circuit unit to perform a boost operation, as in the above embodiment.

[0139] 14 shows an example of the control timing of switching elements Q1 to Q12 during forward operation and boost control. If the dead time is not taken into consideration, switching element Q2 will have a phase difference of 180° with respect to switching element Q1, switching element Q4 will have a phase difference of 180° with respect to switching element Q3, and switching element Q6 will have a phase difference of 180° with respect to switching element Q5, so the control timing of switching elements Q2, Q4, and Q6 is omitted.

[0140] The control unit 4b turns the switching elements Q1 to Q6 on and off at the same drive frequency and the same on-duty (e.g., 50%), fixes the phase difference between the switching elements Q1, Q2 and the switching elements Q3, Q4 to 120°, fixes the phase difference between the switching elements Q3, Q4 and the switching elements Q5, Q6 to 120°, and fixes the phase difference between the switching elements Q5, Q6 and the switching elements Q1, Q2 to 120°.

[0141] Furthermore, the control unit 4b turns on and off the switching elements Q7 to Q12 with the same drive frequency, the same on-duty as the switching elements Q1 to Q6, and with a phase shift of the phase difference θ. In Fig. 14, the control unit 4b phase-shifts the on-off timing of the switching elements Q7 and Q8 by the phase difference θ with respect to the on-off timing of the switching elements Q1 and Q2, phase-shifts the on-off timing of the switching elements Q9 and Q10 by the phase difference θ with respect to the on-off timing of the switching elements Q3 and Q4, and phase-shifts the on-off timing of the switching elements Q11 and Q12 by the phase difference θ with respect to the on-off timing of the switching elements Q5 and Q6.

[0142] Fig. 15 shows the current paths in each mode in the control timing diagram of Fig. 14. In Fig. 15, (A) is a diagram of the period of mode 1B in Fig. 14, and (B) is a diagram of the period of mode 1 in Fig. 14.

[0143] In mode 1B shown in Fig. 15(A), when the mode starts, switching element Q1 is turned on while switching element Q4 is on, so that a resonant current flows on the primary side due to the first phase resonant circuit (resonant inductance Lr1, resonant capacitance Cr1, and excitation inductance of transformer Tr1). On the secondary side, switching elements Q8 and Q10 are on, so that a short circuit is formed by switching elements Q8 and Q10. As a result, a large short-circuit current (boost current) flows in the short circuit, and a large amount of energy is stored in resonant inductance Lr1 on the primary side.

[0144] In mode 1 shown in FIG. 15(B), at the start of the mode, switching element Q8 is turned off and switching element Q7 is turned on, so that synchronous rectification is performed by switching elements Q7, Q10, and Q11, and the large energy stored in resonant inductance Lr1 flows to terminals T3 and T4 as a load current.

[0145] Similarly, in mode 3B, a short-circuit current (boost current) flows through the short circuit formed on the secondary side, causing a large amount of energy to accumulate in the resonant inductance Lr2 on the primary side. In mode 3, synchronous rectification is performed on the secondary side, and the large amount of energy accumulated in the resonant inductance Lr2 flows to terminals T3 and T4 as a load current.

[0146] Similarly, in mode 5B, a short-circuit current (boost current) flows through the short circuit formed on the secondary side, causing a large amount of energy to accumulate in the resonant inductance Lr3 on the primary side. In mode 5, synchronous rectification is performed on the secondary side, and the large amount of energy accumulated in the resonant inductance Lr3 flows to terminals T3 and T4 as a load current.

[0147] In the LLC DC / DC converter 10B, the control unit 4b performs phase shift control during boost control to shift the on / off timings of the switching elements Q7, Q8, Q9, Q10, Q11, and Q12 by a phase difference θ relative to the on / off timings of the switching elements Q1, Q2, Q3, Q4, and Q5, Q6, respectively, to generate a boost period and a synchronous rectification period. In the LLC DC / DC converter 10B, a boost operation that forms a short circuit during the boost period and a synchronous rectification operation that outputs a load current during the synchronous rectification period are performed, thereby performing a boost operation during boost control.

[0148] As described above, the LLC DC / DC converter 10B performs boost control in the same manner as in the above embodiment. That is, the control flows shown in Figures 8 and 9 can also be applied to the LLC DC / DC converter 10B.

[0149] Therefore, in the LLC DC / DC converter 10B, even during boost control, the boost pulse width Tb is changed to increase in accordance with the reduction in the drive frequency fs, so control can be performed using only the drive frequency fs. Also, even during boost control, the drive frequency fs is reduced as the output increases, so the output increases due to the output characteristics of the LLC system as the drive frequency fs decreases. Therefore, compared to a conventional LLC DC / DC converter that fixes the drive frequency during boost control and controls the output only using the boost pulse width, the LLC DC / DC converter 10B can suppress the boost current.

[0150] [Other variations] The LLC type DC / DC converter of the present invention is an LLC type DC / DC converter including a main circuit unit including a transformer circuit, a primary side switching circuit including a primary side switching element provided on the primary side of the transformer circuit, a secondary side switching circuit including a secondary side switching element provided on the secondary side of the transformer circuit, and a resonant circuit including a resonant inductance and a resonant capacitance, and a control unit that generates control signals for performing on / off control of the primary side switching element and the secondary side switching element, wherein the control unit performs boost control to cause the main circuit unit to perform a boost operation, and during boost control, when the main circuit unit transmits power from the primary side switching circuit to the secondary side switching circuit, the control unit can be appropriately configured as long as it turns on the secondary side switching element to form a short circuit in the secondary side switching circuit and changes the boost pulse width to increase in accordance with the reduction in drive frequency.

[0151] For example, the primary side switching circuit and / or the secondary side switching circuit may be configured as a multi-phase bridge circuit with three or more phases. Also, the half-bridge circuit, full-bridge circuit, and multi-phase bridge circuit configuring the primary side switching circuit and / or the secondary side switching circuit may be configured as a multi-level (e.g., three-level) circuit in which each arm includes a plurality of switching elements. Also, the rectifier circuit (e.g., the secondary side switching circuit) may be a rectifier circuit with a bridgeless configuration (also called a bridgeless rectifier circuit or bridgeless circuit) in which the switching elements configuring the upper arm of the rectifier circuit are replaced with diodes.

[0152] Regarding the method of shorting the rectifier circuit during boost control, the conduction loss of the boost current can be reduced by forming a short circuit by turning on two switching elements in a bridge rectifier circuit or one switching element in a bridgeless rectifier circuit.

[0153] In the above embodiment, the period of the phase difference θ by the phase shift control signal that phase-shifts the control timing of the rectifier circuit by the phase difference θ is set as the boost period, but the control unit may separately generate a pulse width (on-duty) control signal for generating the boost period (forming a short circuit) and a pulse width (on-duty) control signal for performing synchronous rectification after the boost period in the control signal for turning on and off the switching element of the rectifier circuit, and may turn off the switching element of the rectifier circuit after the boost period to perform diode bridge rectification. In addition, the control unit may measure the synchronous rectification current after the phase shift, and turn off the switching element that performs synchronous rectification when the synchronous rectification current becomes a certain value or less or 0 from a current in the positive direction. The synchronous rectification current may be measured by measuring the current flowing in each leg, the voltage of each switching element, or another method. [Explanation of symbols]

[0154] 10, 10A, 10B LLC DC / DC Converter 1, 1a, 1b Primary side switching circuit 2, 2a, 2b resonant circuit 3, 3a, 3b Secondary side switching circuit 4, 4a, 4b Control section

Claims

1. a main circuit section including a transformer circuit, a primary side switching circuit including a primary side switching element provided on a primary side of the transformer circuit, a secondary side switching circuit including a secondary side switching element provided on a secondary side of the transformer circuit, and a resonant circuit including a resonant inductance and a resonant capacitance; a control unit that generates a control signal for controlling on / off of the primary side switching element and the secondary side switching element; An LLC type DC / DC converter comprising: The control unit performs boost control to cause the main circuit unit to perform a boost operation, The control unit during the boost control is When the main circuit unit transmits power from the primary side switching circuit to the secondary side switching circuit, The control signal reduces a drive frequency of the primary side switching element and turns on the secondary side switching element to form a short circuit in the secondary side switching circuit.

1. An LLC type DC / DC converter comprising:

2. The control unit during the boost control is When the main circuit unit transmits power from the secondary side switching circuit to the primary side switching circuit, The control signal reduces a drive frequency of the secondary side switching element and turns on the primary side switching element to form a short circuit in the primary side switching circuit.

2. The LLC type DC / DC converter according to claim 1 .

3. The control unit is The drive frequency is controlled to a value equal to or higher than the resonant frequency of the resonant circuit, and when the drive frequency is lower than a predetermined boost start frequency, the boost control is performed.

2. The LLC type DC / DC converter according to claim 1 .

4. The control unit is When the drive frequency is equal to or higher than the boost start frequency, a frequency modulation control is performed to modulate the drive frequency; When the boost control is performed, the drive frequency is used to calculate a boost pulse width of the control signal, and the short circuit is formed during the period of the boost pulse width.

4. The LLC type DC / DC converter according to claim 3.

5. The control unit is The boost start frequency is varied so that the smaller the boost ratio at which the boost operation is performed, the lower the boost start frequency becomes, and the larger the boost ratio, the higher the boost start frequency becomes.

4. The LLC type DC / DC converter according to claim 3.

6. The control unit is When the boost pulse width of the control signal is Tb, the drive frequency is fs, and the gain of the boost ratio for performing the boost operation is G, First equation (coefficient a / driving frequency fs)-constant b Second formula: [(coefficient a / driving frequency fs)-constant b] x gain G Equation 3: (-coefficient a × driving frequency fs) + constant b Fourth formula: [(-coefficient a x driving frequency fs) + constant b] x gain G The boost pulse width is calculated from any one of the following equations, and the short circuit is formed during the boost pulse width.

2. The LLC type DC / DC converter according to claim 1 .

7. The gain G becomes larger as the boost ratio increases, and becomes smaller as the boost ratio decreases.

7. The LLC type DC / DC converter according to claim 6.

8. The primary side switching circuit and the secondary side switching circuit include A half-bridge circuit, a full-bridge circuit, or a multi-phase bridge circuit including three phases, The full bridge circuit and the multi-phase bridge circuit of the secondary side switching circuit include a bridgeless circuit in which the secondary side switching element of the upper arm is replaced with a diode.

2. The LLC type DC / DC converter according to claim 1 .

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