Electric power conversion system

Abstract

To provide an electric power conversion device capable of suppressing the generation of electromagnetic noise while ensuring a wide driving range for an input voltage.SOLUTION: A dual inverter type electric power conversion device 1 switches switching elements 4A to 4F of a first inverter 6 in a first control mode that switches the voltage vector six times during one rotation of a motor 3. The electric power conversion device 1 switches switching elements 5A to 5F of a second inverter 7 in a second control mode that is pulse width modulation that outputs only odd-numbered voltage vectors during one rotation of the motor 3, or pulse width modulation that outputs only even-numbered voltage vectors during one rotation of the motor 3.SELECTED DRAWING: Figure 1

Description

Technical Field

[0001] The present invention relates to a power conversion device using a dual inverter system.

Background Art

[0002] There is known a power conversion device using a dual inverter system in which two inverters are connected to a motor having a plurality of windings with both ends open, and the motor is driven by the differential voltage between the two inverters. As this power conversion device using a dual inverter system, there are roughly classified into a common power supply type in which two inverters are connected to the same DC power supply (see, for example, Patent Document 1), a floating capacitor type in which one inverter is connected to a DC power supply and the other inverter is connected to a capacitor (see, for example, Patent Document 2), and a two power supply type in which two inverters are connected to different DC power supplies (see, for example, Patent Document 3).

Prior Art Documents

Patent Documents

[0003]

Patent Document 1

Patent Document 2

Patent Document 3

Summary of the Invention

Problems to be Solved by the Invention

[0004] However, although these power conversion devices using a dual inverter system can increase the output voltage to the motor with respect to the input voltage, since the two inverters perform switching, there is a possibility that common mode noise increases. In particular, in a power conversion device applied to an in-vehicle electric compressor, not only a wide driving range is required with respect to the input voltage (battery voltage), but also suppression of electromagnetic noise is required.

[0005] An object of the present invention is to provide a power conversion device that can suppress the generation of electromagnetic noise while ensuring a wide driving range with respect to an input voltage.

Means for Solving the Problems

[0006] The present invention connects a first inverter and a second inverter each including a plurality of switching elements to a motor including a plurality of windings with both ends open, and switches the switching elements by a control device to apply a differential voltage between the first inverter and the second inverter to the windings. A power conversion device of a dual inverter method, wherein the control device switches the switching elements of the first inverter in a first control mode in which the voltage vector is switched six times during one rotation of the motor, and the switching elements of the second inverter are switched in a second control mode which is pulse width modulation for outputting only odd voltage vectors or pulse width modulation for outputting only even voltage vectors during one rotation of the motor.

Effects of the Invention

[0007] According to the present invention, it is possible to significantly suppress the generation of electromagnetic noise while ensuring a wide driving range with respect to the input voltage.

Brief Description of the Drawings

[0008]

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[0009] Hereinafter, embodiments of the present invention will be described with reference to the drawings.

[0010] [Premise Technology of the Present Invention] First, the prior art of the present invention will be described with reference to FIGS. 15 to 24.

[0011] FIG. 15 is a circuit diagram of a conventional power conversion device 101. The power conversion device 101 includes an inverter 103 composed of a plurality of switching elements 102A to 102F. One end sides of three-phase windings 105U, 105V, and 105W provided in the motor 104 are respectively connected to one end side of the inverter 103. The other end sides of the windings 105U, 105V, and 105W are connected at the midpoint.

[0012] The other end side of the inverter 103 is connected to a battery 108, which is a DC power source, via a positive power line 106 and a negative power line 107. The inverter 103 drives the motor 104 by applying a voltage to the windings 105U, 105V, and 105W based on the switching control of the switching elements 102A to 102F by the control device 109. Note that an inductor 110 interposed in the positive power line 106 and a capacitor 111 connected between the positive power line 106 and the negative power line 107 constitute a noise filter.

[0013] In this power conversion device 101, for example, let the voltage applied by the inverter 103 to the U phase of the motor 104 be Vu. At this time, in order to calculate the voltage Vu_mtr applied across both ends of the winding 105U of the U phase of the motor 104, using the midpoint voltage Vmid of the motor 104, Vu_mtr = Vu - Vmid is obtained. Vmid is as follows. Vmid = (Vu + Vu + Vu) / 3

[0014] Let the battery voltage applied to the inverter 103 be Vdc. The voltage is a relative value, and 0V can be defined as an arbitrary potential. Here, if half of the battery voltage is defined as 0V, the positive electrode side of the battery 108 can be expressed as Vdc / 2, and the negative electrode side can be expressed as -Vdc / 2. At this time, considering the voltage of Vu, when the upper arm of the U phase is ON and the lower arm is OFF, it is Vdc / 2, and in the reverse case, it is -Vdc / 2.

[0015] In the voltage application state to the motor 104 by the inverter 103, for each of the UVW phases, there are a state where the upper arm is ON and the lower arm is OFF (referred to as High in the drawing) and a state where the upper arm is OFF and the lower arm is ON (referred to as Low in the drawing). The combination pattern is shown in Fig. 16(a). At this time, the applied voltages Vu, Vv, Vw of each phase of the inverter 103 and the neutral point voltage Vmid of the motor 104 are as shown in Fig. 16(b). Also, the voltages Vu_mtr, Vv_mtr, Vw_mtr applied to the respective windings 105U, 105V, 105W of the motor 104 are as shown in Fig. 16(c). In this way, the voltage can be applied to the motor 104 within a range of ±2Vdc / 3.

[0016] The inverter 103 applies a phase voltage to the motor 104 by means of a PWM output that performs rising and falling during a specified period using a carrier signal. An example is shown in Fig. 17(a). As shown in Fig. 17(a), when the motor voltage is applied and represented by a vector diagram, the voltage vector shown in Fig. 17(b) is output. In this way, the inverter voltage that performs PWM output can be represented by a voltage vector diagram. Also, in order to rotate the motor 104 smoothly, it is necessary to apply a sinusoidal voltage modulated by PWM to the UVW phases.

[0017] The amplitude of the voltage that can be applied to the motor 104 is determined by the magnitude of the input voltage Vdc and the modulation rate. The maximum amplitude at which a sinusoidal voltage can be applied to the motor 104 is the size of the circle F1 shown in the voltage vector diagram shown in Fig. 17(b), which is the inscribed circle of the hexagonal F2 of the voltage vector. Note that in order to utilize the voltage up to the inscribed circle, it is necessary to use an inter-phase modulation such as a generally known two-phase modulation.

[0018] The maximum voltage that the inverter 103 can output in PWM is as shown in Fig. 18(a). The voltage vector output in this case is as shown in Fig. 18(b) and is located at the maximum amplitude of the hexagon F2. This means that the pattern V1 shown in Fig. 16(a) is always selected in one PWM cycle. Similarly, for the patterns V2, V3, V4, V5, and V6, the maximum amplitude of the hexagon F2 is also selected. Thus, when V1 to V6 are selected in order, although the voltage applied to the motor 104 is not a sine wave, the maximum amplitude is applied as the amplitude. This voltage application method is called the 6-step method. Since the applied voltage contains many harmonics while the input voltage can be utilized to the maximum extent, it is a control method mainly used when driving the motor in the highest speed region where the ratio of the number of PWMs to the rotational speed deteriorates.

[0019] Fig. 19 is a circuit diagram showing a power conversion device 201 of a common power supply type dual inverter system. As shown in Fig. 19, the power conversion device 201 of the common power supply type dual inverter system is connected to a motor 203 having a plurality of windings 202U, 202V, 202W with both ends open, and a first inverter 206 and a second inverter 207 having a plurality of switching elements 204A to 204F, 205A to 205F. Then, by switching the switching elements 204A to 204F, 205A to 205F with a control device 208, a differential voltage between the first inverter 206 and the second inverter 207 is applied to the windings 202U, 202V, 202W. The input terminals of the first inverter 206 and the second inverter 207 are connected in parallel to a battery 209 to form a common power supply type dual inverter.

[0020] Here, for example, when the voltage across both ends of the U-phase winding 202U is defined as Vu1 for the U-phase output voltage of the first inverter 206 and Vu2 for the U-phase output voltage of the second inverter 207, the following equation holds. Vu_mtr = Vu1 - Vu2 Therefore, the voltage amplitude that can be applied is ±Vdc as shown in Fig. 20. As a result, the voltage amplitude that can be applied to the motor 203 is 1.5 times that of the power conversion device 101 shown in Fig. 15. In this way, when using the common power supply type dual inverter method, the voltage that can be applied to the motor 203 can be increased. Note that in Fig. 20, only the voltage amplitude of the U phase is shown, but the V phase and W phase can be obtained in the same way.

[0021] The voltage vectors of the UVW phases that the power conversion device 201 can output are as shown in Fig. 21. Here, Fig. 21(a) shows the output voltage vector of the first inverter 206. Fig. 21(b) shows the output voltage vectors by the first inverter 206 and the second inverter 207. In the power conversion device 201 of the dual inverter method, the output voltage vector of the second inverter 207 is applied in a form subtracted from the output voltage vector of the first inverter 206. For this reason, the voltage vector that can be applied to the motor 203 can be increased up to the circle F3 shown in Fig. 21(b).

[0022] Note that the outer hexagon F4 shown in Fig. 21(b) is the range of the voltage vectors that the second inverter 207 can output when the output voltage vector of the first inverter 26 is always maximized. In this way, the outer hexagon F4 can be represented from each vertex of the inner hexagon F5.

[0023] Since the electric compressor is an in-vehicle component used in an electric vehicle, electromagnetic noise is an important factor. In the electric compressor, the main factor of electromagnetic noise is the common mode noise of the inverter that drives the motor. As shown in Fig. 22, the common mode noise of the inverter 103 leaks from the parasitic capacitance inside the motor 104 to the housing (such as the motor case). Note that in Fig. 22, a capacitor is shown inside the motor 104, but this does not mean that a capacitor is mounted, but represents the parasitic capacitance provided in the motor 104.

[0024] The voltage applied to the floating capacitance of the U-phase of the motor 104 is Vu. The current Iu_com that leaks from the floating capacitance Cu is as follows, and the time variation of the inverter output voltage becomes common-mode noise. The same applies to the V-phase and W-phase.

[0025]

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[0026] Considering the switching pattern of the inverter 103 shown in FIG. 17(a) described above, since the rise and fall of the output voltage occur 6 times during one rotation of the motor, the common-mode current leaks 6 times. An example of a control method for suppressing this common-mode noise is shown in FIG. 23. In the switching pattern of the inverter 103 shown in FIG. 23, the timings of the rise and fall of each phase of the inverter 103 coincide, and the voltage fluctuations cancel each other out. As a result, the leakage of the common-mode current can be suppressed.

[0027] As shown in FIG. 23, the PWM output method for canceling voltage fluctuations is called the RSPWM (Remote State Pulse Width Modulation) method. When this method is used, the voltage vectors that can be output are limited to the inside of the triangle F6 shown in FIG. 24. Therefore, the region where a sine wave can be applied to the motor 104 becomes as narrow as the size of the inscribed circle F7 of the triangle F6. This means that the magnitude of the voltage that can be output becomes as small as 1 / √3 (≈0.577) times compared to the inscribed circle F1 of the hexagon F2 until now. Thus, while the RSPWM method can reduce common-mode noise, there is a problem that the voltage that can be applied to the motor 103 becomes small and the driving range of the motor 103 with respect to the input voltage becomes narrow.

[0028] Note that the RSPWM method shown in Fig. 24(a) outputs only odd-numbered voltage vectors among the voltage vectors, and thus is called the odd-numbered RSPWM method. Conversely, as shown in Fig. 24(b), there is also a method of outputting only even-numbered voltage vectors, and in this case as well, common-mode noise can be suppressed. This is called the even-numbered RSPWM method.

[0029] [One Embodiment of the Present Invention] Next, a power conversion device 1 according to one embodiment of the present invention will be described with reference to Figs. 1 to 13.

[0030] The power conversion device 1 shown in Fig. 1 is connected to a motor 3 having a plurality of windings 2U, 2V, 2W with open ends, and a first inverter 6 and a second inverter 7 having a plurality of switching elements 4A to 4F, 5A to 5F. Then, by switching these switching elements 4A to 4F, 5A to 5F with a control device 8, a differential voltage between the first inverter 6 and the second inverter 7 is applied to the windings 2U, 2V, 2W.

[0031] One end side of the first inverter 6 is connected to one end side of the windings 2U, 2V, 2W, and the other end side is connected to a battery 9 which is a DC power supply. Also, one end side of the second inverter 7 is connected to the other end side of the windings 2U, 2V, 2W, and the other end side is connected to a capacitor 10. Thereby, the power conversion device 1 constitutes a floating capacitor type dual inverter.

[0032] Specifically, the first inverter 6 includes a U-phase half-bridge circuit 11U, a V-phase half-bridge circuit 11V, and a W-phase half-bridge circuit 11W. In each phase half-bridge circuit 11U, 11V, 11W, switching elements 4A to 4C of the upper arm and switching elements 4D to 4F of the lower arm are arranged respectively. Also, each of the switching elements 4A to 4F incorporates a flywheel diode connected in antiparallel.

[0033] The positive input terminal of the first inverter 6 (collector electrodes of the switching elements 4A to 4C of the upper arm) is connected to the positive power supply line 12 of the battery 9. On the other hand, the negative input terminal of the first inverter 6 (emitter electrodes of the switching elements 4D to 4F of the lower arm) is connected to the negative power supply line 13 of the battery 9.

[0034] Note that the inductor 14 interposed in the positive power supply line 12 and the capacitor 15 connected between the positive power supply line 12 and the negative power supply line 14 constitute a noise filter.

[0035] In the first inverter 6, the emitter electrode of the switching element 4A of the upper arm and the collector electrode of the switching element 4D of the lower arm of the U-phase half-bridge circuit 11U are connected. This connection point is connected to one end side of the U-phase winding 2U of the motor 3. Also, the emitter electrode of the switching element 4B of the upper arm and the collector electrode of the switching element 4E of the lower arm of the V-phase half-bridge circuit 11V are connected. This connection point is connected to one end side of the V-phase winding 2V of the motor 3. Further, the emitter electrode of the switching element 4C of the upper arm and the collector electrode of the switching element 4F of the lower arm of the WU-phase half-bridge circuit 11W are connected. This connection point is connected to one end side of the W-phase winding 2W of the motor 3.

[0036] The second inverter 7 includes a U-phase half-bridge circuit 16U, a V-phase half-bridge circuit 16V, and a W-phase half-bridge circuit 16W. In each phase half-bridge circuit 16U, 16V, 16W, switching elements 5A to 5C of the upper arm and switching elements 5D to 5F of the lower arm are arranged. Also, each of the switching elements 5A to 5F incorporates a flywheel diode connected in antiparallel.

[0037] A capacitor 10 is connected between the positive input terminal of the second inverter 7 (collector electrodes of the switching elements 5A to 5C of the upper arm) and the negative input terminal of the second inverter 7 (emitter electrodes of the switching elements 5D to 5F of the lower arm).

[0038] In the second inverter 7, the emitter electrode of the switching element 5A of the upper arm of the U-phase half-bridge circuit 16U and the collector electrode of the switching element 5D of the lower arm are connected. This connection point is connected to the other end side of the U-phase winding 2U of the motor 3. Also, the emitter electrode of the switching element 5B of the upper arm of the V-phase half-bridge circuit 16V and the collector electrode of the switching element 5E of the lower arm are connected. This connection point is connected to the other end side of the V-phase winding 2V of the motor 3. Further, the emitter electrode of the switching element 5C of the upper arm of the WU-phase half-bridge circuit 16W and the collector electrode of the switching element 5F of the lower arm are connected. This connection point is connected to the other end side of the W-phase winding 2W of the motor 3.

[0039] In the power conversion device 1 of the floating capacitor type dual inverter system configured as described above, the voltage on the second inverter 7 side is determined by the capacitor 10. Also, the voltage of the capacitor 10 is charged from the battery 9 which is the power source on the first inverter 6 side via the motor 3. At that time, the voltage on the second inverter 7 side can be boosted using the motor 3.

[0040] FIG. 2 shows the voltage that can be applied to the U-phase by the power conversion device 1. Here, when the voltage Vdc2 on the second inverter 7 side is boosted to twice the magnitude of the voltage Vdc1 on the first inverter 6 side, the voltage that can be applied to the motor 3 becomes ±3Vdc / 2, which is larger than that of the common power supply type.

[0041] In the case of the floating capacitor type where the voltage on the second inverter 7 side is increased to twice the voltage on the first inverter 6 side, as shown in FIG. 3, the hexagon F8 on the second inverter 7 side becomes larger. For this reason, the voltage vector can be made larger compared to the voltage vector of the common power supply type (see FIG. 21(b)). Note that the hexagon F8 shown in FIG. 3 indicates the voltage vector that the second inverter 7 can select when the first inverter 6 always selects the pattern V1.

[0042] The common-mode noise leaking from the floating capacitance of the motor 3 is an issue to be solved even in the dual-inverter system. In the dual-inverter system, since the number of switching elements increases, the occurrence frequency of common-mode noise increases. Therefore, in the dual-inverter system, the use of the RSPWM method is considered.

[0043] Fig. 5 is a voltage vector diagram showing the output voltage when the first inverter 6 is controlled by the odd-numbered RSPWM method and the second inverter 7 is controlled by the even-numbered RSPWM method. As shown in Fig. 5, the circle F9 of the voltage vectors that can be output to the motor 3 becomes larger compared to the single-inverter system, but smaller compared to the normal dual-inverter system.

[0044] Therefore, the first inverter 6 is controlled by the 6-step method (first control mode), and the second inverter 7 is controlled by the even- or odd-numbered RSPWM method (second control mode). With such a control method, as shown in Fig. 6, the circle F10 of the voltage vectors that can be output to the motor 3 becomes larger. Although the driving range becomes narrower compared to the normal floating capacitor type, the voltage vector becomes larger than that of the common power supply type, and the driving range expands.

[0045] Fig. 7 shows the motor speed, dq-axis current, and three-phase current when the motor 3 is driven by the conventional control method in the power conversion device 1. Here, DC250V is input to the first inverter 6 side, and the second inverter 7 side is boosted to 500V. According to such a conventional control method, the motor 3 is stably driven up to 4000 rpm.

[0046] Fig. 8 shows the output voltage and output current when the motor 3 is driven with the first inverter 6 in the 6-step method and the second inverter 7 in the even- or odd-numbered RSPWM method in the power conversion device 1. Fig. 9 shows the motor speed, dq-axis current, and three-phase current. As shown in Figs. 8 and 9, although the current ripple is large, the motor 3 is driven without problems.

[0047] FIG. 10 is a waveform diagram showing the output voltage of the first inverter 6 when controlled in a six-step method in the power conversion device 1. FIG. 11 is a waveform diagram showing the output voltage of the second inverter 7 when controlled in the RSPWM method in the power conversion device 1. As shown in FIGS. 10 and 11, on the first inverter 6 side, since it is a six-step method, the number of switching times is small. Also, on the second inverter 7 side, since it is the RSPWM method, although the number of switching times is large, the number of fluctuations in the common-mode voltage is small. By such a combination of control methods, while ensuring the driving range, the common-mode noise can be significantly suppressed.

[0048] Next, the calculation steps of the control device 8 will be described with reference to FIGS. 12 and 13.

[0049] The control device 8 performs vector control to drive according to the rotation speed command and torque command of the motor 3, and generates voltage commands Vd and Vq to be applied to the motor 3. This is the same as normal motor control.

[0050] Next, the control device 8 calculates how to apply the voltage commands Vd and Vq to be applied to the motor 3 to the first inverter 6 and the second inverter 7. For example, let the point P1 shown in FIG. 12(a) be the applied voltage command point. In the case of a normal dual inverter, even when the operating point is determined, there is freedom in the applied voltages on the first inverter 6 side and the second inverter 7 side. That is, as shown in FIG. 12(b), when the first inverter 6 side selects the point P2, since the second inverter 7 side can select any voltage vector within the frame W1, P1 can be selected as the synthesized voltage. Or when the first inverter 6 side selects the point P3, since the second inverter 7 side can select any voltage vector within the frame W2, P1 can be selected as the synthesized voltage in the same way. By utilizing this freedom and adjusting the phase difference of the voltages on the first inverter 6 side and the second inverter 7 side, the voltage of the capacitor 10 on the second inverter 7 side can be charged.

[0051] In the control method of this embodiment, the outputtable voltage range is limited. For example, as shown in Fig. 12(a), when the voltage on the first inverter 6 side is equal to the voltage on the second inverter 7 side, there is only one voltage that satisfies the voltage command. Therefore, it is difficult to drive while boosting the voltage on the second inverter 7 side.

[0052] On the other hand, when boosting the second inverter 7 side (for example, 1.5 times that of the first inverter 6 side), as shown in Fig. 13, either point P2 or point P3 can be selected. That is, the applied voltages on the first inverter 6 side and the second inverter 7 side can be adjusted to some extent. In that case, since the phase difference of the voltage can be utilized, the capacitor 10 on the second inverter 7 side can be charged.

[0053] Note that the charging method of the capacitor 10 on the second inverter 7 side is not limited to the method of utilizing the phase difference of the voltage described above, and there are various methods. However, in any method, it is necessary to adjust the voltages on the first inverter 6 side and the second inverter 7 side. Therefore, freedom is required, and in order to adjust this freedom, it is necessary to boost the voltage on the second inverter 7 side. Also, when controlling the first inverter 6 in a 6-step method at startup, the second inverter 7 side cannot be charged. For this reason, it is necessary to execute a charging sequence for charging the second inverter 7 side before startup. At that time, by controlling both the first inverter 6 side and the second inverter 7 side in the RSPWM method, charging can be easily achieved.

[0054] Next, the boosting method for the second inverter 7 side in the power conversion device 1 will be described.

[0055] The current idc1 flowing between the switching elements 4A to 4F of the first inverter 6 and the capacitor 15 is obtained by the following formula. idc1=(sign(Vu1)iu + sign(Vv1)iv + sign(Vw1)iw) / 2 Here, sign(Vu) is a sign function that means "1" when Vu is positive, i.e., when the upper arm is ON, and "-1" when Vu is negative, i.e., when the lower arm is ON. Also, iu, iv, and iw represent the direction of the current flowing from the first inverter 6 side toward the second inverter 7 side.

[0056] In the case of the second inverter 7, since the direction of the flowing current is different, the following equation holds. ids2=-((sign(Vu2)iu + sign(Vv2)iv + sign(Vw2)iw)) / 2 Here, when idc2 is negative, the capacitor 10 of the second inverter 7 will be charged. Therefore, by applying a voltage to adjust the current idc2 flowing into the capacitor on the second inverter 7 side while observing the voltage of the capacitor 10, the voltage on the second inverter 7 side can be boosted to an arbitrary value.

[0057] As described above, according to the above embodiment, a first inverter 6 and a second inverter 7 each having a plurality of switching elements 4A to 4F and 5A to 5F are connected to a motor 3 having a plurality of windings 2U, 2V, and 2W with both ends open. Then, by switching the switching elements 4A to 4F and 5A to 5F by a control device 8, a dual-inverter type power conversion device 1 that applies the differential voltage between the first inverter 6 and the second inverter 7 to the windings 2U, 2V, and 2W is formed. In this configuration, the control device 8 switches the switching elements 4A to 4F of the first inverter 6 in a 6-step method of switching the voltage vector only 6 times during one rotation of the motor 3. As a result, not only can the switching frequency be reduced to suppress common-mode noise, but also the amplitude of the output voltage can be maximized.

[0058] Further, the control device 8 switches the switching elements 5A to 5F of the second inverter 7 by the RSPWM method, which is pulse width modulation that outputs only odd-numbered voltage vectors during one rotation of the motor 3, or pulse width modulation that outputs only even-numbered voltage vectors during one rotation of the motor. Therefore, although the output voltage becomes small, the variation in the common-mode voltage can be reduced and the common-mode noise can be suppressed. By combining such control methods of the first inverter 6 and the second inverter 7, it is possible to significantly suppress the generation of electromagnetic noise while ensuring a wide driving range with respect to the input voltage.

[0059] One end of the first inverter 6 is connected to one end of the windings 2U, 2V, 2W, and the other end is connected to the battery 9. One end of the second inverter 7 is connected to the other end of the windings 2U, 2V, 2W, and the other end is connected to the capacitor 10. Therefore, the present invention can be applied to a floating capacitor type dual inverter.

[0060] As described above, one embodiment of the present invention has been described, but the present invention is not limited to the configuration of the above-described embodiment. For example, as shown in FIG. 14, the present invention can also be applied to a power conversion device 1B of a floating capacitor type dual inverter method in which the negative input terminal of the first inverter 6 and the negative input terminal of the second inverter 7 are connected via a connection line 20 to make the reference potential common.

[0061] Further, the present invention is applicable to any dual inverter method that can make the voltage on the second inverter side higher than the voltage on the first inverter side. Therefore, the present invention can also be applied to a power conversion device of a two-power-source type dual inverter method that includes a power source with a higher voltage on the second inverter side than on the first inverter side.

Description of Reference Numerals

[0062] 1 Power conversion device 2U, 2V, 2W Windings 3 Motor 4A to 4F, 5A to 5F Switching elements 6 First inverter 7 Second Inverter 8 Control Device 9 Battery 10 Capacitor

Claims

1. A power conversion device of a dual inverter type in which a first inverter and a second inverter each having a plurality of switching elements are connected to a motor having a plurality of windings with both ends open, and a differential voltage between the first inverter and the second inverter is applied to the windings by switching the switching elements with a control device, wherein the control device switches the switching elements of the first inverter in a first control mode in which the voltage vector is switched six times during one rotation of the motor, and switches the switching elements of the second inverter in a second control mode which is pulse width modulation for outputting only odd voltage vectors or pulse width modulation for outputting only even voltage vectors during one rotation of the motor. A power conversion device characterized by the above.

2. One end side of the first inverter is connected to one end side of the winding, and the other end side is connected to a DC power supply, One end side of the second inverter is connected to the other end side of the winding, and the other end side is connected to a capacitor. The power conversion device according to claim 1, characterized by the above.

3. The reference potentials of the first inverter and the second inverter are made common. The power conversion device according to claim 2, characterized by the above.

4. One end side of the first inverter is connected to one end side of the winding, and the other end side is connected to a DC power supply, One end side of the second inverter is connected to the other end side of the winding, and the other end side is connected to a second DC power supply. The power conversion device according to claim 1, characterized by the above.

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