Method for the measurement of any waveform and system for implementing said method - Patents.com

Abstract

TIFF2025504229000039.tif96164

Description

[Technical field]

[0001] The technical fields of the invention are teratronic signal processing, photonics and microwave engineering. Teratronic signal processing focuses on the generation, measurement and processing of signals with bandwidths up to hundreds of GHz. The design of high bandwidth systems relies on the fields of photonics and microwave engineering. [Background technology]

[0002] The term "arbitrary" is used to indicate that the arbitrary signal being measured has no inherent structure that is exploited by the measurement system.

[0003] The state-of-the-art technique for detecting any optical waveform, hereafter called the used signal, with a good signal-to-noise ratio is based on intradyne coherent detection. An intradyne coherent receiver consists of a 90° optical hybrid, a pair of balanced photodetectors (BPDs) and an analog-to-digital converter (ADC) and is capable of measuring the in-phase and quadrature components of the used signal. The optical bandwidth of such an intradyne coherent receiver is limited to twice the bandwidth of the limiting component, usually the ADC or the BPD. For further details on coherent receivers, the reader can refer to

[14] or other basic works.

[0004] The bandwidth limitations imposed by electronic components can be overcome by replacing a single coherent receiver with a coherent receiver array and using a low-noise optical frequency comb as a multi-wavelength local oscillator. This concept has been used, for example, in spectrally sliced ​​coherent detection [3] and parallel optical sampling [4]. While spectrally sliced ​​coherent detection has been demonstrated for the measurement of arbitrary optical signals, all demonstrations of parallel optical sampling have so far been limited to signals with inherent structure, e.g., data signals. In the following, we briefly discuss the aforementioned methods and their limitations.

[0005] Frequency-sliced ​​coherent detection was demonstrated for arbitrary optical waveforms in 2010 [3]. In frequency-sliced ​​arbitrary optical waveform measurements, optical slicing filters, e.g., arrayed-waveguide gratings (AWGs), are used to split the working signal into several slightly overlapping spectral bands that can be detected individually by slow electronics. The local oscillator frequencies for detecting the individual spectral slices are placed at the center of each slice. To digitally reconstruct the working signal from the measured narrowband frequency slices, the frequency differences between the local oscillator tones must be precisely known. Therefore, all local oscillator tones are derived from the coherent optical frequency comb by separating the individual tones with a bank of non-overlapping demultiplexing filters. The separately measured frequency bands are then digitally combined to obtain the digital working signal. The correct phase and amplitude for stitching are obtained directly from the spectral overlap regions between adjacent spectral slices. A receiver with an aggregate bandwidth of 228 GHz was demonstrated in 2012 [6]. The advantages of this method are a fairly simple concept, good scalability, high fidelity, and perfect compensation of the linear transfer characteristic of the receiving system. However, in the case of monolithic integration of optical slicing filters and coherent receiver arrays on a photonic integrated circuit (PIC), disadvantages arise, especially when relying on integration platforms with high refractive index contrast such as indium phosphide (InP) or silicon photonics. Slicing filters based on arrayed waveguide gratings suffer from fabrication errors and require additional phase error correction

[15] . Additional temperature stabilization may be required

[15] . In the case of passive filters, all components in the system, e.g., frequency comb generators and slicing filters, need to be precisely aligned in frequency and this alignment must be maintained over a wide range of operating conditions, including ambient temperature. Some of these difficulties can be solved by frequency-tunable coupled-resonator optical waveguide (CROW) filters [2], which are dynamically tunable and compactly integrated on silicon photonic platforms.Nevertheless, such filters introduce the additional complexity of controlling several heating elements per slicing filter. Moreover, precise temperature control is required, and small temperature variations can reduce the accuracy of medium- and long-term calibration. Thus, a fully passive, yet compact structure, when used in systems based on parallel optical sampling, significantly simplifies chip-level integration.

[0006] In [4], the concept of parallel optical sampling for the reception of 56 GBd QPSK data signals is demonstrated. In contrast to spectrally sliced ​​coherent reception, no optical slicing filters are required, but a simple power splitter is sufficient to split the working signal and the multi-wavelength local oscillator into several paths. As a result, the photonic integration of such a system is easier compared to spectrally sliced ​​systems. However, in [4] the authors only report the measurement of quadrature phase shift keying (QPSK) data signals and resort to a multidimensional optimization procedure that exploits the structure of the transmitted data signal to correct the unknown phase parameters of the receiver system

[16] . Thus, the system does not have the ability to measure arbitrary working signals that do not have any structure known a priori. In addition, the local oscillators used for parallel optical sampling need to generate narrow pulses in the time domain to limit crosstalk between adjacent samples.

[0007] Besides the two approaches mentioned above, there are many other concepts of photonic-assisted analog-to-digital conversion [5]. However, these are less relevant for the present application, as the concepts are significantly different from the present invention. Further insights on spectrally sliced ​​arbitrary waveform measurement systems can be found in [1, 2, 6, 7]. Further information on the concepts of parallel optical sampling and optical time division multiplexing (OTDM) can be found in [8-10]. Summary of the Invention [Problem to be solved by the invention]

[0008] The technical problem of interest here is to provide a method for measuring any electromagnetic signal, which makes it possible to reduce the number of filters or the required filter steepness provided by the measurement system, where the term "filter steepness" refers to the steepness of the power transfer function, usually expressed in decibels (dB), at the edge of the filter passband. The method will be generally applicable to any signal that does not have to have any structure known a priori. In addition, the method should be able to compensate for drifts in system parameters, for example caused by temperature variations. Furthermore, the method should provide robust calibration techniques to compensate for the characteristics of the measurement system. [Means for solving the problem]

[0009] The technical problem of this object stated above is solved by the method disclosed herein according to the features of the first independent claim 1. The associated dependent claims, i.e. the dependent claims, reflect advantageous designs or embodiments. Advantageous further embodiments, which can be realised individually or in any combination, are set out in the dependent claims.

[0010] In the following, the terms "have", "comprise" or "include" or any of their grammatical variants are used in a non-exclusive manner. These terms can therefore refer both to the situation where there are no further features besides those introduced by these terms, or to the situation where one or more further features are present. For example, the expressions "A has B", "A includes B" or "A includes B" can refer both to the situation where there are no other elements in A besides B (i.e., A consists exclusively of B), and to the situation where, in addition to B, one or more other elements are present in A, for example element C, elements C and D, or other elements altogether.

[0011] Furthermore, the terms "at least one" and "one or more," as well as grammatical variations of these terms, when used in connection with one or more elements or features, are intended to express that the element or feature may be provided one or more times, will generally be referred to as being used only once, e.g., when the feature or element is first introduced. If the feature or element is subsequently referred to again, without limiting the possibility that the feature or element may be provided singly or multiple times, the corresponding term "at least one" or "one or more" will generally no longer be used.

[0012] Furthermore, the terms "preferably", "particularly", "eg" or similar terms are used below in connection with optional features, without thereby limiting alternative embodiments. Features introduced by these terms are therefore optional features, and are not intended to limit the scope of protection of the claims, in particular the independent claims, by these features. Thus, as understood by a person skilled in the art, the invention can be implemented using other embodiments. Similarly, features introduced by "in one embodiment" or "in a further embodiment" are understood as optional features, without intending to limit the scope of protection of alternative embodiments or the independent claims. Furthermore, all possibilities of combining the features introduced thereby with other features, whether optional or non-optional, will remain unaffected by these introductory expressions.

[0013] [General description and embodiment] The present invention is well suited for photonic integration since it does not require slicing filters with rapidly attenuating passbands, thus avoiding potential manufacturing inaccuracies

[15] , active control [2, 15], or frequency drifts of such filters. The present invention utilizes frequency domain calibration methods to compensate for the transfer characteristics of the receiver system as well as the effects of imperfectly time-interleaved local oscillators. This includes compensation for non-equidistant optical sampling pulses. Furthermore, the method described in the present invention works equally well with strongly distributed local oscillators that are insensitive to the local oscillator pulse shape, i.e., with sampling pulses that overlap in time. For purely time domain methods, overlapping sampling pulses would increase computational complexity. Moreover, the present invention exploits redundant information contained in the measurement signal to estimate at least one, but potentially various, system parameters related to, for example, the relative amplitude and phase between different local oscillator tones, phase variations introduced by propagation through waveguides and / or fibers, or drifts in the resonant frequency spacing of the local oscillators. By exploiting the redundant information in the spectral overlap region, the present invention can measure any signal since no prior knowledge of the structure of the signal itself is required. This capability opens up applications in the area of ​​arbitrary waveform measurement or photonic-assisted analog-to-digital conversion. Besides purely time-interleaved systems, hybrid systems that apply both spectral slicing and time interleaving are also feasible.

[0014] A first embodiment of a method according to the invention for detecting an electromagnetic use signal by means of a measurement system comprises the following steps: a. providing an input interface for an electromagnetic input signal, configured to be supplied with an electromagnetic use signal; providing an electromagnetic reference signal source configured to provide an electromagnetic reference signal including M discrete tones, where M>2; c. providing a subsystem, The subsystem is i. a first division element configured to divide a usage signal into a number N (N>2) of partial usage signals; ii. a second division element configured to divide the reference signal into a number N of partial reference signals, where N is the number of corresponding partial use signals; With TIFF2025504229000002.tif83164, The step includes: The above method is d. at least one partial reference signal includes at least two spectral tones; and e. the entirety of all partial mixture signals contains redundant information that is used to estimate at least one model parameter of the measurement system; It is characterized by:

[0015] Here, the terms "reconstructed digital use signal" and "digital use signal" are used interchangeably herein.

[0016] Having at least one partial reference signal that includes at least two spectral tones (d) makes it possible to reduce the number of slicing filters required or the steepness of such filters and / or to increase the bandwidth of at least one passband of such filters. As a result, the measurement system hardware is simplified and easier to integrate. For example, the first and second splitting elements can be implemented as power splitters with appropriate time delays. This allows a configuration in which no slicing filters are required and all partial reference signals include all spectral tones of the reference signal.

[0017] By utilizing the redundant information contained in the totality of all partial mixed signals to estimate at least one model parameter of the measurement system, the system operates for any input signal that matches the frequency range of the measurement system. More specifically, the measurement system is not limited to input signals with inherent structure or any other kind of a priori known information, such as data signals that follow a specific pattern related to the underlying modulation format, pulse shape, and symbol rate. The frequency range of the measurement system, i.e., the frequency range in which any waveform can be detected, is defined by the frequency range of the reference signal. More precisely, the lower limit of the frequency range of the measurement system is f 1 -B, with an upper limit of f M +B, where f 1 is the lowest frequency reference signal tone that can still produce mixing products with the signal being used, and f M is the highest frequency reference signal tone that can still produce mixing products with the used signal, and B is the bandwidth limit of the combination of the mixing and acquisition elements.

[0018] As used herein, the term "measurement system" refers to a system described in the accompanying invention that is used to measure electromagnetic signals. The bandwidth of the measurement system exceeds the bandwidth of the individual acquisition elements, which may include, for example, electronic analog-to-digital converters. The measurement system does not rely on any structure or a priori information associated with the input signal and is therefore capable of measuring any signal.

[0019] As used herein, an "input interface" is capable of coupling an input signal to a measurement system and providing a signal for use with a subsequently connected component. Examples of input interfaces are: - Electrically conducting means, e.g. cables, coaxial cables, waveguides, probes, etc. - Optical fibers, optical waveguides, photonic wire bonds, grating couplers, edge couplers, etc. - Free-space coupling of optical signals using lenses, mirrors, phased arrays, optical fibers, and lensed optical fibers; free space coupling of electrical signals, e.g. via antennas, focusing elements, or - any other means for coupling signals into a technical system, It is.

[0020] As used herein, the term "signal used" refers to an input signal of a measurement system provided by an input interface, or to an input signal of a measurement provided by an input interface and further modified by a modification element, the modification element being optional (see claim 4).

[0021] As used herein, a "reference signal source" provides a clean reference signal for use as a multi-wavelength local oscillator. The reference signal source may comprise a pump source, a laser, a frequency comb generator, an oscillator, a filter, a resonator having nonlinear electromagnetic properties, a modulator, an amplifier, a mode-locked or gain-switched laser diode, a fiber laser, a solid-state laser, or some other component suitable for generating, transmitting, amplifying, or controlling an electromagnetic signal having multiple discrete spectral tones. In addition to providing a reference signal, the reference signal source can also provide a signal to the correction element, for example to generate a phase-locked stitching signal, or can provide a phase-locked optical carrier to an electro-optical modulator that is an additional part of the correction element. Furthermore, the reference signal source can be synchronized to the acquisition element.

[0022] As used herein, a "first division element" is used to divide a usage signal into N partial usage signals. The first division element provides a transfer characteristic to each of the N output signals. Examples of such transfer characteristics are: ● The group delay of each output, or ● Filter functions including frequency dependent phase and amplitude response for each channel; It is.

[0023] As used herein, a "second splitting element" is used to split the reference signal into N partial reference signals. The second splitting element imparts a transfer characteristic to each of the N output signals. Examples of such transfer characteristics are: ● The group delay of each output, or ● Filter functions including frequency dependent phase and amplitude response for each channel; It is.

[0024] As used herein, the term "mixing element" refers to an element that performs a nonlinear operation. Each mixing element has at least two inputs, one for a partial used signal and one for a corresponding partial reference signal, K n ≥ 1 output, i.e., K n Here, n=1,...N refers to the nth mixing element. All N mixing elements together generate a partial mixing signal. K n The outputs are obtained by nonlinear interaction of the partial used signal and the corresponding partial reference signal, or by nonlinear interaction of any signal components derived from the respective partial used signal and partial reference signal, for example by using a 90° optical hybrid.

[0025] Examples of mixed elements are: - Photodetectors, e.g. photodiodes, photoresistors, phototransistors, etc. - Balanced photodetector, - a phase diversity receiver, for example a coherent receiver comprising a 90° optical hybrid and a pair of balanced photodetectors; - Diode mixer, - any other nonlinear element operating in the correct frequency range, It is.

[0026] As used herein, the term "mixing" refers to the functionality provided by a mixing element.

[0027] As used herein, the term "capture element" refers to an element comprising at least one device, such as an analog-to-digital converter, that digitizes a corresponding kth partial mixed signal generated by an nth mixing element. TIFF2025504229000003.tif10165In addition, each acquisition element is ● An analog front-end, which may include amplifiers, attenuators, filters, connectors, and waveguides; or ● A digital back-end, which may include digital storage, processors, and communication interfaces; It is possible to optionally include electronic building blocks common to digital-to-analog conversion such as:

[0028] TIFF2025504229000004.tif41164

[0029] Most of the model parameters are characterized during a system calibration procedure. However, at least one of the model parameters is either unknown or varies with time and must therefore be determined individually for each measurement or for a set of measurements made within a period of time during which the variation of the at least one unknown model parameter is negligible. For such a measurement or such a set of measurements, the at least one model parameter is determined by using a system model in conjunction with redundant information contained in the digitized partial mixture signal that jointly defines a set of relations from which the at least one model parameter can be extracted. The model parameters can be extracted by directly solving the set of relations using widely known numerical methods. Alternatively, the set of relations can be overdetermined such that the at least one unknown model parameter can be estimated, for example by using parameter estimation techniques that rely on the minimization of a certain cost metric. More specifically, but not by way of limitation, the system model may include:

number

[0030] TIFF2025504229000007.tif41164

[0031] In a further embodiment, the following relationship applies to the variables M and N: M≦N.

[0032] The number M of discrete tones in the reference signal matches the number of mixing elements, in the special case where M=N. Assuming this case in combination with spectrally equally spaced tones, the bandwidth of the mixing elements and the subsequent acquisition elements should exceed the tone spacing at least slightly.

[0033] In the case where N>M, there are more mixing elements N than there are discrete tones M of the reference signal. In this case, the system is significantly overdetermined with respect to the used signal, which reduces the noise of the reconstructed digital used signal. This configuration allows the signal-to-noise ratio of the reconstructed used signal to be improved at the expense of having more mixing and acquisition elements.

[0034] In a further embodiment, the following relationship applies to the variables M and N: M>N.

[0035] In the case M>N, if the used signal covers the entire bandwidth of the reference signal and the mixing element and the subsequent acquisition element have a bandwidth narrower than the spacing of any two tones of the reference signal, the system model may be indeterminate. In this case, signal reconstruction of any signal is not possible due to the spectral superposition of the signal components. However, if the used signal is band-limited such that only MR tones of the reference signal contribute to the measured digital partial mixing product, and N>=MR, it is possible to ignore the additional R tones of the reference signal and a digital signal reconstruction is possible. The case M>N may be very special if, for technical reasons, it is difficult to obtain a reference signal with exactly the desired number of tones. If the used signal is strictly band-limited and M>N, this allows measuring used signals with different center frequencies without adapting the reference signal source, as long as the used signal has a bandwidth within the bandwidth defined by the M reference signal comb lines and does not exceed the maximum measurement system bandwidth. The resonant frequency spacing f FSR In the case of a reference signal consisting of equally spaced spectral tones separated by 1, and assuming identical mixing and acquisition components of bandwidth B, the maximum measurement system bandwidth is 2B + (MR-1) × f FSR The bandwidth defined by the M reference signal tones is given by 2B+(M-1)×f FSR is given by:

[0036] In a further embodiment, an additional correction element is provided, configured such that the use signal provided by the input interface is modified by applying one or more of the correction operations listed below. The correction element has at least one input and at least one output for the use signal, but can have additional optional inputs and outputs that connect it to a reference signal source or another element that is part of the measurement system, for example an acquisition element or a digital signal processor. The connection to the reference signal source allows the correction element to generate a signal or a signal component that is strongly phase-correlated with the reference signal. The connection to the acquisition element allows the signal generated by the correction element to be synchronized to the acquisition element, which helps to eliminate the effects introduced by the correction element. The connection to the digital signal processor can help to digitally eliminate the effects introduced by the correction element.

[0037] The main output signal of the modification element is called the use signal. Below, the modification operations are listed. Either no modification operations or one or more modification operations are performed in any order. 1. Converting the signal used, for example by a modulator that shifts it from one carrier frequency to another; 2. Supplementing the used signal with an auxiliary signal component used for further processing of the used signal, e.g. a stitching signal as described below; 3. Applying a frequency-dependent transfer function (filter) to the signal being used. 4. Amplify the signal used; 5. Attenuate the signal used; 6. Distribute the signals used; and / or 7. Modulate the stitching signal onto the used signal.

[0038] Below is listed an exemplary selection of the various modification operations mentioned above: 1. Given a measurement system with a center frequency fc defined as the average frequency of all reference signal tones, and a measurement bandwidth BW, only signals with frequencies within the "measurement band" around fc in BW can be directly measured. To allow the measurement of signals outside this measurement band, the signals are transformed into the measurement band. Examples of such converters are: a. Electro-optical modulators (Mach-Zehnder modulators, phase modulators, IQ modulators) b. Electroabsorption modulator; c. Acousto-optic modulator; d. electro-optic crystal; e. Radio Frequency Mixer; f. Nonlinear optical devices that rely, for example, on second- or third-order optical nonlinearities; or g. photodetector; It is. 2. The redundant information in the digitized partial mixture signal depends on the used signal. Therefore, in certain cases, it is beneficial to add a stitching signal to the used signal to ensure that at least one model parameter can be reliably determined independently of the input signal of the measurement system. Preferably, the power of the stitching signal is as low as possible so that the used signal is not significantly distorted while at least one model parameter can still be reliably determined. More preferably, the added stitching signal is known, for example by additional monitoring, so that after digital reconstruction, the modification can be digitally undone by subtracting the digital stitching signal. The stitching signal is useful, for example, in the case of a used signal that has few or no spectral components in the spectral range that should contain the redundant information, making it difficult to calculate an estimate of at least one model parameter. The operation of adding a signal can be embodied, for example, by the following components or a combination thereof: ● Directional coupler, or • Multimode interference couplers, or ● Optical filters, or ● Circulator. 3. Before measuring the signal in use, a filter is optionally applied, for example: a. Removing out-of-band noise; b. Limiting the bandwidth to that of the measurement system; or c. Suppressing one or more signal frequencies of the used signal that are not relevant to the measurement. If high power components are suppressed, this can improve the fidelity of the measurement of the remaining weaker used signal components. 4. When the input signal power is low, the noise in the digital use signal can be reduced by amplifying the use signal to a predeterminable higher power level. 5. If the power of the signal is too high, the signal used must be attenuated to protect subsequently connected components from damage. Furthermore, high signal powers may increase distortions of the digitized signal used due to parasitic non-linear behavior of the mixing and acquisition elements. 6. The measurement of ultrashort pulses can result in poor signal-to-noise ratios, since the noise introduced by the acquisition element is multiplied internally over the full range of the analog-to-digital converter. Thus, a high peak-to-average power ratio leads to a low signal-to-noise power ratio, which can increase nonlinear distortion. These problems can be avoided by dispersing the used signal in the correction element, thus reducing its peak-to-average ratio. The dispersion can be completely reversed digitally, and therefore does not adversely affect the digital used signal. 7. The same arguments as in point 2 apply. However, instead of adding a stitching signal, the stitching signal can be generated by appropriately modulating the signals used.

[0039] As used herein, the term "stitching signal" refers to a signal provided by another stitching signal source, or a signal provided by a stitching signal source derived from a reference signal source, or a signal provided by the same physical source that also provides the reference signal. In some embodiments, the stitching signal can be phase-locked to a particular signal component generated by the reference signal source, for example, by any type of feedback or control circuit. The purpose of the stitching signal is to ensure that at least one model parameter of the measurement system can be properly determined without limiting the class of detectable signals used.

[0040] In further embodiments, the modification element modifies the use signal by one or more of the following exemplary operations: a. Add a stitching signal to the signal being used, or b. Convert the signal used.

[0041] The above mentioned modifications, and in particular the use of a high speed electro-optical modulator to convert the used signal to an optical carrier frequency, are particularly important because by utilizing an arbitrary waveform measurement system for the optical signal and a high speed converter that first converts the electrical signal to an optical signal, it is possible to build a wideband analog-to-digital converter with an acquisition bandwidth that far exceeds the acquisition bandwidth of the individual acquisition elements. In this context, the term "analog-to-digital converter" refers to a measurement system that converts an electrical input signal into digital data.

[0042] In a further embodiment, the reconstructed digital input signal is derived from the reconstructed digital input signal by at least partially removing the modification introduced by the modification element, where the terms "reconstructed digital input signal" and "digital input signal" are used interchangeably herein.

[0043] In some embodiments, since the used signal is modified by a modifying element prior to detection, it may be desirable to digitally remove the modification after reconstruction of the used signal.

[0044] In further embodiments, the working signal and the reference signal have center frequencies within the ultraviolet, visible, or infrared optical frequency ranges, including frequencies between 30 PHz and 700 THz, between 400 THz and 700 THz, and between 300 GHz and 400 THz, respectively, but may also be other frequency ranges, for example in the radio frequency, microwave, or millimeter wave spectrum.

[0045] To take advantage of the potentially large optical bandwidth benefits, the optical frequency ranges discussed above are feasible and suitable for implementing measurement systems. As used herein, the term "center frequency" refers to the frequency range f c This means that signals with spectral content in a frequency band of bandwidth B around f can be measured, where B is the bandwidth of the measurement system and f c is the center frequency.

[0046] In yet another embodiment, the working and reference signals have center frequencies in the range of 150 THz to 250 THz, in which case the measurement system can rely on widely available photonic components developed for optical communications in the near infrared spectrum.

[0047] In a further embodiment, the M discrete tones of the reference signal are equally spaced in frequency and are derived from an optical frequency comb.

[0048] Optical frequency combs have been shown to provide low-noise signals with timing jitter comparable to

[17] or even better than that of electronic oscillators.

[18] Examples of optical frequency combs include, but are not limited to: ● Soliton comb generators, e.g., dissipative Kerr soliton (DKS) comb sources; ● mode-locked lasers, e.g. mode-locked laser diodes, fiber lasers, or solid-state lasers; ● Gain switch laser, Comb source by injection locking, Or, a frequency comb source based on an electro-optic modulator, or • Frequency comb sources including third-order nonlinear optical fibers or waveguides for spectral broadening; It is.

[0049] Optionally, additional filter elements are part of the reference signal source to isolate the M tones from the potentially many comb lines. As used herein, the term "derived from" means that the reference signal may be an optical frequency comb, or may be an optical frequency comb that is optimized for an application by shaping its spectrum or by amplifying and removing specific spectral components.

[0050] Considering the current state of the art, the M discrete tones of the reference signal preferably have a resonant frequency interval between 100 MHz and 500 GHz, more preferably between 1 GHz and 200 GHz, and most preferably between 10 GHz and 100 GHz. The latter frequency range is of particular interest since the corresponding systems depend on mixing and acquisition elements with a bandwidth of at least 5 GHz to 50 GHz. Such elements (e.g. photodetectors and analog-to-digital converters) are readily available today. Reference signals with a resonant frequency interval up to 200 GHz or more are also important, since high-speed photodetectors and analog-to-digital converters up to 100 GHz are available today, albeit expensive. In the future, more broadband components may become available, and the preferred frequency range may change accordingly. Reference signals with a resonant frequency interval between 1 GHz and 10 GHz are also important when only low-speed analog-to-digital converters are required. This allows lowering the price and increasing the fidelity of the measurement, at the expense of bandwidth.

[0051] In a further embodiment, the reference signal consists of equally spaced spectral tones corresponding to a regular train of optical pulses in the time domain, where adjacent pulses may overlap in time. By relying on an accurate frequency domain system model, the overlapping time domain pulses of the reference signal do not pose a problem for the measurement system described here, as long as the spectral tones are stable and cover the bandwidth of the signal used. In fact, if the reference signal has a lower peak-to-average power ratio, the effects of nonlinearities are reduced and the fidelity of the measurement can be improved.

[0052] In a further embodiment, at least one model parameter estimated by the redundant information contained in the total number of all partial mixed signals is associated with a characteristic of the partial reference signal, either in absolute value or relative value of one tone of the reference signal to another tone of the reference signal, and with a characteristic of the optical or electrical circuit, either in absolute value or relative value to another circuit. Inherent characteristics of the partial reference signal include, but are not limited to: - amplitude, - Phase, - frequency, - Polarization, and - resonant frequency spacing, It is possible that: Here, some of these unique characteristics may refer individually to each of the M tones of the reference signal, eg, amplitude, phase, and frequency.

[0053] Inherent properties of optical or electrical circuits include, but are not limited to: - Amplitude change, - Phase change, or - Change in polarization, - group delay, - the dispersion experienced by a signal passing through certain paths in a circuit. These paths may be different for different signals, e.g. the phase and / or amplitude of the partial used and partial reference signals are distorted independently in all N channels. It is possible that:

[0054] In further embodiments, the system model includes the amplitudes and / or phases of frequency dependent transfer functions of the entire system or portions of the system such as splitting elements, waveguides, coupling interfaces, optical paths, amplifiers, polarizing beam splitters, phase shifters, filters, demultiplexers, optoelectronic receivers, reference signal pulse shapes, and amplitudes and phases of reference tones.

[0055] By obtaining the system transfer function during a single calibration procedure, the effects of the above mentioned components can be digitally compensated for and the digitally used signal can be accurately reconstructed. Inaccuracies in the model or calibration will reduce the signal-to-noise and distortion ratio of the digitally used signal.

[0056] In a further embodiment, the system is characterized with a dedicated calibration measurement using a known electromagnetic waveform, referred to as a calibration signal, provided by a calibration signal source.

[0057] As used herein, the term "calibration signal" refers to a known signal that can be used to obtain all transfer functions of the measurement system that are part of the system model. The calibration signal preferably covers a large portion of the entire spectral range within which the electromagnetic use signal or a modified version thereof is detected. In certain embodiments, the calibration signal is derived from a tunable and / or broadband signal source. Multiple calibration signal sources can be used in parallel or sequentially to determine all model parameters. An example of a calibration signal is an optical frequency comb with a known pulse shape and appropriate resonant frequency spacing, for example in the MHz or GHz range.

[0058] In further embodiments, the calibration signal is a broadband optical frequency comb with a resonant frequency spacing less than 25 GHz, or less than 10 GHz, or less than 5 GHz, or less than 1 GHz, or less than 500 MHz, or less than 100 MHz.

[0059] The use of a broadband optical frequency comb as a calibration signal has several advantages. First, there are very stable broadband optical frequency combs that can be characterized in advance, for example using frequency-resolved optical gating (FROG) or equivalent optical techniques. Furthermore, the use of a broadband optical frequency comb allows the system to be characterized in a single measurement. If the resonant frequency spacing (FSR) is appropriately chosen, the mixing products of the calibration signal and the individual reference signal tones do not overlap in the frequency domain and can therefore be separated (see Fig. 18 for a concrete example). To further improve the signal-to-noise ratio, it may be advantageous to average many calibration measurements or to use a distributed optical frequency comb as a calibration signal. The latter operation allows the peak-to-average power ratio of the calibration signal to be reduced, thus reducing the effects of nonlinearities and noise added by the acquisition elements.

[0060] The low resonant frequency spacing of the calibration signal in the low GHz, MHz, or kHz range allows the system to be characterized with high frequency resolution in a single measurement. As used above, the term "broadband" means that the calibration frequency comb covers the entire or a substantial portion of the detection bandwidth of the measurement system.

[0061] In a further embodiment, each mixing element has K=2 outputs connected to inputs of K=2 associated capture elements, each comprising an analog-to-digital converter.

[0062] With K=2 or more outputs per mixing element, it is possible to measure the partially used signal with respect to multiple reference phases and thus reconstruct the field amplitude and the corresponding phase that can be represented by a complex-valued field amplitude. An example of such a mixing element is an in-phase / quadrature (IQ) receiver based for example on a 90° optical hybrid followed by a pair of balanced photodiodes. Alternatively, other implementations are possible, such as a 3×3 multimode interference coupler (MMI) followed by a photodiode.

[0063] In a further embodiment, the effective number of bits (ENOB) of each of the acquisition elements is better than 4 bits, more preferably better than 5 bits, and most preferably better than 6 or 7 bits. The analog bandwidth at which such ENOB of each individual acquisition element is obtained is preferably higher than 5 GHz, more preferably higher than 10 GHz, and most preferably higher than 20 GHz or 30 GHz. By using acquisition elements with high ENOB, the measurement fidelity is increased. However, high bandwidth acquisition elements usually have a lower ENOB than acquisition elements with lower bandwidth. Thus, a system with a large number of low bandwidth, high ENOB acquisition elements can have better measurement fidelity than a system with a small number of high bandwidth, low ENOB acquisition elements.

[0064] In a further embodiment, the signal-to-noise-and-distortion ratio (SINAD) of the reconstructed full-scale sinusoidal test signal exceeds 32 dB for an acquisition bandwidth of at least 90 GHz, where for such a signal, the effective number of bits ENOB is related to the SINAD by:

number

[0065] State-of-the-art electronic analog-to-digital converters reach a bandwidth of 110 GHz with an ENOB of 5

[19] . The measurement system described in this invention has the potential to overcome this limitation by paralleling multiple low-bandwidth, high-ENOB ADCs.

[0066] In a further embodiment, the signal-to-noise and distortion ratio (SINAD) of the reconstructed full-scale sinusoidal test signal is greater than 25 dB for an acquisition bandwidth of at least 290 GHz. In a further embodiment, the signal-to-noise and distortion ratio (SINAD) of the reconstructed full-scale sinusoidal test signal is greater than 20 dB for an acquisition bandwidth of at least 490 GHz.

[0067] In a further embodiment, the signal-to-noise and distortion ratio (SINAD) of the reconstructed full-scale sinusoidal test signal exceeds 15 dB for an acquisition bandwidth of at least 1.9 THz. Note that the scalability of the measurement system allows the total bandwidth to be expanded beyond the limitations of a single high-speed analog-to-digital converter.

[0068] An embodiment of a measurement system for performing one of the aforementioned methods, comprising: An embodiment of the measurement system includes: a. input interface; b. an electromagnetic reference signal source; and c. A subsystem comprising: i. a first dividing element; ii. a second dividing element; iN mixture elements, the n-th mixture element is K n N mixing elements generating N partial mixed signals; TIFF2025504229000009.tif10164iv.Digital signal processor, A subsystem comprising: With: d. the input interface is coupled to the first dividing element; e. an electromagnetic reference signal source coupled to the second splitter element; f. the nth output of the first splitting element is coupled to the nth mixing element; g. the n-th output of the second splitting element is coupled to the n-th mixing element; TIFF2025504229000010.tif16164c.Each acquisition element is coupled to a digital signal processor, h. at least one partial reference signal includes at least two spectral tones; i. The sum of all the partial mixed signals contains redundant information; and j. the digital signal processor is configured to extract at least one model parameter of the measurement system from the redundant information; An embodiment of a measurement system for performing one of the aforementioned methods, characterized in that

[0069] The term "coupled" herein describes a connection between different elements. Such connections can be, for example, - optical or electrical waveguides, - optical fibre or electrical cables, - Optical or electrical wire bonds, or - a free space link, optionally comprising a focusing element, reflector, antenna or similar component; This can be achieved by:

[0070] In a further embodiment of the above-mentioned device, the mixing element comprises one or more detection elements, whereby examples of the detection elements are: ● Photodetectors, ● Rectifier, • Nonlinear elements such as diodes, transistors, or other devices with nonlinear signal transfer characteristics; ● Nonlinear optical elements that utilize, for example, second- or third-order optical nonlinearity. ● Bolometer, or similar device, It is.

[0071] In a further embodiment of the above-mentioned apparatus, the mixing element comprises one or more combining elements, where the combining elements superimpose two or more signals. Examples of binding elements are: ● Directional coupler, ● Multimode interference couplers, and / or ● 90° optical hybrid, • An active or passive electronic circuit designed to superimpose two or more signals; It is.

[0072] In a further embodiment of the above device, the mixing element is embodied by a 90° hybrid followed by a pair of balanced photodetectors. The mixing element thus represents a coherent receiver capable of measuring the in-phase and quadrature components and thus the complete optical field.

[0073] In a further embodiment of the aforementioned device, the mixing element comprises a microwave mixer. The above embodiment operates directly with microwave signals and therefore makes it possible to avoid problems associated with optical signals, such as optical phase drift.

[0074] In a further embodiment of the above mentioned measurement system, all acquisition elements and the analog-to-digital converters used within all acquisition elements are synchronized such that the time relationship between all digitized partial mixed signals is known.

[0075] Synchronization of all acquisition elements is required to determine the time alignment of the digitized partial mixed signals, which facilitates the reconstruction of the digital used signal.

[0076] In a further embodiment, the first and second splitting elements are implemented as power splitters with separate path delays. The advantage of using a power splitter as the splitting element is that it is a compact and simple passive hardware that is robust to manufacturing tolerances and environmental effects and does not require active control. The power splitter can also be compactly manufactured in high index contrast integrated platforms such as indium phosphide (InP) or silicon photonics.

[0077] In a further embodiment, the ENOB of the measurement system is at most 1 bit lower than the ENOB of a single acquisition element. Ideally, the ENOB of the measurement system is the same as the ENOB of the underlying acquisition element. However, in practice, the ENOB of the measurement system is lower than the ENOB of the underlying acquisition element, for example due to additional noise and distortion added by the mixing element. Inaccurate system calibration or additional distortion occurring in the detection system may further degrade the signal quality. Preferably, this ENOB (SINAD) penalty is less than 2 bits (12 dB), more preferably less than 1 bit (6 dB), and most preferably less than 0.5 bits (3 dB). [Brief description of the drawings]

[0078] Further details and features of the invention result from the following description of preferred embodiments, in particular in conjunction with the dependent claims, where each feature can be realized alone or in combination with one another. The invention is not limited to the examples or forms of embodiment.

[0079] Examples or forms of embodiment are illustrated diagrammatically in the following figures, in which identical reference signs indicate identical or functionally identical elements or elements that correspond to each other in terms of functionality:

[0080] Further features and advantages of the present invention, given for purposes of illustration and not of limitation, result from the description of the accompanying drawings, in which:

[0081] [Figure 1] FIG. 2 is a schematic diagram of a subsystem for a further embodiment of a measurement system for performing a method for detecting an electromagnetic use signal according to claim 1; [Diagram 2] FIG. 1 is a schematic diagram of a first embodiment of a measurement system for performing the method for detecting an electromagnetic use signal according to claim 1; [Diagram 3] FIG. 5 is a schematic diagram of a further embodiment of a measurement system for performing the method for detecting electromagnetic use signals according to claim 4; [Figure 4] FIG. 5 is a schematic diagram of a further embodiment of a measurement system for performing the method for detecting electromagnetic use signals according to claim 4; [Diagram 5] FIG. 5 is a schematic diagram of a further embodiment of a measurement system for performing the method for detecting electromagnetic use signals according to claim 4; [Figure 6] FIG. 5 is a schematic diagram of a further embodiment of a measurement system for performing the method for detecting electromagnetic use signals according to claim 4; [Figure 7] FIG. 6 is a schematic diagram of a further embodiment of a measurement system for performing a method for detecting an electromagnetic use signal according to claim 4 or 5; [Figure 8] FIG. 6 is a schematic diagram of a further embodiment of a measurement system for performing a method for detecting an electromagnetic use signal according to claim 4 or 5; [Figure 9] FIG. 6 is a schematic diagram of a further embodiment of a measurement system for performing a method for detecting an electromagnetic use signal according to claim 4 or 5; [Figure 10] FIG. 6 is a schematic diagram of a further embodiment of a measurement system for performing a method for detecting an electromagnetic use signal according to claim 4 or 5; [Figure 11] FIG. 6 is a schematic diagram of a further embodiment of a measurement system for performing a method for detecting an electromagnetic use signal according to claim 4 or 5; [Figure 12] FIG. 6 is a schematic diagram of a further embodiment of a measurement system for performing a method for detecting an electromagnetic use signal according to claim 4 or 5; [Figure 13] FIG. 2 is a schematic diagram of a further embodiment of a measurement system; [Figure 14] FIG. 14 is an exemplary diagram for measuring a 600 GHz wideband optical waveform using the measurement system shown in FIG. 13. [Figure 15] FIG. 13 is a schematic diagram of a further embodiment of a measurement system with an optional correction element that adds a stitching signal. [Figure 16] FIG. 16 is an exemplary diagram for measuring a single light tone using the measurement system shown in FIG. 15. [Figure 17] Schematic schematic diagram of the system configuration used for calibration. [Figure 18] 16 is an exemplary spectrum for a digitized partial mixture signal obtained when measuring a calibration signal according to claim 15. [Figure 19] 16 is an exemplary diagram of a transfer function obtained for a further embodiment calibrated according to claim 15. FIG. [Figure 20] FIG. 13 is a schematic diagram of a further embodiment for a dividing element; [Figure 21] FIG. 2 is a schematic diagram of an input and an output of a further embodiment of a measurement system; DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0082] TIFF2025504229000011.tif131164

[0083] 2 shows a measurement system 2000 related to claim 1, comprising an input interface 2100, a reference signal source 2200 and a subsystem 2500. The electromagnetic input signal 1000 passes through the input interface 2100 and is thereafter referred to as the usage signal 2401. The reference signal source 2200 generates a reference signal 2201 comprising M discrete spectral tones. The above subsystem receives as input the usage signal 2401 and the reference signal 2201 and outputs a digitized usage signal 2551. It is noted that the subsystem 2500, the input interface 2100 and / or the reference signal source 2200 may be merged into a monolithic assembly in which different components are integrated together, rather than being implemented as separate hardware items.

[0084] 3 shows, in comparison with FIG. 2, a measurement system 2000 further comprising a correction element 2300 capable of correcting the used signal 2401. For more details of the correction element, see claim 4.

[0085] 4 shows the measurement system 2000 further comprising a connection between the modification element 2300 and the reference signal source 2200, indicated by a double-headed arrow, in comparison with FIG. 3. This connection indicates that the modification element 2300 can use information or signal components provided by the reference signal source 2200 or can control parameters such as frequency, phase, amplitude, etc. of the reference signal 2201. In particular, this double-headed arrow makes it possible to add a stitching signal to the used signal 2401 in the modification element 2300, phase-locked to the reference signal 2201, or to shift the used signal 2401 in the frequency domain by mixing it with a tone phase-locked to a component of the reference signal 2201.

[0086] Figure 5 shows, in comparison with Figure 4, the measurement system 2000 further comprising a connection between the correction element 2300 and the subsystem 2500, also indicated by a double-headed arrow, which allows information to be exchanged between said correction element and said subsystem, which can be used to digitally undo the effects of the signal corrections made by the correction element.

[0087] Figure 6 shows the measurement system 2000 further comprising a connection between the reference signal source 2200 and the subsystem 2500 in comparison with Figure 5. This connection, also indicated by a double-headed arrow, allows the acquisition elements in the subsystem 2500 to be synchronized with the reference signal source 2200. This connection has the advantage that the number of time-varying model parameters can be reduced. More specifically, the relative time delay from the first sample of the processing block acquired by the acquisition element to the first pulse of the reference signal is not required since this time is fixed by the synchronization mechanism.

[0088] Fig. 7 shows a further embodiment of the measurement system 2000 shown in Fig. 4. The reference signal source 2200 comprises a pump source 2210 and an additional reference signal element 2220, which converts the output signal of the pump source into a reference signal 2201. The additional reference signal element 2220 can be, for example, an electro-optical modulator or any other device based on optical second- or third-order optical nonlinearity, for example a resonator, a highly nonlinear waveguide, or an optical fiber. The correction element 2300 comprises a conversion element 2310, which converts the signal provided by the input interface 2100 into a signal in the same frequency range as the reference signal 2201 provided by the reference signal source 2200. A specific example of the conversion element 2310 is an electro-optical modulator, which modulates the use signal provided by the input interface onto a carrier provided by the reference signal source 2200.

[0089] Figure 8 shows a further embodiment of the measurement system 2000 shown in Figure 4. The reference signal source 2200 comprises a pump source 2210 and a reference signal element 2220 that converts the output signal of the pump source into a reference signal 2201. The modification element 2300 comprises a stitching signal element 2320 and a combining element 2330. The stitching signal element 2320 obtains a signal from the reference signal source, which is converted into a stitching signal and then combined by the combining element 2330 with the used signal obtained from the input interface 2100.

[0090] FIG. 9 shows a further embodiment of the measurement system 2000 shown in FIG. 4, in which the conversion element 2310 as described in FIG. 7 and the stitching signal element 2320 and combining element 2330 as described in FIG. 8 are combined into a single correction element 2300.

[0091] Figure 10 shows a further embodiment of the measurement system 2000 shown in Figure 3, which is similar to the measurement system described in Figure 7. However, in the case of Figure 10, the correction element 2300 is independent of the reference signal source, and the correction element comprises a pump source 2340 and a conversion element 2310. This configuration can be more flexible compared to the configuration of Figure 7, since the combining element 2300 and the reference signal source 2200 are independent.

[0092] Figure 11 shows a further embodiment of the measurement system 2000 shown in Figure 3, which is similar to the measurement system described in Figure 8. However, in the case of Figure 11, the modification element 2300 is independent of the reference signal source, and the modification element comprises a stitching signal element 2320 and a combining element 2330. This configuration can be more flexible than the configuration of Figure 8, since the combining element 2300 and the reference signal source 2200 are independent.

[0093] Figure 12 shows a further embodiment of the measurement system 2000 shown in Figure 3, which is similar to the measurement system described in Figure 9. However, in the case of Figure 12, the modification element 2300 is independent of the reference signal source, and the modification element comprises a stitching signal element 2320, a combining element 2330, a pump source 2340, and a conversion element 2310. This configuration can be more flexible than the configuration of Figure 9, since the combining element 2300 and the reference signal source 2200 are independent.

[0094] FIG. 13 shows a further embodiment of the measuring device 2000. In this case, the input signal 1000 is an optical signal in the wavelength range between 1200 nm and 1700 nm so that the implementation can rely on widely available components for optical communication. The use signal 2401 is provided by the input interface 2100, optionally amplified by an optical amplifier 2010 and band-pass filtered by an optical bandpass 2011 for better signal quality before being coupled to the splitting element 2510 of the subsystem 2500. The reference signal source 2200 provides an optical frequency comb with M discrete tones as the optical reference waveform 2201. The first splitting element 2510 splits the use signal evenly into N partial use signals 2511-1 to 2511-N. The second splitting element 2520 splits the reference signal evenly into N paths and further optionally applies a time delay τ in the nth path to obtain a partial reference signal 2521-n. n where n=1, ...N. In a particular embodiment, for all time delays τ n are different from each other. Each of the mixing elements 2530-1 to 2530-N includes a 90° optical hybrid 2532 and two balanced photodetectors 2533 to acquire a partial mixed signal 2531. Each mixing element 2530-n outputs two partial mixed signals 2531-nl and 2531-n-2 corresponding to the in-phase and quadrature components of a respective in-phase / quadrature (IQ) receiver, where n=1,...,N. Each acquisition element 2540-nk includes an analog-to-digital converter 2542. All the analog-to-digital converters 2542 operate synchronously. TIFF2025504229000012.tif22164

[0095] In the following, a more detailed system model for the embodiment of Fig. 13 is given. In the following description, lower case letters refer to optical signals, upper case letters refer to (mostly) baseband signals, underlines indicate complex values, tildes indicate Fourier transformed quantities, and primes distinguish between estimated and true values. Note that some symbols using upper case letters also capture optical and electrical effects at baseband. Complex amplitude of reference signal 2201 a LO (t) is modeled as a frequency comb consisting of M equally spaced phase-locked tones, where LO,m The mth tone has complex amplitude A LO,m has.

number

[0096] The frequency difference between two adjacent comb lines is fixed, and the resonant frequency spacing (FSR) of the comb source is f r is given by: The center frequency of the system is

number

[0097] TIFF2025504229000015.tif8164

number

[0098] TIFF2025504229000017.tif21164

number

number

[0099] TIFF2025504229000021.tif20164

number

[0100] moreover,

number

number

number

[0101] The matrix equation (11) in this case is I 1 ~I N and Q 1 ~Q N The digitized partial mixed signal 2541 for the in-phase (I) and quadrature (Q) components, referred to as a S This can be inverted along with equation (9) to obtain a transformation relating the used signal 2401, called .times. ... A preferred choice of delay introduced in splitting element 2510 is:

number

number

[0102] TIFF2025504229000029.tif38164 Model parameters relating to the characteristics of the detection system are therefore found by minimizing a cost function that measures the error between redundant signal components.

number

number

[0103] Fig. 14 shows an application example for the measurement system 2000 shown in Fig. 13 and evaluated according to the approach described in the previous paragraph. In this example, the working signal 2401 has a bandwidth of 600 GHz and includes several optical data signals modulated in amplitude and phase. The data signals were generated by individually modulating seven optical carriers provided by seven free-running lasers. As a reference signal, a dissipative Kerr soliton comb with a resonant frequency spacing FSR of 150 GHz is used. As mixing elements, four parallel coherent receivers are utilized. Fig. 13 shows the spectrum of the reconstructed digital working signal 2552, the constellation diagram for the constructed data signals, and the signal-to-noise power ratio of each of the constructed data signals. These results prove that the measurement method according to claim 1 is feasible.

[0104] Figure 15 shows a practical realization of the measurement system 2000 according to Figure 3. In this case, a combining element 2330 in a correction element 2300 adds four single tone signals generated by separate pump sources 2340 to the used signal 2401. The pump sources 2340 are also part of the correction element.

[0105] Figure 16 shows the measurement results of a single tone recorded by a measurement system configured according to Figure 15. Note that the subsystems of the measurement system are configured according to Figure 13. The used signal 2401 consists of a single tone and does not feature any spectral components in the spectral range that should contain redundant information, thereby making the estimation of at least one model parameter difficult. To overcome this problem, the correction element 2300 adds a stitching signal to the used signal. In this case, the stitching signal contains four weak tones in the spectral range that is assumed to contain redundant information. Figure 16 shows - digitally reconstructed signal in the time domain 2551, - the spectrum of the reconstructed digital use signal 2552, including the tones related to the input signal A, as well as the four tones b, c, d, e related to the stitching signal, - the spectrum of the reconstructed digital input signal 2554 obtained by digitally removing the corrections introduced by the correction elements, is shown from top to bottom.

[0106] Note that the frequencies on the abscissa of both the display of the reconstructed digital usage signal 2552 (second column of FIG. 16) and the display of the reconstructed digital input signal 2554 (third column of FIG. 16) are shifted to cover the range of 0 to 500 GHz.

[0107] Noise spurs around the tones associated with the input signal are associated with a low carrier-to-noise ratio of the reference signal and can be avoided by using a better reference signal source.

[0108] 17 shows a configuration that can be used to calibrate a measurement system 2000. A calibration signal source 1100 is used to generate a calibration signal 1101. In this case, the measurement system 2000 processes the measured calibration signal 1101 to obtain all parameters of the system model. TIFF2025504229000033.tif18164

[0109] FIG. 18 shows an example of a calibration measurement for a measurement system 2000 according to FIG. 17, where a frequency comb according to claim 15 is used as the calibration signal 1101. In this case, the measurement system 2000 is internally configured according to FIG. 13. FIG. 18 shows a spectral detail of the digitized partial mixing signal 2541-nk. Since a frequency comb is used as the calibration signal 1101, the mixing products of the calibration signal and the different tones of the reference signal 2201 do not overlap. This makes it possible to characterize the entire measurement system 2000 with a single-shot measurement by evaluating the amplitude and phase of the received comb tones of FIG. 13 and comparing them with the amplitude and phase of the comb tones emitted by the calibration signal source 1100, whose emitted pulse shape was measured independently by FROG. The single-shot measurement is important because some model parameters may change between subsequent measurements.

[0110] Figure 19 shows some calibration results obtained from the measurement setup shown in Figure 17 and the calibration signal 1101 of Figure 18. According to equation (6) four transfer functions are shown describing the measurement system 2000 according to Figure 13. Figure 19 shows that a detailed calibration of the amplitude and phase is obtained and the phase relationship between the different detection paths is obtained.

[0111] 20 shows further embodiments A to C of the first dividing element 2510 and similar embodiments D to F of the second dividing element 2520. The first dividing element 2510 divides the use signal 2401 into N partial use signals 2511-1 to 2511-N. The second dividing element 2520 divides the reference signal 2201 into N partial reference signals 2521-1 to 2521-N.

[0112] A first embodiment A of the first splitting element 2510 comprises a power splitter 2512 with an additional delay 2513. Note that not all delays 2513 are the same, and the delays 2513 may simply be introduced by the connection between the first splitting element 2510 and the mixing element 2530. However, the delays 2513 are shown to illustrate that these delays are important design parameters for the above system and must be selected appropriately (see equation (12) for examples of preferred delays).

[0113] The second embodiment B comprises an optical filter module, which splits the used signal into N paths and further applies an amplitude transfer function, instead of a power splitter. This is similar to a spectrum slice system, but the requirements for the filter steepness and passband are relaxed due to the defined time delay 2513. This approach can improve the scalability of the measurement system 2000 when implemented as an optical integrated circuit. The combination of a filter and a power splitter can help to avoid low signal levels at the acquisition element 2540 when a large number of mixing elements 2530 are used.

[0114] The third embodiment C of the first splitting element 2510 includes both a filter module 2514 and a power splitter 2512 with a defined delay 2513, thus describing a hierarchical system. Compared to B, fewer filter outputs are required, since at least one of the filter outputs is further split by the power splitter 2512. Compared to A, the measurement system 2000 generates fewer out-of-band mixing products, improving power efficiency. Note that the order of the filter module 2514 and the power splitter 2512 can also be changed, allowing for a multi-stage hierarchical structure.

[0115] The 1D, 2E, and 3F embodiments of the second division element 2520 have similar structures compared to the 1A, 2B, and 3C embodiments of the first division element 2510. However, parameters such as time delay 2513, filter transfer function 2514 are designed independently.

[0116] 21 shows that the measurement system 2000 outputs either a reconstructed digital input signal A or a reconstructed digital input signal B obtained by at least partially digitally correcting the correction applied by the correction element. It can also output their spectra C, D, or a combination of the above mentioned signals.

[0117] References [1]Scott, RP, Fontaine, NK, Heritage, JP, Yoo, SJB: 'Dynamic optical arbitrary waveform generation and measurement', Opt. Express, 2010, 18, (18), pp. 18655-18670. https: / / doi.org / 10.1364 / OE.18.018655 [2]Fang, D., et al.: 'Optical Arbitrary Waveform Measurement (OAWM) on the Silicon Photonic Platform', in: Optical Fiber Communication Conference (OFC'21), San Diego (CA), USA, June 6 - 11, 2021; The Optical Society of America (OSA); paper F2E.G. https: / / doi.org / 10.5445 / IR / 1000135495 [3]Fontaine, N.K., Scott, R.P., Zhou, L., Soares, F.M., Heritage, J.P., Yoo, S.J.B.: 'Real-time full-field arbitrary optical waveform measurement', Nature Photonics, 2010, 4, (4), pp. 248-254. https: / / doi.org / 10.1038 / nphoton.2010.28 [4]Fischer, J.K., Ludwig, R., Molle, L., et al.: 'High-Speed Digital Coherent Receiver Based on Parallel Optical Sampling', Journal of Lightwave Technology, 2011, 29, (4), pp. 378-385. https: / / doi.org / 10.1109 / JLT.2010.2090132 [5]Valley, G.C.: 'Photonic analog-to-digital converters', Opt. Express, 2007, 15, (5), 1955--1982. https: / / doi.org / 10.1364 / OE.15.001955 [6]Fontaine, N.K., Raybon, G., Guan, B., et al.: '228-GHz Coherent Receiver using Digital Optical Bandwidth Inter-leaving and Reception of 214-GBd (856-Gb / s) PDM-QPSK'; in European Conference and Exhibition on Optical Communication, OSA Technical Digest (online) (Optical Society of America, 2012), paper Th.3.A.1. https: / / doi.org / 10.1364 / ECEOC.2012.Th.3.A.1 [7]Fontaine, N.K., Scott, R.P., Heritage, J.P., Yoo, S.J.B.: 'Near quantum-limited, single-shot coherent arbitrary optical waveform measurements', Opt. Express, 2009, 17, (15), pp. 12332-12344. https: / / doi.org / 10.1364 / OE.17.012332 [8]Chen, X., Xie, X., Kim, I., Li, G., Zhang, H., Zhou, B.: 'Coherent Detection Using Optical Time-Domain Sampling', IEEE Photonics Technology Letters, 2009, 21, (5), pp. 286-288. https: / / doi.org / 10.1109 / OFC.2008.4528094 [9]Okamoto, K., Ito, F.: 'Dual-Channel Linear Optical Sampling for Simultaneously Monitoring Ultrafast Intensity and Phase Modulation', Journal of Lightwave Technology, 2009, 27, (12), pp. 2169-2175. https: / / doi.org / 10.1109 / JLT.2008.2009646

[10] Sunnerud, H., Skold, M., Westlund, M., Andrekson, P.A.: 'Characterization of Complex Optical Modulation Formats at 100 Gb / s and Beyond by Coherent Optical Sampling', Journal of Lightwave Technology, 2012, 30, (24), pp. 3747-3759. https: / / doi.org / 10.1109 / JLT.2012.2204957

[11] Ummethala, S., Kemal, J.N., Alam, A.S., et al.: 'Hybrid Electro-Optic Modulator Combining Silicon Photonic Slot Waveguides with High-k Radio-Frequency Slotlines' (20.10.2020). https: / / doi.org / 10.1364 / OPTICA.411161

[12] Kieninger, C., Fullner, C., Zwickel, H., et al.: 'Silicon-organic hybrid (SOH) Mach-Zehnder modulators for 100 GBd PAM4 signaling with sub-1 dB phase-shifter loss', Opt. Express, OE, 2020, 28, (17), p. 24693. https: / / doi.org / 10.1364 / OE.390315

[13] Ummethala, S., Harter, T., Koehnle, K., et al.: 'THz-to-optical conversion in wireless communications using an ultra-broadband plasmonic modulator', Nature Photonics, 2019, 13, (8), pp. 519-524. https: / / doi.org / 10.1038 / s41566-019-0475-6

[14] K. Kikuchi, "Fundamentals of Coherent Optical Fiber Communications," Journal of Lightwave Technology 34, 157-179 (2016). https: / / doi.org / 10.1109 / JLT.2015.2463719

[15] N. K. Fontaine, R. P. Scott, and S. Yoo, "Dynamic optical arbitrary waveform generation and detection in InP photonic integrated circuits for Tb / s optical communications," Optics Communications 284, 3693-3705 (2011). https: / / doi.Org / 10.1016 / j.optcom.2011.03.045

[16] L. D. Coelho, 0. Gaete, and N. Hanik, "An algorithm for global optimization of optical communication systems,", pp. 541-550. https: / / doi.org / 10.1016 / j.aeue.2009.02.009

[17] J. Liu, E. Lucas, A. S. Raja, J. He, J. Riemensberger, R. N. Wang, M. Karpov, H. Guo, R. Bouchand, and T. J. Kippenberg, "Photonic microwave generation in the X- and K-band using integrated soliton microcombs, " Nature Photonics 14, 486-491 (2020). https: / / doi.org / 10.1038 / s41566-020-0617-x

[18] A. Khilo et al., "Photonic ADC overcoming the bottleneck of electronic jitter," Opt. Express 20, 4454-4469 (2012). https: / / doi.org / 10.1364 / OE.20.004454

[19] Keysight Technologies, "Infiniium UXR-Series Oscilloscopes," Data Sheet (2019) [Explanation of symbols]

[0118] 100, ... 175 Method embodiments according to the respective corresponding claims from claims 1 to 19 200, ... 235 Embodiments of a measurement system 2000 according to the invention corresponding to claims 20 to 29 1000 Input Signals 1100 Calibration Signal Source 1101 Calibration signal 2000 Measurement System 2010 Amplifier 2011 Bandpass Filter 2100 Input Interface 2200 Reference signal source 2201 Reference Signal 2210 Pump Source 2220 Reference Signal Element 2300 Modifications 2310 Conversion Element 2320 Stitching Signal Elements 2321 Stitching Signal 2330 Joint Element 2340 Pump Source 2401 Signal used 2500 Subsystem 2510 First division element 2511 Total number of all N partial use signals 2511-nnth partial use signal, where n=1,...N 2512 power splitter 2513 Time Delay 2514 Filter 2520 Second division element 2521 The total number of all N partial reference signals 2521-nnth partial reference signal, where n=1,...N 2530 The total number of all N mixed elements 2530 - the nnth mixed element, where n=1,...N TIFF2025504229000034.tif181642531-n The total number of all partial mixture signals generated by the nth mixture element, where n=1,...N 2531-nk The kth partial mixed signal generated by the nth mixing element, where n=1,...N, k=1,2,...K n 2532 90° Optical Hybrid 2533 Balanced Photodetector 2540 Total number of all retrieved elements 2540-nk The kth acquisition element used to digitize the kth partial mixed signal generated by the nth mixing element, where n=1,...N, k=1,2,...K n TIFF2025504229000035.tif101642541-n All K corresponding to the n-th mixed element n The total number of digitized partial mixed signals, where n=1,...N 2541-nk kth digitized partial mixture signal corresponding to nth mixture element, where n=1, 2, ... N, k=1, 2, ... K n 2542 Analog-to-Digital Converter (ADC) 2550 Digital Signal Processor 2551 Reconstructed digital use signal / Digital use signal 2552 Digital Signal Spectrum 2553 Reconstructed digital input signal / Digital input signal Spectrum of 2554 digital input signal

Claims

1. A method (100) for detecting an electromagnetic use signal (2401) by a measurement system (2000), comprising: The method (100) comprises the steps of: a. providing an input interface (2100) for an electromagnetic input signal (1000), adapted to be supplied with an electromagnetic use signal (2401); b. providing an electromagnetic reference signal source (2200) configured to provide an electromagnetic reference signal (2201) including M discrete tones (M>2); and c. providing a subsystem (2500), The subsystem (2500) i. a first division element (2510) configured to divide said use signal (2401) into a number N (N>2) of partial use signals (2511); ii. a second division element (2520) configured to divide said reference signal (2201) into a number N of partial reference signals (2521), where N is the number of corresponding partial use signals (2511); Equipped with Steps include: The method (100) comprises: d. at least one partial reference signal (2531-n) includes at least two spectral tones; and e. the entirety of all of the partial mixed signals (2531) contains redundant information that is used to determine the at least one model parameter of the measurement system (2000); Characterized by: Method (100).

2. The method (105) of claim 1, wherein M≦N.

3. The method of claim 1 , wherein M>N.

4. The method (110) according to any one of claims 1 to 3, further comprising providing a modifying element (2300) configured such that the usage signal (2401) provided by the input interface (2100) is modified.

5. The modification element (2300) performs the following operations: a. adding a stitching signal (2321) to the use signal (2401); and / or b. Transforming the signal used (2401); The method of claim 4, further comprising modifying the usage signal by one or more of the following:

6. The method (116) according to claim 4 or 5, wherein the digital input signal (2553) is reconstructed from a reconstructed digital usage signal by at least partially removing the modification caused by the modification element.

7. The method (120) according to any one of the preceding claims, wherein the working signal and the reference signal have centre frequencies within the range between 100 THz and 600 THz.

8. the use signal and the reference signal have a center frequency within the range between 150 THz and 250 THz; The method (121) according to any one of claims 1 to 7.

9. The method (125) of any one of claims 1 to 8, wherein the M discrete tones of the reference signal (2201) are equally spaced in frequency and are derived from an optical frequency comb.

10. The method (126) of any one of claims 1 to 9, wherein the M discrete tones of the reference signal (2201) have a resonant frequency spacing between 10 GHz and 100 GHz.

11. The method (130) of any one of claims 1 to 10, wherein adjacent pulses of the reference signal (2201) overlap in time.

12. The method (135) according to any one of claims 1 to 11, wherein the at least one model parameter estimated by the redundant information contained in the entirety of all the partial mixed signals (2351) comprises a characteristic of the partial reference signal (2521) or a characteristic of an optical or electrical circuit.

13. The method (140) according to any one of the preceding claims, wherein the system model comprises the magnitude and / or phase of a frequency dependent transfer function of the whole system or of part of the system.

14. The method (145) of any one of claims 1 to 13, wherein the system features are extracted in dedicated calibration measurements using a known electromagnetic waveform, called a calibration signal (1101), provided by a calibration signal source (1100).

15. The method of claim 14 , wherein the calibration signal is a broadband optical frequency comb having a resonant frequency spacing of less than 5 GHz.

16. The method (155) according to any one of the preceding claims, wherein each mixing element (021) has K=2 outputs connected to inputs of K=2 associated acquisition elements (2540), each comprising an analog-to-digital converter.

17. The method (160) of any one of claims 1 to 16, wherein the effective number of bits (ENOB) of each of the capture elements (2540) is better than six.

18. The method (165) of any one of claims 1 to 17, wherein the signal-to-noise-and-distortion ratio (SINAD) of the reconstructed full-scale sinusoidal test signal is greater than 32 dB for an acquisition bandwidth of at least 90 GHz.

19. The method (170) of any one of claims 1 to 18, wherein the signal-to-noise-and-distortion ratio (SINAD) of the reconstructed full-scale sinusoidal test signal is greater than 25 dB for an acquisition bandwidth of at least 290 GHz.

20. A measurement system (200) configured to be able to perform one of the methods (100, . . . 175) according to any one of claims 1 to 20, comprising: The measurement system (200) a. An input interface (2100); b. an electromagnetic reference signal source (2200) configured to provide an electromagnetic reference signal (2201) including M discrete tones (M>2); c. A subsystem (2500), i. a first division element (2510); ii. a second division element (2520); and iii. N mixing elements (2530), the n-th mixing element (2530-n) being K n N mixing elements (2530) generating N partial mixed signals (2531-n); v. a digital signal processor (2550); A subsystem (2500) comprising: Equipped with d. the input interface (2100) is coupled to the first division element (2510); e. the electromagnetic reference signal source (2200) is coupled to the second splitting element (2520); f. the n-th output of the first splitting element (2510) is coupled to the n-th mixing element (2530-n); g. the n-th output of the second splitting element (2520) is coupled to the n-th mixing element (2530-n); i. each acquisition element (2540-n-k) is coupled to said digital signal processor (2550); j. at least one partial reference signal (2531) includes at least two spectral tones; k. The sum of all said partial mixed signals (2531) contains redundant information; and l. the digital signal processor (2550) is configured to extract at least one model parameter of the measurement system (2000) from the redundant information; The measurement system (200).

21. The measurement system (205) of claim 21 , wherein the mixing element comprises one or more sensing elements.

22. the mixing element comprises one or more combining elements; A combining element superimposes two or more signals; 23. The measurement system (210) of claim 21 or 22.

23. Measurement system (215) according to any one of claims 21 to 23, wherein the mixing element is embodied by a 90° hybrid followed by a pair of balanced photodetectors.

24. The measurement system (220) of any one of claims 21 to 24, wherein the mixing element comprises a microwave mixer.

25. 26. The measurement system (225) of any one of claims 21 to 25, wherein all acquisition elements and analog-to-digital converters used within all acquisition elements are synchronized such that the time relationships between all digitized partial mixed signals are known.

26. The measurement system (230) of any one of claims 21 to 26, wherein the first and second splitting elements are implemented as power splitters in conjunction with separate path delays.

27. A measurement system (235) according to any one of claims 21 to 27, having an ENOB at most one bit lower than the ENOB of a single acquisition element (2540).

28. Use of one of the methods (100,..175) according to any one of claims 1 to 19 for detecting an electromagnetic use signal.

29. Use of one of the measurement systems (200,..235) according to any one of claims 20 to 29 for detecting an electromagnetic use signal by one of the methods according to any one of claims 1 to 19.

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