Motor control device for electric compressor

Abstract

To provide a motor control device of an electric compressor capable of suppressing inaccurate position estimation of a motor (rotor) due to superimposition of a harmonic voltage on a voltage command.SOLUTION: A motor control device 100 includes a PWM calculator 204 that generates a PWM signal for controlling a motor 910 on the basis of an addition value obtained by adding a harmonic voltage command value to a voltage command value for driving the motor 910, and a dead time corrector 213 that corrects a dead time for a power conversion unit 10. The dead time corrector 213 calculates the corrected harmonic current by correcting the phase of the harmonic current based on the output delay Tdel and the frequency ω h of the harmonic current, and corrects the dead time based on the corrected harmonic current and the corrected primary current obtained by correcting the primary current based on the output delay Tdel and the frequency ω of the primary current.SELECTED DRAWING: Figure 4

Description

[Technical Field]

[0001] The present disclosure relates to a motor control device for an electric compressor. [Background technology]

[0002] For example, Japanese Patent Laid-Open Publication No. 8-308286 (Patent Document 1) discloses detecting the operation of a motor based on an induced electromotive force induced in the motor.

[0003] Japanese Patent Laid-Open Publication No. 7-245981 (Patent Document 2) discloses a system that detects the magnetic pole position of a motor using a d-axis voltage command value on which an alternating voltage is superimposed. [Prior art documents] [Patent documents]

[0004] [Patent Document 1] Japanese Patent Application Publication No. 8-308286 [Patent Document 2] Japanese Patent Application Publication No. 7-245981 Summary of the Invention [Problem to be solved by the invention]

[0005] When driving a motor for a relatively large-capacity electric compressor, the motor may frequently switch between on and off (ON / OFF operation) when the refrigerant flow rate of the electric compressor is low. In this case, the electric compressor may be damaged due to an oil shortage or other issues. Therefore, in order to avoid this ON / OFF operation, it is possible to control the motor at a relatively low speed.

[0006] However, since the induced electromotive force of a motor is small in the low speed region, there is a drawback in that the position of the motor cannot be accurately estimated using the method described in Patent Document 1.

[0007] In contrast, the technique of Patent Document 2 makes it possible to estimate the position of the motor (rotor) relatively accurately even in low-speed regions. However, by adding a high-frequency alternating voltage (harmonic voltage) to the voltage command, a high-frequency alternating current (harmonic current) is superimposed on the AC current flowing through the motor. This causes irregular timing at which the AC current switches between positive and negative, which can lead to inaccurate dead time correction. In this case, there is a risk that the motor position will be estimated inaccurately.

[0008] An object of the present technology is to provide a motor control device for an electric compressor that can suppress inaccuracies in motor (rotor) position estimation caused by superimposing harmonic voltages on a voltage command. [Means for solving the problem]

[0009] According to one aspect of the present disclosure, there is provided a motor control device for an electric compressor, the motor control device including: a PWM signal generating unit that generates a PWM signal to control the motor based on a sum obtained by adding a harmonic voltage command value for generating a harmonic current used to estimate a rotor position of the motor to a driving voltage command value for generating a primary current that drives the motor of the electric compressor; a position estimator that estimates the rotor position based on the harmonic current; and a compensator that corrects dead time for a power converter that outputs an AC voltage based on the PWM signal to the motor. Given that a corrected primary current is defined as an output delay indicating a delay in output of the PWM signal resulting from generation of the PWM signal and a primary current whose phase has been corrected based on the frequency of the primary current, the compensator calculates the corrected harmonic current by correcting the phase of the harmonic current based on the output delay and the frequency of the harmonic current, and corrects the dead time based on the corrected harmonic current and the corrected primary current.

[0010] In the motor control device for an electric compressor according to one aspect of the present disclosure, as described above, the corrector corrects the dead time based on the corrected harmonic current, the phase of which is corrected based on the output delay and the frequency of the harmonic current, and the corrected primary current, the phase of which is corrected based on the output delay and the frequency of the primary current. This allows for more appropriate correction of the phase shift of the harmonic current due to the output delay than, for example, when the phase of the harmonic current is corrected based on the frequency of the primary current rather than based on the frequency of the harmonic current. As a result, the current value after the output delay can be accurately estimated, thereby preventing inappropriate dead time correction. This prevents inaccurate motor (rotor) position estimation caused by superimposing harmonic voltages on the voltage command.

[0011] The corrector may calculate a corrected combined current by correcting the phase of the combined current, which is a combination of the corrected harmonic current and the primary current on the γ-δ axes, which are the estimated axes of the dq axes, based on the output delay and the frequency of the primary current, and then correct the dead time using the corrected combined current. With this configuration, the dead time can be corrected based on the corrected harmonic current corrected based on the frequency of the harmonic current, which is further corrected based on the frequency of the primary current. As a result, the phase shift of the harmonic current caused by the output delay can be more appropriately corrected than when correction based on the frequency of the primary current is not performed. Furthermore, by correcting the corrected harmonic current and the primary current together based on the frequency of the primary current, the processing of the motor control device can be simplified compared to when the corrected harmonic current and the primary current are corrected separately based on the frequency of the primary current.

[0012] The motor may be a three-phase AC motor. The corrector may calculate a corrected primary current and a corrected harmonic current for each phase, and correct the dead time for each phase based on the sum of the corrected harmonic current and the corrected primary current for each phase. With this configuration, the dead time can be corrected using the AC current value for each phase that has not been subjected to coordinate transformation, thereby simplifying the dead time correction process. [Effects of the Invention]

[0013] According to the present technology, it is possible to prevent the position estimation of the motor (rotor) from becoming inaccurate due to the superposition of harmonic voltages on the voltage command. [Brief explanation of the drawings]

[0014] [Figure 1] 1 is a block diagram showing the configuration of a motor system according to a first embodiment. [Figure 2] FIG. 2 is a diagram showing a detailed configuration of the motor system according to the first embodiment. [Figure 3] FIG. 2 is a diagram showing the d, q axes and the γ, δ axes. [Figure 4] FIG. 2 is a diagram illustrating a detailed configuration of a controller according to the first embodiment. [Figure 5] FIG. 10 is a diagram showing a composite current of the γ-δ axes before and after correction based on an output delay. [Figure 6] 10 is a vector diagram showing a combined current of a primary current and a corrected harmonic current. [Figure 7] 1 is a vector diagram showing a resultant current mapped from the γ-δ axes to the α-β axes. [Figure 8] FIG. 10 is a diagram illustrating a detailed configuration of a controller according to a second embodiment. [Figure 9] FIG. 9 is a diagram illustrating a detailed configuration of a correction unit in FIG. 8. [Figure 10] FIG. 10 is a diagram illustrating a detailed configuration of the dead time corrector in FIG. DETAILED DESCRIPTION OF THE INVENTION

[0015] [First embodiment] A first embodiment of the present disclosure will be described in detail with reference to the drawings. In the drawings, the same or corresponding parts are designated by the same reference numerals and their description will not be repeated.

[0016] FIG. 1 is a diagram showing the overall configuration of a motor system 1 including a motor control device 100 for an electric compressor 900 according to a first embodiment. The motor system 1 is mounted on, for example, an electric vehicle. However, the use of the motor system 1 is not limited to vehicle use. The motor system 1 may also be used in a stationary system (for example, an air conditioning system). The motor system 1 includes the motor control device 100, a motor 910 for the electric compressor 900, a power source 800, and a main controller 700.

[0017] The power source 800 supplies power to the motor control device 100. The power source 800 is, for example, a DC power source (DC system) such as a storage battery or a solar cell. The power source 800 may also be an AC power source (AC system). In the case of an AC power source, the AC power must be rectified using a rectifier to convert it into DC.

[0018] The motor control device 100 drives (controls) the motor 910. The motor control device 100 includes a power conversion unit 10 and a controller 20. The power conversion unit 10 performs power conversion operations on the power supplied from the power source 800. The controller 20 controls the power conversion unit 10 in accordance with control commands from the main controller 700. The control commands from the main controller 700 to the controller 20 include a speed command Fm* (command related to the angular acceleration of the motor 910). The power conversion unit 10 is an example of a "power converter" in the present disclosure.

[0019] Motor 910 is a three-phase AC rotating electric machine or a three-phase brushless DC rotating electric machine, such as an IPM (Interior Permanent Magnet) motor. Motor 910 is not provided with a position sensor (resolver) that detects the position of rotor 911 (described below, FIG. 3). Therefore, motor control device 100 executes sensorless control of motor 910.

[0020] Fig. 2 is a diagram showing an example of the configuration of the motor system 1. However, in Fig. 2, the main controller 700 (Fig. 1) is not shown.

[0021] In this example, the power source 800 is a storage battery. The power source 800 outputs DC power to the power conversion unit 10 via DC terminals Tp and Tn of the power conversion unit 10. The power source 800 is provided with a monitoring unit (including a voltage sensor, a current sensor, etc.) 810 that monitors the state of the power source 800. The monitoring unit 810 outputs the monitored voltage, current, etc. to the controller 20.

[0022] Power conversion unit 10 converts DC power from power source 800 into AC power (AC voltage) in accordance with a control command from controller 20, and outputs the AC power (AC voltage) to motor 910. More specifically, power conversion unit 10 includes, for example, a voltage sensor 12 and an inverter 13.

[0023] The voltage sensor 12 detects the voltage between the power line PL and the power line NL, and outputs the detected voltage to the controller 20.

[0024] Inverter 13 is, for example, a two-level three-phase full-bridge circuit. In accordance with a control command from controller 20, inverter 13 converts DC power between power lines PL and NL into AC power and outputs the AC power to AC terminals Tu, Tv, and Tw. In this example, inverter 13 includes six switching elements Q1 to Q6 and six freewheel diodes D1 to D6. Each of switching elements Q1 to Q6 is a metal-oxide-semiconductor field-effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), a bipolar transistor, or the like. The freewheel diodes D1 to D6 are connected in antiparallel to switching elements Q1 to Q6, respectively. Switching elements Q1 and Q2 are connected in series to form a U-phase arm of the full-bridge circuit. Switching elements Q3 and Q4 are connected in series to form a V-phase arm of the full-bridge circuit. Switching elements Q5 and Q6 are connected in series to form a W-phase arm of the full-bridge circuit. The U-phase arm, V-phase arm, and W-phase arm are connected to AC terminals Tu, Tv, and Tw, respectively. Each phase arm is connected between power line PL and power line NL. When MOSFETs are used as switching elements Q1 to Q6, parasitic diodes of the MOSFETs serve as freewheel diodes D1 to D6.

[0025] The motor 910 includes a rotor 911 (FIG. 3) having a permanent magnet and a stator 912 around which coils are wound. In this example, the stator 912 has a U-phase coil, a V-phase coil, and a W-phase coil. One end of each phase coil is star-connected to the neutral point. The other end of each phase coil is connected to a connection point of the switching elements of each phase arm of the inverter 13.

[0026] The motor 910 is provided with current sensors 913 and 914. The current sensor 913 detects a U-phase current Iu flowing through the motor 910. The current sensor 914 detects a V-phase current Iv flowing through the motor 910. Each of the current sensors 913 and 914 outputs the detected current to the controller 20. Note that the U-phase current Iu and the W-phase current Iw, or the V-phase current Iv and the W-phase current Iw, may be output to the controller 20.

[0027] The controller 20 controls the inverter 13 based on a speed command Fm* from the main controller 700 (FIG. 1) and detection results from various sensors (such as the monitoring unit 810, the voltage sensor 12, and the current sensors 913 and 914). For example, the controller 20 outputs a switching signal SW to each of six switching elements Q1 to Q6 included in the inverter 13. The switching signal SW (FIG. 1) is a PWM (Pulse Width Modulation) signal.

[0028] The controller 20 includes, as its main components, a processor 21 and a memory 22. The processor 21 includes processing circuitry such as a central processing unit (CPU) and a microprocessing unit (MPU). The memory 22 includes volatile storage devices such as dynamic random access memory (DRAM) and static random access memory (SRAM), and nonvolatile storage devices such as a hard disk drive (HDD), a solid state drive (SSD), and a flash memory. The memory 22 stores a system program including an operating system (OS), a control program including computer-readable code, and various parameters for controlling the power conversion operation of the power conversion unit 10. The processor 21 performs various arithmetic operations by reading the system program, the control program, and the parameters, and then loading and executing them in the memory 22. The arithmetic operations performed by the controller 20 may be performed using an application-specific integrated circuit (ASIC), a field-programmable gate array (FPGA), or the like.

[0029] In inverter 13, a dead time is provided to prevent the upper arm switching elements (Q1, Q3, Q5) and the lower arm switching elements (Q2, Q4, Q6) in each phase from being turned on simultaneously. Furthermore, the dead time is corrected in inverter 13 to prevent a voltage error (a voltage error between a PWM command and the inverter output voltage) from occurring due to the dead time. If the dead time is not corrected correctly, an unintended excessive or insufficient voltage will be applied to motor 910. This voltage error causes a ripple in the current flowing through motor 910, and this current ripple also causes a ripple in the estimated position of rotor 911 (estimated position ripple).

[0030] It is not essential that controller 20 and main controller 700 of motor control device 100 are provided separately. Controller 20 may be configured to calculate speed command Fm* by itself.

[0031] <Calculating the angle difference> 3 is a diagram illustrating the relationship between the magnetic pole position of rotor 911 and coordinate axes while motor 910 is in operation. As shown in FIG. 3, the d-axis is an axis extending from rotation axis C of rotor 911 toward the north pole of rotor 911. The d-axis rotates counterclockwise at angular frequency (angular velocity) ω of rotor 911. The q-axis is an axis orthogonal to the d-axis (an axis extending in a direction 90 degrees electrical angle ahead of the d-axis).

[0032] When sensorless control of the motor 910 is performed, it is difficult for the controller 20 to accurately determine the d-axis and q-axis of the rotor 911. Therefore, a γ-δ rotating coordinate system is used instead of a dq rotating coordinate system defined by the d-axis and q-axis. The γ-δ rotating coordinate system is defined by the γ-axis and δ-axis, which are estimated d-axis and q-axis. The γ-axis is an axis extending from the rotation axis C toward the estimated north pole of the rotor 911. The δ-axis is an axis orthogonal to the γ-axis (an axis extending in a direction 90 electrical degrees ahead of the γ-axis).

[0033] The γ-axis current and δ-axis current in the γ-δ rotating coordinate system are written as Iγ and Iδ, respectively. The γ-axis current command and δ-axis current command required to generate torque in the motor 910 are written as Iγ* and Iδ*, respectively. The γ-axis current Iγ is a current used to generate a magnetic field in the motor 910. The δ-axis current Iδ is a current corresponding to the torque of the motor 910. The controller 20 sets the δ-axis current command Iδ* to zero and sets the γ-axis current command Iγ* to a variable value, thereby preventing the motor 910 from generating torque and generating a magnetic field at a specified position.

[0034] Hereinafter, the angular difference between the γ axis and the d axis (γ-δ rotating coordinate system with respect to the dq rotating coordinate system) will be referred to as the “estimation error Δθ.” The d-axis self-inductance and q-axis self-inductance of the coil of stator 912 will be referred to as Ld and Lq, respectively.

[0035] <Function block> 4 is a functional block diagram of the controller 20 in the first embodiment. The controller 20 includes a speed control unit 201, a current control unit 202, a harmonic amplitude adjuster 203, a PWM calculator 204, a coordinate converter 205, a band stop filter (BSF) 206, a band pass filter (BPF) 207, an amplitude estimator 208, a position estimator 209, a low-pass filter (LPF) 210, a phase corrector 211, a coordinate converter 212, and a dead time corrector 213. The controller 20 also includes a subtractor 214, subtractors 215 and 216, a multiplier 217, an adder 218, adders 219, 220, and 221, multipliers 222 and 223, and adders 224 and 225. The PWM calculator 204 is an example of the "PWM signal generator" of the present disclosure.

[0036] The subtractor 214 calculates an angular velocity error by subtracting the signal Fmlpf from the velocity command Fm* input from the main controller 700 (FIG. 1) to the motor control device 100, for example. The signal Fmlpf is a signal obtained by removing noise from the current estimated velocity (angular velocity) Fm output from the position estimator 209 to the LPF 210 by the LPF 210. The subtractor 214 outputs the calculated angular velocity error to the velocity control unit 201.

[0037] The speed control unit 201 generates a torque command value such that the angular velocity error input from the subtractor 214 approaches 0, and generates a γ-axis current command Iγ* and a δ-axis current command Iδ* for generating the generated torque command value according to PI control. The speed control unit 201 outputs the γ-axis current command Iγ* and the δ-axis current command Iδ* to subtractors 215 and 216, respectively.

[0038] The subtractor 215 calculates a γ-axis current deviation ΔIγ (=Iγ*−Iγ), which is the deviation between the γ-axis current Iγ (hereinafter, sometimes referred to as the γ-axis primary current) from the coordinate converter 205, from which harmonic components have been removed by passing through the BSF 206, and the γ-axis current command Iγ* from the speed control unit 201, and outputs the γ-axis current deviation ΔIγ to the current control unit 202.

[0039] The subtractor 216 calculates a δ-axis current deviation ΔIδ (=Iδ*−Iδ), which is the deviation between the δ-axis current Iδ (hereinafter sometimes referred to as the δ-axis primary current) from the coordinate converter 205, from which harmonic components have been removed by passing through the BSF 206, and the δ-axis current command Iδ* from the speed control unit 201, and outputs the δ-axis current deviation ΔIδ to the current control unit 202.

[0040] The current control unit 202 performs a proportional-plus-integral (PI) calculation on the γ-axis current deviation ΔIγ from the subtractor 215, and outputs the calculation result as a γ-axis voltage command Vγ* to the adder 218. The current control unit 202 performs a proportional-plus-integral (PI) calculation on the δ-axis current deviation ΔIδ from the subtractor 216, and outputs the calculation result as a δ-axis voltage command Vδ* to the PWM calculator 204. Each of the γ-axis voltage command Vγ* and the δ-axis voltage command Vδ* is a "driving voltage command value" in the present disclosure.

[0041] The voltage of the power source 800 is input to the harmonic amplitude adjuster 203 as an input voltage Vdc from, for example, the monitoring unit 810 (FIG. 2). The harmonic amplitude adjuster 203 calculates (adjusts) the amplitude Vh of the harmonic voltage command value based on the input voltage Vdc. The harmonic voltage command value is used to generate a harmonic current used to estimate the rotor position of the motor 910.

[0042] The multiplier 217 multiplies the amplitude Vh input from the harmonic amplitude adjuster 203 by the harmonic angular frequency ω h and time t (ω h cosine value of t (cosω ht). The multiplied value calculated by the multiplier 217 is added to the γ-axis voltage command Vγ* by the adder 218. This makes it possible to suppress torque pulsation caused by adding the multiplied value (harmonic component) to the γ-axis voltage command that is unrelated to the torque of the motor 910. The added value (Vγ*+Vhcosω h t) is input to the PWM calculator 204. The multiplied value may be added to the δ-axis voltage command Vδ*.

[0043] The PWM calculator 204 converts the γ-axis voltage command Vγ* and the δ-axis voltage command Vδ* on the dq(γδ) two-phase coordinate into a U-phase voltage command, a V-phase voltage command, and a W-phase voltage command on the UVW three-phase coordinate in accordance with a known coordinate transformation formula (dq two-phase → UVW three-phase transformation formula (inverse Park transformation formula)) using the estimated position Hm of the rotor 911 input from the position estimator 209. The PWM calculator 204 further generates a switching signal SW from the voltage commands for the three phases. More specifically, the PWM calculator 204 generates a PWM signal as the switching signal SW based on a comparison between the voltage command for each phase and a predetermined carrier wave. The PWM calculator 204 outputs the generated switching signal SW to the adders 219 to 221.

[0044] The adder 219 adds the switching signal SW corresponding to the U phase and the correction value Uc of the pulse width (pulse width of the PWM signal) corresponding to the U phase output from the dead time corrector 213. The adder 220 adds the switching signal SW corresponding to the V phase and the correction value Vc of the pulse width corresponding to the V phase output from the dead time corrector 213. The adder 221 adds the switching signal SW corresponding to the W phase and the correction value Wc of the pulse width corresponding to the W phase output from the dead time corrector 213. The above correction values ​​will be described later.

[0045] Each of the adders 219 to 221 outputs the sum of the switching signal SW and the correction value to the power conversion unit 10 (inverter 13, FIG. 2).

[0046] The coordinate converter 205 calculates the γ-axis current Iγ and the δ-axis current Iδ based on the U-phase current Iu detected by the current sensor 913 and the V-phase current Iv detected by the current sensor 914. The coordinate converter 205 calculates the γ-axis current Iγ and the δ-axis current Iδ in accordance with a known coordinate transformation formula (UVW three-phase to dq two-phase transformation formula (Park transformation formula)) using the estimated position Hm of the rotor 911 input from the position estimator 209. The coordinate converter 205 outputs the γ-axis current Iγ and the δ-axis current Iδ to the BSF 206 and the BPF 207, respectively.

[0047] The BPF 207 removes sub-harmonic components from each of the input γ-axis current Iγ and δ-axis current Iδ. The BPF 207 extracts the γ-axis harmonic current iγ′ and the δ-axis harmonic current iδ′ from which the sub-harmonic components have been removed, and outputs the γ-axis harmonic current iγ′ and the δ-axis harmonic current iδ′ to the amplitude estimator 208.

[0048] Here, the γ-axis harmonic current iγ' and the δ-axis harmonic current iδ' are expressed by the following equations (1) and (2), respectively. Note that L0 is (Ld+Lq) / 2, and L1 is (Lq-Ld) / 2. Also, Δθ represents the difference (estimation error) between the estimated angle of the rotor 911 and the actual angle of the rotor 911.

[0049] iγ'=Vh(L0+L1cos2Δθ)sinω h t / ω h (L0 2 -L1 2 )···(1) iδ'=Vh(L1sin2Δθ)sinω h t / ω h (L0 2 -L1 2 )···(2) The amplitude estimator 208 estimates (calculates) the amplitude Aγ of the γ-axis harmonic current iγ′ from the input γ-axis harmonic current iγ′, and estimates (calculates) the amplitude Aδ of the δ-axis harmonic current iδ′ from the input δ-axis harmonic current iδ′. The amplitude estimator 208 outputs the amplitude Aγ to the multiplier 222. The amplitude estimator 208 outputs the amplitude Aδ to each of the multiplier 223 and the position estimator 209. The amplitude estimator 208 is configured by, for example, a synchronous detector.

[0050] The multiplier 222 multiplies the amplitude Aγ by sin(ω h t+ω h T del ) and calculate the corrected γ-axis harmonic current iγ'h. del is a delay time indicating a delay in the output of a PWM signal due to the generation of the PWM signal (PWM control). Since feedback processing is performed to generate a PWM signal based on the detected values ​​of the current sensors 913 and 914, a delay time occurs between the acquisition of the detected values ​​and the output of the PWM signal. The multiplication by the multiplier 222 is del and harmonic angular frequency ω h This means that the γ-axis harmonic current iγ' is corrected by the phase component corresponding to the product of

[0051] The multiplier 223 multiplies the amplitude Aδ by sin(ω h t+ω h T del ) to calculate the corrected δ-axis harmonic current iδ'h. The multiplication by the multiplier 223 is performed by multiplying the output delay T del and harmonic angular frequency ω h This means that the δ-axis harmonic current iδ' is corrected by the phase component corresponding to the product of

[0052] 5 is a vector diagram conceptually showing the correction in the multipliers 222 and 223. del ω corresponding to h T del As a result of the phase correction by this amount, the magnitude and phase of the resultant vector of the γ-axis harmonic current and the δ-axis harmonic current change.

[0053] 4 again, the adder 224 adds the corrected γ-axis harmonic current iγ'h output from the multiplier 222 and the γ-axis primary current output from the BSF 206. The adder 225 adds the corrected δ-axis harmonic current iδ'h output from the multiplier 223 and the δ-axis primary current output from the BSF 206. Each of the adders 224 and 225 outputs the sum to the phase corrector 211.

[0054] The phase corrector 211 calculates a composite current (dashed line in FIG. 6) obtained by combining the primary composite current (solid line in FIG. 6) and the corrected harmonic current (dashed line in FIG. 6) based on the sum output from the adders 224 and 225. The primary composite current is a composite current of the γ-axis primary current and the δ-axis primary current. The corrected harmonic current is a composite current of the corrected γ-axis harmonic current iγ'h and the corrected δ-axis harmonic current iδ'h.

[0055] The phase corrector 211 converts the resultant current (dashed line in FIG. 6) on the γ-δ axes (rotating Cartesian coordinates) into a resultant current on the α-β axes (fixed Cartesian coordinates) (see the left diagram in FIG. 7) according to the known inverse Park transformation formula, using the estimated position Hm output from the position estimator 209. This means that the resultant current on the γ-δ axes is mapped onto the α-β axes.

[0056] As shown in FIG. 7, the phase corrector 211 adjusts the phase of the composite current on the α-β axis based on the angular frequency ω (=Fmlfp) of the primary current and the output delay T del and the product ωT del In other words, the angle θ (angle difference between the α-axis and the γ-axis) indicating the estimated rotor position is corrected by θ + ωT del The current obtained by correcting the composite current is an example of the "corrected composite current" of the present disclosure.

[0057] Referring back to FIG. 4, the phase corrector 211 outputs the α-axis current and the β-axis current that make up the corrected composite current on the α-β axes to the coordinate converter 212.

[0058] The coordinate converter 212 calculates U-phase, V-phase, and W-phase currents according to the known inverse Clarke transformation using the α-axis current and β-axis current output from the phase corrector 211. The coordinate converter 212 outputs the calculated U-phase, V-phase, and W-phase currents to the dead time corrector 213.

[0059] The dead time corrector 213 corrects the dead time for the inverter 13 based on the signs (positive or negative) of the U-phase, V-phase, and W-phase currents output from the coordinate converter 212. The dead time corrector 213 determines correction values ​​Uc, Vc, and Wc of the pulse width of the PWM signal in correcting the dead time based on the positive or negative of the U-phase, V-phase, and W-phase currents, respectively.

[0060] As one example, when the calculated U-phase (or V-phase or W-phase) current is positive, the dead time corrector 213 sets the correction value Uc output to the adder 219 to a positive value in order to increase the pulse width of the PWM signal. As another example, when the V-phase (or U-phase or W-phase) current is negative, the dead time corrector 213 sets the correction value Vc output to the adder 220 to a negative value in order to decrease the pulse width of the PWM signal. The positive correction value and the negative correction value may have the same absolute value. Alternatively, the positive correction value and the negative correction value may be fixed values ​​stored in advance in the memory 22 (FIG. 2) or the like.

[0061] As mentioned above, the output delay T del By separately correcting the harmonic current and the primary current using the above, it is possible to perform dead time correction by appropriately taking the output delay into account. Therefore, it is possible to prevent the dead time correction from failing (becoming inappropriate) due to the output delay. As a result, it is possible to reduce torque pulsation due to estimated position ripple, and also to realize improved controllability by reducing the estimated position ripple.

[0062] [Second embodiment] A second embodiment of the present disclosure will be described with reference to Figures 8 to 10. In the second embodiment, a correction unit 300 is provided in a controller 120. Note that the same components as those in the first embodiment are denoted by the same reference numerals as those in the first embodiment, and repeated description will not be given.

[0063] 8 is a diagram showing the configuration of a controller 120 according to the second embodiment. The controller 120 differs from the controller 20 of the first embodiment in that a correction unit 300 is provided instead of the phase corrector 211, coordinate converter 212, dead time corrector 213, multipliers 222 and 223, and adders 224 and 225 of the first embodiment. The controller 120 also differs from the controller 20 of the first embodiment in that the γ-axis current iγ output from the coordinate converter 205 is output to the BSF 206 but is not output to the BPF 207.

[0064] The U-phase current Iu detected by the current sensor 913 and the V-phase current Iv detected by the current sensor 914 are output to the correction unit 300.

[0065] The correction unit 300 determines correction values ​​Uc, Vc, and Wc corresponding to the U phase, V phase, and W phase, respectively, based on the input U phase current Iu and V phase current Iv. The correction unit 300 outputs the correction values ​​Uc, Vc, and Wc to adders 219, 220, and 221, respectively.

[0066] 9 is a diagram showing a detailed configuration of the correction unit 300. The correction unit 300 includes a BPF 301, an amplitude estimator 302, an amplitude estimator 303, a dead time corrector 304, a subtractor 305, subtractors 306, 307, and 308, multipliers 309, 310, and 311, and multipliers 312, 313, and 314.

[0067] The subtractor 305 calculates a W-phase current Iw (Iw=-Iu-Iv) based on the U-phase current Iu and the V-phase current Iv, and outputs the calculated W-phase current Iw to the BPF 301. In addition to the W-phase current Iw, the BPF 301 also receives the U-phase current Iu and the V-phase current Iv as input.

[0068] The U-phase current Iu, the V-phase current Iv, and the W-phase current Iw are also input to subtractors 306, 307, and 308, respectively.

[0069] BPF 301 extracts harmonic current components from each of U-phase current Iu, V-phase current Iv, and W-phase current Iw, and outputs the extracted harmonic current components of each phase to amplitude estimator 302. The U-phase harmonic current components output from BPF 301 are input to subtractor 306. The V-phase harmonic current components output from BPF 301 are input to subtractor 307. The W-phase harmonic current components output from BPF 301 are input to subtractor 308.

[0070] Subtractor 306 calculates the U-phase primary current component by subtracting the U-phase harmonic current component from the U-phase current Iu, and outputs the calculated primary current component to amplitude estimator 303. Subtractor 307 calculates the V-phase primary current component by subtracting the V-phase harmonic current component from the V-phase current Iv, and outputs the calculated primary current component to amplitude estimator 303. Subtractor 308 calculates the W-phase primary current component by subtracting the W-phase harmonic current component from the W-phase current Iw, and outputs the calculated primary current component to amplitude estimator 303.

[0071] The amplitude estimator 302 estimates the amplitude of the input harmonic current component of each phase. The amplitude estimator 302 outputs the estimated amplitude of the U-phase harmonic current component to the multiplier 309. The amplitude estimator 302 outputs the estimated amplitude of the V-phase harmonic current component to the multiplier 310. The amplitude estimator 302 outputs the estimated amplitude of the W-phase harmonic current component to the multiplier 311. The amplitude estimator 302 is configured by, for example, a synchronous detector.

[0072] The multiplier 309 multiplies the amplitude of the input U-phase harmonic current component by sin(ω h t+ω h T del ) and multiplied to obtain the corrected U-phase harmonic current Iu h The multiplier 309 calculates the corrected U-phase harmonic current Iuh to the dead time corrector 304.

[0073] The multiplier 310 multiplies the amplitude of the input V-phase harmonic current component by sin(ω h t+ω h T del ) and multiply it by the corrected V-phase harmonic current Iv h The multiplier 310 calculates the corrected V-phase harmonic current Iv h to the dead time corrector 304.

[0074] The multiplier 311 multiplies the amplitude of the input W-phase harmonic current component by sin(ω h t+ω h T del ) and multiplied to obtain the corrected W-phase harmonic current Iw h The multiplier 311 calculates the calculated corrected W-phase harmonic current Iw h to the dead time corrector 304.

[0075] The amplitude estimator 303 estimates the amplitude of the input primary current component of each phase. The amplitude estimator 303 outputs the estimated amplitude of the U-phase primary current component to the multiplier 312. The amplitude estimator 303 outputs the estimated amplitude of the V-phase primary current component to the multiplier 313. The amplitude estimator 303 outputs the estimated amplitude of the W-phase primary current component to the multiplier 314. The amplitude estimator 303 is also configured by, for example, a synchronous detector.

[0076] The multiplier 312 multiplies the amplitude of the input U-phase primary current component by cos(θu+ω m T del ) and the corrected U-phase primary current Iu l The multiplier 312 calculates the corrected U-phase primary current Iu l to the dead time corrector 304. The multiplication by the multipliers 312 to 314 results in an output delay T del and the angular frequency of the primary current ω m This means that the phases of the primary currents of the U, V, and W phases are corrected by the phases corresponding to the product of θu and ω. mare expressed by the following equations (3) and (4), respectively.

[0077] θu=Hm+arctan(iδ / iγ)···(3) ω m =(Fmlpf / 60) × 2π × number of pole pairs (4) The multiplier 313 multiplies the amplitude of the input V-phase primary current component by cos(θv+ω m T del ) and the corrected V-phase primary current Iv l The multiplier 313 calculates the corrected V-phase primary current Iv l is output to the dead time corrector 304. Note that θv is expressed by the following equation (5).

[0078] θv=θu-2π / 3 (5) The multiplier 314 multiplies the amplitude of the input W-phase primary current component by cos(θw+ω m T del ) and the corrected W-phase primary current Iw l The multiplier 314 calculates the corrected W-phase primary current Iw l is output to the dead time corrector 304. Note that θw is expressed by the following equation (6).

[0079] θw=θu+2π / 3 (6) 10 is a diagram showing a detailed configuration of the dead time corrector 304. The dead time corrector 304 includes a sign determiner 315, a correction value determiner 316, and adders 317, 318, and 319.

[0080] The adder 317 calculates the corrected U-phase harmonic current Iu h and corrected U-phase primary current Iu l and outputs the calculated sum to the sign determiner 315. The adder 318 calculates the corrected V-phase harmonic current Iv h and corrected V-phase primary current Iv l The adder 319 calculates the sum of the corrected W-phase harmonic current Iw hand corrected W-phase primary current Iw l The sum of these is calculated and output to the sign determiner 315.

[0081] The sign determiner 315 determines the sign (positive or negative) of the input sum of each phase. The sign determiner 315 outputs a signal Su indicating information on the sign of the sum corresponding to the U phase to the correction value determiner 316. The sign determiner 315 outputs a signal Sv indicating information on the sign of the sum corresponding to the V phase to the correction value determiner 316. The sign determiner 315 outputs a signal Sw indicating information on the sign of the sum corresponding to the W phase to the correction value determiner 316.

[0082] The correction value determination unit 316 determines correction values ​​Uc, Vc, and Wc for the pulse width of the PWM signal based on the input signals Su, Sv, and Sw, respectively. Note that the control of the correction value determination unit 316 is the same as the control of the dead time corrector 213 in the first embodiment, and therefore a detailed description thereof will be omitted.

[0083] The other configurations and processes are the same as those in the first embodiment, and therefore will not be described repeatedly.

[0084] As described above, in the second embodiment, the dead time is corrected based on the sign of the sum of the primary current and the harmonic current in each of the three-phase AC currents. This allows the dead time to be corrected without requiring complex processing such as coordinate conversion.

[0085] <Modification> In the first embodiment, the phase of the composite current of the corrected harmonic current and the primary current is calculated based on the frequency of the primary current and the output delay T del However, the present disclosure is not limited to this. For example, the harmonic current after correction can be calculated based on the frequency of the primary current and the output delay T del The primary current is corrected based on the frequency and output delay T del The corrected value may be combined with the corrected value based on the above.

[0086] In the second embodiment, an example was described in which the dead time correction value was determined based on the sign of the sum of the corrected harmonic current and the corrected primary current, but the present disclosure is not limited to this. For example, the dead time correction value may be determined based on the magnitude relationship between the corrected harmonic current and the corrected primary current with the sign reversed (hereinafter referred to as the reverse current). For example, when the corrected primary current is negative and the corrected harmonic current is greater than the reverse current, it may be determined that the combined current of the corrected harmonic current and the corrected primary current will be positive. Also, when the corrected primary current is positive and the corrected harmonic current is smaller than the reverse current, it may be determined that the combined current will be negative. In other cases, it may be determined that the sign of the corrected primary current matches the sign of the combined current.

[0087] The configurations described in the above embodiment and the various modifications described above may be implemented in any combination.

[0088] The embodiments disclosed herein should be considered to be illustrative in all respects and not restrictive. The scope of the present disclosure is defined by the claims, not by the description of the above embodiments, and is intended to include all modifications within the meaning and scope of the claims. [Explanation of symbols]

[0089] 10 power conversion unit (power converter), 100 motor control device, 204 PWM calculator (PWM signal generator), 209 position estimator, 213, 304 dead time corrector (corrector), 900 electric compressor, 910 motor.

Claims

1. a PWM signal generating unit that generates a PWM signal to control the motor based on a sum obtained by adding a harmonic voltage command value for generating a harmonic current used to estimate a rotor position of the motor to a driving voltage command value for generating a primary current that drives the motor of the electric compressor; a position estimator that estimates the rotor position based on the harmonic current; a corrector that corrects a dead time for a power converter that outputs an AC voltage based on the PWM signal to the motor, When an output delay indicating a delay in output of the PWM signal due to generation of the PWM signal and a corrected primary current obtained by correcting the phase of the primary current based on the frequency of the primary current are defined as: The corrector is calculating a corrected harmonic current by correcting the phase of the harmonic current based on the output delay and the frequency of the harmonic current; A motor control device for an electric compressor that corrects the dead time based on the corrected harmonic current and the corrected primary current.

2. The corrector is calculating a corrected combined current by correcting the phase of a combined current obtained by combining the corrected harmonic current and the primary current on a γ-δ axis that is an estimated axis of the d-q axis, based on the output delay and the frequency of the primary current; The motor control device for an electric compressor according to claim 1 , wherein the dead time is corrected using the corrected combined current.

3. the motor is a three-phase AC motor, The corrector is calculating the corrected primary current for each phase and the corrected harmonic current for each phase; 2. The motor control device for an electric compressor according to claim 1, wherein the dead time is corrected for each of the phases based on a sum of the corrected harmonic current and the corrected primary current in each of the phases.

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