Power conversion device

Abstract

To provide a power conversion device capable of suppressing a temperature rise by reducing a current ripple of a smoothing capacitor by relatively simple control.SOLUTION: The power conversion device 1 generates a three phase AC output from the DC power supply 29 and applies the three phase AC output to the motor 8. The power converter includes a smoothing capacitor 32 connected in parallel to a DC power source 29, an invertor 28 for generating a three phase alternating current output by switching upper and lower arm switching elements 18A to 18F of each phase connected to the smoothing capacitor 32, and a control device 21 for controlling the upper and lower arm switching elements of the invertor based on a PWM pulse of each phase for each predetermined switching period. The control device 21 includes the resonance suppressor 35 that performs frequency component dispersion control for dispersing the frequency component of the DC link current Idc.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] The present invention relates to a power conversion device that generates a three-phase AC output from a DC power supply using an inverter connected to a smoothing capacitor and applies the output to a motor. [Background technology]

[0002] Conventionally, power conversion devices for driving motors built into electric compressors for vehicle air conditioning systems have consisted of a three-phase inverter using upper and lower arm switching elements for each of the UVW phases, and have controlled the switching elements of each phase using PWM (Pulse Width Modulation) to generate a voltage waveform (three-phase AC output) that is close to a sine wave from a DC power source (battery) and apply it to the motor to drive it.

[0003] In such electric compressors, a smoothing capacitor is connected in parallel to the DC power supply and placed near the inverter. This smoothing capacitor is used to smooth out the current with large distortion components caused by the inverter's PWM and return it to the DC power supply, and for this purpose, a capacitor with a relatively large capacity is required. However, as the capacitor's capacity increases, its volume also increases, which causes the electric compressor itself to become larger, so a smoothing capacitor with as small a capacity as possible is preferable.

[0004] Furthermore, there is wiring inductance between the DC power supply and the smoothing capacitor, or a normal mode choke coil is separately connected. These inductances and the smoothing capacitor described above form an input LC filter to reduce the effects of conducted noise from other devices connected to the DC power supply and conducted noise returning from the electric compressor to the DC power supply (see, for example, Patent Document 1).

[0005] Here, large film capacitors or electrolytic capacitors are used as smoothing capacitors. In the case of film capacitors, increasing the capacitance requires increasing the volume. Also, although electrolytic capacitors are smaller than film capacitors, they have poor current ripple resistance characteristics, so multiple electrolytic capacitors must be connected in parallel to disperse the current ripple, ultimately resulting in a larger capacitor. Furthermore, even in film capacitors, if the input ripple current is large, loss in the capacitor causes a temperature rise, leading to breakdown. Therefore, in order to prevent the smoothing capacitor from becoming larger, the ripple current must be suppressed. [Prior art documents] [Patent documents]

[0006] [Patent Document 1] Patent No. 7157412 Summary of the Invention [Problem to be solved by the invention]

[0007] In the smoothing capacitor described above, frequency components of the high-frequency current generated by the inverter that are higher than the resonant frequency Fin of the input LC filter flow into the smoothing capacitor, while frequency components lower than the resonant frequency Fin flow out to the DC power supply. Furthermore, the high-frequency current generated by the inverter does not include the resonant frequency Fin of the input LC filter. The reason for this is that if the resonant frequency Fin of the input LC filter is included, resonance will occur, and the current flowing through the smoothing capacitor (capacitor current) will increase, which may result in overvoltage or the smoothing capacitor may rise in temperature due to ripple current.

[0008] Therefore, for example, if the carrier frequency Fc (carrier wave frequency) of the inverter is 20 kHz, the resonant frequency Fin of the input LC filter will be designed to be 14 kHz or the like, which is a sufficiently large difference from the carrier frequency Fc.

[0009] Meanwhile, the frequency components of the current flowing between the smoothing capacitor and the inverter, i.e., the DC link current Idc, contain multiple sidebands whose frequencies vary with the motor's rotation speed (rotational frequency Fm) around the carrier frequency Fc, and when the motor's rotation speed increases, the sidebands reach the resonant frequency Fin of the input LC filter.When the sidebands reach the resonant frequency Fin, this causes a sudden increase in the ripple current in the smoothing capacitor, which can lead to a rise in the temperature of the smoothing capacitor.

[0010] In the aforementioned Patent Document 1, in order to suppress such a temperature rise in the smoothing capacitor, the switching timing of the inverter is adjusted, thereby slightly reducing the ripple of the current (capacitor current) flowing through the smoothing capacitor. However, the method of adjusting the switching timing required very precise calculations based on the magnitude of the current and the magnitude of the voltage command value.

[0011] In addition, the ON / OFF timing of the switching is changed at any time, making ON / OFF control difficult.Furthermore, although the temperature of the smoothing capacitor does not rise, the distortion of the current flowing through the motor increases, causing the motor temperature to rise and generating noise.

[0012] The present invention has been made to solve the above-mentioned conventional technical problems, and aims to provide a power conversion device that can reduce the current ripple in the smoothing capacitor and suppress temperature rise through relatively simple control. [Means for solving the problem]

[0013] A power conversion device of the present invention generates a three-phase AC output from a DC power supply and applies the output to a motor. The power conversion device includes: a smoothing capacitor connected in parallel to the DC power supply; an inverter that generates the three-phase AC output by switching upper and lower arm switching elements of each phase connected to the smoothing capacitor; and a control device that controls the upper and lower arm switching elements of the inverter based on PWM pulses for each phase at a predetermined switching period. The control device is characterized by having a resonance suppression unit that performs frequency component dispersion control to distribute the frequency components of a DC link current flowing between the smoothing capacitor and the inverter.

[0014] A power conversion device according to a second aspect of the present invention is characterized in that in the above-mentioned invention, the resonance suppression unit does not change the order of rising and falling edges of the PWM pulses of each phase in one switching period during frequency component distribution control.

[0015] The power conversion device of the third invention is characterized in that in the above invention, the resonance suppression unit shifts the position on the time axis without changing the pulse width of the phase whose pulse width is intermediate in one switching period during frequency component dispersion control.

[0016] The power conversion device of the fourth invention is characterized in that in the above invention, the resonance suppression unit changes the amount of shift of the position on the time axis of the phase with the intermediate pulse width depending on the motor rotation speed in frequency component distribution control.

[0017] The power conversion device of the fifth invention is characterized in that in the third invention, the resonance suppression unit changes the amount of shift of the position on the time axis of the phase with the intermediate pulse width in accordance with the motor torque in frequency component distribution control.

[0018] A power conversion device according to a sixth aspect of the present invention is characterized in that the resonance suppression unit executes frequency component distribution control based on whether or not the motor has reached a target rotation speed.

[0019] A power conversion device according to a seventh aspect of the present invention is characterized in that, in the above-mentioned aspects, the resonance suppression unit uses frequency component dispersion control to make the phase currents uneven between the first half and the second half of one switching cycle, thereby dispersing the frequency components of the DC link current and suppressing sidebands of the DC link current that become the resonance frequency of the LC filter including the smoothing capacitor. [Effects of the Invention]

[0020] A power conversion device of the present invention generates a three-phase AC output from a DC power supply and applies the output to a motor. The power conversion device includes a smoothing capacitor connected in parallel to the DC power supply, an inverter that generates a three-phase AC output by switching upper and lower arm switching elements of each phase connected to the smoothing capacitor, and a control device that controls the upper and lower arm switching elements of the inverter based on PWM pulses of each phase at a predetermined switching period. The control device has a resonance suppression unit that performs frequency component dispersion control to distribute the frequency components of a DC link current flowing between the smoothing capacitor and the inverter.

[0021] This resonance suppression unit disperses the frequency components of the DC link current by, for example, using frequency component dispersion control as in the seventh aspect of the invention, making the phase currents uneven between the first and second halves of one switching cycle and preventing concentration of the frequency components of the DC link current. When the frequency components of the DC link current are dispersed, the sidebands present around the carrier frequency become smaller. Therefore, even if the frequency of the sidebands changes depending on the motor rotation speed, the sidebands of the DC link current that are the resonant frequency of the LC filter including the smoothing capacitor can be suppressed and their magnitude can be reduced.

[0022] This suppresses resonance of the LC filter, reduces current ripple in the smoothing capacitor, and suppresses temperature rise in the smoothing capacitor, thereby effectively avoiding damage to the smoothing capacitor due to temperature rise.

[0023] In this case, if the resonance suppression unit does not change the order of rise and fall of the PWM pulses of each phase during one switching period in frequency component distributed control, as in the second aspect of the present invention, the control can be simplified, motor vibration noise and control noise can be reduced, and temperature rise can also be suppressed.

[0024] Furthermore, as in the third aspect of the present invention, if the resonance suppression unit shifts the position on the time axis in frequency component distribution control without changing the pulse width of the phase whose pulse width is intermediate in one switching period, resonance suppression can be achieved without varying the average output voltage of the motor.

[0025] Furthermore, as in the fourth aspect of the present invention, if the resonance suppression unit is configured to change the amount of shift on the time axis of the phase with the intermediate pulse width in accordance with the rotation speed of the motor in frequency component dispersion control, resonance can be suppressed in an appropriate manner in response to sideband waves whose frequency changes depending on the rotation speed (rotational frequency) of the motor, thereby achieving effective reduction in vibration noise and heat generation.

[0026] Furthermore, as the motor torque increases, the current flowing through the smoothing capacitor also increases. Therefore, if the resonance suppression unit, as in the fifth aspect of the invention, is configured to change the amount of shift on the time axis of the phase with the intermediate pulse width in frequency component dispersion control according to the motor torque, it becomes possible to more effectively reduce vibration noise and suppress heat generation.

[0027] Furthermore, if the resonance suppression unit performs frequency component distribution control based on whether the motor has reached the target rotation speed, as in the sixth aspect of the present invention, the motor can be stably controlled up to the target rotation speed without disturbances in the current or acceleration / deceleration rate. [Brief explanation of the drawings]

[0028] [Figure 1] 1 is an electric circuit diagram of a power conversion device according to an embodiment of the present invention (Embodiment 1). [Figure 2]3A and 3B are diagrams illustrating PWM count values ​​of each UVW phase and PWM pulses of each UVW phase for explaining the basic operation of motor control. [Figure 3] FIG. 3 is an enlarged view of a portion of a frame X in FIG. 2. [Figure 4] 2 is a diagram illustrating the resonance of an LC filter configured by the normal mode choke coil and smoothing capacitor in FIG. 1. FIG. [Figure 5] 2 is a diagram illustrating a DC link current flowing between the smoothing capacitor and the inverter in FIG. 1. FIG. [Figure 6] FIG. 2 is a diagram showing frequency components of a DC link current when the rotation frequency of the motor of FIG. 1 is 400 Hz. [Figure 7] This figure compares the frequency components of the DC link current when the motor rotation frequency is 200 Hz with those in FIG. 6. [Figure 8] 2 is a diagram illustrating an example of frequency component dispersion control executed by a resonance suppression unit of the control device of FIG. 1. FIG. [Figure 9] FIG. 1 is a diagram illustrating a DC link current in general motor control. [Figure 10] 2 is a diagram illustrating a DC link current in motor control of the power conversion device of FIG. 1. FIG. [Figure 11] 2 is a diagram showing a comparison of frequency components of a DC link current in general motor control and in motor control of the power conversion device of FIG. 1. FIG. [Figure 12] FIG. 10 is a block diagram illustrating another example of frequency component distribution control executed by the power conversion device of FIG. 1 (second embodiment). [Figure 13] 13 is a diagram showing a comparison of PWM count values ​​in the case of general motor control and the case of the power conversion device of FIG. 12. FIG. [Figure 14] FIG. 1 is a diagram showing the capacitor current of a smoothing capacitor and its FFT waveform in the case of general motor control. [Figure 15] 13 is a diagram showing a capacitor current of a smoothing capacitor and its FFT waveform in the case of the power conversion device of FIG. 12. FIG. DETAILED DESCRIPTION OF THE INVENTION

[0029] Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. [Example]

[0030] A power conversion device 1 according to an embodiment of the present invention drives a motor 8 of a so-called inverter-integrated electric compressor that constitutes a refrigerant circuit of a vehicle air conditioner mounted on a vehicle such as an electric automobile.

[0031] (1) Power conversion device 1 1, the power conversion device 1 of this embodiment includes a three-phase inverter 28 and a control device 21. The inverter 28 is a circuit that converts the DC voltage of a DC power source (vehicle battery: for example, 350 V) 29 into a three-phase AC voltage (three-phase AC output) and applies it to the motor 8. In this case, the motor 8 of this embodiment is an IPMSM (Interior Permanent Magnet Synchronous Motor).

[0032] The inverter 28 includes a U-phase half-bridge circuit 19U, a V-phase half-bridge circuit 19V, and a W-phase half-bridge circuit 19W. Each of the half-bridge circuits 19U to 19W for each phase includes upper-arm switching elements 18A to 18C and lower-arm switching elements 18D to 18F. Furthermore, each of the switching elements 18A to 18F includes a flywheel diode 31 connected in anti-parallel. In this embodiment, each of the upper and lower arm switching elements 18A to 18F is configured as an insulated gate bipolar transistor (IGBT) incorporating a MOS structure in the gate portion.

[0033] In this embodiment, a normal mode choke coil 30 is connected to the upper arm power supply line (positive bus) 10 of the DC power supply 29, and a smoothing capacitor 32 is connected between the upper arm power supply line 10 and the lower arm power supply line (negative bus) 15 of the DC power supply 29 in the downstream of this normal mode choke coil 30.

[0034] The normal mode choke coil 30 and smoothing capacitor 32 form an input LC filter. In this case, when the carrier frequency Fc is set to the aforementioned 20 kHz, the resonant frequency of the input LC filter formed by the normal mode choke coil 30 and smoothing capacitor 32 is set to the aforementioned 14 kHz. Note that the normal mode choke coil 30 does not necessarily have to be provided. In that case, the LC filter is formed by the wiring inductance of the upper arm power supply line 10 and the smoothing capacitor 32.

[0035] The collectors of the upper arm switching elements 18A to 18C of the inverter 28 are connected to an upper arm power supply line (positive bus) 10 in the rear stage of the smoothing capacitor 32. On the other hand, the emitters of the lower arm switching elements 18D to 18F of the inverter 28 are connected to a lower arm power supply line (negative bus) 15 in the rear stage of the smoothing capacitor 32.

[0036] In this case, the emitter of upper arm switching element 18A and the collector of lower arm switching element 18D of U-phase half-bridge circuit 19U are connected in series, the emitter of upper arm switching element 18B and the collector of lower arm switching element 18E of V-phase half-bridge circuit 19V are connected in series, and the emitter of upper arm switching element 18C and the collector of lower arm switching element 18F of W-phase half-bridge circuit 19W are connected in series.

[0037] The connection point (U-phase voltage Vu) between the upper arm switching element 18A and the lower arm switching element 18D of the U-phase half-bridge circuit 19U is connected to the U-phase armature coil of the motor 8, the connection point (V-phase voltage Vv) between the upper arm switching element 18B and the lower arm switching element 18E of the V-phase half-bridge circuit 19V is connected to the V-phase armature coil of the motor 8, and the connection point (W-phase voltage Vw) between the upper arm switching element 18C and the lower arm switching element 18F of the W-phase half-bridge circuit 19W is connected to the W-phase armature coil of the motor 8.

[0038] (2) Control device 21 Next, the control device 21 is composed of a microcomputer having a processor, and in this embodiment, a built-in vector control unit 25 derives a d-axis voltage command value Vd and a q-axis voltage command value Vq, and based on these, controls the ON / OFF state (switching) of each of the switching elements 18A to 18F of the inverter 28. Specifically, it controls the gate voltage applied to the gate of each of the switching elements 18A to 18F.

[0039] The control device 21 of the embodiment has a vector control unit 25, a phase voltage command calculation unit 33, a line-to-line modulation calculation unit 34, a PWM signal generation unit 36, a gate driver 37, and current sensors 26A, 26B, and 26C consisting of current transformers for measuring the motor currents (phase currents) of each phase flowing through the motor 8, namely, U-phase current Iu, V-phase current Iv, and W-phase current Iw.

[0040] In this embodiment, each of the current sensors 26A, 26B, and 26C is connected to the line modulation calculation unit 34. Furthermore, the current sensor 26A measures the U-phase current Iu, the current sensor 26B measures the V-phase current Iv, and the current sensor 26C measures the W-phase current Iw, but the U-phase current Iu may be measured by the current sensor 26A and the V-phase current Iv may be measured by the current sensor 26B, and the W-phase current Iw may be calculated from these.

[0041] Furthermore, in addition to measuring the motor current of each phase with current sensors 26A to 26C as in the embodiment, there is also a method in which the current value of lower arm power supply line 15 is detected by a shunt resistor and estimated by phase voltage command calculation unit 33 from the current value and the operating state of motor 8, so the method for detecting and estimating the current of each phase is not particularly limited.

[0042] (2-1) Vector control unit 25 12 in detail, the vector control unit 25 calculates a q-axis current command value Iqref by PI calculation from the difference between the speed command value ωrmref and the mechanical angular velocity ωrm of the motor 8. Then, the vector control unit 25 calculates a d-axis voltage command value Vd and a q-axis voltage command value Vq from the d-axis current command value Idref and the d-axis current Id, and the q-axis current command value Iqref and the q-axis current Iq, and outputs them to the control device 21.

[0043] Here, to rotate the motor 8 (IPMSM), it is necessary to generate a torque τ. This torque τ is calculated by the following formula (I). Note that Id and Iq are the d-axis current and q-axis current, and Ld and Lq are the d-axis inductance and q-axis inductance. kE is the power generation constant of the motor 8, and P is the number of pole pairs.

[0044]

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[0045] The d-axis and q-axis are two-dimensional axes, with the magnetic pole position of motor 8 (IPMSM) being the d-axis and the coordinate perpendicular to it being the q-axis. In this case, the output torque τ is proportional to the q-axis current Iq, so a simple way to control torque τ is to control the q-axis current Iq. Furthermore, the d-axis voltage command value Vd and q-axis voltage command value Vq for passing the d-axis current Id and q-axis current Iq through motor 8 can be calculated using the following formula (II):

[0046]

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[0047] Here, ωre is the electrical angle rotation speed of motor 8, p is the differential term, and R is the phase resistance value. If the d-axis current Id and q-axis current Iq to be applied to motor 8 and the electrical angle rotation speed ωre of motor 8 are determined from this formula (II) and the above-mentioned formula (I), the d-axis voltage command value Vd and q-axis voltage command value Vq to be applied to motor 8 can be determined.

[0048] As described above, the motor 8 (IPMSM) is driven by the half-bridge three-phase inverter 28. If the voltages applied by the inverter 28 to the three-phase motor 8 are the U-phase voltage Vu, V-phase voltage Vv, and W-phase voltage Vw, the d-axis voltage command value Vd, the q-axis voltage command value Vq, and the U, V, and W-phase voltages Vu, Vv, and Vw satisfy the relationship shown in the following formula (III).

[0049]

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[0050] (2-2) Phase voltage command calculation unit 33 The phase voltage command calculation unit 33 calculates the U-phase voltage command value Vu using the following formula (IV) based on the d-axis voltage command value Vd and the q-axis voltage command value Vq obtained from the vector control unit 25. * , V-phase voltage command value Vv * , W-phase voltage command value Vw * In other words, a dq axis-three phase transformation is performed. Note that Vm and θm in formula (IV) can be calculated from formula (V). Also, θ is the magnetic pole position based on the U phase, and θm is the voltage phase difference relative to the magnetic pole position.

[0051]

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[0052] (2-3) Line modulation calculation unit 34 The line modulation calculation unit 34 calculates and outputs the phase voltage command value Vu for each phase calculated and output by the phase voltage command calculation unit 33. * , Vv * , Vw * From this, the PWM count values ​​Nu1 (U-phase PWM count value), Nv1 (V-phase PWM count value), and Nw1 (W-phase PWM count value) of each phase normalized (corrected to 0 to 1) by the DC voltage Vdc are calculated using the following formula (VI). These PWM count values ​​Nu1, Nv1, and Nw1 are three-phase modulation command values ​​before line-to-line modulation.

[0053]

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[0054] Furthermore, the line modulation calculation unit 34 performs line modulation using the PWM count values ​​Nu1, Nv1, and Nw1 of the respective phases using formula (VII), where Nu, Nv, and Nw are the PWM count values ​​of the respective phases after line modulation (U-phase PWM count value Cu, V-phase PWM count value Cv, and W-phase PWM count value Cw).

[0055]

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[0056] In addition, Nmod in formula (VII) is a line-to-line modulation value for performing line-to-line modulation (two-phase modulation). When this line-to-line modulation value Nmod is zero (Nmod=0), the PWM count values ​​Nu, Nv, and Nw are not modulated, i.e., they are the same as the PWM count values ​​Nu1, Nv1, and Nw1 before line-to-line modulation. In this embodiment, Nmod=0 is set, and no line-to-line modulation is performed.

[0057] In the embodiment, the line modulation calculation unit 34 includes the resonance suppression unit 35 of the present invention, and the frequency component dispersion control executed by the resonance suppression unit 35 will be described in detail later.

[0058] (2-4) PWM signal generation section 36 The PWM signal generating unit 36 ​​receives the PWM count values ​​Nu, Nv, and Nw calculated by the line modulation calculating unit 34, and compares the magnitude with that of the carrier signal (carrier wave) cnt to generate and output PWM pulses (PWM signals) that serve as drive command signals for the U-phase half-bridge circuit 19U, the V-phase half-bridge circuit 19V, and the W-phase half-bridge circuit 19W of the inverter 28. That is, one cycle (carrier cycle) of the carrier signal cnt is one switching cycle Ts, and each of the switching elements 18A to 18F is controlled based on the PWM pulse of each phase for each switching cycle Ts.

[0059] Furthermore, based on the PWM count values ​​Nu, Nv, and Nw, the PWM signal generating unit 36 ​​generates a PWM pulse Su for switching the switching elements 18A and 18D of the U-phase half-bridge circuit 19U, a PWM pulse Sv for switching the switching elements 18B and 18E of the V-phase half-bridge circuit 19V, and a PWM pulse Sw for switching the switching elements 18C and 18F of the W-phase half-bridge circuit 19W.

[0060] When each PWM pulse Su, Sv, Sw rises to "1," the upper arm switching elements 18A to 18C of each phase turn ON, and when it falls to "0," the lower arm switching elements 18D to 18F turn ON (there is a dead time to prevent the upper and lower arms from being turned ON simultaneously). Here, the period during which each PWM pulse Su, Sv, Sw rises to "1" is defined as the pulse width of the PWM pulse.

[0061] (2-5) Gate driver 37 The gate driver 37 generates a gate voltage Sup of the switching element 18A of the U-phase half-bridge circuit 19U and a gate voltage Sun of the switching element 18D based on the PWM pulse Su output from the PWM signal generating unit 36, and similarly generates a gate voltage Svp of the switching element 18B of the V-phase half-bridge circuit 19V and a gate voltage Svn of the switching element 18E based on the PWM pulse Sv output from the PWM signal generating unit 36, and similarly generates a gate voltage Swp of the switching element 18C of the W-phase half-bridge circuit 19W and a gate voltage Swn of the switching element 18F based on the PWM pulse Sw output from the PWM signal generating unit 36.

[0062] Here, the gate voltages Sup and Sun of the U-phase half-bridge circuit 19U indicate the switching timing with a dead time provided to prevent the upper and lower poles from being turned on simultaneously, and by providing a certain dead time using the gate driver 37, the upper and lower poles are reliably prevented from being turned on simultaneously.

[0063] Similarly, the gate voltages Svp and Svn of the V-phase half-bridge circuit 19V and the gate voltages Swp and Swn of the W-phase half-bridge circuit 19W also provide dead times.

[0064] Each of the switching elements 18A to 18F of the inverter 28 is driven to turn on and off based on gate voltages Sup, Sun, Svp, Svn, Swp, and Swn output from a gate driver 37. That is, when the gate voltage is in the ON state (a predetermined voltage value), the switching element operates ON, and when the gate voltage is in the OFF state (zero), the switching element operates OFF. If the switching elements 18A to 18F are the IGBTs described above, the gate driver 37 is a circuit for applying a gate voltage to the IGBTs based on a PWM pulse, and is composed of a photocoupler, a logic IC, a transistor, etc.

[0065] The voltage at the connection point between the upper arm switching element 18A and the lower arm switching element 18D of the U-phase half-bridge circuit 19U is applied (output) to the U-phase armature coil of the motor 8 as a U-phase voltage Vu (phase voltage), the voltage at the connection point between the upper arm switching element 18B and the lower arm switching element 18E of the V-phase half-bridge circuit 19V is applied (output) to the V-phase armature coil of the motor 8 as a V-phase voltage Vv (phase voltage), and the voltage at the connection point between the upper arm switching element 18C and the lower arm switching element 18F of the W-phase half-bridge circuit 19W is applied (output) to the W-phase armature coil of the motor 8 as a W-phase voltage Vw (phase voltage).

[0066] An example of the basic operation of the PWM signal generation unit 36 ​​for controlling the motor 8 as described above is shown in Figures 2 and 3. Figure 2 shows the PWM count values ​​Nu, Nv, and Nw for each of the UVW phases and the PWM pulses Su, Sv, and Sw for each of the UVW phases, and Figure 3 is an enlarged view of the framed X portion of Figure 2. As mentioned above, one cycle (carrier cycle) of the carrier signal cnt in Figure 3 is one switching cycle Ts, and in this embodiment, the falling edge of the carrier signal cnt corresponds to the first half of one switching cycle Ts, and the rising edge of the carrier signal cnt corresponds to the second half.

[0067] During PWM switching, switching loss occurs in the switching elements 18A-18F. The loss in the switching elements 18A-18F affects the efficiency of the inverter 28 and also causes a temperature rise in the switching elements 18A-18F. Both efficiency and temperature rise are important factors in the power conversion device 1, and it is preferable to reduce switching loss. However, because electric compressors in vehicle air conditioners are required to be quiet, the PWM switching frequency, i.e., the frequency of the carrier signal cnt (carrier frequency Fc), is often set to a high value such as 20 kHz. This increases the number of switching operations and increases switching loss.

[0068] (3) High-frequency current generated by the inverter Next, we will explain the influence of the high-frequency current generated by the inverter using Figure 4. The dashed line in Figure 4 represents the current I L The solid line shows the current (capacitor current) Ic that flows through the smoothing capacitor, with the horizontal axis representing frequency and the vertical axis representing its magnitude. Figure 4 shows that at 10 kHz, both currents increase, and this amplified current is output to both the smoothing capacitor and the power supply. This frequency of 10 kHz is the resonant frequency Fin of the normal mode choke coil inductance L1 and the smoothing capacitor capacitance C1 in this example.

[0069] Therefore, if the current generated by the inverter contains frequency components of the resonant frequency Fin of the LC filter made up of the normal mode choke coil and smoothing capacitor, excessive noise will flow into the power supply, and the smoothing capacitor will be destroyed by overcurrent.In addition, current with frequency components lower than the resonant frequency Fin leaks to the power supply side, while higher frequency components flow into the smoothing capacitor and do not leak to the power supply side.

[0070] From the above, it can be said that it is desirable that the DC link current Idc flowing between the smoothing capacitor and the inverter does not include frequency components of the resonant frequency Fin. Therefore, in this embodiment, the carrier frequency Fc (carrier wave frequency) is set to 20 kHz, and the resonant frequency Fin of the input LC filter composed of the normal mode choke coil 30 and smoothing capacitor 32 is set to 14 kHz.

[0071] (4) DC link current Idc Next, the DC link current Idc of the power conversion device 1 of Fig. 1 will be described with reference to Fig. 5. The high-frequency current flowing through the smoothing capacitor 32 is determined by the characteristics of the DC link current Idc. The DC link current Idc changes depending on the magnitude of the current flowing through the motor 8 and the states of the switching elements 18A to 18F.

[0072] The currents flowing through the U-phase, V-phase, and W-phase are Iu, Iv, and Iw, respectively, and are assumed to be 2 A, -3 A, and 1 A. In State 1 of FIG. 5, all of the lower arm switching elements 18D to 18F are ON, so the current flowing through the motor 8 flows back through the lower arm switching elements 18D to 18F. As a result, the DC link current Idc becomes 0.

[0073] In State 2 of FIG. 5, upper arm switching element 18A of only the U phase is ON, and this current flows through the path of the DC link current Idc (indicated by the open arrow in the figure), so Idc = 2 A. At this time, 1 A of the W phase is circulated between the V phase and the W phase. In this state, the U-phase current Iu appears in Idc. In State 3 of FIG. 5, upper arm switching elements 18A and 18C of the U and W phases are ON, and these two currents flow through the path of the DC link current Idc (indicated by the open arrow in the figure), so Idc = 3 A. In this state, the negative value of the W-phase current Iw appears in Idc. In State 4 of FIG. 5, all upper arm switching elements 18A to 18C are ON, and all currents circulate through the motor 8.

[0074] In this way, the DC link current Idc changes successively depending on the switching states of the switching elements 18A to 18F, and the phase currents Iu, Iv, and Iw flowing through the UVW phases take on positive or negative values. This DC link current Idc is expressed by the following equation (VIII). Note that sign(x) is a sign function, which is 1 when x is positive and -1 when x is negative. In this case, x is also 1 when x is 0.

[0075]

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[0076] (4-1) High-frequency component of DC link current Idc Here, the current flowing through the motor 8 is an approximately sinusoidal wave and does not have high-frequency components, but the DC link current Idc is affected by PWM and therefore has high-frequency components, i.e., sidebands. Among the sidebands (high-frequency components) contained in the DC link current Idc, those with large amplitude spectra can be obtained by the following formula (IX). This formula (IX) is described in the following paper. Here, Fx in formula (IX) is the large-amplitude sideband frequency. Tomoyuki Ogawa, Shinji Wakao, Jat Taufiq, Keiichiro Kondo, and Nobuo Terauchi, "Theoretical Analysis of Sideband Harmonics of DC Current in Railway Vehicle Traction Inverters," IEEJ Transactions on Industrial Applications, Vol. 126, No. 7, pp. 1049-1057 (2006).

[0077]

number

[0078] (4-2) Sidebands of DC link current Idc That is, the DC link current Idc has sidebands with a rotational frequency Fm centered around the carrier frequency Fc, as shown in Fig. 6. Note that Fig. 6 shows the case where the carrier frequency Fc = 20 kHz and the rotational frequency Fm = 400 Hz. According to equation (IX), the large-amplitude sideband frequency Fx is 18,800 Hz when L = 1, 16,400 Hz when L = 4, and 14,000 Hz when L = 7. As a result, the resonant frequency Fin = 14 kHz of the input LC filter composed of the normal mode choke coil 30 and smoothing capacitor 32 overlaps with the large-amplitude sideband frequency Fx when L = 7, causing the excessive current amplitude mentioned above to flow into the smoothing capacitor 32.

[0079] Figure 7 compares the results for a rotational frequency Fm of 200 Hz with those for 400 Hz. When Fm = 200 Hz, the large-amplitude sideband frequency Fx is 19,400 Hz when L = 1, 18,200 Hz when L = 4, 17,000 Hz when L = 7, 15,800 Hz when L = 10, 14,600 Hz when L = 13, and 13,400 Hz when L = 16. Therefore, the large-amplitude sideband frequency Fx does not overlap with the resonant frequency Fin = 14 kHz. The large-amplitude sideband frequency Fx shown here matches the frequency of the spectrum shown in the upper part of Figure 7, confirming that there is no large-amplitude sideband frequency Fx at 14 kHz. On the other hand, when the rotation frequency Fm rises to 400 Hz, the large amplitude sideband frequency Fx is 14,000 Hz when L=7 as described above, and increases as shown by the X1 box at the bottom of Figure 7. Note that the X2 box in the figure represents the component of the large amplitude sideband frequency Fx (18,800 Hz) when L=1.

[0080] (5) Resonance suppressor 35 7, the amplitude spectrum of the large-amplitude sideband frequency Fx is largest when L=1, and as L increases, the amplitude spectrum rapidly decreases. On the other hand, when L=1, according to Equation (IX), even if the rotation frequency Fm increases, the large-amplitude sideband frequency Fx is unlikely to become lower than the carrier frequency Fc, making it easy to design the resonance frequency Fin. On the other hand, when L of the large-amplitude sideband frequency Fx increases, the large-amplitude sideband frequency Fx becomes lower than the carrier frequency Fc when the rotation frequency Fm is high. Therefore, although the amplitude spectrum of the large-amplitude sideband frequency Fx when L=7 is very small compared to L=1, it overlaps with the resonance frequency Fin, which is extremely dangerous. Therefore, in the power conversion device 1 of the present invention, the resonance suppression unit 35 performs frequency component dispersion control, which will be described below.

[0081] In this embodiment, the resonance suppression unit 35 executes frequency component dispersion control when the rotational speed of the motor 8 reaches a value at which the sideband wave of L=7 reaches the resonance frequency Fin (14 kHz), such as the rotational frequency Fm=400 Hz described above. Furthermore, when the target rotational speed (target rotational frequency Fm) of the motor 8 is, for example, 400 Hz, the resonance suppression unit 35 executes frequency component dispersion control based on whether the target rotational speed has been reached. Here, the target rotational speed refers to the rotational speed output by the control device 21 and includes rotational speeds that pass through transiently. Although this embodiment focuses on the sideband wave of L=7, as described in Equation (IX), frequency component dispersion control is executed when all frequencies at L=1, 4, 7, 10, 13, and 16 are near the resonance frequency Fin.

[0082] That is, when the large-amplitude sideband frequency Fx (in this embodiment, L=7) is near the resonance frequency Fin, the resonance suppression unit 35 executes frequency component dispersion control to adjust the switching control and disperse and reduce the sidebands (especially the large-amplitude sideband frequency Fx). The frequency component dispersion control executed by the resonance suppression unit 35 will be specifically described below with reference to FIG. 8.

[0083] (6) Frequency component dispersion control by the resonance suppression unit 35 In this embodiment, the resonance suppression unit 35 shifts the position on the time axis of the pulse of the phase in which the width of the PWM pulses Su, Sv, and Sw in one switching period Ts is intermediate in frequency component dispersion control. This will be explained with reference to Figure 8. Figure 8(a) shows the case of general motor control, and Figure 8(b) shows the case of the power conversion device 1 of the present invention.

[0084] In typical motor control, as shown in FIG. 8(a), the pulse widths of the positive and negative phase currents Iu, Iv, and Iw appearing in the DC link current Idc are equal in the first and second halves of a switching period Ts. This equal width results in a concentration of the frequencies of the pulse-width components, which increases the amplitude of the large-amplitude sideband frequency Fx with L=7. Therefore, the resonance suppression unit 35 shifts the position on the time axis of the phase whose pulse width is intermediate within one switching period Ts. In the example of FIG. 8(a), the U-phase PWM pulse Su has the widest width, the W-phase PWM pulse Sw has the narrowest width, and the V-phase PWM pulse Sv has the intermediate width. Therefore, the position of the V-phase PWM pulse Sv on the time axis is shifted as shown in FIG. 8(b).

[0085] In this case, the resonance suppression unit 35 reduces the V-phase PWM count value Nv in the first half of the switching period Ts and increases the V-phase PWM count value Nv in the second half of the switching period Ts, thereby shifting the position of the V-phase PWM pulse Sv on the time axis. In FIG. 8(b), the PWM pulse Sv is shifted to be earlier in time (to the right in the figure) than in FIG. 8(a). As a result, the pulse widths of the positive and negative phase currents Iu, Iv, and Iw appearing in the DC link current Idc become uneven between the first and second halves of the switching period Ts, and therefore the pulse width of the DC link current Idc becomes asymmetric between the first and second halves of the switching period Ts.

[0086] When the pulse width of the DC link current Idc becomes asymmetric between the first and second halves of the switching period Ts, the frequency components of the DC link current Idc are dispersed, and the sideband waves are also dispersed, the magnitude of each sideband wave becomes smaller, and the amplitude of the large-amplitude sideband wave frequency Fx (L=7) that reaches the resonant frequency Fin (14 kHz) also becomes smaller.

[0087] Fig. 9 shows the DC link current Idc in the case of conventional motor control, Fig. 10 shows the DC link current Idc when the above-mentioned frequency component distributed control is implemented, and Fig. 11 shows the frequency components of the DC link current Idc in the case of conventional motor control (solid line) and when the frequency component distributed control is implemented (dash-dotted line). As is clear from Fig. 11, the frequency component at 14 kHz of the resonant frequency Fin is smaller in the case of the dash-dotted line (frequency component distributed control) than in the case of the solid line (unilateral motor control).

[0088] In this way, the resonance suppression unit 35 of the present invention uses frequency component dispersion control to make the positive and negative phase currents Iu, Iv, and Iw appearing in the DC link current Idc uneven between the first and second halves of one switching cycle Ts, thereby avoiding concentration of frequency components in the DC link current Idc and dispersing them. Dispersing the frequency components of the DC link current Idc reduces the size of each sideband, including the large-amplitude sideband frequency Fx, which exists around the carrier frequency Fc. Therefore, even if the frequency of the large-amplitude sideband frequency Fx varies depending on the rotation speed of the motor 8, the large-amplitude sideband frequency Fx (L=7) of the DC link current Idc, which corresponds to the resonant frequency Fin of the LC filter formed by the smoothing capacitor 32 and the normal-mode choke coil 30, can be suppressed and its magnitude can be reduced.

[0089] This suppresses the resonance of the LC filter, reduces the current ripple of the smoothing capacitor 32, and suppresses the temperature rise of the smoothing capacitor 32, thereby effectively avoiding damage to the smoothing capacitor 32 due to the temperature rise.

[0090] Here, in the frequency component distribution control, the resonance suppression unit 35 does not change the order of rise and fall of the PWM pulses Su, Sv, and Sw of each phase during one switching period Ts, as shown in FIG. 8(b), so that the order remains the same as in FIG. 8(a). That is, the PWM pulse Su rises first, the PWM pulse Sv rises next, and the PWM pulse Sw rises last. The order of the PWM pulse Sw falling first, then the PWM pulse Sv falls, and finally the PWM pulse Su falls remains the same in both FIG. 8(a) and FIG. 8(b). This simplifies the control, reduces vibration noise and control noise of the motor 8, and also suppresses temperature rise.

[0091] 8(b), the resonance suppression unit 35 decreases the V-phase PWM count value Nv in the first half of a switching period Ts and increases it in the second half so as not to change the width (rising period) of the V-phase PWM pulse Sv, which has the intermediate pulse width within one switching period Ts. This makes it possible to achieve resonance suppression without varying the average output voltage to the motor 8, and also because the average value Idcave of the DC link current Idc is the same in FIGS. 8(a) and 8(b).

[0092] Furthermore, as described above, when the rotational frequency Fm of the motor 8 reaches, for example, 400 Hz, the large-amplitude sideband frequency Fx for L=7 reaches the resonance frequency Fin. Therefore, in the frequency component dispersion control, the resonance suppression unit 35 changes the shift amount of the phase position on the time axis where the pulse width is intermediate, depending on the rotational speed of the motor 8. That is, in this embodiment, the resonance suppression unit 35 does not execute frequency component dispersion control when the rotational frequency Fm is 200 Hz, but executes it when the rotational frequency Fm reaches 400 Hz. In this way, the shift amount may be increased depending on the rotational frequency Fm. For example, when Fm=666 Hz, the large-amplitude sideband frequency Fx for L=4 becomes 14,000 Hz. In this case, the amplitude spectrum is higher than when L=7, so the shift amount is increased. This allows resonance to be suppressed in accordance with the large-amplitude sideband frequency Fx, which changes depending on the rotational speed (rotational frequency Fm) of the motor 8, thereby effectively reducing vibration noise and heat generation.

[0093] Here, as the torque of the motor 8 increases, the amplitude of the phase currents Iu, Iv, and Iw of the motor 8 increases, and as a result, the positive or negative phase currents Iu, Iv, and Iw appearing in the DC link current Idc also increase. As a result, the current flowing through the smoothing capacitor 32 also increases. Therefore, in frequency component dispersion control, the resonance suppression unit 35 changes the amount of shift on the time axis of the V phase, where the pulse width is intermediate, in accordance with the torque of the motor 8. This more effectively reduces vibration noise and suppresses heat generation.

[0094] When frequency component distributed control is implemented, the average output current and average output voltage can be maintained without fluctuation, but the pulse width is shifted and the switching timing is changed, causing a slight deviation in the PWM ripple of the motor current. As a result, there is a risk of errors or delays in the convergence of the feedback control system until the motor 8 reaches the target rotation speed. Therefore, as described above, the resonance suppression unit 35 executes frequency component distributed control based on whether the motor 8 has reached the target rotation speed. This makes it possible to stably control the motor 8 up to the target rotation speed. [Example]

[0095] Next, other embodiments of the power conversion device 1 of the present invention will be described with reference to Figures 12 to 15. In each figure, the same reference numerals as in Figures 1 to 11 denote the same or similar functions.

[0096] In the above-described embodiment, the resonance suppression unit 35 that executes frequency component dispersion control is provided in the line modulation calculation unit 34, but in this embodiment, it is provided separately in the control unit 21 as a unit that adds a dither signal to the d-axis voltage command value Vd output from the vector control unit 25 to the control unit 21. In other words, the resonance suppression unit 35 in this embodiment is also included in the control unit 21. The rest is the same as in FIG. 1.

[0097] This dither signal is a sine wave, and its frequency is preferably about 1 / 10 of the carrier frequency Fc. That is, in frequency component distribution control, the resonance suppression unit 35 of this embodiment adds a dither signal to the d-axis voltage command value Vd input from the vector control unit 25 to the phase voltage command calculation unit 33. When the dither signal is added to the d-axis voltage command value Vd, the U-phase PWM count value Nu oscillates as shown by the dashed line in Fig. 13. Note that the thick solid line in Fig. 13 shows the case where no dither signal is added.

[0098] As a result, in this embodiment, the PWM pulse width is adjusted across multiple switching periods Ts (10 switching periods in this embodiment), the frequency components of the DC link current Idc are dispersed, and the sidebands are also dispersed as described above, reducing the magnitude of each sideband and the large-amplitude sideband frequency (L=7) that reaches the resonant frequency Fin (14 kHz).In this embodiment, the value of the DC link current Idc over 10 switching periods is equal to that in the case where frequency component dispersion control is not performed.

[0099] The upper part of Fig. 14 shows the capacitor current Ic in general motor control, and the lower part shows its FFT waveform. The upper part of Fig. 15 shows the capacitor current Ic in motor control of the power conversion device in Fig. 12, and the lower part shows its FFT waveform. Comparing the two figures clearly shows that the frequency component at 14 kHz of the resonant frequency Fin is smaller in the lower part of Fig. 15 (frequency component distributed control) than in the lower part of Fig. 14.

[0100] As a result, this embodiment also suppresses the resonance of the LC filter, reducing the current ripple of the smoothing capacitor 32 and suppressing the temperature rise of the smoothing capacitor 32, thereby effectively avoiding damage to the smoothing capacitor 32 due to the temperature rise.

[0101] In the embodiment, the present invention is applied to the power conversion device 1 that controls the drive of the motor 8 of the electric compressor, but the present invention is not limited to this and is effective when various devices are controlled by an inverter. [Explanation of symbols]

[0102] 1 Power conversion device 8 motors 18A~18F Upper and lower arm switching elements 19U U-phase half-bridge circuit 19V V-phase half-bridge circuit 19W W-phase half-bridge circuit 21 Control device 25 Vector control unit 28 Inverter 29 DC power supply 30 Normal mode choke coil 32 smoothing capacitor 33 Phase voltage command calculation unit 34 Line modulation calculation section 35 Resonance suppressor 36 PWM signal generation section 37 Gate Driver

Claims

1. A power conversion device that generates a three-phase AC output from a DC power supply and applies the output to a motor, a smoothing capacitor connected in parallel to the DC power supply; an inverter that generates the three-phase AC output by switching upper and lower arm switching elements of each phase connected to the smoothing capacitor; a control device that controls upper and lower arm switching elements of the inverter based on PWM pulses of each phase for each predetermined switching period; The control device is a power conversion device characterized in that it has a resonance suppression unit that executes frequency component dispersion control to distribute frequency components of a DC link current flowing between the smoothing capacitor and the inverter.

2. The resonance suppression unit is 2. The power conversion device according to claim 1, wherein the order of rising and falling edges of the PWM pulses of each phase in one switching period is not changed in the frequency component distribution control.

3. The resonance suppression unit is 3. The power conversion device according to claim 2, wherein in the frequency component dispersion control, the position on the time axis is shifted without changing the pulse width of a phase whose pulse width in one switching period is intermediate.

4. The resonance suppression unit is 4. The power conversion device according to claim 3, wherein the frequency component dispersion control changes the shift amount of the position on the time axis of the phase whose pulse width is intermediate, depending on the rotation speed of the motor.

5. The resonance suppression unit is 4. The power conversion device according to claim 3, wherein the frequency component dispersion control changes the amount of shift of the position on the time axis of the phase whose pulse width is intermediate, depending on the torque of the motor.

6. The power conversion device according to claim 1 , wherein the resonance suppression unit executes the frequency component dispersion control based on whether or not the motor has reached a target rotation speed.

7. 7. The power conversion device according to claim 1, wherein the resonance suppression unit disperses the frequency components of the DC link current by making the phase currents uneven between the first half and the second half of one switching cycle through the frequency component dispersion control, thereby suppressing sidebands of the DC link current that correspond to a resonance frequency of an LC filter including the smoothing capacitor.

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