Frequency-stabilized microwave source using an IQ mixer to detect amplitude modulation of the reflected signal

Abstract

An IQ mixer is used in a Pound-stabilized microwave source to detect amplitude modulation of the signal reflected from the reference resonator. By properly configuring the IQ mixer so that the LO and RF inputs are maintained in quadrature at the Q mixer and therefore in phase at the I mixer, lower levels of amplitude modulation can be detected at lower modulation frequencies consistent with optimal selection of resonator coupling and maximum phase-to-amplitude conversion.

Description

[Technical Field]

[0001] Priority claims This application claims the benefit of priority to U.S. Patent Application No. 18 / 095,840, filed January 11, 2023, which is incorporated herein by reference in its entirety.

[0002] The present disclosure relates to frequency stabilized microwave sources using "pound stabilization" or "pound servo" techniques, and more particularly to the use of an IQ mixer to detect amplitude modulation of a reflected signal. [Background technology]

[0003] Stable microwave frequency sources are used in applications such as radar and communications systems. The most stable microwave sources, such as hydrogen masers or cryogenic sapphire oscillators, are utilized as time bases for precision clocks. Such sources and clocks are bulky and unsuitable for portable devices that may require synchronization even when separated by large distances. Therefore, many such devices synchronize by utilizing time information provided by GPS signals. However, GPS reception may be unavailable for periods of time, requiring small, low-power oscillators with sufficient stability to maintain synchronization during such periods.

[0004] A simple oscillator may include a loop containing at least a resonator, an amplifier, and a phase shifter, which, when combined, may satisfy the oscillation conditions of a gain and phase shift around the loop that are greater than 1 and a multiple of 2π, respectively. A resonator with a highly stable center frequency will not immediately result in a highly stable oscillation frequency unless the phase shift through the amplifier and phase shifter is also highly stable.

[0005] RVPound, "Frequency Stabilization of Microwave Oscillators" (Proceedings of the IRE, vol. 35, no. 12, pp. 1405-1415, December 1947), describes a technique that utilizes the reflection of a microwave oscillator signal from a reference resonator to improve oscillator stability. In U.S. Pat. No. 2,681,998, Pound combined this technique with modulation of the oscillator signal to avoid limitations induced by microwave detector flicker noise. Microwave oscillator stabilization techniques that involve the reflection of a modulated signal from a resonator have become commonly referred to as "Pound stabilization" or "Pound servo" techniques. More recently, a single resonator is often utilized both as a component of a microwave oscillator loop and as a stabilizing element used to improve oscillator loop stability. See also the following references: N. Luiten, AG Mann, NJ McDonald and DGBlair, “Latest results of the UWAcryogenic sapphire oscillator”, Proceedings of the 1995 IEEE International Frequency Control Symposium (49th Annual Symposium), 1995, pp. 433-437; "Resonator-oscillator with exceptional stability: an update", Proceedings of IEEE 48th Annual Symposium on Frequency Control, 1994, pp. 441-446.

[0006] Referring now to FIG. 1, a frequency-stabilized microwave source 100 uses pound stabilization techniques to stabilize a microwave oscillator, i.e., a voltage-controlled oscillator (VCO) 102, to a reference resonator 104 to generate a frequency-stabilized output signal 106 at an output port 108. The VCO generates an output signal 110, a portion of which is taken as the frequency-stabilized output signal 106. The signal frequency can be tuned by a pound servo voltage 112 applied to a tuning port 114 of the VCO. The output signal 110 passes through a phase shifter 116 and a circulator 118 before hitting the reference resonator 104. The frequency of the VCO 102 is adjusted to the center frequency (f Res ) Typically the resonator will be a cavity resonator which may include a dielectric element, but the technique can be applied using any resonator with suitable center frequency and stability.

[0007] A portion of the output signal 110 that strikes the resonator is reflected according to known resonator characteristics as a reflected signal 120. This reflected energy is directed by a circulator 118 onto an amplitude detector 122, most commonly a simple diode detector. A modulation source 124 generates a modulation frequency f to apply phase modulation to the output signal 110 that strikes the resonator via a phase shifter 116. m As explained in more detail below, the phase modulation can result in an amplitude modulation of the reflected signal 120 seen by the amplitude detector 122. Such amplitude modulation produces a fluctuating signal at the detector output, the fluctuations occurring at the same modulation frequency f m It has.

[0008] A synchronous detector 126, such as a synchronous rectifier, mixes the output of amplitude detector 122 with a modulating signal 125 to produce a modulating frequency f mThe amplitude detector converts fluctuations in the amplitude detector output at VCO 102 into a DC frequency error voltage 128 that represents the magnitude of the fluctuations. This DC frequency error voltage is then amplified and preferably integrated by integrator 130 to provide a pound servo gain to generate a pound servo voltage 112 that is supplied to tuning port 114 of VCO 102. A synchronous rectifier may also be known in the art by terms such as a synchronous detector, lock-in detector, lock-in amplifier, or lock-in mixer.

[0009] To understand the operation of a frequency-stabilized microwave source, consider the incident spectrum 200 of a signal impinging on a reference resonator, as shown in FIG. 2A, and the variation of the complex reflection coefficient 202 of the reference resonator with frequency as separate amplitude and phase plots 204 and 206, respectively, as shown in FIG. 2B. For purposes of explanation, consider the VCO frequency (f c ) is intentionally shifted from the center frequency of the resonator (f c ≠f Res The first pair of phase modulation sidebands 210 and 212 are at frequencies f U =f c +f m and f L =f c -f m and are located symmetrically around the carrier wave.

[0010] The upper and lower sidebands 210 and 212, respectively, are at the center frequency f of the resonator. Res Since the reference resonator is not symmetrically positioned around f, it reflects off the reference resonator with different amplitudes and phases. As a result, the reflected carrier is modulated in both phase and amplitude. c =f Res Only in the special case of , both the upper and lower sidebands 210 and 212 are reflected equally, and the frequency f m As will become apparent, the reflected carrier signal has a frequency f m The degree of amplitude modulation at c -f Res Shows.

[0011] To understand the conversion from phase modulation to amplitude modulation, consider phasor diagram 300 and phasor diagram 302 shown in Figures 3A-3E. Figures 3A-3C provide a simplified representation of the phase modulation shown in phasor diagram 300 of a phase-modulated carrier 304 as it strikes a reference resonator. The phase modulation is expressed as ±2πf at the edges of the carrier phasor. m The sideband phasors 306 and 308 are represented by two counter-rotating phasors 306 and 308 at a rate of radians per second. When the sideband phasors 306 and 308 are aligned with the carrier 304, as shown in Figure 3A, the sum of the phasors is zero. However, when the sideband phasors 306 and 308 are not aligned with the carrier 304, as shown in Figures 3B and 3C, the incident lower sideband 310 and upper sideband 312 are added to the carrier 304, advancing and retarding the carrier phase, respectively. The result is a modulation frequency f without amplitude modulation. m This is a periodic variation of the carrier phase of a signal. Careful examination of Figures 3A-3C shows that higher-order sidebands are necessary to accurately describe pure phase modulation. This is because the length (amplitude) of the carrier phasor changes slightly in Figures 3B and 3C compared to Figure 3A. However, the simplification of ignoring higher-order sidebands is sufficient to explain the process.

[0012] If the incident signal is a carrier 304 plus two equal amplitude sidebands 306 and 308 that go to zero, as shown in FIG. 3A, the reflected sidebands 316 and 318 will have different amplitudes and different phases, as shown in FIG. 3D.

[0013] Consider a reflected signal 320 in which the sidebands are modified in both amplitude 322 and phase 324 according to the frequency dependence of the resonator reflection coefficient. As shown in Figure 3E (not to scale), the reflected upper sideband 326 is reflected with a much larger amplitude than the reflected lower sideband 328. Now, the sidebands can no longer cancel in either direction, and their sum follows an approximately elliptical locus 330. Thus, both the length and phase angle of the reflected carrier change significantly, causing f c ≠f ResWhenever the reflected carrier frequency f m In , both amplitude and phase modulation is present.

[0014] Furthermore, when the frequency error has the opposite sign, the relative changes of the reflected sidebands 326 and 328 are swapped, with the lower sideband 328 dominating, thus reversing the total movement around the elliptical locus 330. Thus, the phase of the amplitude modulated signal seen at the detector output is reversed relative to the original modulated signal, and the sign of the DC signal recovered by the synchronous detector is also reversed, resulting in a frequency error f c -f Res When is less than the bandwidth, the synchronous detector output is f C -f Res is proportional to.

[0015]

number

number

number

number

[0016] It should be understood that the width of the dip in the amplitude reflection coefficient 204, or alternatively the peak in the phase-to-amplitude conversion 214, depends on the coupling coefficient of the resonator, not the transmission bandwidth of the resonance. When the coupling of the resonator is near critical, the conversion of phase modulation to amplitude modulation is maximized, and f m The optimum range for f can be much smaller than the bandwidth of the resonator. For example, Res For a critically coupled resonator with a transmission bandwidth of 10 GHz and 400 kHz, the peak width of the phase-to-amplitude conversion, measured by a 3 dB change from the maximum value, is less than 1 kHz.

[0017] Another embodiment of a frequency stabilized microwave source 400 is shown in Figure 4 as an improvement over the microwave source 100 shown in Figure 1, with like numbers used to identify like elements. The phase shift 116 is removed and the modulation source 124 is added to the pound servo voltage 112 via summing node 402 and input to the VCO tuning port 114, resulting in frequency modulation of the VCO 102. The frequency modulation and phase modulation are simply

number

[0018] Referring now to FIG. 5, a frequency stabilized microwave source 500 includes a reference resonator 502 also having a resonant frequency f Res , and a loop amplifier 510, thereby eliminating the need for a separate VCO. Figure 5, like Figure 4, illustrates that frequency modulation achieves phase modulation of the signal incident on the reference resonator.

[0019] A phase change in any of the oscillator loops 504 results in a frequency change. Therefore, frequency can be modulated by providing a modulation signal 516 from a modulation source 518 to any of the phase shifters in the oscillator loops 504. The pounded servo feedback and modulation signal can be applied to a common phase shifter or separate phase shifters. If a common phase shifter can provide both the frequency modulation and tuning capabilities required by the pounded servo feedback, it can be used. The phase shift range over which the precision required for modulation is achieved may be insufficient to achieve the tuning required for the pounded servo feedback than a separate phase shifter.

[0020] If a common phase shifter is used, the pound servo voltage 520 and the modulation signal 516 are summed by a summing node 522 and input to a common modulation phase shifter, either a loop phase shifter 508 placed before the loop amplifier 510, or a modulation phase shifter 524 placed after the loop amplifier 510. In this configuration, the other phase shifter is eliminated. It may be preferable to place the common phase shifter before the loop amplifier 510 to utilize the full amplifier output power.

[0021] If separate phase shifters are used, the pound servo voltage 520 is input to the loop phase shifter 508, and the modulation signal 516 is input to the modulation phase shifter 524. In this configuration, the summing node 522 is eliminated. The position of the phase shifter within the loop may be inverted and is determined by practical considerations such as whether the phase shifter can handle the amplified output power.

[0022] The phase modulation induced by the frequency modulation of signal 512 can result in an amplitude modulation of the reflected signal 530 that is directed by circulator 532 to amplitude detector 534. Such amplitude modulation gives rise to a fluctuating signal at the detector output. The fluctuating signal has the same modulation frequency f m The synchronous detector 536 detects the modulation frequency f mThe amplitude detector converts the variation in the output of the amplitude detector at 508 into a DC frequency error voltage 538 that represents the magnitude of the variation. This DC voltage is then amplified and preferably integrated 540 to produce a pound servo voltage 520 that is either fed to a summing node 522 or fed directly to the loop phase shifter 508.

[0023] 5, the frequency stable output signal 542 is taken from a different output 544 positioned after the reference resonator 502 of the oscillator loop 504, which typically results in reduced noise in the oscillator output due to the filtering action of the reference resonator 502. However, the output may equally well be taken before the reference resonator, as in FIGS. 1 and 4, without otherwise affecting the functionality of the microwave source. Summary of the Invention

[0024] The following is a summary to provide a basic understanding of some aspects of the disclosure. This summary is not intended to identify key or critical elements of the disclosure or to delineate the scope of the disclosure. Its sole purpose is to present some concepts of the disclosure in a simplified form as a prelude to the more detailed description and defined claims that are presented later.

[0025] This disclosure describes the use of an IQ mixer in a pound servo loop to detect amplitude modulation of a signal reflected from a reference resonator. By properly configuring the IQ mixer so that the LO and RF inputs are maintained in quadrature at the Q mixer and therefore in phase at the I mixer, lower levels of amplitude modulation can be detected at lower modulation frequencies consistent with optimal selection of resonator coupling and maximum phase-to-amplitude conversion.

[0026] In frequency-stabilized microwave sources that use pound stabilization or pound servo techniques to stabilize the oscillation frequency, the amplitude detector is implemented as an IQ mixer, where the signal reflected from the reference resonator provides the RF input and the I output is a modulation frequency f that represents the amplitude modulation (AM) of the reflected carrier. m The demodulated signal is applied to a synchronous detector. A phase error voltage indicative of the phase difference between the incident and reflected carriers provided at the Q output is integrated and fed back to the LO phase shifter, where it is combined with a portion of the signal impinging on the resonator to produce a signal at the LO input, thereby maintaining the LO and RF inputs in quadrature at the Q mixer and therefore in phase at the I mixer. As a result, the synchronous detector detects the frequency error f c -f Res This generates a frequency error voltage proportional to , which in turn reduces the frequency error of the oscillator loop towards zero.

[0027] An IQ mixer configured in this manner can be used as an AM detector for any frequency-stabilized microwave source that utilizes Pound stabilization or Pound servo techniques. For example, these techniques may utilize a reference resonator as a frequency-determining element in an oscillator loop and modulate the phase of a signal incident on the resonator by applying a modulation signal to a phase shifter at any point in the oscillator loop. One or more phase shifters may be provided in the oscillator loop, thereby allowing the modulation signal to be combined with or separated from the Pound servo feedback signal. Furthermore, these techniques may include configurations in which the reference resonator is separate from the VCO that generates the microwave signal, and phase modulation is achieved by a phase shifter between the VCO output and the reference resonator, or by combining the modulation signal with the Pound servo feedback signal applied to tune the VCO.

[0028] In one embodiment, a quadrature compensation voltage may be added to the phase error voltage of the Q output at the input to the integrator to compensate for non-ideal characteristics of the quadrature phase shift between the LO signals at the individual I and Q mixers.

[0029] These and other features and advantages of the present disclosure will become apparent to those skilled in the art from the following detailed description of the preferred embodiments, taken in conjunction with the accompanying drawings. [Brief explanation of the drawings]

[0030] [Figure 1] As explained above, Pound stabilization is a known configuration of frequency stabilised microwave source that uses Pound stabilisation to stabilise the oscillator against an independent reference resonator. [Figure 2] Figure 2A is a plot of the incident spectrum as described above, and Figure 2B is a plot of the reflected spectrum from the reference cavity as described above. [Figure 3] 3A-3E are phasor diagrams illustrating the conversion from phase modulation to amplitude modulation as described above. [Figure 4] As explained above, Pound stabilization is another known configuration of a frequency stabilised microwave source, which uses Pound stabilisation to stabilise the oscillator against an independent reference resonator. [Figure 5] As explained above, it is a known configuration of a frequency stabilised microwave source in which ground stabilisation is performed by a reference resonator which also forms part of the oscillator loop. [Figure 6] 1 is a plot of the noise floor of a diode and a mixer as an AM detector. [Figure 7] 1 is a schematic diagram and symbolic representation of an IQ mixer. [Figure 8] FIG. 1 is a schematic diagram of a core of a Pound-stabilized microwave source including an IQ mixer configured for amplitude modulation detection. [Figure 9] 1 is an alternative configuration of the IQ mixer to compensate for non-ideal characteristics of the individual I and Q mixers. [Figure 10] 1 is a first embodiment of a Pound-stabilized frequency-stabilized microwave source in which an IQ mixer is configured to perform AM detection. [Figure 11]1 is a second embodiment of a Pound-stabilized frequency-stabilized microwave source in which an IQ mixer is configured to perform AM detection. [Figure 12] 1 is a first embodiment of a Pound-stabilized frequency-stabilized microwave source in which an IQ mixer is configured to perform AM detection. DETAILED DESCRIPTION OF THE INVENTION

[0031] This disclosure describes the use of an IQ mixer, instead of a simple diode, in a pound servo loop to detect amplitude modulation of a signal reflected from a reference resonator. By properly configuring the IQ mixer so that the LO and RF inputs are maintained in quadrature at the Q mixer, lower levels of amplitude modulation can be detected at lower modulation frequencies consistent with optimal selection of resonator coupling and maximum phase-to-amplitude conversion.

[0032] In frequency-stabilized microwave sources using Pound stabilization or Pound servo techniques, diode detectors offer a very simple and low-cost way to recover amplitude modulation in the reference resonator's reflected signal. However, signal-to-noise ratio (SNR) must be considered when selecting a detector. Diodes are nonlinear devices, and their conversion ratio, i.e., the signal voltage generated for a given change in input amplitude (sometimes written in mV / dB), depends on the input carrier amplitude and the load resistance seen by the diode, as shown in Figure 1 of Serge Grop and Enrico Rubiola, "Flicker Noise of Microwave Power Detectors," 2009 IEEE International Frequency Control Symposium Joint with the 22nd European Frequency and Time forum, 2009, pp. 40-43. Diode detectors also generate noise that limits the minimum detectable signal. This noise has both a flicker or 1 / f component that depends on the incident carrier power, and a white floor, often set by the load resistance.

[0033] Inherent within the synchronous detector is a low-pass filter (LPF) element, which allows the synchronous detector to filter the modulation frequency f m Therefore, the advantage of the Pound stabilization scheme is that it is only sensitive to frequencies close to f m Only noise power within the bandwidth of the synchronous detector close to f contributes to the noise of the diode detector output. Therefore, for a diode detector with noise 600 as shown by the dashed line in FIG. 6, f greater than about 10 kHz is required to achieve the lowest detector noise. m It would be appropriate to choose

[0034] When selecting a diode detector, the designer must also consider the amount of power present in the diode detector due to the power incident on the resonator and the resonator reflection coefficient due to the resonator coupling coefficient β. If very low power is reflected back into the diode, as occurs when the resonator is critically coupled, the diode's conversion rate will be low, and if a high load resistor is selected to improve the conversion rate, the resistor's thermal noise will be high. Alternatively, if the resonator coupling is set to a non-optimal value, resulting in higher power hitting the diode, the conversion rate may be higher and a lower load resistor may be used. However, as the power incident on the diode increases, flicker, or 1 / f noise, increases, so the modulation frequency f may be increased to improve the signal-to-noise ratio. m It may be necessary to increase f. Therefore, the use of a diode detector is less than optimal, as determined by maximizing the phase-to-amplitude conversion, and, as mentioned above, is close to the critical coupling in the resonator, and f m is determined to be small.

[0035] A mixer can be used as a phase detector at microwave frequencies by arranging the input LO and RF signals so that they are in quadrature (90° relative phase). See, for example, Stephan R. Kurtz, "Mixers as Phase Detectors," Tech-note, Watkins-Johnson Company, Vol. 5, No. 1, January / February 1978. Conversely, when the input LO and RF signals are in phase (relative phase is 0°), the mixer responds to amplitude modulation of the RF signal. At intermediate phase angles between the LO and RF, the mixer is sensitive to modulation of both phase angle and amplitude, with the relative sensitivity varying as the tangent of the phase angle. Because a mixer can be arranged to detect amplitude modulation, it can also be considered an alternative to a diode detector.

[0036] Given the wide variability in devices and operating conditions, it is always possible to find exceptions to the broad comparison of the relative performance of diodes and mixers as AM detectors. Generally, the conversion ratios of diodes and mixers are similar, but the noise level of mixers may be significantly lower. Figure 6 shows an example of the noise power (noise floor) 600 and 602 of a diode detector and a mixer detector, respectively, in a typical microwave AM detection application. The diode noise floor is reported, for example, in Grop and Rubiola's paper. The mixer noise floor is reported in CA Barnes et al., "Residual PM Noise Evaluation of Radio Frequency Mixers," 2011 Joint Conference of the IEEE International Frequency Control and the European Frequency and Time Forum (FCS) Proceedings, 2011, pp. 1-5.

[0037] The low noise power achievable using a mixer, as shown by the solid line 602 in Figure 6, provides a significant advantage. As is evident from Figure 6, the modulation frequency f mneeds only to be increased by about 100 Hz to reach minimum mixer noise, compared to the 10 kHz mentioned above for the diodes. This is consistent with the optimum choice for maximum phase to amplitude conversion.

[0038] While these advantages are clear, the in-phase relationship between the LO and RF signals at the I mixer must be maintained at a constant value close to 0° to minimize detection of phase modulation in the reflected signal. If the in-phase relationship is not maintained, the resulting inaccuracy in amplitude detection due to unwanted detection of phase modulation can negate the advantage of low noise power compared to that of a diode detector. The phase difference between the LO and RF signals drifts over time and temperature, which means a degree of unreliability due to phase modulation and the resulting drift in frequency error.

[0039] FIG. 7 shows a schematic diagram 700 and symbol 702 of a circuit called an IQ mixer 704. The IQ mixer 704 includes a pair of conventional mixers (i.e., an I mixer 706 and a Q mixer 708) and circuitry 710 for providing a fixed 90° phase shift between the LO signals at each mixer. Circuitry 710 is typically a microwave transmission line structure, such as a branch-line coupler, also known as a quadrature hybrid, or a power splitter, that splits a common LO signal 712 input at an LO input port 713 into two equal signals 714 and 716, the phases of which are 90° apart. An input RF signal 720 at an RF input port 715 is also split by a microwave power splitter 717 into equal components 722 and 724, with each component fed to one of the mixers in the IQ configuration. Note that the 90° shift could equally well be applied to the RF input instead of the LO input. If the lower mixer in FIG. 7 were sensitive to phase modulation (PM), depending on the phase relationship between the LO and RF signals applied to the configuration, it would be identified as a “Q” (quadrature) mixer 708. Meanwhile, the upper mixer 706 is an “I” (in-phase), or AM-sensitive, mixer 706. The Q mixer 708 produces a downconverted Q signal 730 at its intermediate frequency (IF) port 732, and the I mixer 706 produces a downconverted I output 734 at its IF port 736. IQ mixers operating at microwave frequencies are readily available as integrated circuits housed within a single package. These devices therefore provide a means of simultaneously measuring the modulation of both AM and PM signals, requiring only one phase adjustment between a common LO and RF signal. For convenience, an IQ mixer can be represented by the simpler schematic symbol 702 shown on the right side of FIG. 7.

[0040] This disclosure relates to a method for detecting amplitude modulation of a reflected signal. The present invention can be applied to any implementation of a frequency-stabilized microwave source using pound stabilization or pound servo techniques. The present disclosure replaces the diode detector with a specially configured IQ mixer to achieve a lower AM detection noise floor at lower modulation frequencies. The configured IQ mixer addresses and mitigates any phase drift over time and temperature, which in turn reduces the reliability of the phase modulation.

[0041] A frequency stabilized microwave source using pound stabilization or pound servo techniques consists of an oscillator loop that provides a gain of >1 and a phase shift of a multiple of 2π, and a carrier frequency f in the upper and lower sidebands. c , modulation frequency f m a modulation source configured to modulate a microwave carrier signal at a resonant frequency f Res a reference resonator with a modulation frequency f m and a pound servo loop that detects the difference between the carrier frequency and the resonant frequency via AM of the reflected signal at the resonator frequency and feeds back an error signal to drive the carrier frequency to the resonant frequency, thereby stabilizing the carrier signal at the resonator frequency. As mentioned above, in different configurations, modulation can be performed directly in the phase domain or indirectly in the frequency domain. Furthermore, in different configurations, oscillation can be provided by an arrangement in which the resonator is separate from the VCO or by an arrangement in which a reference resonator is included in the oscillation loop. The present disclosure uses an IQ mixer in the pound servo loop to detect the amplitude modulation of the signal reflected from the reference resonator.

[0042] 8, any such Pound-stabilized microwave source according to the present disclosure includes a core circuit 800 in which a microwave signal 802 passes through a circulator 804 and is reflected off a reference resonator 806. A modulation source 808 generates a modulation frequency f to apply phase modulation, either directly or indirectly via frequency modulation, to the microwave signal 802 impinging on the resonator. mThe frequency f present in the reflected carrier signal generates a modulated signal 810. m The degree of amplitude modulation at the carrier frequency f c and the resonator frequency f Res The frequency error between c -f Res The reflected signal 812 is redirected by the circulator 804 to an AM detector 814. A synchronous detector 816 detects the modulation frequency f m , is converted into a DC frequency error voltage 818 that represents the magnitude of the variation. This DC voltage is then amplified and preferably integrated by an integrator 820 to provide a pound servo gain to produce a pound servo voltage 822, which is fed back (to the tuning port of a VCO or to a phase shifter in the oscillator loop) to adjust the carrier frequency f c The resonator frequency f Res Drive to.

[0043] According to the present disclosure, the AM detector 814 is implemented as an IQ mixer 830 to detect amplitude modulation (AM) in the reflected signal 812. This achieves the low noise floor 602 associated with the mixer shown in FIG. 6 and thus the lower modulation frequency f m (For example, to achieve minimum mixer noise, m (need only to be greater than approximately 100 Hz), which can accommodate optimal selection of resonator coupling and maximum phase-to-amplitude conversion. Additionally, feedback is provided from the output of the Q mixer to the LO input of the IQ mixer to maintain the phase at approximately 90 degrees at the LO input to the Q mixer, thereby maintaining the phase at the LO input to the I mixer near 0 degrees. This is also referred to as "maintaining the LO and RF inputs in quadrature at the Q mixer." This is important to minimize the detection of any phase modulation in the reflected signal 812, so that the signal detected at the output of the I mixer is due solely to amplitude modulation of the reflected signal. If the LO input to the Q mixer is not maintained at 90 degrees, the phase modulation of the reflected signal 812 will induce errors in the ground servo voltage 822.

[0044] IQ mixer 830 includes an I mixer 832 and a Q mixer 842; I mixer 832 has an LO input port 834 and an RF input port 836, and an IF output port 840; Q mixer 842 has an LO input port 844 and an RF input port 846, and an IF output port 850. Microwave power splitter 852 splits LO signal 854 into first LO signal 856 and second LO signal 858 having relative phases 90° apart, which are applied to LO input ports 834 and 844, respectively. Microwave power splitter 860 splits reflected signal 812 into first LO signal 862 and second LO signal 864 having the same phase, which are applied to RF input ports 836 and 846, respectively. The I and Q mixers generate I and Q downconverted signals 870 and 872, respectively, at IF output ports 840 and 850, respectively.

[0045] In operation, a portion 880 of the signal 802 directed to the reference resonator 806 is coupled and transmitted to the LO port of the IQ mixer as LO signal 854 via a local oscillator (LO) phase shifter 882. For purposes of explanation, it is assumed that the voltage applied to the LO phase shifter 882 is zero volts, and that other phase-shifting means, such as a length of appropriate transmission line (not shown), or a combination of lumped circuit elements, can be provided so that the LO signal 854 and the RF signal 812, respectively, arrive at the input ports of the IQ mixer with some small phase difference (Δφ ≠ 0). Practical phase shifters typically function with control voltages of only one polarity, and it is a simple matter of summing with an offset to satisfy this condition.

[0046] The I mixer 832 in this scenario, while primarily responsive to the amplitude modulation of the reflected signal 812, also has small sensitivity to the phase modulation of the reflected signal, and the DC frequency error voltage 818 at the synchronous detector output is made up of two component voltages resulting from the amplitude and phase modulation of the reflected carrier, namely V AM and V PHTherefore, the phase modulation applied to the carrier and reflected from the resonator (still retained as phase modulation) can be considered as the sum of a small V PH =V Error V Error In response to V, the ground servo voltage 822 adjusts the frequency of the oscillator loop, and the amplitude-to-phase conversion effect of the resonator reflection is V AM =-V Error which causes the synchronous detector output to go to zero volts and the oscillator loop to operate at frequency f c =f Res +f Error It remains as it is.

[0047] Here, a small phase error Δφ can be thought of as a "DC modulation" of the phase between the LO signal 858 and the RF signal 864 at the input to the Q mixer 842. The downconverted Q signal 872 therefore carries a DC signal Vφ, which is the phase error voltage and is proportional to Δφ. It is important to note that Vφ arises not from modulation sidebands, but from the phase difference of the carrier (incident and reflected) at the mixer. The sidebands are located at frequencies f m Produces a varying signal at the Q output having

[0048] If Vφ is amplified and fed back with the correct sign to LO phase shifter 882 via LPF or amplifying integrator 890, the phase error Δφ will decrease while simultaneously tuning Q mixer 842 for pure phase sensitivity and tuning I mixer 832 for pure amplitude sensitivity. If I mixer 832 loses phase sensitivity, the voltage V Error decreases, resulting in a frequency error f Error decreases.

[0049] The fluctuating signal at the Q output resulting from phase modulation of the carrier is effectively removed from consideration by the low-pass filtering action of the integrator (or LPF). From the above discussion, it will be apparent that a simple low-pass filter and amplifier between the Q output 850 and the LO phase shift 882 will serve to produce a similar result. However, due to the infinite DC gain provided by the integrator, the phase error Δφ will be reduced to zero. The time constant of the integrator in the LO phase feedback loop can be long, since it only needs to compensate for relatively slow thermal or aging effects, and therefore f m It is very effective at removing modulated signals.

[0050] As a result, the downconverted I signal 870 has a frequency f m The demodulated signal has a magnitude and sign that represents the amplitude modulation of the reflected carrier. Thus, the synchronous detector detects the frequency error f c -f Res This generates a frequency error voltage 818 proportional only to the frequency error of the oscillator loop, which in turn reduces the frequency error of the oscillator loop towards zero.

[0051] The discussion so far has assumed that ideally, the independent I and Q mixers in an IQ mixer would respond as if the LO signals were exactly 90° out of phase (quadrature). This ideal condition may not be realized in practice, and some small quadrature error usually exists. As a result, even if the LO phase shifter is adjusted so that the Q mixer is completely phase sensitive, the I mixer will retain some slight sensitivity to phase modulation, resulting in a phase modulation-induced frequency error (f PMError ≠0). Diode detectors also have a small sensitivity to phase modulation. Essentially, both diode detectors and AM sensitive mixers produce a carrier frequency f c has some low-pass bandwidth that contributes to blocking f c The minimum detector bandwidth and therefore the maximum suppression value of mTherefore, both the diode and the mixer amplitude detector must meet the same fundamental rejection limits and the same minimum f for the phase modulated signal. PMError is affected by.

[0052] 8, achieving this fundamental limit may require compensation for any small quadrature errors in the IQ mixer. This compensation is achieved by adding a small quadrature compensation voltage 900 via a summing node 902 at the input to the integrator 890, as shown in FIG. 9. The f resulting from the quadrature errors in the IQ mixer 830 Error Since the sign of π ...

[0053] In either case, the carrier frequency f c The requirement that remains constant does not necessarily mean that f PMError This does not extend to requiring f = 0, PMError f = constant. PMError The stability of is a separate consideration and does not impair the ability of the mixer amplitude detector to detect smaller frequency deviations in the pound servo loop due to its low noise floor.

[0054] Thus, the disclosed technique has been demonstrated to maintain the correct phase relationship between the LO and RF signals in an IQ mixer such that the Q output maintains maximum sensitivity to amplitude modulation and minimum sensitivity to phase modulation, thereby enabling the practical use of the mixer in place of a diode as a detector in a Pound stabilization system and realizing the benefits of the mixer's low noise floor.

[0055] As will be explained, the IQ mixer can be implemented in any Pound-stabilized or Pound-servo frequency-stabilized microwave source. Figures 10-12 show three of the known Pound-stabilized microwave source topologies shown in Figures 1, 4, and 5, respectively.

[0056] 10, a frequency stabilized microwave source 1000 uses pound stabilization techniques to stabilize a microwave oscillator, i.e., stabilize a VCO 1002, relative to a reference resonator 1004 separate from the VCO, and generate a frequency stabilized output signal 1006 at an output port 1008. The VCO generates an output signal 1010, a portion of which is taken as the frequency stabilized output signal 1006. The signal frequency can be tuned by a pound servo voltage 1012 applied to a tuning port 1014 of the VCO. The output signal 1010 passes through a phase shifter 1016 and a circulator 1018 before hitting the reference resonator 1004. The frequency of the VCO 1002 is adjusted by the pound servo voltage 1012 applied to the tuning port 1014 of the VCO. Res ) should be tunable to match the center frequency of the resonator. Typically, the resonator will be a cavity that may include a dielectric element, but the technique can be applied using any resonator with suitable center frequency and stability.

[0057] A portion of the output signal 1010 that strikes the resonator is reflected according to known characteristics of the resonator as a reflected signal 1020. This reflected energy is directed by a circulator 1018 onto an amplitude detector 1022, which is implemented as an I / Q mixer 1023. A modulation source 1024 generates a modulation frequency (f m ) to generate a modulation signal 1025 and apply a phase modulation to the output signal 1010 impinging on the resonator using a phase shifter 1016. The phase modulation may result in an amplitude modulation of the reflected signal 1020 seen by an amplitude detector 1022. Such amplitude modulation gives rise to a fluctuating signal at the detector output, the fluctuation occurring at the same modulation frequency f m The IQ mixer 1023 is configured to minimize or eliminate any contribution to the demodulated signal 1032 caused by phase modulation in the reflected signal 1020.

[0058] The IQ mixer 1023 receives the reflected signal 1020 at its RF input 1030 and outputs at its I output 1034 a modulation frequency f m1010. An LO phase shifter 1042 receives a portion 1044 of signal 1010. A phase error voltage 1036 indicative of the phase difference between the incident and reflected carriers is provided at the Q output port 1038 of the carrier mixer, which is integrated 1040 and fed back as a nearly DC signal to the LO phase shifter 1042, which then modifies the effective phase shift to produce a signal 1046 at the LO input 1048 of the mixer, so that the LO and RF inputs are maintained in quadrature at the Q mixer and therefore in phase at the I mixer.

[0059] The synchronous detector 1050 converts the demodulated signal 1032 into a DC frequency error voltage 1052 that represents the magnitude of the fluctuations. By maintaining the inputs to the Q mixer in quadrature, the synchronous detector detects any amplitude modulation of the reflected signal, and therefore the frequency error f c -f Res 1054, which in turn reduces the frequency error of the oscillator loop towards zero. This DC voltage is then amplified and preferably integrated by an integrator 1054 to provide a pound servo gain to generate a pound servo voltage 1012 that is fed to the tuning port 1014 of the VCO 1002.

[0060] Another embodiment of a frequency stabilized microwave source 1100 is shown in Figure 11 as an improvement over microwave source 1000 shown in Figure 10, with like numbers used to identify like elements. The phase shift 1016 is removed and modulation source 1024 is added to the pound servo voltage 1012 via summing node 1102 and input to VCO tuning port 1014, resulting in frequency modulation of VCO 1002. The frequency modulation and phase modulation are simply

number

[0061] Referring now to FIG. 12, a frequency stabilized microwave source 1200 is shown in which a reference resonator 1202 also has a resonant frequency f Res 12 is arranged to form part of an oscillator loop 1204 that includes a bandpass filter (BPF) 1206 tuned to pass , and a loop amplifier 1210, thereby eliminating the need for a separate VCO. Figure 12, like Figure 11, illustrates that frequency modulation achieves phase modulation of the signal incident on the reference resonator.

[0062] A phase change in any of the oscillator loops 1204 results in a frequency change. Therefore, frequency can be modulated by providing a modulation signal 1216 from a modulation source 1218 to any of the phase shifters in the oscillator loop 1204. The pound servo feedback and modulation signal can be applied to a common phase shifter or separate phase shifters located somewhere in the oscillator loop 1204. If a common phase shifter can provide both the frequency modulation and tuning required by the pound servo feedback, it can be used. The phase shift range over which the precision required for modulation is achieved may be insufficient to achieve the tuning required for the pound servo feedback than with separate phase shifters. As shown, a common phase shifter 1224 is located at the output of the loop amplifier 1210. The pound servo voltage 1220 and modulation signal 1216 are summed by a summing node 1222 and input to the common phase shifter 1224.

[0063] The phase modulation induced by the frequency modulation of signal 1212 can result in amplitude modulation of the reflected signal 1230, which is directed by a circulator 1232 to an amplitude detector 1234, implemented as an IQ mixer 1236. At this amplitude detector 1234, the phase error voltage at the Q output is integrated 1238 and summed with a portion of signal 1212 by an LO phase shifter 1240, which feeds back to the LO input to keep the LO and RF inputs of the Q mixer in quadrature. Such amplitude modulation produces a fluctuating signal at the detector output. The fluctuating signal has the same modulation frequency f m The IQ mixer 1236 is configured to minimize or eliminate any contribution to the demodulated signal 1241 at the I output caused by phase modulation in the reflected signal 1230.

[0064] The synchronous detector 1242 detects the modulation frequency f m converts fluctuations in the I output signal at f into a DC frequency error voltage 1244 that represents the magnitude of the fluctuations. By maintaining the inputs to the Q mixer in quadrature, the synchronous detector detects any amplitude modulation of the reflected signal and therefore the frequency error f c -f Res This generates a frequency error voltage proportional only to the frequency error of the oscillator loop, which in turn reduces the frequency error of the oscillator loop towards zero. This DC voltage is then amplified and preferably integrated by integrator 1246 to generate the pound servo voltage 1220 that is supplied to summing node 1222.

[0065] The frequency stable output signal 1250 is taken from a different output 1252 positioned after the reference resonator 1202 of the oscillator loop 1204, which typically results in reduced noise in the oscillator output due to the filtering action of the reference resonator 1202. However, the output may equally well be taken before the reference resonator without affecting the functionality of the microwave source.

[0066] While several exemplary embodiments of the present disclosure have been shown and described, numerous variations and alternative embodiments will occur to those skilled in the art. Such variations and alternative embodiments are contemplated and can be made without departing from the scope of the invention as defined in the appended claims.

Claims

1. 1. A frequency stabilized microwave source comprising: Microwave carrier frequency f ｃ and configured to generate a signal including a carrier wave of said frequency f in response to a tuning voltage. ｃ an oscillator having a tuning port for controlling a pound stabilization loop, the pound stabilization loop comprising: Modulation frequency f ｍ a modulation source configured to generate a modulated signal of f ｃ +f ｍ and f ｃ -f ｍ a modulation source configured to add an upper sideband and a lower sideband of a center frequency f configured to receive at least a portion of the signal and generate a reflected carrier, a reflected upper sideband, and a reflected lower sideband; Ｒｅｓ a reference resonator having a modulation frequency f ｍ and the amplitude modulation (AM) of the reflected carrier at ｃ -f Ｒｅｓ an indicator of In response to the reflected signal, a modulation frequency f ｍ an AM detector for generating a demodulated signal representative of the amplitude modulation of the reflected carrier at The demodulated signal is mixed with the modulated signal to obtain the frequency error f ｃ -f Ｒｅｓ a synchronous detector generating a frequency error voltage indicative of the AM detector comprises an IQ mixer having an LO input port and an RF input port, and an I output port and a Q output port, the RF input port receiving the reflected signal and the demodulated signal being produced at the I output port; The IQ mixer generates a phase error voltage at the Q output port indicative of a phase difference between the carrier wave and the reflected carrier wave, and the AM detector further comprises: an LO phase shifter responsive to the phase error voltage for sampling the signal and generating a phase shifted signal; the phase-shifted signal is applied to the LO input port of the IQ mixer, so that the demodulated signal produced at the I output port is not affected by phase modulation of the reflected signal; Frequency stabilized microwave source.

2. 2. The frequency stabilized microwave source of claim 1, wherein the oscillator comprises a voltage controlled oscillator (VCO) that generates the signal, and the reference resonator is separate from the VCO.

3. 3. The frequency stabilized microwave source of claim 2, further comprising a phase shifter at the output of the VCO, the phase shifter responsive to the modulating signal to apply phase modulation to the signal to add the upper sideband and the lower sideband.

4. a summing node at the tuning port of the VCO, the summing node responding to the modulation signal and the frequency error voltage to produce a carrier frequency f ｃ 3. The frequency stabilized microwave source of claim 2, wherein the frequency modulates

5. 3. The frequency stabilized microwave source of claim 2, further comprising an integrator configured to integrate the frequency error voltage supplied to the tuning port of the VCO.

6. The oscillator includes the reference resonator as a frequency determining element in an oscillator loop, and further includes a first phase shifter in the oscillator loop, the first phase shifter being responsive to the tuning voltage to shift the frequency f ｃ 10. The frequency stabilized microwave source of claim 1, further comprising a voltage control port for controlling

7. 7. The frequency stabilized microwave source of claim 6, further comprising a second phase shifter in the oscillator loop responsive to the modulating signal to apply a phase modulation to the signal to add the upper sideband and the lower sideband.

8. 8. The frequency stabilized microwave source of claim 7, wherein the first phase shifter and the second phase shifter are a common phase shifter, and further comprising a summing node that sums the modulation signal and the frequency error voltage.

9. The frequency stabilized microwave source of claim 8 , further comprising an integrator configured to integrate the frequency error voltage provided to the summing node.

10. 10. The frequency stabilized microwave source of claim 1, further comprising a low pass filter (LPF) or an integrator between the Q output port and the LO phase shifter.

11. 11. The frequency stabilized microwave source of claim 10, further comprising a summing node at the input to the LPF or integrator, the summing node configured to sum the phase error voltage at the Q output port and a quadrature compensation voltage having a magnitude configured to reduce sensitivity of the demodulated signal to phase modulation of the reflected signal.

12. The I / Q mixer an I mixer having an LO input port, an RF input port, and said I output port; a Q mixer having an LO input port, an RF input port, and the Q output port; a first microwave power splitter positioned between an LO input port of the IQ mixer and the LO input ports of the I mixer and the Q mixer, the first microwave power splitter receiving the phase-shifted signal and generating a first LO signal and a second LO signal; a second microwave power splitter located between an RF input port of the IQ mixer and the RF input ports of the I mixer and the Q mixer, the second microwave power splitter receiving the reflected signal and generating a first RF signal and a second RF signal having the same phase; the first microwave power splitter and the second microwave power splitter are configured to generate a first relative phase and a second relative phase between the first LO signal and the first RF signal, and the second LO signal and the second RF signal, respectively, wherein a difference between the first relative phase and the second relative phase is 90°; 2. The frequency stabilized microwave source of claim 1, wherein the phase-shifted signal is applied to the LO input port of the IQ mixer, whereby the second LO signal at the LO input port of the Q mixer and the second RF signal at the RF input port are maintained in quadrature phase so that the demodulated signal produced at the I output port is not affected by phase modulation of the reflected signal.

13. The modulation frequency f ｍ is 5Hz<f ｍ 2. The frequency stabilized microwave source of claim 1, wherein the frequency is <5 KHz.

14. 1. A frequency stabilized microwave source comprising: Microwave carrier frequency f ｃ and configured to generate a signal including a carrier wave of said frequency f in response to a tuning voltage. ｃ an oscillator having a tuning port for controlling a pound stabilization loop, the pound stabilization loop comprising: Modulation frequency f ｍ a modulation source configured to generate a modulated signal of f ｃ +f ｍ and f ｃ -f ｍ a modulation source configured to add an upper sideband and a lower sideband of a center frequency f configured to receive at least a portion of the signal and generate a reflected carrier, a reflected upper sideband, and a reflected lower sideband; Ｒｅｓ a reference resonator having a modulation frequency f ｍ and the amplitude modulation (AM) of the reflected carrier at ｃ -f Ｒｅｓ an indicator of In response to the reflected signal, a modulation frequency f ｍ an AM detector for generating a demodulated signal representative of the amplitude modulation of the reflected carrier at The demodulated signal is mixed with the modulated signal to obtain the frequency error f ｃ -f Ｒｅｓ a synchronous detector generating a frequency error voltage indicative of The AM detector comprises an IQ mixer, the IQ mixer comprising: an I mixer having an LO input port, an RF input port, and an I output port; a Q mixer having an LO input port, an RF input port, and a Q output port; a first microwave power splitter that receives the phase-shifted signals and generates a first LO signal and a second LO signal at the input ports of the I mixer and the Q mixer, respectively; a second microwave power splitter that receives the reflected signal and generates a first RF signal and a second RF signal at the RF ports of the I mixer and the Q mixer, respectively; the first microwave power splitter and the second microwave power splitter are configured to generate a first relative phase and a second relative phase between the first LO signal and the first RF signal, and the second LO signal and the second RF signal, respectively, wherein a difference between the first relative phase and the second relative phase is 90°; the I mixer producing the demodulated signal at its I output port; The Q mixer generates a phase error voltage at the Q output port indicative of a phase difference between the carrier wave and the reflected carrier wave, and the AM detector further comprises: an LO phase shifter responsive to the phase error voltage for sampling the signal and generating the phase-shifted signal; the phase-shifted signal is applied to the LO input port of the Q mixer, whereby the second LO signal at the LO input port and the second RF signal at the RF input port of the Q mixer are maintained in quadrature such that the demodulated signal produced at the I output port is not affected by phase modulation of the reflected signal. Frequency stabilized microwave source.

15. 15. A frequency stabilized microwave source according to claim 14, wherein the oscillator comprises a voltage controlled oscillator (VCO) that generates the signal, and the reference resonator is separate from the VCO.

16. The oscillator includes the reference resonator as a frequency determining element in an oscillator loop, and further includes a first phase shifter in the oscillator loop, the first phase shifter being responsive to the tuning voltage to shift the carrier frequency f ｃ 15. The frequency stabilised microwave source of claim 14, having a voltage controlled port for controlling

17. 15. The frequency stabilized microwave source of claim 14, further comprising a low pass filter (LPF) or an integrator between the Q output port and the LO phase shifter.

18. 18. The frequency stabilized microwave source of claim 17, further comprising a summing node at an input to the LPF or integrator, the summing node configured to sum the phase error voltage at the Q output port and a quadrature compensation voltage having a magnitude configured to reduce sensitivity of the demodulated signal to phase modulation of the reflected signal.

19. An amplitude modulation (AM) detector for a ground-stabilized microwave source, the center frequency of which is f Ｒｅｓ is a reference resonator having a carrier frequency f ｃ and a carrier wave of modulation frequency f ｍ f in ｃ +f ｍ and f ｃ -f ｍ and the AM detector is configured to reflect a signal including an upper sideband and a lower sideband of 1. An IQ mixer having LO and RF input ports, and I and Q output ports, The RF input port receives a reflected signal including a reflected carrier and a reflected upper sideband and a reflected lower sideband, a portion of the reflected upper sideband and the reflected lower sideband having a modulation frequency f ｍ and the amplitude modulation (AM) of the reflected carrier at ｃ -f Ｒｅｓ and ,represents ,an indicator of The IQ mixer is responsive to a reflected signal to output a modulated signal at the I output port. ｍ and configured to generate a demodulated signal representative of the amplitude modulation of the reflected carrier at an IQ mixer configured to generate a phase error voltage at the Q output port indicative of a phase difference between the carrier wave and the reflected carrier wave; an LO phase shifter responsive to the phase error voltage for sampling the signal and generating a phase shifted signal; the phase-shifted signal is applied to the LO input port of the IQ mixer, so that the demodulated signal produced at the I output port is not affected by phase modulation of the reflected signal; AM detector.

20. The I / Q mixer an I mixer having an LO input port, an RF input port, and said I output port; a Q mixer having an LO input port, an RF input port, and the Q output port; a first microwave power splitter positioned between an LO input port of the IQ mixer and the LO input ports of the I mixer and the Q mixer, the first microwave power splitter receiving the phase-shifted signal and generating a first LO signal and a second LO signal; a second microwave power splitter located between an RF input port of the IQ mixer and the RF input ports of the I mixer and the Q mixer, the second microwave power splitter receiving the reflected signal and generating a first RF signal and a second RF signal; the first microwave power splitter and the second microwave power splitter are configured to generate a first relative phase and a second relative phase between the first LO signal and the first RF signal, and the second LO signal and the second RF signal, respectively, wherein a difference between the first relative phase and the second relative phase is 90°; 20. The AM detector of claim 19, wherein the phase-shifted signal is applied to the LO input port of the IQ mixer, whereby the second LO signal at the LO input port and the second RF signal at the RF input port of the Q mixer are maintained in quadrature phase so that the demodulated signal produced at the I output port is not affected by phase modulation of the reflected signal.

21. 20. The AM detector of claim 19, further comprising a low pass filter (LPF) or an integrator between the Q output port and the LO phase shifter.

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