Frequency-stabilized and phase-noise-suppressed microwave source using an IQ mixer to detect amplitude modulation and phase perturbations of the reflected signal

Abstract

An IQ mixer is used in a Pound-stabilized microwave source to detect amplitude modulation of the signal reflected from the reference resonator. By properly configuring the IQ mixer so that the LO and RF inputs are kept in quadrature at the Q mixer and therefore in phase at the I mixer, lower levels of amplitude modulation can be detected at lower modulation frequencies consistent with optimal selection of resonator coupling and maximum phase-to-amplitude conversion. With the Q mixer held in quadrature, it acts as a wideband phase noise detector. A portion of the Q mixer output is bandpass filtered and summed with the Pound servo voltage of the I mixer to achieve both center frequency stabilization and wideband phase noise suppression.

Description

[Technical Field]

[0001] Priority claims This application claims the benefit of priority to U.S. Application No. 18 / 095,855, filed January 11, 2023, which is incorporated herein by reference in its entirety.

[0002] This disclosure relates to frequency-stabilized microwave sources using "Pound stabilization" or "Pound servo" techniques, and more particularly to the use of an IQ mixer to detect amplitude modulation and phase perturbations in a reflected signal. This dual use of an IQ mixer in a Pound servo system simultaneously enables both center frequency stabilization and wideband phase noise suppression. [Background technology]

[0003] Stable microwave frequency sources are used in applications such as radar and communications systems. The most stable microwave sources, such as hydrogen masers or cryogenic sapphire oscillators, find use as time bases for precision clocks. Such microwave sources and clocks are bulky and unsuitable for portable devices that may require synchronization over large distances. Therefore, many such devices synchronize by utilizing time information provided by GPS signals. However, GPS reception may be unavailable during certain periods, and a small, low-power oscillator with sufficient stability to maintain synchronization during such periods is needed.

[0004] A simple oscillator may include a loop containing at least a resonator, an amplifier, and a phase shifter, which together may satisfy the oscillation condition that the gain and phase shift around the loop are greater than 1 and a multiple of 2π, respectively. A resonator with a highly stable center frequency will not immediately lead to a highly stable oscillation frequency unless the phase shift through the amplifier and phase shifter is also highly stable.

[0005] RVPound, "Frequency Stabilization of Microwave Oscillators," in Proceedings of the IRE, vol. 35, no. 12, pp. 1405-1415, December 1947, describes a technique that utilizes the reflection of a microwave oscillator signal from a reference resonator to improve oscillator stability. In U.S. Pat. No. 2,681,998, Pound combined this technique with modulation of the oscillator signal to avoid limitations caused by flicker noise in microwave detectors. Microwave oscillator stabilization techniques that involve the reflection of a modulated signal from a resonator have become commonly referred to as "Pound stabilization" or "Pound servo" techniques. More recently, a single resonator is often utilized as both a component of a microwave oscillator loop and as a stabilizing element used to improve oscillator loop stability. See N. Luiten, A.G. Mann, N.J. McDonald and D.G. Blair, "Latest results of the UW A cryogenic sapphire oscillator," Proceedings of the 1995 IEEE International Frequency Control Symposium (49th Annual Symposium), 1995, pp. 433-437, and N. Luiten, A.G. Mann, M.E. Costa and D.G. Blair, "Cryogenic sapphire resonator-oscillator with exceptional stability: an update," Proceedings of the IEEE 48th Annual Symposium on Frequency Control, 1994, pp. 441-446.

[0006] 1, a frequency-stabilized microwave source 100 uses Pound stabilization techniques to stabilize a microwave oscillator, or voltage-controlled oscillator (VCO) 102, to a reference resonator 104 to generate a frequency-stabilized output signal 106 at an output port 108. The VCO generates an output signal 110, a portion of which is tapped off as the frequency-stabilized output signal 106. The signal frequency may be tuned by a Pound servo voltage 112 applied to a tuning port 114 of the VCO. The output signal 110 passes through a phase shifter 116 and a circulator 118 before being incident on the reference resonator 104. The frequency of the VCO 102 is adjusted to a center frequency f Res Typically, the resonator will be a cavity resonator that may include a dielectric element, but the technique may be applied using any resonator with suitable center frequency and stability.

[0007] A portion of the output signal 110 incident on the resonator is reflected according to known characteristics of the resonator as a reflected signal 120. This reflected energy is directed by a circulator 118 onto an amplitude detector 122, most commonly a simple diode detector. A modulation source 124 generates a modulation frequency f 124 to impart a phase modulation to the output signal 110 incident on the resonator by a phase shifter 116. m As will be explained in more detail below, the phase modulation may result in an amplitude modulation in the reflected signal 120 that is detected by the amplitude detector 122. Any such amplitude modulation will result in a fluctuating signal at the detector output, the fluctuations occurring at the same modulation frequency f m It has.

[0008] A synchronous detector 126, such as a synchronous rectifier, mixes the output of amplitude detector 122 with a modulating signal 125 to produce a modulating frequency f mThe amplitude detector converts fluctuations in the amplitude detector output at 124 into a DC frequency error voltage 128 that represents the magnitude of the fluctuations. This DC frequency error voltage is then amplified and preferably integrated via an integrator 130 to provide a Pound servo gain to generate the Pound servo voltage 112 that is fed to the tuned port 114 of the VCO 102. A synchronous rectifier may also be known in the art by terms such as a synchronous detector, a lock-in detector, a lock-in amplifier, or a lock-in mixer.

[0009] To understand the operation of a frequency-stabilized microwave source, consider the incident spectrum 200 of a signal incident on a reference resonator, as shown in Figure 2A, and the variation with frequency of the complex reflection coefficient 202 of the reference resonator, as shown in Figure 2B, as separate amplitude and phase plots 204 and 206, respectively. For ease of explanation, consider the VCO frequency f of a carrier wave 208. c is intentionally offset from the center frequency of the resonator, and f c ≠f Res The first pair of phase modulation sidebands 210 and 212 are at frequencies f U =f c +f m and f L =f c -f m are arranged symmetrically around the carrier wave.

[0010] The upper and lower sidebands 210 and 212 are located at the resonance center f Res Because the reference resonators are not symmetrically positioned around f, they are reflected from the reference resonator with different amplitudes and phases. As a result, the reflected carrier is modulated in both phase and amplitude. c =f Res Only in the special case of , both the upper and lower sidebands 210 and 212 are reflected equally, and the frequency f m As will become apparent, the reflected carrier signal will have no amplitude modulation at frequency f m The degree of amplitude modulation at c -f Res is an indicator of.

[0011] To understand the conversion from phase to amplitude modulation, consider the phasor diagrams 300 and 302 shown in Figures 3A-3E. Figures 3A-3C provide a simplified representation of the phase modulation, as shown in phasor diagram 300, of a phase-modulated carrier 304 as incident on a reference resonator. The phase modulation is expressed as ±2πf at the ends of the carrier phasor. m The sideband phasors 306 and 308 are represented by two counter-rotating phasors 306 and 308 at a rate of radians per second. As shown in Figure 3A, when the sideband phasors 306 and 308 are aligned with the carrier 304, the sum of the phasors is zero. However, as shown in Figures 3B and 3C, when the sideband phasors 306 and 308 are not aligned with the carrier 304, the incident lower and upper sidebands 310 and 312 are added to the carrier 304, advancing and retarding the phase of the carrier, respectively. As a result, the modulation frequency f m A periodic variation of the carrier phase at 1 / 2 Hz results. Careful examination of Figures 3A-3C reveals that higher-order sidebands are required to accurately describe the pure phase modulation, since the length (amplitude) of the carrier phasor changes slightly in Figures 3B and 3C compared to Figure 3A, but the simplification of ignoring the higher-order sidebands is sufficient to explain the process.

[0012] If the incident signal is a carrier 304 and two equal amplitude sidebands 306 and 308 that add to zero, as shown in FIG. 3A, the reflected sidebands 316 and 318 will have different amplitudes and different phases, as shown in FIG. 3D.

[0013] Consider a reflected signal 320 in which the sidebands vary in both amplitude 322 and phase 324 according to the frequency dependence of the resonator reflection coefficient. As shown in Figure 3E (not to scale), the reflected upper sideband 326 is reflected with a much larger amplitude than the reflected lower sideband 328. The sidebands can no longer cancel each other in any orientation, and their sum follows a roughly elliptical locus 323. Thus, both the length and phase angle of the reflected carrier wave change significantly, resulting in f c ≠f Res Whenever the reflected carrier frequency fm There is both amplitude and phase modulation in

[0014] Furthermore, when the frequency error has an opposite sign, the relative changes of the reflected sidebands 326 and 328 swap, with the lower sideband 328 dominating and reversing the sum motion around the elliptical locus 323. Thus, the phase of the amplitude modulated signal seen at the detector output reverses relative to the original modulated signal, and the sign of the DC signal recovered by the synchronous detector also reverses, resulting in a frequency error f below the resonator bandwidth. c -f Res For , the synchronous detector output is f c -f Res is proportional to.

[0015]

number

number

number

number

[0016] It should be understood that the width of the dip in the amplitude reflection coefficient 204, or alternatively the peak of the phase to amplitude conversion 214, depends on the coupling coefficient of the resonator, not the transmission bandwidth of the resonance. When the resonator coupling is near critical, the conversion of phase to amplitude modulation is at a maximum, f m The optimum range for f can be much smaller than the resonator bandwidth. For example, Res For a critically coupled resonator with a transmission bandwidth of 10 GHz and 400 kHz, the width of the phase-to-amplitude conversion peak, as measured by a 3 dB change from maximum, is less than 1 kHz.

[0017] FIG. 4 shows another embodiment of a frequency-stabilized microwave source 400 as a variation of the microwave source 100 shown in FIG. 1, with like numbers used to identify like elements. Phase shifter 116 is eliminated, and modulation source 124 is summed with Pound servo voltage 112 via summing node 402 and input to VCO tuning port 114 to cause frequency modulation of VCO 102. Because frequency modulation and phase modulation are simply related by a factor of 1 / jω, phase modulation of the carrier occurs with a 90° phase shift relative to the modulation source. However, this is easily accounted for by adjusting the relative phase of the signals in synchronous detector 126.

[0018] Referring now to FIG. 5, a frequency stabilized microwave source 500 includes a reference resonator 502 having a resonant frequency f Res 5 also forms part of an oscillator loop 504 that includes a bandpass filter (BPF) 506 tuned to pass .gt., and a loop amplifier 510, arranged to eliminate the need for a separate VCO. Figure 5, like Figure 4, achieves phase modulation of the signal incident on the reference resonator through frequency modulation.

[0019] Any phase change in the oscillator loop 504 causes a frequency change. Therefore, the frequency can be modulated by providing a modulation signal 516 from a modulation source 518 to any phase shifter in the oscillator loop 504. The Pound servo feedback and modulation signals can be applied to a common phase shifter or separate phase shifters. If a common phase shifter can provide the performance for both the frequency modulation and the tuning required by the Pound servo feedback, it can be used. If the phase shift range at which the precision required for the modulation is achieved is insufficient to achieve the tuning required for the Pound servo feedback, a separate phase shifter may be required.

[0020] If a common phase shifter is used, the Pound servo voltage 520 and the modulation signal 516 are summed by a summing node 522 and input to a common modulation phase shifter, either a loop phase shifter 508 before the loop amplifier 510, or a modulation phase shifter 524 after the loop amplifier 510. In this configuration, the other phase shifter is eliminated. It may be preferable to place the common phase shifter before the loop amplifier 510 to use the full output power of the amplifier.

[0021] If separate phase shifters are used, the Pound servo voltage 520 is input to the loop phase shifter 508, and the modulation signal 516 is input to the modulation phase shifter 524. The summing node 522 is eliminated in this configuration. The positions of the phase shifters in the loop may be reversed, dictated by practical considerations such as whether they can handle the amplified output power.

[0022] Phase modulation induced through frequency modulation in signal 512 can result in amplitude modulation in reflected signal 530, which is directed by circulator 532 to amplitude detector 534. Any such amplitude modulation produces a fluctuating signal at the detector output, which fluctuates at the same modulation frequency f m The synchronous detector 536 detects the modulation frequency f mThe amplitude detector converts the variation in the output of the amplitude detector at 508 into a DC frequency error voltage 538 that represents the magnitude of the variation. This DC voltage is then amplified and preferably integrated 540 to produce the Pound servo voltage 520 that is fed to the summing node 522 or directly to the loop phase shifter 508.

[0023] 5, the frequency stabilized output signal 542 is taken from a different output 544 located after the reference resonator 502 in the oscillator loop 504, which typically results in a reduction of noise in the oscillator output due to the filtering action of the reference resonator 502. However, the output may also be taken before the reference resonator as in FIGS. 1 and 4 without affecting the functionality of the microwave source. Summary of the Invention

[0024] The following is a summary to provide a basic understanding of some aspects of the disclosure. This summary is not intended to identify key or critical elements of the disclosure or to delineate the scope of the disclosure. Its sole purpose is to present some concepts of the disclosure in a simplified form as a prelude to the more detailed description and claims that are presented later.

[0025] This disclosure describes the use of an IQ mixer in a Pound servo loop to detect amplitude modulation of a signal reflected from a reference resonator. By properly configuring the IQ mixer so that the LO and RF inputs are maintained in quadrature at the Q mixer and therefore in phase at the I mixer, lower levels of amplitude modulation can be detected at lower modulation frequencies consistent with optimal selection of resonator coupling and maximum phase-to-amplitude conversion. With the Q mixer in quadrature, it can be utilized as a wideband phase noise detector summed with the Pound servo loop to achieve wideband phase noise suppression in addition to center frequency stabilization.

[0026] In frequency-stabilized microwave sources that use Pound stabilization or Pound servo techniques to stabilize the oscillation frequency, the amplitude detector detects the RF input, where the signal reflected from the reference resonator provides the RF input, and the I output is the modulation frequency f, which represents the amplitude modulation (AM) of the reflected carrier. m The LO phase shifter is implemented as an IQ mixer that produces a demodulated signal at √Hz, which is applied to a synchronous detector. A phase error voltage, indicative of the phase difference between the incident and reflected carriers provided at the Q output, is integrated and fed back to the LO phase shifter, where it is combined with a portion of the signal incident on the resonator to produce a signal at the LO input such that the LO and RF inputs remain in quadrature at the Q mixer and therefore in phase at the I mixer. As a result, the synchronous detector detects the frequency error f c -f Res The Q mixer generates a frequency error voltage proportional to the LO and RF inputs. The LO and RF inputs are held in quadrature at the Q mixer, which acts as a phase perturbation detector. A portion of the Q mixer output is bandpass filtered and summed with the integrated frequency error voltage (Pound servo voltage). This combined voltage is applied to the oscillator tune port, providing noise rejection for both center frequency error / close-in phase noise and wideband phase noise. The frequency error is corrected by the Pound servo, and the wideband phase noise is suppressed by the Q mixer output.

[0027] To suppress wideband phase noise without degrading close-in phase noise, the BPF has a low-frequency cutoff frequency (f) designed to minimize muting (e.g., reduction or suppression) of the modulated signal. cLow >f m , high frequency cutoff f designed to allow gain margin >0dB and phase margin >0deg for closed loop stability cHigh , as well as sufficient gain in the passband to reach the noise floor of the IQ mixer.

[0028] An IQ mixer configured in this manner can be used as an AM and PM detector in any frequency-stabilized microwave source that utilizes Pound stabilization or Pound servo techniques. For example, these techniques may utilize a reference resonator as a frequency-determining element in an oscillator loop and modulate the phase of a signal incident on the resonator by applying a modulation signal to a phase shifter at any point in the oscillator loop. One or more phase shifters may be provided in the oscillator loop, whereby the modulation signal may be combined with or separate from the Pound servo feedback signal. Furthermore, these techniques may include configurations in which the reference resonator is separate from the VCO that generates the microwave signal, with phase modulation achieved by a phase shifter between the VCO output and the reference resonator or by combining the modulation signal with a Pound servo feedback signal applied to tune the VCO.

[0029] In one embodiment, to compensate for non-idealities in the quadrature phase shift between the LO signals in the individual I and Q mixers, a quadrature compensation voltage can be summed with the phase error voltage at the Q output at the input to the integrator. When LO phase compensation is used, the wideband noise suppression signal of the Q mixer needs to be sampled before compensation injection so that changes in the compensation voltage are not injected into the noise suppression voltage.

[0030] These and other features and advantages of the present disclosure will become apparent to those skilled in the art from the following detailed description of the preferred embodiments, taken in conjunction with the accompanying drawings. [Brief explanation of the drawings]

[0031] [Figure 1] As mentioned above, Pound stabilization is a known configuration of a frequency stabilised microwave source that is used to stabilise an oscillator against an independent reference resonator. [Figure 2A] As mentioned above, this is a plot of the incident spectrum. [Figure 2B] As mentioned above, this is a plot of the reflection spectrum from the reference cavity. [Figure 3A]As mentioned above, this is a phasor diagram illustrating the conversion from phase to amplitude modulation. [Figure 3B] As mentioned above, this is a phasor diagram illustrating the conversion from phase to amplitude modulation. [Figure 3C] As mentioned above, this is a phasor diagram illustrating the conversion from phase to amplitude modulation. [Figure 3D] As mentioned above, this is a phasor diagram illustrating the conversion from phase to amplitude modulation. [Figure 3E] As mentioned above, this is a phasor diagram illustrating the conversion from phase to amplitude modulation. [Figure 4] As mentioned above, Pound stabilization is another known configuration of a frequency stabilised microwave source that is used to stabilise an oscillator against an independent reference resonator. [Figure 5] As mentioned above, Pound stabilization is a known configuration of a frequency stabilised microwave source that is implemented using a reference resonator that also forms part of the oscillator loop. [Figure 6A] 1 is a plot of the noise floor of a diode and a mixer as an AM detector. [Figure 6B] 10 is a single-sideband phase noise plot of oscillator phase noise comparing free-running, Pound stabilization with IQ mixer, and Pound stabilization with IQ mixer plus phase noise suppression configurations. [Figure 7] 1 is a schematic diagram and symbolic representation of an IQ mixer. [Figure 8] 1 is a schematic diagram of a core portion of a Pound stabilized microwave source including an IQ mixer configured to perform amplitude modulation detection. [Figure 9] 1 is an alternative configuration of the IQ mixer to compensate for non-idealities of the individual I and Q mixers. [Figure 10] 1 is a first embodiment of a Pound-stabilized frequency-stabilized microwave source in which an IQ mixer is configured to perform AM detection. [Figure 11] 1 is a second embodiment of a Pound-stabilized frequency-stabilized microwave source in which the IQ mixer is configured to perform AM detection. [Figure 12]1 is a first embodiment of a Pound-stabilized frequency-stabilized microwave source in which an IQ mixer is configured to perform AM detection. DETAILED DESCRIPTION OF THE INVENTION

[0032] This disclosure describes the use of an IQ mixer in place of a simple diode in a Pound servo loop to detect amplitude modulation of a signal reflected from a reference resonator. By properly configuring the IQ mixer so that the LO and RF inputs are maintained in quadrature at the Q mixer, lower levels of amplitude modulation can be detected at lower modulation frequencies consistent with optimal selection of resonator coupling and maximum phase-to-amplitude conversion. With the Q mixer in quadrature, it can be utilized as a wideband phase noise detector summed with the Pound servo loop to achieve wideband phase noise suppression in addition to center frequency stabilization.

[0033] In frequency-stabilized microwave sources using Pound stabilization or Pound servo techniques, diode detectors offer a very simple and low-cost method for recovering amplitude modulation in the reference resonator's reflected signal. However, signal-to-noise ratio (SNR) must be considered when selecting a detector. Diodes are nonlinear devices, and their conversion ratio—the signal voltage, sometimes expressed in mV / dB, generated for a given change in input amplitude—depends on the input carrier amplitude and the load resistance seen by the diode, as shown in Figure 1 of Serge Grop and Enrico Rubiola, “Flicker Noise of Microwave Power Detectors,” 2009 IEEE International Frequency Control Symposium Joint with the 22nd European Frequency and Time Forum, 2009, pp. 40–43. Diode detectors also generate noise, which limits the minimum detectable signal. The noise has both a flicker or 1 / f component that depends on the incident carrier power and a white floor, often set by the load resistance.

[0034] Inherent within the synchronous detector is a low-pass filter (LPF) element that allows the synchronous detector to filter the modulation frequency f m It is only sensitive to the surrounding frequency bandwidth. Therefore, the advantage of Pound stabilization is that m Only noise power within the surrounding synchronous detector bandwidth contributes to noise at the diode detector output. Therefore, for a diode detector with noise 600 as shown by the dashed line in FIG. 6, f > about 10 kHz is required to achieve the lowest detector noise. m It would be appropriate to choose

[0035] When selecting a diode detector, the designer must also consider the amount of power present in the diode detector resulting from the power incident on the resonator and the resonator reflection coefficient due to the resonator coupling coefficient β. If very low power is reflected into the diode, as is the case when the resonator is critically coupled, the diode conversion ratio will be low, and if a high load resistance is selected to improve the conversion ratio, the resistance will contribute higher thermal noise. Alternatively, if the resonator coupling is set to a less optimal value, resulting in higher power impinging on the diode, the conversion ratio may be higher and a lower load resistance may be used. However, as the power incident on the diode increases, flicker or 1 / f noise increases, so the modulation frequency f must be increased to improve the signal-to-noise ratio. m Therefore, the use of diode detectors is dictated by maximizing the phase-to-amplitude conversion, i.e., a small f near the critical coupling in the resonator as mentioned above. m indicates a choice that is inferior to the optimal choice.

[0036] A mixer may be used as a phase detector at microwave frequencies by arranging the input LO and RF signals to be in quadrature (90° relative phase); see, for example, Stephan R. Kurtz, "Mixers as Phase Detectors," Tech-note, Watkins-Johnson Company, Vol. 5, No. 1, January / February 1978. Conversely, when the input LO and RF signals are in phase (0° relative phase), the mixer responds to amplitude modulation of the RF signal. At intermediate phase angles between the LO and RF, the mixer is sensitive to both phase and amplitude modulation, with the relative sensitivity varying as the tangent of the phase angle. Because a mixer may be arranged to detect amplitude modulation, it may also be considered an alternative to a diode detector.

[0037] Given the wide variability in devices and operating conditions, it is always possible to find exceptions to the rough comparison of the relative performance of diodes and mixers as AM detectors. Generally, the conversion ratios of diodes and mixers are similar, but the noise level of mixers can be significantly lower. Figure 6a shows examples of noise power (noise floors) 600 and 602 for diode and mixer detectors, respectively, in a typical microwave AM detection application. Diode noise floors are reported, for example, in Grop and Rubiola's paper. Mixer noise floors are reported in C.A. Barnes et al., "Residual PM Noise Evaluation of Radio Frequency Mixers," 2011 Joint Conference of the IEEE International Frequency Control and the European Frequency and Time Forum (FCS) Proceedings, 2011, pp. 1-5.

[0038] The lower noise power that can be achieved by using a mixer, shown by the solid line 602 in Figure 6a, provides a significant advantage. As is evident from Figure 6, to reach minimum mixer noise, the modulation frequency f m need only be greater than about 100 Hz compared to 10 kHz for the diodes mentioned above, which is consistent with the optimum selection for maximum phase-to-amplitude conversion.

[0039] While these advantages are clear, the in-phase relationship between the LO and RF signals at the I mixer must be maintained at a constant value close to 0° to minimize the detection of phase modulation in the reflected signal. If the in-phase relationship is not maintained, the advantage of low noise power compared to diode detectors can be negated by contamination of the amplitude detection due to the unwanted detection of the accompanying phase modulation. The phase difference between the LO and RF signals drifts over time and temperature, which means the degree of phase modulation contamination and the resulting frequency error drift.

[0040] Figure 6b shows representative single-sideband phase noise plots of the resulting oscillator phase noise for three different configurations. The free-running oscillator configuration is represented by trace 604, with neither Pound stabilization nor phase noise suppression enabled. Trace 606 shows the phase noise when Pound stabilization with an IQ mixer is enabled. The integrated center frequency error correction provided by the Pound stabilization network using an IQ mixer reduces the phase noise at the crossover frequency f, the frequency at which the closed-loop gain of the Pound servo passes through 0 dB. cross-over To the left of 610, effective suppression of the close-in phase noise 608 is provided. m must be outside the bandwidth of the Pound servo (i.e., f m >f cross-over ). If a trace of a Pound stabilization network using a diode AM detector were shown, the close-in phase noise would be higher (less suppressed). Trace 612 shows the phase noise when both Pound stabilization and wideband phase noise suppression by the IQ mixer are enabled. When enabled, the wideband phase noise 614 to the right of the crossover frequency 610 is reduced. Trace 616 shows the phase noise at the modulation frequency f due to the introduction of the modulating signal. m 6 shows a Pound spur at carrier frequency f. The low frequency end of the Q-channel wideband phase noise suppression network is rolled off to prevent muting of the desired Pound stabilization modulation tone and degradation of the excellent close-in phase noise performance of Pound stabilization. As shown, close-in phase noise 608 is c (0 Hz on the x-axis) and the crossover frequency 610, and the wideband phase noise 614 is above the crossover frequency 610. In this example, m = 100Hz and f cross-over = 10 Hz. A BPF that implements wideband phase noise suppression has a low frequency cutoff f designed to minimize muting of the modulated signal. cLow >f m , high frequency cutoff f designed to allow gain margin >0dB and phase margin >0deg for closed loop stability cHigh, as well as sufficient gain to reach the noise floor 602 of the IQ mixer. In this example, f cLow is about 1 kHz, and f cHigh is about 10MHz.

[0041] FIG. 7 shows a schematic diagram 700 and symbol 702 of a circuit called an IQ mixer 704. The IQ mixer 704 includes a pair of conventional mixers, an I mixer 706 and a Q mixer 708, and a circuit element 710 that provides a fixed 90° phase shift between the LO signals at each mixer. Circuit element 710 is typically a microwave transmission line structure, such as a branch-line coupler, also known as a quadrature hybrid, or a power divider, that splits an input common LO signal 712 at an LO input port 713 into two equal signals 714 and 716 whose relative phases are 90° apart. An input RF signal 720 at an RF input port 715 is split by the microwave power divider into equal components 722 and 724, with each component fed to one of the mixers in the IQ configuration. It should be noted that, equivalently, the 90° shift could be applied to the RF input instead of the LO input. Depending on the phase relationship between the LO and RF signals applied to the arrangement, if the lower mixer in FIG. 7 is sensitive to phase modulation (PM), it is identified as a “Q” (quadrature) mixer 708, while the upper mixer is an “I” (in-phase), or AM-sensitive, mixer 706. The Q mixer 708 produces a down-converted Q signal 730 at its intermediate frequency (IF) port 732, and the I mixer 706 produces a down-converted I output 734 at its IF port 736. IQ mixers for operation at microwave frequencies are readily available as integrated circuits housed in a single package. These devices therefore provide a means of simultaneously measuring both AM and PM signal modulation, with only one phase adjustment required between the common LO and RF signals. For convenience, an IQ mixer can be represented by the simpler circuit symbol 702 shown on the right side of FIG. 7.

[0042] This disclosure relates to how the amplitude modulation of the reflected signal is detected and can be applied to any implementation of a frequency-stabilized microwave source that uses Pound stabilization or Pound servo techniques. This disclosure replaces the diode detector with a (specially configured) IQ mixer to achieve a lower AM detection noise floor at lower modulation frequencies. The so-configured IQ mixer addresses and mitigates any phase drift over time and temperature, and therefore any drift in the phase modulation contamination.

[0043] A frequency stabilized microwave source using Pound stabilization or Pound servo technology includes an oscillator loop that provides a gain >1 and a phase shift of a multiple of 2π, with upper and lower sidebands along with a modulation frequency f m At carrier frequency f c a modulation source configured to modulate a microwave carrier signal at a resonant frequency f to reflect a portion of the microwave signal; Res a reference resonator with a modulation frequency f m The present disclosure includes a Pound servo loop that detects the difference between the carrier frequency and the resonant frequency via AM of the reflected signal at the reference resonator and feeds back an error signal to drive the carrier frequency to the resonant frequency, thereby stabilizing the carrier signal at the resonator frequency. As mentioned above, in different configurations, modulation may be performed directly in the phase domain or indirectly in the frequency domain. Furthermore, in different configurations, oscillation may be provided by an arrangement in which the resonator is separate from the VCO or by an arrangement in which a reference resonator is included in the oscillation loop. The present disclosure uses an IQ mixer in the Pound servo loop to detect the amplitude modulation of the signal reflected from the reference resonator.

[0044] 8, any such Pound stabilized microwave source according to the present disclosure includes a core circuit 800 in which a microwave signal 802 passes through a circulator 804 and is reflected off a reference resonator 806. A modulation source 808 generates a modulation frequency f to impart phase modulation, either directly or indirectly through frequency modulation, to the microwave signal 802 incident on the resonator. mThe frequency f present in the reflected carrier signal generates a modulated signal 810. m The degree of amplitude modulation at the carrier frequency f c and the resonator frequency f Res The frequency error between c -f Res The reflected signal 812 is redirected by the circulator 804 to an AM detector 814. A synchronous detector 816 detects the modulation frequency f m The fluctuations in the amplitude detector output at the carrier frequency f are converted into a DC frequency error voltage 818 that represents the magnitude of the fluctuations. This DC voltage is then amplified and preferably integrated via an integrator 820 to provide the Pound servo gain, c The resonator frequency f Res , which generates a Pound servo voltage 822 that is fed back (to the tuned port of a VCO or to a phase shifter in the oscillator loop) to drive

[0045] According to the present disclosure, the AM detector 814 is implemented as an IQ mixer 830 to detect amplitude modulation (AM) in the reflected signal 812. This achieves the low noise floor 602 associated with the mixer shown in FIG. 6 and therefore the lower modulation frequency f m (e.g., f m The use of a 100 Hz frequency (which only needs to be greater than approximately 100 Hz to achieve minimum mixer noise) is promoted, which is consistent with optimal selection of resonator coupling and maximum phase-to-amplitude conversion. Furthermore, feedback is provided from the output of the Q mixer to the LO inputs of the IQ mixers to maintain the phase at the LO input to the Q mixer at approximately 90 degrees and the phase at the LO input to the I mixer at approximately 0 degrees. This is also referred to as "maintaining the LO and RF inputs in quadrature at the Q mixer." This is important to minimize detection of any phase modulation in the reflected signal 812 so that the detected signal at the output of the I mixer is due only to amplitude modulation in the reflected signal. If the LO input to the Q mixer is not maintained at 90 degrees, phase modulation in the reflected signal 812 will induce errors in the pound servo voltage 822.

[0046] IQ mixer 830 includes an I mixer 832 having an LO input port 834 and an RF input port 836 and an IF output port 840, and a Q mixer 842 having an LO input port 844 and an RF input port 846 and an IF output port 850. A microwave power splitter 852 splits an LO signal 854 into a first LO signal 856 and a second LO signal 858 whose relative phases are separated by 90°, which are applied to LO input ports 834 and 844, respectively. A microwave power splitter 860 splits a reflected signal 812 into a first RF signal 862 and a second RF signal 864 having the same phase, which are applied to RF input ports 836 and 846, respectively. The I and Q mixers generate I and Q down-converted signals 870 and 872, respectively, at IF output ports 840 and 850, respectively.

[0047] In operation, a portion 880 of the signal 802 directed to the reference resonator 806 is decoupled and sent to the LO port of the IQ mixer as LO signal 854 via a local oscillator (LO) phase shifter 882. For ease of explanation, assume that the voltage applied to LO phase shifter 882 is zero volts, and that other phase-shifting means, such as appropriate transmission line path lengths or a combination of lumped circuit elements (not shown), are provided, so that the LO signal 854 and the RF signal 812 arrive at the input ports of the IQ mixer with some small phase difference Δφ≠0. Practical phase shifters typically function with control voltages of only one polarity, but it is a simple matter to add an offset to satisfy this condition.

[0048] The I mixer 832 in this scenario is primarily responsive to the amplitude modulation of the reflected signal 812, but also has a small sensitivity to phase modulation in the reflected signal, and the DC frequency error voltage 818 at the synchronous detector output is a mixture of two component voltages V resulting from the amplitude and phase modulation of the reflected carrier. AM and V PH Therefore, the phase modulation applied to the carrier and reflected from the resonator (still as a phase modulation) can be considered as the sum of a small V PH =V Error V ErrorIn response to this, the Pound servo voltage 822 adjusts the frequency of the oscillator loop so that the amplitude-to-phase conversion effect of the resonator reflections causes the synchronous detector output to go to zero volts and the oscillator loop to operate at a frequency f c =f Res +f Error To stay in V AM =-V Error Sufficient amplitude modulation is generated to provide

[0049] Here, a small phase error Δφ can be thought of as a "DC modulation" of the phase between the LO signal 858 and the RF signal 864 at the input to the Q mixer 842. Thus, the down-converted Q signal 872, which is a phase error voltage, carries a DC signal Vφ that is proportional to Δφ. It is important to note that Vφ arises from the phase difference between the carriers (incident and reflected) at the mixer, not from modulation sidebands. The sidebands are at frequencies f m Produces a varying signal at the Q output having

[0050] If Vφ is amplified and fed back with the correct sign to LO phase shifter 882 via LPF or amplifying integrator 890, the phase error Δφ is reduced, simultaneously tuning Q mixer 842 for pure phase sensitivity and I mixer 832 for pure amplitude sensitivity. If I mixer 832 loses phase sensitivity, the voltage V Error decreases, resulting in a frequency error f Error is reduced.

[0051] The fluctuating signal at the Q output due to phase modulation of the carrier is effectively eliminated from consideration by the low-pass filtering action of the integrator (or LPF). From the above discussion, it is clear that a simple low-pass filter and amplifier between the Q output 850 and the LO phase shifter 882 would serve to produce a similar result. However, due to the infinite DC gain provided by the integrator, the phase error Δφ is reduced to zero. The time constant of the integrator in the LO phase feedback loop can be long since it only needs to compensate for relatively slow thermal or aging effects, and therefore f mIt is extremely effective in removing modulated signals in

[0052] As a result, the downconverted I signal 870 has a frequency f m The demodulated signal has a magnitude and sign that represents the amplitude modulation of the reflected carrier. As such, the synchronous detector detects the frequency error f c -f Res , thereby reducing the frequency error in the oscillator loop towards zero.

[0053] With the Q mixer held in quadrature by the aforementioned LO phase servo, the Q mixer 842 acts as a wideband oscillator phase noise detector based on the reflection characteristics of the resonator. A portion 873 of the Q mixer output 872 is sampled and fed to a bandpass filter 874. The output of the bandpass filter is a wideband phase correction voltage 875 that is summed with the Pound servo integrator frequency error voltage 822 at summing junction 876. The combined voltage 877 has correction terms for both the center frequency error and the wideband phase noise and is applied to the tuned port of the oscillator. The low frequency cutoff f of the bandpass filter cLow is designed to minimize muting of the desired Pound modulation of the modulated signal 810, and f cLow >f m Without sufficient low-frequency roll-off, the BPF may degrade close-in phase noise suppression. The high-frequency cutoff of the bandpass filter, f cHigh is designed to allow a gain margin >0dB and a phase margin >0deg for closed loop stability. The BPF 874 is configured to provide sufficient gain in the passband to reach the noise floor of the IQ mixer.

[0054] The discussion so far has assumed that the independent I and Q mixers in an IQ mixer respond ideally as if the phase difference between their LO signals were exactly 90° (quadrature). This ideal condition may not be realized in practice, and some small quadrature error usually exists. As a result, even if the LO phase shifter is adjusted so that the Q mixer is perfectly phase sensitive, the I mixer will retain some slight sensitivity to phase modulation, which will cause a phase modulation-induced frequency error f in the Pound servo. PMError ≠0 occurs. Diode detectors are also slightly sensitive to phase modulation. Essentially, both diode detectors and AM sensitive mixers produce a carrier frequency f c The minimum detector bandwidth, and therefore the maximum f c The removal of the modulation frequency f m Therefore, both the diode and mixer amplitude detectors have the same fundamental limitations in rejecting phase-modulated signals and the same minimum f PMError is affected by.

[0055] 9, achieving this fundamental limit may require compensation for any small quadrature errors in the IQ mixer. This compensation is achieved by adding a small quadrature compensation voltage 900 via summing node 902 at the input to integrator 890, as shown in FIG. 9. The f due to quadrature errors in IQ mixer 830 Error Since the sign of πf reverses as the quadrature error passes through zero, it is straightforward to determine the magnitude of the required quadrature compensation voltage 900 by adjusting it to reduce the sensitivity of the demodulated signal to phase modulation in the reflected signal. If LO phase compensation is used, the output of the Q mixer should be sampled to provide signal 873 before compensation injection at summing node 902 so that changes in compensation voltage 900 are not injected into noise suppression voltage 875.

[0056] In either case, the carrier frequency f cThe requirement that f be kept constant does not necessarily PMError = 0, and PMError f = constant. PMError The stability of is another consideration and does not detract from the ability of the mixer amplitude detector to detect smaller frequency deviations in the Pound servo loop due to its lower noise floor.

[0057] Thus, it is demonstrated that the disclosed technique maintains the correct phase relationship between the LO and RF signals in the IQ mixer, with the Q output being maintained to be maximally sensitive to amplitude modulation and minimally sensitive to phase modulation, thereby enabling the practical use of a mixer instead of a diode as a detector in a Pound stabilization system and realizing the benefits of the mixer's lower noise floor.

[0058] As described, the IQ mixer can be implemented in any Pound-stabilized or Pound-servo frequency-stabilized microwave source. Figures 10-12 show three of the known Pound-stabilized microwave source topologies shown in Figures 1, 4, and 5, respectively.

[0059] 10, a frequency stabilized microwave source 1000 uses Pound stabilization techniques to stabilize a microwave oscillator, or VCO 1002, to a reference resonator 1004 separate from the VCO to generate a frequency stabilized output signal 1006 at an output port 1008. The VCO generates an output signal 1010, a portion of which is tapped off as the frequency stabilized output signal 1006. The signal frequency may be tuned by a Pound servo voltage 1012 applied to a tuning port 1014 of the VCO. The output signal 1010 passes through a phase shifter 1016 and a circulator 1018 before being incident on the reference resonator 1004. The frequency of the VCO 1002 is adjusted to a center frequency f ResTypically, the resonator will be a cavity resonator that may include a dielectric element, but the technique may be applied using any resonator with suitable center frequency and stability.

[0060] A portion of the output signal 1010 incident on the resonator is reflected according to known characteristics of the resonator as a reflected signal 1020. This reflected energy is directed by a circulator 1018 to an amplitude detector 1022, implemented as an I / Q mixer 1023. A modulation source 1024 generates a modulation frequency f m The phase modulation may result in an amplitude modulation in the reflected signal 1020 that is detected by the amplitude detector 1022. Any such amplitude modulation will result in a fluctuating signal at the detector output, the fluctuations occurring at the same modulation frequency f m The IQ mixer 1023 is configured to minimize or remove any contribution to the demodulated signal 1032 caused by phase modulation in the reflected signal 1020.

[0061] The IQ mixer 1023 receives the reflected signal 1020 at its RF input 1030 and outputs at its I output 1034 a modulation frequency f that represents the AM of the reflected carrier. m 1032 at LO input 1048. LO phase shifter 1042 receives a portion 1044 of signal 1010. A phase error voltage 1036 indicative of the phase difference between the incident and reflected carriers is provided at the Q output 1038 of the mixer, which is integrated 1040 and fed back as a nearly DC signal to LO phase shifter 1042, which modifies the effective phase shift through LO phase shifter 1042 to produce signal 1046 at the LO input 1048 of the mixer, maintaining the LO and RF inputs in quadrature at the Q mixer and therefore in phase at the I mixer.

[0062] The synchronous detector 1050 converts the demodulated signal 1032 into a DC frequency error voltage 1052 that represents the magnitude of the fluctuations. By keeping the inputs to the Q mixer in quadrature, the synchronous detector detects only any amplitude modulation of the reflected signal and therefore the frequency error f c -f Res , thereby reducing the frequency error in the oscillator loop towards zero. This DC voltage is then amplified and preferably integrated via integrator 1054 to provide a Pound servo gain and generate Pound servo voltage 1012. Samples 1056 of the Q mixer output 1036 are amplified and filtered by BPF 1058. The filtered output, wideband phase correction voltage 1060, is combined with Pound servo voltage 1012 at summing junction 1062. The combined output of summing junction 1062 is combined close-in and wideband error signal 1064 that is fed to tune port 1014 of VCO 1002.

[0063] Another embodiment of a frequency-stabilized microwave source 1100 is shown in FIG. 11 as a variation of the microwave source 1000 shown in FIG. 10, with like numbers used to identify like elements. The phase shifter 1016 is eliminated, and a modulation source 1024 is added to the VCO's tuned port 1014 via summing node 1102 along with the Pound servo voltage 1012 and the wideband phase correction voltage 1060 to cause frequency modulation of the VCO 1002. Because frequency modulation and phase modulation are simply related by a factor of 1 / jω, phase modulation of the carrier occurs with a 90° phase shift relative to the modulation source. However, this is easily accounted for by adjusting the relative phase of the signals at the synchronous detector 1050. The IQ mixer 1023 is configured and operates as previously described.

[0064] Referring now to FIG. 12, a frequency stabilized microwave source 1200 includes a reference resonator 1202 having a resonant frequency f Res12 also forms part of an oscillator loop 1204 that includes a bandpass filter (BPF) 1206 tuned to pass a reference resonator 1204 and a loop amplifier 1210, arranged to eliminate the need for a separate VCO. Figure 12, like Figure 11, achieves phase modulation of the signal incident on the reference resonator through frequency modulation.

[0065] Any phase change in the oscillator loop 1204 causes a frequency change. Therefore, frequency can be modulated by providing a modulation signal 1216 from a modulation source 1218 to any phase shifter in the oscillator loop 1204. The Pound servo feedback, phase noise suppression signal, and modulation signal can be applied to a common phase shifter or separate phase shifters located anywhere in the oscillator loop 1204. A common phase shifter can be used if it can provide performance for both frequency modulation and the tuning required by the Pound servo feedback. Separate phase shifters may be required if the phase shift range within which the required precision for modulation is achieved is insufficient to achieve the tuning required for the Pound servo feedback. As shown, a common phase shifter 1224 is located at the output of the loop amplifier 1210. The Pound servo voltage 1220, wideband phase correction voltage 1260, and modulation signal 1216 are summed by a summing node 1222 and input to the common phase shifter 1224.

[0066] Phase modulation induced through frequency modulation of signal 1212 can result in amplitude modulation in the reflected signal 1230 that is directed by circulator 1232 to amplitude detector 1234, implemented as an IQ mixer 1236, where the phase error voltage at the Q output is integrated 1238 and summed with a portion of signal 1212 by LO phase shifter 1240 and fed back to the LO input to keep the LO and RF inputs of the Q mixer in quadrature. Any such amplitude modulation causes a fluctuating signal at the detector output that is modulated at the same modulation frequency f mThe IQ mixer 1236 is configured to minimize or eliminate any contribution to the demodulated signal 1241 at the I output caused by phase modulation in the reflected signal 1230.

[0067] The synchronous detector 1242 detects the modulation frequency f m converts variations in the I output signal at c -f Res , thereby reducing the frequency error in the oscillator loop towards zero. This DC voltage is then amplified and preferably integrated 1246 to produce the Pound servo voltage 1220 that is fed to the summing node 1222.

[0068] A portion 1262 of the Q output of the IQ mixer 1234 is sampled and provided to a bandpass filter 1264 , which bandpass filters and amplifies the signal to produce a wideband phase correction voltage 1260 that is provided to the summing node 1222 .

[0069] The frequency stabilized output signal 1250 is taken from a different output 1252 located after the reference resonator 1202 in the oscillator loop 1204, which typically results in a reduction of noise in the oscillator output due to the filtering action of the reference resonator 1202. However, the output may equally well be taken from before the reference resonator without affecting the functionality of the microwave source.

[0070] While several exemplary embodiments of the present disclosure have been shown and described, numerous variations and alternative embodiments will occur to those skilled in the art. Such variations and alternative embodiments are contemplated and can be made without departing from the scope of the present disclosure, as defined in the appended claims.

Claims

1. 1. A frequency stabilized microwave source comprising: Microwave carrier frequency f ｃ configured to generate a signal including a carrier at frequency f ｃ an oscillator having a tuning port responsive to a tuning voltage for controlling A Pound stabilization loop comprising: Modulation frequency f ｍ a modulation source configured to generate a modulated signal at f ｃ +f ｍ and f ｃ -f ｍ a modulation source configured to add upper and lower sidebands in center frequency f Ｒｅｓ a reference resonator configured to receive at least a portion of the signal and to generate a reflected carrier and reflected upper and lower sidebands, the portions of the reflected upper and lower sidebands having a modulation frequency f ｍ represents the amplitude modulation (AM) of the reflected carrier at ｃ -f Ｒｅｓ a reference resonator, which is an indicator of Modulation frequency f ｍ an AM detector responsive to the reflected signal to generate a demodulated signal representative of the amplitude modulation of the reflected carrier at The frequency error f ｃ -f Ｒｅｓ a synchronous detector that mixes the demodulated signal with the modulated signal to generate a frequency error voltage indicative of an integrator that integrates the frequency error voltage to generate a pound servo voltage that controls the tuning voltage to reduce the frequency error toward zero; the AM detector comprises an IQ mixer having LO and RF input ports and I and Q output ports, the RF input port receiving the reflected signal and the demodulated signal being produced at the I output port; the IQ mixer includes a Pound stabilization loop that generates a phase error voltage at the Q output port that is indicative of the phase difference between the carrier and the reflected carrier; an LO phase shifter responsive to the phase error voltage for sampling the signal and generating a phase shifted signal; The phase shifted signal is generated such that the demodulated signal produced at the I output port is insensitive to phase modulation in the reflected signal and is at a carrier frequency f ｃ an LO phase shifter applied to the LO input port of the IQ mixer to suppress close-in phase noise; a bandpass filter (BPF) coupled to the Q output of the IQ mixer for filtering the phase error voltage to produce a wideband phase correction voltage that is added to the Pound servo voltage to suppress wideband phase noise.

2. The adjacent phase is the carrier frequency f ｃ and crossover frequency f ｃｒｏｓｓ－ｏｖｅｒ and the wideband phase noise is between the crossover frequency f ｃｒｏｓｓ－ｏｖｅｒ 2. The frequency stabilized microwave source of claim 1, wherein the frequency stabilized microwave source is located above the

3. To suppress the wideband phase noise without degrading the close-in phase noise, the BPF has a low frequency cutoff f designed to minimize muting of the modulated signal. ｃＬｏｗ >f ｍ , a high frequency cutoff f designed to enable a gain margin >0 dB and a phase margin >0 deg for closed loop stability ｃＨｉｇｈ 10. The frequency stabilized microwave source of claim 1, characterized by: a low frequency cutoff and a high frequency cutoff; and sufficient gain in a passband between the low frequency cutoff and the high frequency cutoff to reach the noise floor of the IQ mixer.

4. 10. The frequency stabilized microwave source of claim 1, further comprising a low pass filter (LPF) or integrator between the Q output port and the LO phase shifter.

5. a summing node at the input to the LPF or integrator, the summing node configured to sum the phase error voltage at the Q output port and a quadrature compensation voltage having a magnitude configured to reduce the sensitivity of the demodulated signal to phase modulation in the reflected signal; 5. The frequency stabilized microwave source of claim 4, wherein the band-pass filter samples the phase error voltage at the Q output port before the summing node.

6. The IQ mixer comprises: an I mixer having an LO input port, an RF input port, and said I output port; a Q mixer having an LO input port, an RF input port, and the Q output port; a first microwave power divider between an LO input port of the IQ mixer and the LO input ports of the I and Q mixers, which receives the phase-shifted signals and generates first and second LO signals; a second microwave power divider between the RF input port of the IQ mixer and the RF input ports of the I and Q mixers, the second microwave power divider receiving the reflected signal and generating first and second RF signals having the same phase; the first and second microwave power dividers are arranged to generate first and second relative phases between the first LO signal and the first RF signal, and between the second LO signal and the second RF signal, respectively, wherein a difference between the first relative phase and the second relative phase is 90°; 2. The frequency stabilized microwave source of claim 1, wherein the phase-shifted signals are applied to the LO input port of the IQ mixer such that the second LO signal and the second RF signal at the LO input port and RF input port of the Q mixer, respectively, remain in quadrature and the demodulated signal produced at the I output port is insensitive to phase modulation in the reflected signal.

7. 5Hz<f ｍ 2. The frequency stabilized microwave source of claim 1, wherein the frequency is <5 KHz.

8. 2. The frequency stabilized microwave source of claim 1, wherein the oscillator comprises a voltage controlled oscillator (VCO) having the tuning port that generates the signal, and the reference resonator is separate from the VCO.

9. 9. The frequency stabilized microwave source of claim 8, further comprising a phase shifter at an output of the VCO, the phase shifter responsive to the modulating signal to apply a phase modulation to the signal to add the upper and lower sidebands.

10. A summing node at the tuning port of the VCO, ｃ 9. The frequency stabilized microwave source of claim 8, further comprising a summing node responsive to the modulation signal and the frequency error voltage for frequency modulating a

11. The oscillator includes the reference resonator as a frequency determining element in an oscillator loop, and further comprises a first phase shifter in the oscillator loop, the first phase shifter shifting the frequency f ｃ 2. The frequency stabilized microwave source of claim 1, wherein the tuning port is responsive to the tuning voltage to control

12. 12. The frequency stabilized microwave source of claim 11, further comprising: a second phase shifter in the oscillator loop responsive to the modulating signal to apply a phase modulation to the signal to add the upper and lower sidebands.

13. 13. The frequency stabilized microwave source of claim 12, wherein the first and second phase shifters are a common phase shifter, and further comprising a summing node that sums the modulation signal and the frequency error voltage.

14. center frequency f Ｒｅｓ is a reference resonator having a carrier frequency f ｃ The carrier wave and modulation frequency f of the modulating signal ｍ of ｃ +f ｍ and f ｃ -f ｍ 1. An amplitude modulation (AM) and phase modulation (PM) detector for a Pound stabilized microwave source configured to reflect a signal including upper and lower sidebands of:

1. An IQ mixer having LO and RF input ports and I and Q output ports, The RF input port is configured to receive a reflected signal including a reflected carrier and reflected upper and lower sidebands, a portion of the reflected upper and lower sidebands being modulated at a modulation frequency f ｍ represents the AM of the reflected carrier at ｃ -f Ｒｅｓ is an indicator of The IQ mixer outputs a modulation frequency f ｍ and configured to generate a demodulated signal representative of the amplitude modulation of the reflected carrier at an IQ mixer configured to generate a phase error voltage at the Q output port indicative of a phase difference between the carrier and the reflected carrier; an LO phase shifter responsive to the phase error voltage for sampling the signal and generating a phase shifted signal; an LO phase shifter, wherein the phase-shifted signal is applied to the LO input port of the IQ mixer such that the demodulated signal produced at the I output port is insensitive to phase modulation in the reflected signal; The frequency error f ｃ -f Ｒｅｓ a synchronous detector that mixes the demodulated signal with the modulated signal to generate a frequency error voltage indicative of an integrator for integrating the frequency error voltage to generate a Pound servo voltage; a bandpass filter (BPF) coupled to the Q output of the IQ mixer for filtering the phase error voltage to produce a wideband phase correction voltage; In the Pound stabilized microwave source, the carrier frequency f ｃ The adjacent phase noise is suppressed, and the carrier frequency f ｃ a summing node that sums the Pound servo voltage and the wideband phase correction voltage to suppress upper wideband phase noise; 1. An AM and PM detector comprising:

15. The adjacent phase is the carrier frequency f ｃ and crossover frequency f ｃｒｏｓｓ－ｏｖｅｒ and the wideband phase noise is between the crossover frequency f ｃｒｏｓｓ－ｏｖｅｒ 15. The AM and PM detector of claim 14, wherein the AM and PM detector is located above the

16. To suppress the wideband phase noise without degrading the close-in phase noise, the BPF has a low frequency cutoff f designed to minimize muting of the modulated signal. ｃＬｏｗ >f ｍ , a high frequency cutoff f designed to enable a gain margin >0 dB and a phase margin >0 deg for closed loop stability ｃＨｉｇｈ and sufficient gain in a passband between the low frequency cutoff and the high frequency cutoff to reach the noise floor of the IQ mixer.

17. 15. The AM and PM detector of claim 14, further comprising a low pass filter (LPF) or integrator between the Q output port and the LO phase shifter.

18. a summing node at the input to the LPF or integrator configured to sum the phase error voltage at the Q output port and a quadrature compensation voltage having a magnitude configured to reduce the sensitivity of the demodulated signal to phase modulation in the reflected signal; 18. The AM and PM detector of claim 17, wherein the band-pass filter samples the phase error voltage at the Q output port before the summing node.

19. The IQ mixer comprises: an I mixer having an LO input port, an RF input port, and said I output port; a Q mixer having an LO input port, an RF input port, and the Q output port; a first microwave power divider between an LO input port of the IQ mixer and the LO input ports of the I and Q mixers, which receives the phase-shifted signals and generates first and second LO signals; a second microwave power divider between the RF input port of the IQ mixer and the RF input ports of the I and Q mixers, the second microwave power divider receiving the reflected signal and generating first and second RF signals; the first and second microwave power dividers are arranged to generate first and second relative phases between the first LO signal and the first RF signal and between the second LO signal and the second RF signal, respectively, wherein a difference between the first relative phase and the second relative phase is 90°; 15. The AM and PM detector of claim 14, wherein the phase-shifted signals are applied to the LO input port of the IQ mixer such that the second LO signal and the second RF signal at the LO input port and RF input port of the Q mixer, respectively, remain in quadrature and the demodulated signal produced at the I output port is insensitive to phase modulation in the reflected signal.

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