FILTER DEVICE, ANTENNA DEVICE, AND ANTENNA MODULE

Abstract

Provided is a filter device with which good characteristics can be obtained even when an attenuation band achieved by parallel resonance and a pass band achieved by series resonance are brought close to each other. A filter device (100) according to the present disclosure has a pass band of an f1 band, and an attenuation band of an f2 band. The filter device (100) comprises: a terminal (P1); a terminal (P2); an inductor (L1) connected to the terminal (P1); and an LC series resonator (RS) that includes a capacitor (C1) and an inductor (L2), disposed in a first path (TL1) among the first path (TL1) and a second path (TL2) that are provided in parallel between the inductor (L1) and the terminal (P2). The inductor (L1) and the inductor (L2) are magnetically coupled to each other.

Description

[Technical field]

[0001] The present disclosure relates to a filter device, an antenna device, and an antenna module, and more particularly to a technique for improving attenuation and transmission characteristics. [Background technology]

[0002] High-frequency circuits are provided with filter devices such as band-stop filters and band-pass filters. As an example of a filter device provided in a high-frequency circuit, a filter device is disclosed in Japanese Patent No. 6531824 (Patent Document 1). The filter device includes a first inductor and a first capacitor that form a first series circuit, and a second inductor that is connected in parallel to the first series circuit. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Patent No. 6531824 Summary of the Invention [Problem to be solved by the invention]

[0004] However, in the filter device disclosed in Patent Gazette No. 6531824 (Patent Document 1), when the attenuation band due to parallel resonance and the pass band due to series resonance are brought close to each other, it is difficult to maintain both the attenuation characteristics and the pass characteristics at high levels.

[0005] The present disclosure has been made to solve such problems, and its purpose is to provide a filter device that can obtain good characteristics even when the attenuation band due to parallel resonance and the pass band due to series resonance are placed close to each other. [Means for solving the problem]

[0006] A filter device according to the present disclosure has a pass band in a first frequency band and an attenuation band in a second frequency band lower than the first frequency band. The filter device includes a first terminal, a second terminal, a first inductor connected to the first terminal, and a series resonator including a first capacitor and a second inductor arranged in the first path of a first path and a second path provided in parallel between the first inductor and the second terminal. The first inductor and the second inductor are magnetically coupled to each other. Effect of the Invention

[0007] In the filter device according to the present disclosure, a series resonator is disposed in the first path of the first and second paths provided in parallel between the first inductor and the second terminal, and the first and second inductors are magnetically coupled to each other. With this configuration, the filter device according to the present disclosure can obtain high attenuation and pass characteristics even when the attenuation band due to parallel resonance and the pass band due to series resonance are close to each other. [Brief description of the drawings]

[0008] [Figure 1] 1 is a circuit diagram of a filter device according to a first embodiment. [Diagram 2] 1 is a diagram illustrating a configuration of an antenna device according to a first embodiment. [Diagram 3] 5 is a diagram illustrating the reactance characteristics of the filter device according to the first embodiment. FIG. [Figure 4] 2 is an equivalent circuit diagram of the filter device according to the first embodiment. [Diagram 5] 5 is a diagram illustrating an example of an insertion loss of the filter device according to the first embodiment. FIG. [Figure 6] 4 is a diagram illustrating an example of reactance characteristics of the filter device according to the first embodiment. FIG. [Figure 7] 5 is a diagram illustrating an example of reactance characteristics when a coupling coefficient is changed in the filter device according to the first embodiment. FIG. [Figure 8]1 is a perspective view of a filter device according to a first embodiment. [Figure 9] 1 is an exploded plan view showing a configuration of a filter device according to a first embodiment. [Figure 10] 2 is an exploded plan view showing a configuration in which the winding direction of inductor L1 and the winding direction of inductor L2 are opposite to each other in the filter device according to the first embodiment. FIG. [Figure 11] 11 is a diagram showing a configuration of an antenna module according to a second embodiment. FIG. [Figure 12] FIG. 11 is a diagram illustrating isolation characteristics between antenna devices according to the second embodiment. [Figure 13] 10 is a diagram showing the radiation efficiency of each antenna device in the second embodiment. FIG. [Figure 14] 11 is an external view of an antenna module according to a second embodiment. FIG. [Figure 15] FIG. 11 is a circuit diagram of a filter device according to a third embodiment. [Figure 16] FIG. 11 is a schematic diagram of a filter device according to a third embodiment. [Figure 17] 13 is a diagram illustrating an example of insertion loss and reactance characteristics of the filter device according to the third embodiment. [Figure 18] FIG. 11 is a circuit diagram of a filter device according to a fourth embodiment. [Figure 19] FIG. 11 is a schematic diagram of a filter device according to a fourth embodiment. [Figure 20] FIG. 13 is a circuit diagram of a filter device according to a fifth embodiment. [Figure 21] 13 is a diagram showing an example of an insertion loss of the filter device according to the fifth embodiment. FIG. [Figure 22] FIG. 13 is a diagram illustrating an example of reactance characteristics of the filter device according to the fifth embodiment. [Diagram 23] FIG. 13 is a circuit diagram of a filter device according to a sixth embodiment. [Figure 24] FIG. 23 is a diagram showing an example of an insertion loss of the filter device according to the sixth embodiment. [Diagram 25]FIG. 23 is a diagram illustrating an example of reactance characteristics of the filter device according to the sixth embodiment. [Figure 26] FIG. 11 is a circuit diagram of a filter device according to a modified example. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0009] Hereinafter, embodiments of the present disclosure will be described in detail with reference to the drawings. In the drawings, the same or corresponding parts are designated by the same reference characters and their description will not be repeated.

[0010] [Embodiment 1] <Basic configuration of filter device and antenna device> Fig. 1 is a circuit diagram of a filter device 100 according to the first embodiment. Fig. 2 is a diagram showing a configuration of an antenna device 150 according to the first embodiment. The filter device 100 is a trap filter that is used in the antenna device 150 to prevent and attenuate high-frequency signals in a specific frequency band from passing through. The filter device 100 is also called a band elimination filter.

[0011] The antenna device 150 includes a feeding circuit RF1, a filter device 100, and an antenna 155. The antenna device 150 is mounted on a communication device such as a mobile terminal such as a mobile phone, a smartphone, or a tablet, or a personal computer with a communication function.

[0012] The power feed circuit RF1 supplies a high-frequency signal in the f1 frequency band to the antenna 155. The antenna 155 is, for example, a monopole antenna, and can radiate the f1 frequency signal supplied from the power feed circuit RF1 into the air as radio waves. The f1 frequency band is, for example, n41 (2.5-2.7 GHz).

[0013] The filter device 100 is useful when the antenna device 150 is used near an antenna in the 2.4 GHz band (2.4-2.5 GHz) of Wi-Fi (registered trademark). The filter device 100 is configured to attenuate high-frequency signals in the 2.4 GHz frequency band (f2 band) and pass high-frequency signals in the f1 frequency band. FIG. 3 is a diagram for explaining the reactance characteristics of the filter device 100 in the first embodiment. As shown in FIG. 3, the filter device 100 has an attenuation band due to parallel resonance in the f2 frequency band and a pass band due to series resonance in the f1 frequency band.

[0014] As shown in Fig. 3, the f1 band and the f2 band are adjacent frequency bands. Whether or not the frequency bands are adjacent can be determined, for example, by using a bandwidth and a center frequency for that bandwidth. For example, if the ratio of the bandwidth between the frequency end of the f1 band and the frequency end of the f2 band and the center frequency for that bandwidth is within a predetermined range, the f1 band and the f2 band are determined to be adjacent. Note that whether or not the frequency bands are adjacent may be determined by other methods.

[0015] 2 has a terminal P1 and a terminal P2. The terminal P1 is a terminal for connecting the filter device 100 to a transmission line on the power feed circuit RF1 side. The terminal P2 is a terminal for connecting the filter device 100 to a transmission line on the antenna 155 side.

[0016] When the power feed circuit RF1 supplies a high-frequency signal to the antenna 155 via the filter device 100, the terminal P1 serves as an input terminal and the terminal P2 serves as an output terminal. When a high-frequency signal received by the antenna 155 is transmitted to a circuit on the power feed circuit RF1 side via the filter device 100, the terminal P1 serves as an output terminal and the terminal P2 serves as an input terminal.

[0017] 1, the filter device 100 includes an inductor L1, an inductor L2, and a capacitor C1. A first path TL1 and a second path TL2 are provided between the inductor L1 and a terminal P2. The first path TL1 is provided with an LC series resonator RS in which the inductor L2 and the capacitor C1 are connected in series. The second path TL2 is a short path.

[0018] The inductor L1 and the inductor L2 are magnetically coupled to each other. As a result, a mutual inductance M occurs between the inductor L1 and the inductor L2. Due to the mutual inductance M, an inductance occurs in each of the first path TL1 and the second path TL2, thereby forming a parallel resonator. FIG 4 is an equivalent circuit diagram of the filter device 100 according to the first embodiment.

[0019] The circuit diagram shown in Fig. 4(a) illustrates the circuit of the filter device 100 when the coils constituting the inductor L1 and the inductor L2 have the same winding direction. The equivalent circuit diagram shown in Fig. 4(b) illustrates the equivalent circuit of the filter device 100 shown in Fig. 4(a), and shows a mutual inductance +M in the first path TL1 and a mutual inductance -M in the second path TL2.

[0020] Here, the series resonant frequency of the LC series resonator RS is f0 = 1 / (2π(L2 × C1) 1 / 2 ) At this series resonance frequency f0, the LC series resonator RS has a combined reactance X of the inductor L2 and the capacitor C1 that is 0 (zero) (X=0). Therefore, at the series resonance frequency f0 where the combined reactance X of the LC series resonator RS is 0 (zero), the filter device 100 functions as a parallel resonator with mutual inductances -M and +M. The resonance frequency of this parallel resonator coincides with the series resonance frequency f0 of the LC series resonator RS, and is the parallel resonance frequency of the attenuation band (f2 band) of the filter device 100.

[0021] In conventional filter devices, all components such as inductors and capacitors affect the parallel resonance frequency in the attenuation band. Therefore, in conventional filter devices, it was necessary to take all components into consideration when designing the parallel resonance frequency in the attenuation band. However, in filter device 100, it is possible to design the parallel resonance frequency in the attenuation band (f2 band) simply by considering inductor L2 and capacitor C1 that configure LC series resonator RS. Therefore, filter device 100 has an extremely advantageous configuration in terms of structural design.

[0022] Specifically, the filter device 100 was simulated with the inductor L1 set to 1.0 nH, the inductor L2 set to 2.0 nH, the capacitor C1 set to 2.2 pF, and the coupling coefficient K set to 0.5. When the inductor L2 is set to 2.0 nH and the capacitor C1 set to 2.2 pF, the series resonance frequency f0 of the LC series resonator RS is calculated to be 2.4 GHz, which is equal to the parallel resonance frequency (center frequency) of the attenuation band (f2 band) of the filter device 100, which is 2.4 GHz. The series resonance frequency (center frequency) of the pass band (f1 band) of the filter device 100 is 2.77 GHz. In the filter device 100, it is preferable that the inductance of the inductor L1 is smaller than the inductance of the inductor L2. This can reduce the loss of the entire filter device 100.

[0023] Fig. 5 is a diagram showing an example of the insertion loss of filter device 100 according to the first embodiment. In Fig. 5, the horizontal axis represents frequency, and the vertical axis represents insertion loss. Fig. 6 is a diagram showing an example of the reactance characteristic of filter device 100 according to the first embodiment. In Fig. 6, the horizontal axis represents frequency, and the vertical axis represents reactance. Here, the insertion loss is the ratio of the power output to the power input to filter device 100.

[0024] 5, in addition to the line Ln1 indicating the insertion loss of the filter device 100, a line Ln2 indicating the insertion loss of a comparative filter device is shown. Although not shown, the comparative filter device has a configuration in which an inductor Lb is connected in parallel to an LC series resonator constituted by an inductor La and a capacitor Ca. The comparative filter device was simulated with an inductor Lb of 0.069 nH, an inductor La of 42.19 nH, a capacitor Ca of 0.1 pF, and a coupling coefficient K2 between the inductor Lb and the inductor La of 0.5. In the comparative filter device as well, the parallel resonance frequency (center frequency) of the attenuation band (f2 band) is 2.4 GHz, and the series resonance frequency (center frequency) of the pass band (f1 band) is 2.77 GHz.

[0025] 5 indicates the position of the parallel resonance frequency (center frequency) of 2.4 GHz, and the insertion loss of the line Ln1 at the mark m1 is 16.6 dB, whereas the insertion loss of the line Ln2 is 2.75 dB. Therefore, the filter device 100 has a sufficient attenuation characteristic in the attenuation band (f2 band), whereas the comparative filter device does not have a sufficient attenuation characteristic.

[0026] 5 indicates the position of the series resonance frequency (center frequency) of 2.77 GHz, and the insertion loss of the line Ln1 at the mark m2 is 0.068 dB, whereas the insertion loss of the line Ln2 is 0.404 dB. Therefore, the filter device 100 has higher pass characteristics in the pass band (f1 band) than the comparative filter device.

[0027] In Fig. 6, in addition to the line Ln3 showing the reactance characteristics of the filter device 100, a line Ln4 showing the reactance characteristics of a comparative filter device is shown. Mark m3 shown in Fig. 6 indicates the position of the parallel resonance frequency (center frequency) of 2.4 GHz, and the reactance of the line Ln3 at the mark m3 changes significantly compared to the reactance of the line Ln4. Therefore, the filter device 100 has sufficient attenuation characteristics in the attenuation band (f2 band), but the comparative filter device does not have sufficient attenuation characteristics.

[0028] 6 indicates the position of the series resonance frequency (center frequency) of 2.77 GHz, and the reactance of the line Ln3 at the mark m4 is approximately 0. Therefore, the filter device 100 has sufficient pass characteristics in the pass band (f1 band).

[0029] In the filter device 100, when the attenuation band (f2 band) due to parallel resonance and the pass band (f1 band) due to series resonance are brought close to each other, sufficient attenuation characteristics and pass characteristics are obtained as shown in Figs. 5 and 6. On the other hand, in the comparative filter device, when the attenuation band (f2 band) due to parallel resonance and the pass band (f1 band) due to series resonance are brought close to each other, sufficient attenuation characteristics and pass characteristics are not obtained as shown in Figs. 5 and 6. Furthermore, in the comparative filter device, the inductor La is large at 42.19 nH, while the inductor Lb must be extremely small at 0.069 nH in order to bring the attenuation band (f2 band) due to parallel resonance and the pass band (f1 band) due to series resonance close to each other. Therefore, it is difficult to actually realize a configuration in which the coupling coefficient K2 between the inductors Lb and La is 0.5.

[0030] As described above, in the filter device 100, the parallel resonance frequency of the attenuation band (f2 band) is determined by the inductor L2 and the capacitor C1 that configure the LC series resonator RS. Therefore, the filter device 100 can change the series resonance frequency of the pass band (f1 band) by changing the coupling coefficient between the inductor L1 and the inductor L2, and can make the series resonance frequency closer to the attenuation band (f2 band) due to parallel resonance than to the pass band (f1 band) due to series resonance. In other words, the filter device 100 can realize a narrow-band filter device in which the attenuation characteristics change sharply in the vicinity of the parallel resonance frequency of the attenuation band (f2 band).

[0031] Fig. 7 is a diagram showing an example of the reactance characteristic when the coupling coefficient K is changed in the filter device 100 according to the first embodiment. In Fig. 7, the horizontal axis represents frequency, and the vertical axis represents reactance.

[0032] 7, in addition to the line Ln3 which shows the reactance characteristics of the filter device 100 with the coupling coefficient K being 0.5, a line Ln5 which shows the reactance characteristics of the filter device 100 with the coupling coefficient K being 0.3 is shown. Note that the filter device 100 was simulated with the inductor L1 set to 1.0 nH, the inductor L2 set to 2.0 nH, and the capacitor C1 set to 2.2 pF, all other than the coupling coefficient K.

[0033] At the mark m3 shown in FIG. 7, the reactance of the line Ln5 changes sharply from the reactance of the line Ln3. When the coupling coefficient K=0.5, the reactance of the line Ln3 is approximately 0 (zero) Ω at the mark m4, and the series resonance frequency of the pass band (f1 band) is 2.77 GHz. On the other hand, when the coupling coefficient K=0.3, the reactance of the line Ln4 is approximately 0 (zero) Ω at the mark m5, and the series resonance frequency of the pass band (f1 band) is 2.51 GHz, which is closer to the parallel resonance frequency (center frequency) of 2.4 GHz. In other words, the filter device 100 can make the series resonance frequency (center frequency) closer to the parallel resonance frequency (center frequency) by reducing the coupling coefficient K. Note that, since the mutual inductance M itself is also reduced when the coupling coefficient K is reduced, the coupling coefficient K in the filter device 100 is preferably 0.1 or more.

[0034] <An example of an element that integrates a filter device> Next, an example in which the filter device 100 according to the first embodiment is formed as an integrated element will be described with reference to the drawings. Fig. 8 is a perspective view of the filter device according to the first embodiment. Fig. 9 is an exploded plan view showing the configuration of the filter device 100 according to the first embodiment.

[0035] The filter device 100 is formed integrally as, for example, a chip component, and includes an inductor L1 and an LC series resonator RS shown in FIG. 1 in an insulator 1 (housing) in which dielectric layers are laminated, and external electrodes 2a to 2d are formed on the outside of the insulator 1. The terminal P1 is connected to the external electrode 2a (first external electrode), and the terminal P2 is connected to the external electrode 2b (second external electrode). The filter device 100 has a short side direction in the X direction, a long side direction in the Y direction, and a height direction in the Z direction, and the lamination direction of the dielectric layers is the Z direction. The external electrodes 2c and 2d are GND electrodes that are not connected to an internal circuit. Furthermore, the filter device 100 shown in FIG. 8 has a four-terminal configuration in which the external electrodes 2a to 2d are formed on the outside of the insulator 1, but may have a two-terminal configuration in which only the external electrodes 2a and 2b are formed on the outside of the insulator 1.

[0036] The filter device 100 is formed by a lamination process, and is formed by stacking a plurality of substrates of dielectric layers Ly1 to Ly9 (hereinafter simply referred to as dielectric layers Ly1 to Ly9) shown in Fig. 9. Each of the dielectric layers Ly1 to Ly9 is a ceramic green sheet, and a wiring pattern is formed by printing a conductive paste (for example, Ni paste) by a screen printing method.

[0037] A wiring pattern r1 that constitutes a part of the inductor L1 is formed on the dielectric layer Ly1. One end of the wiring pattern r1 is connected to the terminal P1, and the other end is connected to the via conductor h1a.

[0038] The dielectric layer Ly2 has a wiring pattern r2a that constitutes a part of the inductor L1, with one end of the wiring pattern r2a connected to the via conductor h1a and the other end connected to the wiring pattern r2b and the wiring pattern r2c of the second path TL2. The wiring pattern r2b constitutes a part of the inductor L2, with the via conductor h2a connected to the end opposite to the wiring pattern r2a. The wiring pattern r2c of the second path TL2 has the via conductor h2b connected to the end opposite to the wiring pattern r2a.

[0039] A wiring pattern r3 constituting a part of the inductor L2 is formed on the dielectric layer Ly3, and one end of the wiring pattern r3 is connected to the via conductor h2a and the other end is connected to the via conductor h3a. A via conductor h3b connected to the via conductor h2b is provided on the dielectric layer Ly3.

[0040] A wiring pattern r4 constituting a part of the inductor L2 is formed on the dielectric layer Ly4, and one end of the wiring pattern r4 is connected to the via conductor h3a and the other end is connected to the via conductor h4a. A via conductor h4b connected to the via conductor h3b is provided on the dielectric layer Ly4.

[0041] A wiring pattern r5 constituting a part of the inductor L2 is formed on the dielectric layer Ly5, and one end of the wiring pattern r5 is connected to the via conductor h4a and the other end is connected to the via conductor h5a. A via conductor h5b connected to the via conductor h4b is provided on the dielectric layer Ly5.

[0042] An electrode pattern p1 constituting a part of the capacitor C1 is formed on the dielectric layer Ly6 at a position not overlapping with the inductors L1 and L2 when viewed from the lamination direction. The electrode pattern p1 is connected to the terminal P2 and also to a via conductor h6b. The via conductor h6b is connected to the via conductor h5b, and electrically connects the electrode pattern p1 to the wiring pattern r2c of the second path TL2. The dielectric layer Ly6 is provided with a via conductor h6a that connects to the via conductor h5a.

[0043] An electrode pattern p2 constituting a part of the capacitor C1 is formed on the dielectric layer Ly7 at a position not overlapping with the inductors L1 and L2 when viewed from the lamination direction. The electrode pattern p2 is connected to the via conductor h7a and electrically connected to the inductor L2, but is not directly electrically connected to the electrode pattern p1. The dielectric layer Ly7 is provided with a via conductor h7b connected to the via conductor h6b.

[0044] An electrode pattern p3 constituting a part of the capacitor C1 is formed on the dielectric layer Ly8 at a position not overlapping with the inductors L1 and L2 when viewed from the stacking direction. The electrode pattern p3 is connected to the terminal P2 and also to the via conductor h7b. The electrode pattern p1 and the electrode pattern p3 are electrically connected through the via conductor h7b. The dielectric layer Ly8 is provided with a via conductor h8 that is connected to the via conductor h7a.

[0045] An electrode pattern p4 constituting a part of the capacitor C1 is formed on the dielectric layer Ly9 at a position not overlapping with the inductors L1 and L2 when viewed from the lamination direction. The electrode pattern p4 is connected to the via conductor h8 and electrically connected to the electrode pattern p2, but is not directly electrically connected to the electrode patterns p1 and p3.

[0046] The wiring pattern r1 formed on the dielectric layer Ly1 and the wiring pattern r2a formed on the dielectric layer Ly2 form a winding shape when viewed from the stacking direction, constituting the inductor L1. The wiring pattern r2b formed on the dielectric layer Ly2 and the wiring patterns r3 to r5 formed on the dielectric layers Ly3 to Ly5 form a winding shape when viewed from the stacking direction, constituting the inductor L2. The inductors L1 and L2 are disposed to face each other, and at least a part of the opening of the inductor L1 overlaps with the opening of the inductor L2 when viewed from the stacking direction. Therefore, if the overlapping part of the opening of the inductor L1 and the opening of the inductor L2 is increased when viewed from the stacking direction, the coupling coefficient between the inductors L1 and L2 increases, and the mutual inductance M due to magnetic coupling increases.

[0047] 9, the filter device 100 is stacked in the order of inductor L1, inductor L2, and capacitor C1 when viewed from the stacking direction, but may be stacked in another order, such as inductor L2, inductor L1, and capacitor C1. Note that in FIG. 9, by switching the stacking order of inductor L1 and inductor L2 and arranging capacitor C1 on the inductor L1 side, the number of via conductors h2b to h5b that form part of the second path TL2 can be reduced, and the length of the second path TL2 can be shortened.

[0048] 1, which is a short path, is a path that connects between inductor L1 and inductor L2 to capacitor C1, and it is preferable that the equivalent series inductance ESL (equivalent series inductance) that is a parasitic inductance generated in this path is smaller than the mutual inductance M. In other words, by making the inductance of the second path TL2 smaller than the mutual inductance M between inductor L1 and inductor L2, the second path TL2 can be regarded as a short path.

[0049] 9, the second path TL2 is composed of via conductors h2b-h5b that connect between the laminated layers and a wiring pattern r2c. The second path TL2, which is a short path, may include some resistance component (R component). However, by reducing the resistance component (R component), the Q value of the filter device 100 can be improved.

[0050] If the filter device 100 is formed by a lamination process, the dielectric materials can be changed between the inductors L1, L2 and the capacitor C1, and to achieve this, it is necessary to separate the layers (dielectric layers Ly1-Ly5) on which the inductors L1, L2 are formed from the layers (dielectric layers Ly6-Ly9) on which the capacitor C1 is formed, as shown in Fig. 9. On the other hand, if the filter device 100 is formed by photolithography, the capacitor C1 can be formed side-by-side with the inductor L1 or inductor L2 without separating the layers on which the inductors L1, L2 are formed from the layers on which the capacitor C1 is formed.

[0051] 9, the wiring patterns r1, r2a, r2b, r3 to r5 are formed so that the winding direction of the inductor L1 is the same as that of the inductor L2. Therefore, the filter device 100 has a structure that makes it easy to increase the coupling coefficient between the inductor L1 and the inductor L2.

[0052] However, the filter device 100 is not limited to the case where the winding directions of the inductor L1 and the inductor L2 are the same, and the winding directions of the inductor L1 and the inductor L2 may be reversed. Fig. 10 is an exploded plan view showing a configuration in which the winding directions of the inductor L1 and the inductor L2 are reversed in the filter device 100 according to the first embodiment. Note that the filter device 100 shown in Fig. 10 is stacked in the order of the inductor L2, the inductor L1, and the capacitor C1 when viewed from the stacking direction.

[0053] A wiring pattern r1 that constitutes a part of the inductor L2 is formed on the dielectric layer Ly1. One end of the wiring pattern r1 is connected to the via conductor h1, and the other end is connected to the via conductor h2a of the dielectric layer Ly2.

[0054] A wiring pattern r2 constituting a part of the inductor L2 is formed on the dielectric layer Ly2, and one end of the wiring pattern r2 is connected to a via conductor h2a and the other end is connected to a via conductor h3a of the dielectric layer Ly3. A via conductor h2b that is connected to the via conductor h1 is provided on the dielectric layer Ly2.

[0055] A wiring pattern r3 constituting a part of the inductor L2 is formed on the dielectric layer Ly3, and one end of the wiring pattern r3 is connected to a via conductor h3a and the other end is connected to a via conductor h4a of the dielectric layer Ly4. A via conductor h3b that is connected to the via conductor h2b is provided on the dielectric layer Ly3.

[0056] The dielectric layer Ly4 is provided with a wiring pattern r4a constituting a part of the inductor L1, with one end of the wiring pattern r4a being connected to the via conductor h5a of the dielectric layer Ly4 and the other end being connected to the via conductor h4a and the wiring pattern r4b of the second path TL2. The wiring pattern r4b of the second path TL2 is connected to a via conductor h4c at an end opposite to the end connected to the wiring pattern r4a. The dielectric layer Ly4 is provided with a via conductor h4b connected to the via conductor h3b.

[0057] A wiring pattern r5 constituting a part of the inductor L1 is formed on the dielectric layer Ly5, and one end of the wiring pattern r5 is connected to the via conductor h5a and the other end is connected to the terminal P1. A via conductor h5b connected to the via conductor h4b is provided on the dielectric layer Ly5.

[0058] An electrode pattern p1 constituting a part of the capacitor C1 is formed on the dielectric layer Ly6 at a position not overlapping with the inductors L1 and L2 when viewed from the stacking direction. The electrode pattern p1 is connected to the terminal P2 and also to a via conductor h6a. The via conductor h6a is connected to the via conductor h4c, and electrically connects the electrode pattern p1 to the wiring pattern r4b of the second path TL2. The dielectric layer Ly6 is provided with a via conductor h6b that connects to the via conductor h5b.

[0059] An electrode pattern p2 constituting a part of the capacitor C1 is formed on the dielectric layer Ly7 at a position not overlapping with the inductors L1 and L2 when viewed from the lamination direction. The electrode pattern p2 is connected to the via conductor h7b and electrically connected to the inductor L2, but is not directly electrically connected to the electrode pattern p1. The dielectric layer Ly7 is provided with a via conductor h7a connected to the via conductor h6a.

[0060] An electrode pattern p3 constituting a part of the capacitor C1 is formed on the dielectric layer Ly8 at a position not overlapping with the inductors L1 and L2 when viewed from the stacking direction. The electrode pattern p3 is connected to the terminal P2 and also to a via conductor h7a. The electrode pattern p1 and the electrode pattern p3 are electrically connected to each other through the via conductor h7a. The dielectric layer Ly8 is provided with a via conductor h8 that is connected to the via conductor h7b.

[0061] An electrode pattern p4 constituting a part of the capacitor C1 is formed on the dielectric layer Ly9 at a position not overlapping with the inductors L1 and L2 when viewed from the lamination direction. The electrode pattern p4 is connected to the via conductor h8 and is electrically connected to the electrode pattern p2, but is not electrically connected to the electrode patterns p1 and p3.

[0062] The wiring pattern r5 formed on the dielectric layer Ly5 and the wiring pattern r4a formed on the dielectric layer Ly4 form an inductor L1 in a winding shape when viewed from the stacking direction. The wiring patterns r1 to r3 formed on the dielectric layers Ly1 to Ly3 form an inductor L2 in a winding shape when viewed from the stacking direction. The inductors L1 and L2 are disposed to face each other, and the opening of the inductor L1 at least partially overlaps with the opening of the inductor L2 when viewed from the stacking direction.

[0063] In the inductor L1 shown in Fig. 10, the wiring pattern r5 and the wiring pattern r4a are wound counterclockwise from the dielectric layer Ly5 to the dielectric layer Ly1, whereas in the inductor L2, the wiring patterns r1 to r3 are wound clockwise, which is the opposite. Therefore, unlike the equivalent circuit diagram shown in Fig. 4(b), a mutual inductance -M occurs in the first path TL1 and a mutual inductance +M occurs in the second path TL2.

[0064] Furthermore, when the winding direction of inductor L1 and the winding direction of inductor L2 are opposite, the wiring pattern r2a constituting part of inductor L1 and the wiring pattern r2a constituting part of inductor L2 are not formed in the same layer as the dielectric layer Ly2 in Fig. 9. Therefore, the degree of freedom of inductor L1 and inductor L2 is relatively high, and the inductance of each inductor L1 and inductor L2 can be easily adjusted.

[0065] As described above, the filter device 100 according to the first embodiment is a filter device having a pass band in the f1 band (first frequency band) and an attenuation band in the f2 band (second frequency band) lower than the f1 band. The filter device 100 includes a terminal P1 (first terminal), a terminal P2 (second terminal), an inductor L1 (first inductor) connected to the terminal P1, and an LC series resonator RS including a capacitor C1 (first capacitor) and an inductor L2 (second inductor) arranged in the first path TL1 of a first path TL1 and a second path TL2 provided in parallel between the inductor L1 and the terminal P2. The inductor L1 and the inductor L2 are magnetically coupled to each other.

[0066] As a result, the filter device 100 according to the first embodiment can achieve both high attenuation and pass characteristics even when the attenuation band due to parallel resonance and the pass band due to series resonance are close to each other.

[0067] It is preferable that the inductance of the second path TL2 is smaller than the mutual inductance M between the inductor L1 and the inductor L2. This allows the second path TL2 to be regarded as a short path, making it easier to design the parallel resonant frequency.

[0068] It is preferable that the inductance of the inductor L1 is smaller than the inductance of the inductor L2, thereby making it possible to reduce the loss in the entire filter device 100.

[0069] The terminals P1 and P2 are electrically connected to a first external electrode and a second external electrode, respectively, provided on the housing, and the inductor L1 and the LC series resonator RS are preferably provided inside the housing. This allows the filter device 100 to be integrally formed, for example, as a chip component. Furthermore, by making the filter device 100 smaller, the number of components of an antenna device incorporating the filter device 100 can be reduced, and the amount of solder used can also be reduced.

[0070] The housing is an insulator, and the inductor L1 and the LC series resonator RS are configured by a plurality of conductor patterns in the insulator. The inductor L1 is electrically connected to the terminal P1, and includes one or more layers of wiring patterns r1, r2a (first conductor patterns). The inductor L2 is electrically connected to the terminal P2, and includes one or more layers of wiring patterns r2b, r3 to r5 (second conductor patterns). It is preferable that the capacitor C1 is electrically connected to the wiring pattern r2c drawn from the wiring patterns r2a, r2b. This allows the filter device 100 to be integrally formed as a chip component of a laminated structure. In addition, by drawing out the wiring pattern r2c from the middle of the wiring patterns of the inductors L1, L2, the number of layers forming the second path TL2 can be reduced, and therefore the filter device 100 can be formed as a low-profile, low-cost (and environmentally friendly) component.

[0071] In the insulator, a substrate having wiring patterns r2b, r3-r5 (second conductor patterns) is laminated on a substrate having wiring patterns r1, r2a (first conductor patterns) so that the inductors L1 and L2 face each other, and it is preferable that the opening of the inductor L1 at least partially overlaps the opening of the inductor L2 when viewed from the lamination direction of the insulator. This increases the coupling coefficient between the inductors L1 and L2, and increases the mutual inductance M due to magnetic coupling.

[0072] It is preferable that the capacitor C1 is disposed on a layer different from the layer on which the inductor L1 and the inductor L2 are disposed, so that the capacitor C1 can be made of different dielectric materials from the inductor L1 and the inductor L2.

[0073] The capacitor C1 is preferably disposed on the inductor L1 side when viewed from the lamination direction of the insulators, which allows the length of the second path TL2 connecting the capacitor C1 and the inductor L1 to be shortened.

[0074] Antenna device 150 according to the first embodiment is capable of emitting radio waves in the f1 band. Antenna device 150 includes antenna 155, power feeding circuit RF1 that supplies a high-frequency signal to antenna 155, and filter device 100 provided between antenna 155 and power feeding circuit RF1.

[0075] As a result, the antenna device 150 according to the first embodiment can pass the f1 band and attenuate the radio waves in the f2 band even when the f1 band and the f2 band are close to each other.

[0076] [Embodiment 2] In the first embodiment, the antenna device 150 having the antenna 155 has been described. In the second embodiment, an antenna module 200 including an antenna device 160 will be described in addition to the antenna device 150 in the first embodiment. Note that the description of the configuration of the antenna module 200 in the second embodiment that overlaps with that of the antenna device 150 in the first embodiment will not be repeated.

[0077] <Basic configuration of antenna module> 11 is a diagram showing a configuration of an antenna module 200 according to the second embodiment. The antenna module 200 includes an antenna device 150 and an antenna device 160. The antenna device 160 includes a power feeding circuit RF2 and an antenna 165. The antenna module 200 is mounted on a communication device such as a mobile terminal such as a mobile phone, a smartphone, or a tablet, or a personal computer equipped with a communication function.

[0078] The power feed circuit RF1 supplies a high-frequency signal in the f1 frequency band to the antenna 155. The antenna 155 can radiate the f1 frequency signal supplied from the power feed circuit RF1 into the air as a radio wave. The f1 frequency band is, for example, n41 (2.5-2.7 GHz).

[0079] The filter device 100 according to the second embodiment is configured to attenuate high-frequency signals in the f2 frequency band. The f2 frequency band is, for example, the 2.4 GHz band (2.4-2.5 GHz) of Wi-Fi (registered trademark).

[0080] In the filter device 100 according to the second embodiment, the f1 band is a pass band and the f2 band is an attenuation band. The f1 band is a frequency band lower than the f2 band.

[0081] The power feeding circuit RF2 supplies a high-frequency signal in the f2 frequency band to the antenna 165. The antenna 165 is capable of radiating the high-frequency signal in the f2 band supplied from the power feeding circuit RF2 into the air as a radio wave.

[0082] In the antenna device 150, a high-frequency signal in the f2 band radiated from the antenna device 160 provided in the same antenna module 200 can become noise. Therefore, the filter device 100 is provided to remove the high-frequency signal in the f2 band that can become noise in the antenna device 150 by increasing the insertion loss due to parallel resonance.

[0083] Antenna 155 and antenna 165 are mounted on the same substrate 170. Although antenna 155 and antenna 165 are provided on the same substrate 170 in Fig. 11, they may be provided on different substrates as long as they are provided within the same antenna module 200. In the second embodiment, power feeding circuit RF1 is not limited to supplying only high frequency signals in the f1 band, and may supply high frequency signals in other bands.

[0084] Fig. 12 is a diagram showing the isolation characteristics between antenna devices 150 and 160 according to the second embodiment. In Fig. 12, the horizontal axis represents frequency, and the vertical axis represents isolation.

[0085] A line Ln6 indicates the isolation between the antenna device 150 and the antenna device 160 of the antenna module 200 according to the second embodiment. A line Ln7 indicates the isolation between the antenna device 150 and the antenna device 160 in which the filter device 100 is not provided, as a comparative example. That is, the ratio of the power received by the power feeding circuit RF1 of the antenna device 150 via the antenna to the power input from the power feeding circuit RF2 of the antenna device 160 is the isolation.

[0086] 12, at a frequency of 2.4 GHz in the f2 band, the isolation (Ln6) of the antenna module 200 is improved by 10 dB or more over the isolation (Ln7) of the antenna module of the comparative example. That is, in the second embodiment, the filter device 100 attenuates the frequency in the f2 band, thereby improving the isolation between the antenna device 150 and the antenna device 160.

[0087] FIG. 13 is a diagram showing the radiation efficiency of each of the antenna devices 150 and 160 according to the second embodiment. In FIG. 13, the horizontal axis indicates frequency, and the vertical axis indicates radiation efficiency. Line Ln8 indicates the radiation efficiency of the antenna device 150 of the antenna module 200 according to the second embodiment. Line Ln9 indicates the radiation efficiency of the antenna device 160 of the antenna module 200 according to the second embodiment. Line Ln8a indicates the radiation efficiency of the antenna device 150 not provided with the filter device 100 as a comparative example. Line Ln9a indicates the radiation efficiency of the antenna device 160 in the case where the filter device 100 is not provided in the antenna device 150 as a comparative example. Here, the radiation efficiency means the ratio of the power radiated from the antenna to the power supplied from the feeding circuit. That is, in FIG. 13, the power radiated from the antenna increases toward the top of the graph for the same supplied power.

[0088] 13, at a frequency of 2.4 GHz in the f2 band, the radiation efficiency (Ln9) of the antenna device 160 of the antenna module 200 is improved by approximately 6 dB compared to the radiation efficiency (Ln9a) of the antenna device 160 of the comparative example. That is, in the second embodiment, the filter device 100 is provided in the antenna device 150, thereby improving the radiation efficiency of the antenna device 160.

[0089] <Antenna structure example> FIG. 14 is an external view of the antenna module 200 in the second embodiment. As shown in FIG. 14, the antenna module 200 includes an antenna device 150 and an antenna device 160. The antenna device 150 includes an antenna 155 which is a monopole antenna, a filter device 100, and a feed circuit RF1. The antenna device 160 includes an antenna 165 which is a monopole antenna, and a feed circuit RF2. The antennas 155 and 165 are not limited to monopole antennas, and may be an inverted-F antenna, a loop antenna, an array antenna, or the like. The antenna 155 is connected to the feed circuit RF1 via the filter device 100. The antenna 165 is connected to the feed circuit RF2.

[0090] As described above, the antenna module 200 according to the second embodiment is capable of emitting radio waves in the f1 and f2 bands. The antenna module 200 includes an antenna device 150 capable of emitting radio waves in the f1 band and an antenna device 160 capable of emitting radio waves in the f2 band. The antenna device 150 is the antenna device according to the first embodiment.

[0091] As a result, the antenna module 200 of embodiment 2 can improve the isolation between the antenna device 150 and the antenna device 160, improve the radiation characteristics of f1 band radio waves in the antenna device 150, and also improve the radiation characteristics of f2 band radio waves in the antenna device 160.

[0092] [Embodiment 3] In the first embodiment, as shown in Fig. 1, the filter device 100 has been described, which has a first path TL1 in which an LC series resonator RS is provided between the inductor L1 and the terminal P2, and a second path TL2 which is a short path. In the third embodiment, a filter device in which an inductor is provided in parallel with the short path of the filter device 100 in the first embodiment will be described. Note that in the filter device of the third embodiment, the same components as those in the filter device 100 of the first embodiment are denoted by the same reference numerals and detailed description will not be repeated. Also, in the antenna device 150 of the first embodiment and the antenna module 200 of the second embodiment, the filter device of the third embodiment may be used instead of the filter device 100.

[0093] Fig. 15 is a circuit diagram of a filter device 100A according to the third embodiment. As shown in Fig. 15, the filter device 100A includes an inductor L1, an inductor L2, an inductor L3, and a capacitor C1. A first path TL1 and a second path TL2 are provided between the inductor L1 and a terminal P2. The first path TL1 is provided with an LC series resonator RS in which the inductor L2 and the capacitor C1 are connected in series. The second path TL2 is a short path. Furthermore, an inductor L3 is provided in parallel with the second path TL2, which is a short path.

[0094] Inductor L1 and inductor L2 are magnetically coupled to each other, but inductor L3 is not magnetically coupled to inductor L1 or inductor L2. Fig. 16 is a schematic diagram of a filter device 100A in the third embodiment. The filter device 100A is integrally formed as a chip component, for example, as shown in Fig. 16, and includes inductor L1, inductor L2, inductor L3, and capacitor C1 shown in Fig. 15 in an insulator 1 (housing) in which dielectric layers are laminated. External electrodes 2a and 2b are formed on the outside of the insulator 1, and terminal P1 is connected to external electrode 2a (first external electrode), and terminal P2 is connected to external electrode 2b (second external electrode).

[0095] 16, the wiring from the connection between the inductor L1 and the inductor L2 to the external electrode 2b corresponds to the second path TL2 which is a short path, and the parasitic inductance of about 2nH of the wiring provided in parallel to the second path TL2 corresponds to the inductor L3. That is, the inductor L3 is located at a position where it does not overlap with the inductor L1 or the inductor L2 when viewed from the coil opening direction. The external electrode 2a is electrically connected to a land electrode 20a of a circuit board on which the filter device 100A is mounted, and the external electrode 2b is electrically connected to a land electrode 20b of a circuit board on which the filter device 100A is mounted.

[0096] FIG. 17 is a diagram showing an example of the insertion loss and reactance characteristics of the filter device 100A according to the third embodiment. The constants of the filter device 100A are L1=2.0nH, L2=2.0nH, C1=2.2pF, k=0.6, and L3=2nH, and the insertion loss and reactance characteristics are shown by solid lines in FIG. 17. As a comparative example, the constants of the filter device 100 in FIG. 1 are L1=2.0nH, L2=2.0nH, C1=2.2pF, and k=0.6, and the insertion loss and reactance characteristics are shown by dashed lines in FIG. 17. FIG. 17(a) is a diagram showing an example of the insertion loss of the filter device 100A according to the third embodiment. In FIG. 17(a), the horizontal axis is frequency, and the vertical axis is insertion loss. FIG. 17(b) is a diagram showing an example of the reactance characteristics of the filter device 100A according to the third embodiment. In FIG. 17(b), the horizontal axis represents frequency and the vertical axis represents reactance.

[0097] 17(a) and 17(b), the filter device 100A has a resonant frequency of about 2.4 GHz, which is the same as that of the filter device 100 of the first embodiment. That is, it can be seen that even if the inductor L3 (a third path having a parasitic inductance of about 2 nH) is provided in parallel with the second path TL2, which is a short path, as in the filter device 100A, there is no change in the insertion loss and reactance characteristics. On the other hand, the filter device 100A can reduce the ESL by providing the inductor L3. In this way, it is possible to form an additional path regardless of the parasitic inductance, and it is possible to reduce the ESL.

[0098] [Embodiment 4] In the first embodiment, as shown in Fig. 1, the filter device 100 has been described, which has a first path TL1 in which an LC series resonator RS is provided between the inductor L1 and the terminal P2, and a second path TL2 which is a short path. In the fourth embodiment, a filter device in which a capacitor is provided in parallel with the short path of the filter device 100 in the first embodiment will be described. Note that in the filter device of the fourth embodiment, the same components as those in the filter device 100 of the first embodiment are denoted by the same reference numerals and detailed description will not be repeated. Also, in the antenna device 150 of the first embodiment and the antenna module 200 of the second embodiment, the filter device of the fourth embodiment may be used instead of the filter device 100.

[0099] Fig. 18 is a circuit diagram of a filter device 100B according to the fourth embodiment. As shown in Fig. 18, the filter device 100B includes an inductor L1, an inductor L2, a capacitor C1, and a capacitor C3. A first path TL1 and a second path TL2 are provided between the inductor L1 and a terminal P2. The first path TL1 is provided with an LC series resonator RS in which the inductor L2 and the capacitor C1 are connected in series. The second path TL2 is a short path. Furthermore, a capacitor C3 is provided in parallel with the second path TL2, which is a short path.

[0100] The inductor L1 and the inductor L2 are magnetically coupled to each other. Fig. 19 is a schematic diagram of a filter device 100B according to the fourth embodiment. The filter device 100B is integrally formed as a chip component as shown in Fig. 19, for example, and includes the inductor L1, the inductor L2, the capacitor C1, and the capacitor C3 shown in Fig. 18 in an insulator 1 (housing) in which dielectric layers are laminated. External electrodes 2a and 2b are formed on the outside of the insulator 1, and a terminal P1 is connected to the external electrode 2a (first external electrode), and a terminal P2 is connected to the external electrode 2b (second external electrode).

[0101] 19, the wiring from the connection portion between the inductor L1 and the inductor L2 to the external electrode 2b corresponds to the second path TL2 which is a short path, and the parasitic capacitance formed between the second path TL2 and the external electrode 2b corresponds to the capacitor C3. The external electrode 2a is electrically connected to a land electrode 20a of a circuit board for mounting the filter device 100B, and the external electrode 2b is electrically connected to a land electrode 20b of the circuit board for mounting the filter device 100B.

[0102] If the values ​​of L1, L2, C1, and k are the same, the resonant frequency of the filter device 100B is about 2.4 GHz, which is the same as that of the filter device 100A shown in FIG. 17(a) and FIG. 17(b), regardless of C3, and has the same resonant frequency as the filter device 100 of the first embodiment. The reactance characteristic is also the same as that of the filter device 100. That is, it can be seen that even if a capacitor C3 (parasitic capacitance) is provided in parallel with the second path TL2, which is a short path, as in the filter device 100B, there is no change in the insertion loss and reactance characteristics. That is, normally, the inductor is placed as far away from the external electrode as possible as shown in FIG. 16, but since the characteristics do not vary due to the parasitic capacitance C3, it is also possible to place the inductor in a position close to the outside as shown in FIG. 19.

[0103] [Embodiment 5] In the third embodiment, as shown in Fig. 15, a filter device 100A in which an inductor L3 is provided in parallel with the second path TL2, which is a short path, has been described. In the fifth embodiment, a filter device in which an inductor is provided in parallel with the entire path, rather than in parallel with the short path as in the filter device 100A in the third embodiment, will be described. Note that in the filter device of the fifth embodiment, the same components as those in the filter device 100 of the first embodiment are denoted by the same reference numerals and detailed description will not be repeated. Also, in the antenna device 150 of the first embodiment and the antenna module 200 of the second embodiment, the filter device of the fifth embodiment may be used instead of the filter device 100.

[0104] FIG. 20 is a circuit diagram of a filter device 100C according to the fifth embodiment. As shown in FIG. 20, the filter device 100C includes an inductor L1, an inductor L2, an inductor L3 (third inductor), and a capacitor C1. A first path TL1 and a second path TL2 are provided between the inductor L1 and the terminal P2. The first path TL1 is provided with an LC series resonator RS in which the inductor L2 and the capacitor C1 are connected in series. The second path TL2 is a short path. Furthermore, an inductor L3 is provided in parallel with the entire path between the terminal P1 and the terminal P2. That is, the inductor L3 is connected in parallel with the inductor L1 and the inductor L2. Note that the inductor L1 and the inductor L2 are magnetically coupled to each other, but the inductor L3 is not magnetically coupled to the inductor L1 and the inductor L2.

[0105] FIG. 21 is a diagram showing an example of the insertion loss of the filter device 100C according to the fifth embodiment. In FIG. 21, the horizontal axis indicates frequency, and the vertical axis indicates insertion loss. The graph shown in FIG. 21(a) is an example of the insertion loss of the filter device 100 according to the first embodiment, and the constants of the filter device 100 are L1=2nH, L2=2nH, C1=2.2pF, and k=0.6. The graph shown in FIG. 21(b) is an example of the insertion loss of the filter device 100C according to the fifth embodiment, and in addition to the constants of FIG. 21(a), L3=2.5nH. FIG. 22 is a diagram showing an example of the reactance characteristic of the filter device 100C according to the fifth embodiment. In FIG. 22, the horizontal axis indicates frequency, and the vertical axis indicates reactance. The graph shown in FIG. 22(a) is an example of the reactance characteristic of the filter device 100 according to the first embodiment, and the graph shown in FIG. 22(b) is an example of the reactance characteristic of the filter device 100C according to the fifth embodiment.

[0106] 21 and 22, the resonant frequency of the filter device 100 is about 2.4 GHz (mark m6), while the resonant frequency of the filter device 100C is about 2.7 GHz (mark m7), which is shifted to the high frequency side by about 0.3 GHz. That is, as in the filter device 100C, the resonant frequency can be adjusted by connecting the inductor L3 in parallel to the entire path. The filter device 100C may be an integrated chip, or may have a structure including a circuit board by adding another inductor element to the filter device 100. By adding another inductor element to the filter device 100, it becomes possible to arbitrarily adjust the resonant frequency.

[0107] [Embodiment 6] In the fourth embodiment, as shown in Fig. 18, a filter device 100B in which a capacitor C3 is provided in parallel with the second path TL2, which is a short path, has been described. In the sixth embodiment, a filter device in which a capacitor is provided in parallel with the entire path, rather than providing a capacitor C3 in parallel with the short path as in the filter device 100B in the fourth embodiment, will be described. Note that in the filter device of the sixth embodiment, the same components as those in the filter device 100 of the first embodiment are denoted by the same reference numerals and detailed description will not be repeated. Also, in the antenna device 150 of the first embodiment and the antenna module 200 of the second embodiment, the filter device of the sixth embodiment may be used instead of the filter device 100.

[0108] Fig. 23 is a circuit diagram of a filter device 100D according to the sixth embodiment. As shown in Fig. 23, the filter device 100D includes an inductor L1, an inductor L2, a capacitor C1, and a capacitor C3 (second capacitor). A first path TL1 and a second path TL2 are provided between the inductor L1 and the terminal P2. The first path TL1 is provided with an LC series resonator RS in which the inductor L2 and the capacitor C1 are connected in series. The second path TL2 is a short path. Furthermore, a capacitor C3 is provided in parallel with the entire path between the terminal P1 and the terminal P2. That is, the capacitor C3 is connected in parallel with the inductor L1 and the inductor L2.

[0109] Furthermore, in the filter device 100D, an inductor L4 and a capacitor C5 are provided in parallel to the second path TL2, which is the short path described in the third and fourth embodiments. As described in the third and fourth embodiments, the filter device 100D has no change in insertion loss and reactance characteristics even when the inductor L4 and the capacitor C5 are provided. The filter device 100D may be configured to include either the inductor L4 or the capacitor C5. Similarly, in the filter device 100C of the fifth embodiment, the inductor L4 and the capacitor C5 may be provided in parallel to the second path TL2, which is the short path described in the third and fourth embodiments.

[0110] FIG. 24 is a diagram showing an example of the insertion loss of the filter device 100D according to the sixth embodiment. In FIG. 24, the horizontal axis indicates frequency, and the vertical axis indicates insertion loss. The graph shown in FIG. 24(a) shows an example of the insertion loss of the filter device 100 according to the first embodiment, where the constants are L1=2.0 nH, L2=2.0 nH, C1=2.2 pF, and k=0.6. The graph shown in FIG. 24(b) shows an example of the insertion loss of the filter device 100D according to the sixth embodiment, where, in addition to the constants, C3=4 pF, C5=2 pF, and L4=2 nH. FIG. 25 is a diagram showing an example of the reactance characteristic of the filter device 100D according to the sixth embodiment. In FIG. 25, the horizontal axis indicates frequency, and the vertical axis indicates reactance. The graph shown in FIG. 25(a) shows an example of the reactance characteristic of the filter device 100 according to the first embodiment, and the graph shown in FIG. 25(b) shows an example of the reactance characteristic of the filter device 100D according to the sixth embodiment.

[0111] 24 and 25, the filter device 100 has one resonant frequency at approximately 2.4 GHz (mark m6), while the filter device 100D has two resonant frequencies at approximately 2.2 GHz (mark m8) and approximately 4.9 GHz (mark m9). That is, by connecting the capacitor C3 in parallel to the entire path as in the filter device 100D, it is possible to add an attenuation region in the L-characteristic region shown in FIG. 25. Note that, since the resonant frequency of the filter device 100D shifts to the low frequency side by providing the capacitor C3, the resonant frequency can be arbitrarily adjusted by adding another capacitor element to the filter device 100 to the entire path as described in the fifth embodiment.

[0112] [Variations] 1, the filter device 100 according to the first embodiment has a configuration in which the inductor L1, the inductor L2, and the capacitor C1 are provided in this order between the terminal P1 and the terminal P2. However, the filter device 100 may switch the order of the inductor L2 and the capacitor C1, or may connect the inductor L1 side to the terminal P2. The filter device 100 can determine whether to connect the inductor L1 to the terminal P2 side (the antenna 155 side) or to connect the inductor L1 to the terminal P1 side (the power feeding circuit RF1 side) based on the antenna impedance.

[0113] Fig. 26 is a circuit diagram of a filter device 100a according to a modified example. As shown in Fig. 26, the filter device 100a has an inductor L2, a capacitor C1, and an inductor L1 provided in this order between a terminal P1 and a terminal P2.

[0114] In the filter device 100a, a first path TL1 and a second path TL2 are provided between the terminal P1 and the inductor L1. The first path TL1 is provided with an LC series resonator RS in which an inductor L2 and a capacitor C1 are connected in series in this order. The second path TL2 is a short path. The inductor L1 and the inductor L2 are magnetically coupled to each other. The filter device 100a can obtain the same effect as the filter device 100, except for the influence of the parasitic capacitance and parasitic inductance that occur in the second path TL2, which is a short path, due to the inductor L2 being swapped in order.

[0115] Although the filter devices 100 and 100a have been described as being designed taking into consideration only the inductor L1, the inductor L2, and the capacitor C1, an actual filter device must be designed taking into consideration stray capacitance, parasitic inductance, and the like.

[0116] The filter device 100, 100a may include other components such as an antenna 155, a matching circuit for matching impedance with the feeding circuit RF1, etc., and a phase shifter for switching the phase of a high-frequency signal.

[0117] The embodiments disclosed herein should be considered to be illustrative and not restrictive in all respects. The scope of the present invention is defined by the claims, not by the description of the embodiments described above, and is intended to include all modifications within the scope and meaning equivalent to the claims. [Explanation of symbols]

[0118] 1 insulator, 2a to 2d external electrodes, 100, 100a filter device, 150, 160 antenna device, 155, 165 antenna, 170 substrate, 200 antenna module, C1 capacitor, ESL equivalent series inductance, K, K2 coupling coefficient, L1, L2 inductor, P1, P2 terminal, RF1, RF2 power supply circuit, TL1 first path, TL2 second path.

Claims

1. 1. A filter device having a pass band in a first frequency band and an attenuation band in a second frequency band lower than the first frequency band, A first terminal; A second terminal; a first inductor connected to the first terminal; a series resonator including a first capacitor and a second inductor disposed in the first path of a first path and a second path provided in parallel between the first inductor and the second terminal; The first inductor and the second inductor are magnetically coupled to each other.

2. The filter device according to claim 1 , wherein an inductance of the second path is smaller than a mutual inductance between the first inductor and the second inductor.

3. 3. The filter device according to claim 1, wherein an inductance of the first inductor is smaller than an inductance of the second inductor.

4. the first terminal and the second terminal are electrically connected to a first external electrode and a second external electrode, respectively, provided on a housing; 3. The filter device according to claim 1, wherein the first inductor and the series resonator are provided within the housing.

5. the housing is an insulator, the first inductor and the series resonator are configured by a plurality of conductor patterns in the insulator, The first inductor is a first conductor pattern electrically connected to the first external electrode and including one or more layers of the first conductor pattern; The second inductor is a second conductor pattern including one or more layers electrically connected to the second external electrode; The filter device according to claim 4 , wherein the first capacitor is electrically connected to a wiring extended from the first conductor pattern or the second conductor pattern.

6. Within the insulator, a substrate on which the second conductor pattern is formed is laminated on a substrate on which the first conductor pattern is formed, and the first inductor and the second inductor are disposed so as to face each other; The filter device according to claim 5 , wherein the opening of the first inductor at least partially overlaps with the opening of the second inductor when viewed from a lamination direction of the insulators.

7. The filter device according to claim 6 , wherein the first capacitor is disposed on a layer different from a layer on which the first inductor and the second inductor are disposed.

8. The filter device according to claim 7 , wherein the first capacitor is disposed on a side of the first inductor when viewed from a lamination direction of the insulators.

9. The filter device according to claim 1 or 2, further comprising a third path connected in parallel to the second path.

10. The filter device according to claim 9 , wherein the third path is not magnetically coupled to the first inductor and the second inductor.

11. The filter device according to claim 10 , wherein the third path does not overlap the first inductor and the second inductor when viewed from a direction in which the first inductor and the second inductor are opened.

12. 3. The filter device according to claim 1, further comprising a third inductor connected in parallel to the first inductor and the second inductor.

13. a third inductor connected in parallel to the first inductor and the second inductor; one end of the third inductor is connected to the first external electrode, and the other end of the third inductor is connected to the second external electrode; The filter device according to claim 4 , wherein the third inductor is provided outside the housing as a separate element.

14. 3. The filter device according to claim 1, further comprising a second capacitor connected in parallel to the first inductor and the second inductor.

15. a second capacitor connected in parallel to the first inductor and the second inductor; one end of the second capacitor is connected to the first external electrode, and the other end of the second capacitor is connected to the second external electrode; The filter device according to claim 4 , wherein the second capacitor is provided outside the housing as a separate element.

16. An antenna device capable of radiating radio waves in the first frequency band, The antenna, a power supply circuit for supplying a high frequency signal to the antenna; 3. An antenna device comprising: the filter device according to claim 1 or 2, which is provided between the antenna and the power supply circuit.

17. 1. An antenna module, comprising: a first antenna device capable of radiating radio waves in the first frequency band; a second antenna device capable of emitting radio waves in the second frequency band, 17. An antenna module, wherein the first antenna device is an antenna device according to claim 16.

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