Magnetic resonance wireless power supply device

Abstract

Provided is a magnetic resonance wireless power supply device that uses PT symmetry and performs non-contact power supply by magnetic resonance of a power supply coil and a power reception coil, the magnetic resonance wireless power supply device being capable of selecting and fixing any one resonance loop out of two resonance loops by a method that does not restrict the design of the power supply coil and the power reception coil regardless of the surrounding environment.　The magnetic resonance wireless power supply device comprises a power supply side resonance circuit including the power supply coil and a power reception side resonance circuit including the power reception coil and uses PT symmetry. In the magnetic resonance wireless power supply device, an inverter 3 and a sensor 4 are connected to the power supply side resonance circuit 3, and a switch timing, which is a time when a switching element inside the inverter 3 is turned on or turned off, is determined on the basis of a pulse generated by a signal after the phase relation is adjusted by a phase adjustment circuit 30, so that any one resonance loop of the two resonance loops is selected and fixed.

Description

Magnetic resonance type wireless power supply device

[0001] The present invention relates to a magnetic resonance type wireless power feeder that utilizes parity-time symmetry (hereinafter referred to as "PT symmetry") to feed power in a contactless manner by magnetically resonating a feeding coil and a receiving coil.

[0002] There are several well-known wireless power transfer technologies, including those based on electromagnetic induction and magnetic resonance. Of these, the electromagnetic induction wireless power transfer technology is used, for example, to charge mobile phones, and has coils arranged one above the other. In other words, based on the same principle as a transformer, power can be transferred only when the distance between the power transfer coil and the power receiving coil (transmission distance) is very short.

[0003] However, because the transmission distance of electromagnetic induction wireless power transfer technology is short, at around a few millimeters, the distance between the power transfer coil and the power receiving coil cannot be large. Furthermore, even a slight misalignment or separation between the power transfer coil and the power receiving coil makes charging and power transfer impossible. In other words, because this technology is sensitive to misalignment, it is difficult to apply it to artificial devices attached inside the human body, such as artificial hearts, or to devices that rotate in multiple directions or have an axis that is misaligned, such as robotic arms.

[0004] Furthermore, magnetic resonance wireless power transfer technology has a long transmission distance of several centimeters to several meters, which allows for a greater distance between the power supply coil and the power receiving coil compared to electromagnetic induction, and it is close to being put to practical use. However, power cannot be transferred unless the distance between the power supply coil and the power receiving coil is kept constant, and if the coils are closer or further away than that distance, or if they are at an angle, the transmission efficiency decreases and the required power cannot be sent. In other words, this method is also vulnerable to positional misalignment, making it difficult to apply to devices that rotate in multiple directions or have an axis misalignment (rotating power supply targets), such as robot arms.

[0005] One wireless power supply method is a technology that utilizes Parity-Time symmetry (hereinafter referred to as "PT symmetry"). This wireless power supply that utilizes PT symmetry is a physical system with a non-Hermitian Hamiltonian, and is a new concept of wireless power supply that was first announced in 2017 (see Non-Patent Document 1).

[0006] When PT symmetry is preserved, the eigenenergy of the Hamiltonian becomes real, so that even though the system is non-Hermitian, it behaves as if the energy of the system is conserved. In this case, the energy transferred between the power supply resonant circuit and the power receiving resonant circuit per unit time does not depend on the coupling coefficient between the two resonant circuits. As a result, even if the transmission distance changes or the position of the power supply coil and the power receiving coil is misaligned, the transmitted power and power transfer efficiency will always remain constant as long as PT symmetry is preserved.

[0007] Furthermore, many methods have been proposed for automatically adjusting the frequency of inverters used as AC power sources in order to suppress fluctuations in transmission power and power transmission efficiency with changes in transmission distance (see Non-Patent Document 2). However, these control methods cannot completely separate the transmission power and power transmission efficiency from the transmission distance because none of these systems are designed as non-Hermitian physical systems and do not preserve PT symmetry.

[0008] Wireless power supply utilizing PT symmetry is a technology in which the AC power supply in conventional magnetic resonance type wireless power supply technology is replaced with an inverter that behaves electrically in the same way as a negative resistor, that is, an inverter that behaves as a negative resistor.

[0009] This is a well-known technology, as disclosed in, for example, Patent Document 1. To explain it in more detail, an inverter that behaves as a negative resistance is an inverter whose switching frequency and voltage amplitude are not fixed in advance, has a circuit configuration in which the switching frequency is determined by the apparent resonant frequency of the wireless power feed circuit as seen from the inverter's output end, and has an inverter in which the switching frequency quickly responds to changes in the apparent resonant frequency of the wireless power feed circuit, which may change due to changes in the coil transmission distance or positional misalignment. Here, the wireless power feed circuit refers to a circuit that includes a power feeder-side resonant circuit, a power receiver-side resonant circuit, and all subsequent circuits connected thereto. Furthermore, since the power feeder-side resonant circuit and the power receiver-side resonant circuit interact with each other, the apparent resonant frequency refers to the actual resonant frequency that takes this interaction into account.

[0010] Furthermore, in wireless power transfer utilizing PT symmetry, the aforementioned inverter is used in a self-oscillating state. There are two modes of oscillation, and PT symmetry can be preserved in either mode. Here, these two modes will be explained in more detail. Figure 1 shows an equivalent circuit of the SP topology, and is the same as Figure 14 of Patent Document 1.

[0011] As shown in FIG. 1, the power supply side resonant circuit 1 and the power receiving side resonant circuit 2 of the SP topology have a mutual inductance k ｍ It can be expressed as a multi-resonant circuit coupled by L. In addition, L in FIG. 1 represents the self-inductance of the power supply coil 11 and the power receiving coil 21. L(1-k ｍ ) represents the leakage inductance of the power supply coil 11 and the power receiving coil 21. １ ' and r ２ ' represents the winding resistance of the power supply coil 11 and the power receiving coil 21. ｃrepresents the iron loss equivalent resistance. C represents the capacitance (electrostatic capacity) of the capacitor. In the circuit shown in Figure 1, there are two resonant loops ("resonant loop of loop I" and "resonant loop of loop II") through which the resonant current flowing in the multi-resonant circuit can circulate. In addition, regarding frequency, the resonant frequency of loop I is ω ｌ , the resonant frequency of loop II is ω ｈ Let's say.

[0012] However, as shown in the experimental results in Non-Patent Document 1, the two oscillation modes (resonance loops) may unintentionally switch during operation. When the oscillation mode (resonance loop) switches, operation becomes unstable and power transmission itself becomes difficult, so a method for reliably selecting one of the oscillation modes (resonance loops) was needed.

[0013] Therefore, as one method for solving such a problem, Patent Document 1 proposes a method in which, in a magnetic resonance type wireless power feeder that utilizes PT symmetry and feeds power contactlessly by magnetically resonating a power feed coil and a power receiving coil, the magnitude of the Q value (selectivity) of the two resonant loops is adjusted by coil design to select one of the resonant loops.

[0014] Japanese Patent Application Laid-Open No. 2022-121324

[0015] Sid Assawaworrarit, Xiaofang Yu & Shanhui Fan,“Robust wireless power transfer using a nonlinear parity-time-symmetric circuit”, Nature, 15 JUNE 2017, volume 546, p.387-390A.P.Sample, D.T.Meyer & J.R.Smith, “Analysis, experimental results, and range adaptation of magnetically coupled resonators for wireless power transfer”, IEEE Trans. Ind. Electron., 2011, vol.58, no.2, pp.544-554J. Zhou, B. Zhang, W. Xiao, D. Qiu, and Y. Chen,“Nonlinear parity-time-symmetric model for constant efficiency wireless power transfer: application to a drone-in-flight wireless charging platform”, IEEE Trans. Ind. Electron., Aug. 2019, vol.66, no.5, pp.4097-4107H. Ishida, T. Kyoden, and H. Furukawa,“Application of parity-time symmetry to low-frequency wireless power transfer system”, IEEJ J. Ind. Appl., 2022, vol.11, no.1, pp.59-68

[0016] However, in a magnetic resonance type wireless power feeder such as that shown in Patent Document 1, a method is adopted in which one of the resonance loops is selected by adjusting the magnitude of the Q value (selectivity) of the two resonance loops through coil design. This imposes restrictions on the design of the feeding coil and receiving coil, and poses an application problem in that the coils cannot be freely shaped or sized.

[0017] The present invention has been made to solve the above-mentioned problems, and has an object to provide a magnetic resonance type wireless power feeder that utilizes PT symmetry to feed power contactlessly by magnetically resonating a feed coil and a receiver coil, and that is capable of selecting and fixing one of two resonant loops in a manner that is not affected by the surrounding environment and does not restrict the design of the feed coil and the receiver coil.

[0018] In order to achieve the above object, the present invention provides a magnetic resonance type wireless power feeder that includes a power feed coil-side resonant circuit and a power receiving coil-side resonant circuit, and that utilizes parity-time symmetry to feed power in a contactless manner by magnetically resonating the power feed coil and the power receiving coil. When the power feed coil and the power receiving coil are considered as a multi-resonant circuit coupled to each other by mutual inductance, there are two resonant loops (a "resonant loop of loop I" and a "resonant loop of loop II") through which a resonant current flowing in the multi-resonant circuit can circulate. The power feed coil-side resonant circuit includes an inverter, and a current or magnetic field of the power feed coil. a sensor for detecting a voltage applied to the power supply side resonant circuit and an AC current flowing through the power supply coil, the inverter including a phase adjustment circuit that adjusts the phase relationship between the AC voltage applied to the power supply side resonant circuit and the AC current flowing through the power supply coil, based on a current phase based on the current in the power supply coil detected by the sensor or a magnetic field phase based on the magnetic field of the power supply coil; and a switch timing, which is the time to turn on or off a switching element in the inverter, is determined based on a pulse generated by a signal after the phase relationship has been adjusted by the phase adjustment circuit, thereby selecting and fixing one of the two resonant loops.

[0019] According to this invention, in a magnetic resonance type wireless power feeder that utilizes PT symmetry to feed power contactlessly by magnetically resonating a feeding coil and a receiving coil, a phase adjustment circuit is provided to adjust the phase relationship between the AC voltage applied to the feeding-side resonant circuit and the AC current flowing through the feeding coil. This makes it possible to select and fix one of two resonant loops in a manner that is not affected by the surrounding environment and does not restrict the design of the feeding coil and the receiving coil.

[0020] The figure shows an equivalent circuit of SP topology. The figure shows a conceptual diagram of two typical circuit configurations of wireless power transfer using magnetic resonance method that utilizes PT symmetry. The magnetic coupling coefficient k ｍ 10 is a graph showing a typical example of the relationship between the resonant frequency ω of loop II and the transmission distance d between two coils (a power supply coil and a power receiving coil). ｈ and the resonant frequency ω of loop I ｌ 5 is a graph showing the relationship between the transmission distance d between two coils (a power supply coil and a power receiving coil) and the power supply side resonant circuit. This is a circuit diagram showing a case where a class D inverter is connected to a power supply side resonant circuit as an inverter used for PT symmetric wireless power supply. This is a schematic diagram showing the phase relationship of each waveform of the power supply side resonant circuit shown in FIG. 5. １ and input voltage v １9 is a circuit diagram showing an equivalent circuit of a wireless power supply connected to an inverter. FIG. 10 is a circuit diagram showing an example of an inverter having a phase adjustment circuit and a power supply side resonant circuit according to the first embodiment of the present invention. FIG. 11 is a schematic diagram showing voltage waveforms at each part of the phase adjustment circuit (preamplifier 36 and phase delay circuit 35) shown in FIG. 9 is a circuit diagram showing an example of an inverter having a time delay circuit and a power supply side resonant circuit according to the first embodiment of the present invention. FIG. 12 is a circuit diagram showing an example of an inverter having an all-pass filter and a power supply side resonant circuit according to the first embodiment of the present invention. FIG. 13 is a circuit diagram showing an example of an inverter having a PLL (phase locked loop) and a power supply side resonant circuit according to the first embodiment of the present invention. FIG. 14 is a block diagram showing an example of a specific internal function of a PLL (phase locked loop) according to the first embodiment of the present invention. FIG. 15 is a circuit diagram showing an example of an inverter having an all-pass filter and a PLL (phase locked loop) and a power supply side resonant circuit according to the first embodiment of the present invention. １ With respect to the input voltage V １ When adjusted to be a leading phase, the AC current i １ and input voltage v １ 16 is a graph showing the measured waveform of the power supply coil 11 provided in the power supply-side resonant circuit 1 and the power receiving coil 21 provided in the power receiving-side resonant circuit 2, i.e., a graph showing the measurement results of the change in resonant frequency when the distance between the power supply coil 11 provided in the power supply-side resonant circuit 1 and the power receiving coil 21 provided in the power receiving-side resonant circuit 2, i.e., the transmission distance d between these two coils, is changed in the phase relationship of Fig. 16. Fig. 17 is a graph showing the measurement results of the transmitted power when the transmission distance d between the two coils is changed.

[0021] The present invention relates to a magnetic resonance type wireless power feeder that utilizes parity-time symmetry (hereinafter referred to as "PT symmetry") to feed power in a contactless manner by magnetically resonating a feeding coil and a receiving coil. Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

[0022] First Embodiment Fig. 2 is a conceptual diagram showing two typical circuit configurations of wireless power transfer using a magnetic resonance method that utilizes PT symmetry. In both circuit examples shown in Fig. 2(a) and (b), a power supply-side resonant circuit 1 is connected to an inverter 3 as an AC power source. On the other hand, a power receiving-side resonant circuit 2 is connected to a load resistor R Ｌ In these circuits, a load of L １ is the self-inductance of the power supply coil 11, C １ is the capacitance of the power supply capacitor, L ２ is the self-inductance of the receiving coil 21, C ２ represents the capacitance of the receiving capacitor. １ and ２ represent the resistance components included in the power supply side resonant circuit 1 and the power receiving side resonant circuit 2, respectively.

[0023] The circuit example shown in Figure 2(a) is called an S-S topology because the coil and capacitor of the power supply side resonant circuit 1 are connected in series, and the coil and capacitor of the power receiving side resonant circuit 2 are also connected in series. The circuit example shown in Figure 2(b) is called an S-P topology because the coil and capacitor of the power supply side resonant circuit 1 are connected in series, and the coil and capacitor of the power receiving side resonant circuit 2 are connected in parallel. An example of wireless power transfer that preserves PT symmetry in an S-S topology is disclosed in Non-Patent Document 3, and an example of wireless power transfer that preserves PT symmetry in an S-P topology is disclosed in Non-Patent Document 4.

[0024] Here, to repeat, taking the SP topology equivalent circuit shown in FIG. 1 as an example, the power supply side resonant circuit 1 and the power receiving side resonant circuit 2 of the SP topology have a mutual inductance k ｍ It can be expressed as a multi-resonant circuit coupled by L. In addition, L in FIG. 1 represents the self-inductance of the power supply coil 11 and the power receiving coil 21. L(1-k ｍ ) represents the leakage inductance of the power supply coil 11 and the power receiving coil 21. １ ' and r ２ ' represents the winding resistance of the power supply coil 11 and the power receiving coil 21.ｃ represents the iron loss equivalent resistance. C represents the capacitance (electrostatic capacity) of the capacitor. In the circuit shown in Figure 1, there are two resonant loops ("resonant loop of loop I" and "resonant loop of loop II") through which the resonant current flowing in the multi-resonant circuit can circulate. Regarding frequency, the resonant frequency of loop I is defined as ω ｌ , the resonant frequency of loop II is ω ｈ Let's say.

[0025] Furthermore, as described above, the wireless power feeder utilizing PT symmetry in the first embodiment of the present invention is the same as the conventional one in that the inverter is used in self-oscillation, there are two modes of oscillation, and PT symmetry can be preserved in either mode. It goes without saying that, as shown in Figures 1 and 2, the power feed-side resonant circuit 1 includes a power feed coil 11, and the power receiver-side resonant circuit 2 includes a power receiver coil 21.

[0026] 2(a) and 2(b), the wireless power supply circuit viewed from the output end of the inverter 3 forms a multi-resonant circuit, and two resonant loops exist, but the exact mathematical formulas for the two resonant loops cannot be derived from the circuit diagram shown in Fig. 2. Strictly speaking, this can be determined from coupled mode theory, and the derivation process is as shown in Non-Patent Document 4 (as shown in Non-Patent Document 4 by one of the applicants of the present application), but detailed explanations and the derivation process will be omitted here, and only the results will be described, as shown in Equations (1) and (2).

[0027]

[0028]

[0029] That is, ω shown in equation (1) ｈ and a resonant loop with a resonant frequency of ω ｌ There is a resonant loop with a resonant frequency of ω. And these two resonant loops have a resonant frequency of ω. ｈ >ω ｌ ω ０is the natural angular resonant frequency when the power supply resonant circuit and the power receiving resonant circuit are far apart and not completely coupled, and can be expressed by equation (3). ２０ and Γ Ｌ are all parameters in coupled mode theory (CMT parameters), and as explained in detail in Patent Document 1 (as shown by one of the applicants of the present application in Patent Document 1 and Non-Patent Document 4), detailed explanation of these parameters will be omitted here because they are not very relevant to the explanation of the present invention and the explanation of their definitions would be too long.

[0030]

[0031] Furthermore, k is a parameter called a coupling rate, which is a parameter relating to the number of times energy is exchanged between the power supply side resonant circuit and the power receiving side resonant circuit per unit time. Furthermore, the coupling rate k is calculated by multiplying the magnetic coupling coefficient k between the power supply coil and the power receiving coil by ｍ Using this, it can be expressed as in equation (4).

[0032]

[0033] For example, in the case of the SP topology of FIG. 2B, if equations (1) and (2) are applied, they can be expressed as the following equations (5) and (6).

[0034]

[0035]

[0036] The above formulas (5) and (6) can also be approximately expressed as formulas (7) and (8), and in Patent Document 1, these approximate formulas (7) and (8) are used in the explanation.

[0037]

[0038]

[0039] From equations (5) and (6), the resonant frequency ω of loop II is ｈ and the resonant frequency ω of loop Iｌ is the magnetic coupling coefficient k ｍ In the case of a complex coil shape, the magnetic coupling coefficient k ｍ It is difficult to formulate the relationship between the distance d (transmission distance) between the power supply coil and the power receiving coil, and it is generally obtained by numerical calculation using a computer. ｍ The relationship that is inversely proportional to the transmission distance d between the coils is universal. ｍ 10 is a graph showing a typical example of the relationship between the power supply coil and the transmission distance d between two coils (a power supply coil and a power receiving coil).

[0040] As shown in FIG. 3, the magnetic coupling coefficient k ｍ The relationship between the transmission distance d between the two coils (the power supply coil and the power receiving coil) and the resonant frequency ω of Loop II can be calculated from Equations (5) and (6). ｈ and the resonant frequency ω of loop I ｌ The relationship between the resonant frequency ω of loop II and the transmission distance d between the two coils (the power supply coil and the power receiving coil) can also be calculated. ｈ and the resonant frequency ω of loop I ｌ 10 is a graph showing the relationship between the transmission distance d between two coils (a power supply coil and a power receiving coil) and the power supply distance d between the two coils (a power supply coil and a power receiving coil).

[0041] As shown in FIG. 4, the resonant frequency ω ｈ As the transmission distance d between the two coils increases, the resonant frequency ω of loop I decreases. ｌ becomes higher as the transmission distance d becomes longer. In the example shown in Figure 4, the resonant frequencies of the two resonant loops match when the transmission distance d between the two coils is around 70 mm. Since PT symmetry is not maintained at transmission distances beyond this point, this is excluded from the discussion of the present invention.

[0042] As mentioned above, in the PT symmetric wireless power transfer, the inverter is used in self-oscillation, so the resonant frequency ω ｈ and the resonant frequency ω of loop I ｌOne of the resonant loops is selected and self-oscillation occurs. However, when the transmission distance d fluctuates during operation, the resonant frequency ω of loop II ｈ and the resonant frequency ω of loop I ｌ If the self-oscillation frequency suddenly changes unexpectedly, the power transmission will be interrupted. Also, since the magnitude of the transmitted power and the power transmission efficiency differ depending on the mode, the resonant frequency ω ｈ and the resonant frequency ω of loop I ｌ There is also the problem that a step occurs in the magnitude of the transmitted power due to the switching.

[0043] Therefore, in this invention, even if the transmission distance d varies, if the resonant loop is fixed to either side, the frequency of self-oscillation changes smoothly (continuously), which is the idea behind enabling stable wireless power supply. ｈ or the resonant frequency ω of loop I ｌ The circuit configuration is provided with a phase adjustment circuit for advancing / delaying the phase of the voltage relative to the current of the power supply coil 11, in order to select and fix one of the resonance loops having the above-mentioned phases.

[0044] First, we will explain the circuit configuration of an inverter that does not have a phase adjustment circuit like the present invention. Class D or E inverters are used as inverters for PT-symmetric wireless power transfer. Figure 5 is a circuit diagram of a case where a Class D inverter is connected to a power supply resonant circuit as an inverter for PT-symmetric wireless power transfer. In reality, there is a power receiving resonant circuit that receives power wirelessly from the power supply resonant circuit 1 shown in Figure 5, and a load resistor connected to the power receiving resonant circuit, but these are not shown in Figure 5. Figure 6 is a schematic diagram showing the phase relationship of each waveform of the power supply resonant circuit 1 shown in Figure 5.

[0045] As explained in the section on the prior art, wireless power supply utilizing PT symmetry is a technology in which the AC power supply in conventional magnetic field resonance type wireless power supply technology is replaced with an inverter that behaves electrically in the same way as a negative resistor, that is, an inverter that behaves as a negative resistor.

[0046] An inverter that behaves as a negative resistance is an inverter whose switching frequency and voltage amplitude are not fixed in advance, has a circuit configuration in which the switching frequency is determined by the apparent resonant frequency of the wireless power transfer circuit as seen from the inverter's output end, and has an inverter whose switching frequency quickly responds to changes in the apparent resonant frequency of the wireless power transfer circuit, which can change due to changes in the transmission distance or positional misalignment between two coils (a power transfer coil and a power receiving coil). The wireless power transfer circuit is a circuit that includes a power transfer side resonant circuit, a power receiving side resonant circuit, and all circuits connected to them. Furthermore, since the power transfer side resonant circuit and the power receiving side resonant circuit interact with each other, the apparent resonant frequency refers to the actual resonant frequency that takes this interaction into account.

[0047] 5 , an inverter 3 and a current sensor 4 are connected to the power supply-side resonant circuit 1, and these inverter 3 and current sensor 4 realize a negative resistance circuit. The elements that make up the inverter 3 include a comparator 31, a gate driver 32, a high-side FET (field-effect transistor) 33, and a low-side FET (field-effect transistor) 34. The high-side FET 33 and the low-side FET 34 are switching elements within the inverter 3.

[0048] The comparator 31 in FIG. 5 detects the AC current i flowing through the power supply coil 11 detected by the current sensor 4. １When the detection signal of is input and an input pulse, which is the output of the comparator 31, is input to the gate driver 32, the gate driver 32 outputs two-phase output pulses (a first-phase output pulse and a second-phase output pulse) with a phase difference of 180° to alternately energize / de-energize (turn on / off) the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3. The input pulses input to the gate driver 32 are used to generate the AC current i flowing through the power supply coil 11. １ Alternatively, it is generated based on the AC magnetic field generated in the power supply coil 11 .

[0049] At this time, the switch timing of the output pulses alternately output from the gate driver 32 to the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3, is determined based on the input pulse to the gate driver 32. Here, the "switch timing" refers to the time when the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3, are turned on or off. Then, the AC current i detected by the current sensor 4 installed between the inverter 3 and the power supply side resonant circuit 1 is １ The detection signal is input to the comparator 31, which generates the input pulse described above.

[0050] That is, the drive current of the power supply coil 11 in the power supply side resonant circuit 1, which is the output of the inverter 3, is fed back to the control side of the inverter 3 to control the output current of the inverter 3, which is called "feedback control." １ The detection signal of AC current i １ In principle, the AC voltage signal is in phase with the AC current i １ and the input voltage v of the power supply resonant circuit １ are in phase (no phase difference), positive feedback causes self-oscillation. This self-oscillation excites the power supply coil, realizing wireless power supply.

[0051] However, in an actual circuit, the AC current i １ and input voltage v １In reality, there is a time delay when the voltage signal propagates through the circuit, so the AC current i １ With respect to the input voltage V １ This phase difference is called the initial phase difference. The phase compensation circuit described in Patent Document 1 compensates for this initial phase difference and reduces the AC current i １ and input voltage v １ This is a circuit for making the signals in phase, and its purpose is significantly different from that of the present invention.

[0052] As mentioned above, the AC current i １ and input voltage v １ Although it is theoretically correct to make the AC current i １ and input voltage v １ If the two resonant loops are in phase, the problem of frequent switching between the two resonant loops occurs. １ and input voltage v １ Experiments have shown that if a phase difference is intentionally introduced, one of the resonant loops will become stable.

[0053] FIG. 7 shows the AC current i １ and input voltage v １ 7(a) shows the experimental results showing the phase relationship of the AC current i １ With respect to the input voltage V １ When the phase is delayed, the AC current i １ and input voltage v １ When the AC currents i １ With respect to the input voltage V １ indicates the case where the phase is advanced.

[0054] As shown in FIG. 7(a), the AC current i １ With respect to the input voltage V １ When is a lagging phase, the resonant frequency ω of loop I is ｌ As shown in FIG. 7(b), the resonant loop of AC current i １ and input voltage v １ When the AC current i １ With respect to the input voltage V １When the phase is leading, the resonant frequency of loop II, ω ｈ The resonant loop is selected.

[0055] Next, the AC current i １ and input voltage v １ As described above, the inverter of the wireless power feeder according to the first embodiment of the present invention generates an AC current i １ The inverter 3 is self-oscillated by positively feeding back the current waveform of the inverter 3. Therefore, this inverter can be regarded as an AC power supply with an unfixed oscillation frequency. As shown in FIG. 5, the inverter 3 is connected to the power supply side resonant circuit 1. The power supply side resonant circuit 2 (see FIGS. 1 and 2) and the load resistance R Ｌ (see FIGS. 1 and 2) is also indirectly connected to the inverter 3.

[0056] The circuit seen from the output end of the inverter 3 forms a resonant circuit, and this resonant circuit is represented as a simple equivalent circuit in Figure 8. Figure 8 is a circuit diagram showing an equivalent circuit of a wireless power supply connected to the inverter. Here, the resistance component of the circuit is R, the inductance component is L, and the capacitance component is C. When the inverter has a resonant frequency ω ｃ When self-oscillation occurs, the magnitudes of the inductive reactance and capacitive reactance must satisfy the following equation (9):

[0057]

[0058] Therefore, the reactance component of the resonant circuit seen from the output terminal of the inverter becomes zero, and it becomes equivalent to a circuit in which only resistance R is connected to the inverter. This is the original equivalent circuit when PT symmetry is preserved. And, as mentioned above, in the original equivalent circuit, since only resistance component R is connected to the inverter, the AC current i １ and input voltage v １ However, in this invention, the AC currents i １ and input voltage v １ This forces a phase difference of

[0059] For example, as shown in FIG. 7(a), the AC current i １ With respect to the input voltage V １ If the AC current i is delayed, the equivalent circuit connected to the inverter becomes capacitive. １ With respect to the input voltage V １ When the resonance frequency ω is delayed, the resonance frequency ω becomes ｃ It can also be said that if the inverter is slightly deviated from this, the equivalent circuit connected to the inverter becomes capacitive.

[0060] On the other hand, as shown in FIG. 7(c), the AC current i １ With respect to the input voltage V １ When the resonant frequency ω is advanced, the resonant frequency ω becomes ｃ If the impedance is slightly different from this, the equivalent circuit connected to the inverter will become inductive. The impedance Z of the equivalent circuit connected to the inverter can be expressed by the following equation (10). The inductive condition is given by equation (11).

[0061]

[0062]

[0063] Furthermore, from equation (9), the inductance component L can be expressed as equation (12), and when equation (12) is substituted into equation (11), the inductive condition becomes equation (13).

[0064]

[0065]

[0066] In this way, the AC current i １ With respect to the input voltage V １ When the resonant frequency ω is advanced, the resonant frequency ω becomes ｃ will be slightly higher than

[0067] As mentioned above, the resonant frequency ω ｃ is the resonant frequency ω of loop II ｈ or the resonant frequency ω of loop I ｌ Also, ω ｈ >ω ｌTherefore, ω>ω shown in equation (13) ｃ The sufficient condition for satisfying the relationship is ω>ω ｈ However, ω is ω ｈ Since ω is a slightly higher frequency than ω ｈ The frequency is almost the same as that of the AC current i １ With respect to the input voltage V １ When the resonant frequency of loop II is advanced, ω ｈ Therefore, the resonant loop is selected.

[0068] On the other hand, the capacitive condition is given by equation (14). From equations (9) and (14), the capacitive condition is given by equation (15).

[0069]

[0070]

[0071] In this way, the AC current i １ With respect to the input voltage V １ When the resonance frequency ω is delayed, the resonance frequency ω becomes ｃ will be slightly lower than

[0072] As mentioned above, the resonant frequency ω ｃ is the resonant frequency ω of loop II ｈ or the resonant frequency ω of loop I ｌ Also, ω ｈ >ω ｌ Therefore, ω<ω shown in equation (15) ｃ The sufficient condition for satisfying the relation is ω<ω ｌ However, ω is ω ｌ Since ω is a slightly lower frequency than ω ｌ The frequency is almost the same as that of the AC current i １ With respect to the input voltage V １ When the resonant frequency of loop I is delayed, ｌ Therefore, the resonant loop is selected.

[0073] Therefore, an example of an actual circuit will be described. As mentioned above, the AC current i １ and input voltage v １It has been confirmed that if a phase difference is intentionally created between the resonant frequency ω of loop II, it can be stabilized in one of the resonant loops (see Figs. 16 to 18 for the experimental results). ｈ In order to select and fix the resonant loop of the AC current i １ With respect to the input voltage V １ To advance the phase, a phase lead circuit can be inserted before the AC current detected by the current sensor is input to the comparator. However, a phase lead circuit also functions as a high-pass filter, preferentially passing high-frequency noise components, which worsens the S / N ratio and makes it unsuitable for stable operation.

[0074] On the other hand, the phase delay circuit also functions as a low-pass filter, blocking high-frequency noise components, making it suitable for stable operation. Of course, since it is a phase delay circuit, it is not possible to create a phase lead circuit. Therefore, in an actual circuit, as shown in Figure 9, a phase adjustment circuit 30 is formed by combining a preamplifier 36 and a phase delay circuit 35, thereby realizing phase lead. This allows one of the two resonant loops to be selected and fixed, thereby achieving stable operation.

[0075] 9 is a circuit diagram showing an example of an inverter having a phase adjustment circuit and a power supply side resonant circuit according to embodiment 1 of the present invention. As shown in Fig. 9, an inverter 3 is connected to a power supply side resonant circuit 1, and the elements constituting the inverter 3 are the same as those in the circuit diagram shown in Fig. 5 in that they include a comparator 31, a gate driver 32, a high-side FET (field effect transistor) 33, and a low-side FET (field effect transistor) 34.

[0076] 9 only shows the power supply side resonant circuit 1 and the inverter 3 and current sensor 4 connected thereto, but the premise is the same as in FIGS. 1 and 2 , that is, a magnetic resonance type wireless power feeder that includes a power supply side resonant circuit 1 including a power supply coil 11 and a power receiving side resonant circuit 2 including a power receiving coil 21, and that uses PT symmetry (Parity-Time symmetry) to feed power contactlessly by magnetically resonating the power supply coil 11 and the power receiving coil 21, and that when the power supply side resonant circuit 1 and the power receiving side resonant circuit 2 are considered to be a multi-resonant circuit coupled to each other by mutual inductance, there are two resonant loops (a “resonant loop of loop I” and a “resonant loop of loop II”) through which the resonant current flowing in the multi-resonant circuit can circulate.

[0077] However, in the case of the circuit of FIG. 9, that is, an example of the circuit according to the first embodiment of the present invention, a phase adjustment circuit 30 including a preamplifier (inverting amplifier circuit) 36 and a phase delay circuit 35 is provided before the input to the comparator 31 provided inside the inverter 3, and the AC current i based on the detection of the current or magnetic field of the feeding coil 11 is １ is input to the comparator 31, the phase adjustment circuit 30 (the inverting amplifier circuit 36 ​​and the phase delay circuit 35) converts the AC current i １ is delayed by more than 180° and input to the comparator 31, thereby performing phase control to control it as an advanced phase. This makes it possible to select and fix the "resonance loop of loop II" out of the two resonant loops.

[0078] That is, if the phase adjustment circuit 30 (preamplifier (inverting amplifier circuit) 36 and phase delay circuit 35) in FIG. 9 is not provided, the comparator 31 of the inverter 3 receives the AC current i flowing through the power supply coil 11 detected by the current sensor 4. １When the detection signal of is input and an input pulse, which is the output of the comparator 31, is input to the gate driver 32, the gate driver 32 outputs two-phase output pulses (a first-phase output pulse and a second-phase output pulse) with a phase difference of 180° to alternately energize / de-energize (turn on / off) the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3. The input pulses input to the gate driver 32 are used to generate the AC current i flowing through the power supply coil 11. １ Alternatively, it is generated based on the AC magnetic field generated in the power supply coil 11 .

[0079] However, in FIG. 9, a preamplifier (inverting amplifier circuit) 36 and a phase delay circuit 35 are provided before the input to the comparator 31, and the AC current i １ is input to the comparator 31, the AC current i １ A signal delayed by more than 180° from the input of the comparator 31 is input to the comparator 31. When an input pulse output from the comparator 31 is input to the gate driver 32, the gate driver 32 outputs two-phase output pulses (a first-phase output pulse and a second-phase output pulse) with a phase difference of 180° for alternately energizing / de-energizing (turning on / off) the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3, and the input pulse input to the gate driver 32 is a pulse that is input to the AC current i １ 5. The inverter 3 shown in FIG. 5 is also similar to the inverter 3 shown in FIG.

[0080] At this time, the switch timing of the output pulses alternately output from the gate driver 32 to the high-side FET 33 and low-side FET 34, which are switching elements inside the inverter 3, is determined based on the input pulse to the gate driver 32. Here, as described above, the "switch timing" refers to the time at which the high-side FET 33 and low-side FET 34, which are switching elements inside the inverter 3, are turned on or off. Then, the AC current i detected by the current sensor 4 installed between the inverter 3 and the power supply side resonant circuit 1 is １ The detection signal is phase-adjusted by a preamplifier (inverting amplifier circuit) 36 and a phase delay circuit 35, and the resulting signal is input to a comparator 31, which generates the aforementioned input pulse.

[0081] That is, in the case of FIG. 9 , the switch timing of the output pulses alternately output from the gate driver 32 to the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3, is determined based on an input pulse generated by inputting to the comparator 31 a signal after phase control has been performed by feedback control of an advanced phase or a delayed phase, based on the current phase or magnetic field phase of the power supply coil 11 detected by the current sensor 4, i.e., a signal after the phase relationship has been adjusted by the phase adjustment circuit 30.

[0082] Here, the preamplifier 36 is an inverting amplifier circuit, so that the AC current i １ When the detection signal is input to the preamplifier 36, the phase of the output signal is inverted by 180° relative to the input signal, as shown in FIG. 10. FIG. 10 is a schematic diagram showing the voltage waveforms of each part of the phase adjustment circuit 30 (preamplifier 36 and phase delay circuit 35) shown in FIG. 9. Thereafter, the phase is further delayed at the output of the phase delay circuit (low-pass filter) 35 using an operational amplifier. For example, if a phase delay of 130° is generated in the phase delay circuit 35, the phase delay including the preamplifier 36 is 180 + 130 = 310°. In other words, a delay of one period (360°) results in a phase advance of 360 - 310 = 50°, so an equivalent phase advance can be achieved. That is, AC current i １is delayed by more than 180° and input to the comparator 31.

[0083] 9 illustrates an example in which an inverter 3 and a current sensor 4 are used as a negative resistance circuit connected to the power supply side resonant circuit 1, but a magnetic sensor may be used instead of the current sensor 4. Because the current phase and the magnetic field phase coincide, the current may be detected by the current sensor 4, the current phase may be detected, and the AC current may be detected based on the current phase, or the magnetic field may be detected by the magnetic sensor, the magnetic field phase may be detected, and the AC current may be detected based on the magnetic field phase. In other words, the sensor that constitutes the negative resistance circuit together with the inverter 3 may be any sensor that can detect the current or magnetic field of the power supply coil 11 in the power supply side resonant circuit 1. This also applies to FIGS. 5 and 8, and to FIG. 11 and subsequent figures described later.

[0084] 9 is a device that detects the current or magnetic field of the power supply coil 11 to oscillate the inverter 3 connected to the power supply side resonant circuit 1, and detects zero crossings of the voltage or current of the power supply coil 11 based on the detected current or magnetic field, thereby causing self-oscillation. ｈ In order to select and fix a resonant loop having a resonant loop of loop II, the AC current i detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 is adjusted to advance the phase of the voltage relative to the current of the power supply coil 11. １ Before being input to the comparator 31 in the inverter 3, a preamplifier (inverting amplifier circuit) 36 and a phase delay circuit 35 are inserted to delay the signal by more than 180°, thereby controlling it as a leading phase. In other words, the leading phase is realized by rotating the delay phase by more than 180°, which solves the problem that leading phase control is vulnerable to noise and therefore causes unstable operation.

[0085] As described above, the power supply side resonant circuit 1 is connected to the inverter 3 and a sensor (current sensor 4 in FIG. 9 ) that detects the current or magnetic field of the power supply coil 11. The inverter 3 is equipped with a phase adjustment circuit 30 that performs phase control by feedback control of an advanced phase or a delayed phase based on the current phase based on the current in the power supply coil 11 detected by the sensor (or the magnetic field phase based on the magnetic field of the power supply coil 11). In other words, the phase adjustment circuit 30 can adjust the phase relationship between the AC voltage applied to the power supply side resonant circuit 1 and the AC current flowing through the power supply coil 11. The switch timing, which is the time to turn on or off a switching element in the inverter 3, is determined based on pulses generated by signals after the phase relationship has been adjusted by the phase adjustment circuit 30, thereby selecting and fixing one of the two resonant loops.

[0086] 9 , a comparator 31 is provided inside the inverter 3, and a phase delay circuit 35 serving as the phase adjustment circuit 30 is provided before the input to the comparator 31. In the inverter 3, the phase delay circuit 35 delays the AC current based on the current or magnetic field of the power supply coil 11 detected by a sensor (current sensor 4 in FIG. 9 ) by more than 180° and inputs it to the comparator 31 for the purpose of selecting the “resonance loop II” of the two resonant loops. This delays the AC current and inputs it to the comparator 31, thereby adjusting the phase relationship by leading the phase of the AC voltage applied to the power supply-side resonant circuit 1 relative to the AC current flowing through the power supply coil 11.

[0087] 9, a low-pass filter using an operational amplifier is used as the phase delay circuit 35 as a specific example of the phase adjustment circuit 30. However, if a large phase delay of about 130° is generated by the low-pass filter, the gain (output amplitude / input amplitude) of the low-pass filter may become too small, resulting in a problem of the amplitude of the output signal becoming too small. In such a case, a method of using a time delay circuit instead of the low-pass filter may be used.

[0088] 11 is a circuit diagram showing an example of an inverter having a time delay circuit and a power supply side resonant circuit according to embodiment 1 of the present invention. As shown in Fig. 11, an inverter 3 is connected to a power supply side resonant circuit 1, and the elements constituting the inverter 3 are the same as those shown in Figs. 5 and 9 in that they include a comparator 31, a gate driver 32, a high-side FET (field effect transistor) 33, and a low-side FET (field effect transistor) 34.

[0089] 11, that is, another example of the circuit according to the first embodiment of the present invention, a preamplifier (inverting amplifier circuit) 36 is provided before the input to the comparator 31 provided inside the inverter 3, and a time delay circuit 37 is provided after the output of the comparator 31, so that phase control is performed by controlling the delay time. In the case of FIG. 11, the phase adjustment circuit 30 includes at least the time delay circuit 37.

[0090] That is, the circuit shown in FIG. 11 has a resonant frequency ω ｈ In order to select and fix a resonant loop having a resonant loop of loop II, the AC current i detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 is adjusted to advance the phase of the voltage relative to the current of the power supply coil 11. １ is input to the comparator 31 in the inverter 3, and a time delay circuit 37 is inserted after the output of the comparator 31, and phase control is performed by controlling the delay time.

[0091] A large phase delay of approximately 130° is generated by connecting multiple time delay circuits 37, which combine NOT logic elements with schmutz triggers and RC integrator circuits, downstream of the comparator 31. In other words, real-time control is achieved using NOT circuits and RC integrator circuits to control the switch signal input to the comparator 31, thereby resolving the problem of phase delay in analog circuits being frequency-dependent. Note that the logic elements do not necessarily have to have schmutz triggers. Also, buffer circuits can be used instead of NOT logic elements.

[0092] 11 , the inverter 3 is provided with a comparator 31, and an AC current based on the current or magnetic field of the power supply coil 11 detected by a sensor (current sensor 4 in FIG. 11 ) is input to the comparator 31. A time delay circuit 37 serving as the phase adjustment circuit 30 is provided after the output of the comparator 31. The inverter 3 adjusts the phase relationship by controlling the delay time using the time delay circuit 37, and one of the two resonant loops is selected and fixed as such.

[0093] In the actual circuit examples of FIGS. 9 and 11, a current sensor 4 is attached to the connection line of the power supply coil 11 to detect the current of the power supply coil 11 (the current of the power supply side resonant circuit 1), thereby detecting the AC current i １ The phase of the inverter 3 is detected. A similar effect can be achieved by attaching a magnetic sensor (not shown) near the power supply coil 11 to detect the magnetic field of the power supply coil 11 (the magnetic field of the power supply-side resonant circuit 1). The magnetic sensor may be a Hall element or a magnetic sensor called a loop coil, in which an electric wire is formed into a loop. In this case, the negative resistance circuit described above would consist of the inverter 3 and the magnetic sensor.

[0094] 9, the phase is controlled by inserting a phase delay circuit (low-pass filter) 35 before the comparator 31, while in Fig. 11, the phase is controlled by inserting a time delay circuit 37 after the comparator 31. As another possible embodiment, a circuit capable of controlling only the phase, called an all-pass filter, may be used instead of the time delay circuit 37 shown in Fig. 11.

[0095] Fig. 12 is a circuit diagram showing an example of an inverter having an all-pass filter and a power supply side resonant circuit according to embodiment 1 of the present invention. As shown in Fig. 12, an inverter 3 is connected to a power supply side resonant circuit 1, and the elements constituting the inverter 3 are the same as those shown in Figs. 5, 9 and 11 in that they include a comparator 31, a gate driver 32, a high-side FET (field effect transistor) 33 and a low-side FET (field effect transistor) 34.

[0096] However, in the case of the circuit of FIG. 12 , that is, in yet another example of the circuit according to the first embodiment of the present invention, a preamplifier (inverting amplifier circuit) 36 is provided before the input to the comparator 31 provided inside the inverter 3, and an all-pass filter 38 is provided after the output of the comparator 31, and phase control is performed by this all-pass filter 38.

[0097] That is, the circuit shown in FIG. 12 has a resonant frequency ω ｈ In order to select and fix a resonant loop having a resonant loop of loop II, the AC current i detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 is adjusted to advance the phase of the voltage relative to the current of the power supply coil 11. １ is input to a comparator 31 in the inverter 3, and an all-pass filter 38 is inserted after the output of the comparator 31, and phase control is performed by this all-pass filter 38.

[0098] 12, the inverter 3 is provided with a comparator 31 inside, and an AC current based on the current or magnetic field of the power supply coil 11 detected by a sensor (current sensor 4 in FIG. 12) is input to the comparator 31. An all-pass filter 38 serving as a phase adjustment circuit 30 is provided after the output of the comparator 31. The inverter 3 controls the adjustment of the phase relationship by the all-pass filter 38, and thereby selects and fixes one of the two resonant loops.

[0099] Furthermore, as shown in Fig. 13, a method of controlling the phase using a PLL (phase-locked loop) 39 instead of the time delay circuit 37 shown in Fig. 11 or the all-pass filter 38 shown in Fig. 12 may be considered. Fig. 13 is a circuit diagram showing an example of an inverter having a PLL (phase-locked loop) and a power supply-side resonant circuit according to the first embodiment of the present invention. As shown in Fig. 13, an inverter 3 is connected to the power supply-side resonant circuit 1, and the elements constituting the inverter 3 are the same as those shown in Figs. 5, 9, 11, and 12, including a comparator 31, a gate driver 32, a high-side FET (field-effect transistor) 33, and a low-side FET (field-effect transistor) 34.

[0100] However, in the case of the circuit of FIG. 13, that is, in yet another example of the circuit according to the first embodiment of the present invention, a preamplifier (inverting amplifier circuit) 36 is provided before the input to the comparator 31 provided inside the inverter 3, and a PLL (phase-locked loop) 39 is provided after the output of the comparator 31, and phase control is performed by this PLL (phase-locked loop) 39.

[0101] That is, the circuit shown in FIG. 13 has a resonant frequency ω ｈ In order to select and fix a resonant loop having a resonant loop of loop II, the AC current i detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 is adjusted to advance the phase of the voltage relative to the current of the power supply coil 11. １ is input to a comparator 31 in the inverter 3, and a PLL (phase locked loop) 39 is inserted after the output of the comparator 31, and phase control is performed by controlling the phase lead / lag using this PLL (phase locked loop) 39.

[0102] 13 , a comparator 31 is provided inside the inverter 3, and an AC current based on the current or magnetic field of the power supply coil 11 detected by a sensor (current sensor 4 in FIG. 13 ) is input to the comparator 31. A PLL (phase-locked loop) 39 serving as a phase adjustment circuit 30 is provided after the output of the comparator 31. The inverter 3 controls the phase relationship by using the PLL (phase-locked loop) 39 to advance or delay the phase of the AC voltage applied to the power supply-side resonant circuit 1 relative to the AC current flowing through the power supply coil 11, thereby selecting and fixing one of the two resonant loops.

[0103] 14 is a block diagram showing an example of specific internal functions of the PLL (phase-locked loop) 39. As shown in FIG. 14, the PLL (phase-locked loop) 39 includes at least a PFD (phase detector) 91 and a VCO (voltage-controlled oscillator) 92. The PFD (phase detector) 91 calculates the phase difference between an input signal serving as a reference frequency and a feedback signal output from the VCO (voltage-controlled oscillator) 92, whose frequency changes depending on the voltage, and inputs this phase difference to the VCO (voltage-controlled oscillator) 92 to synchronize the phases of the input signal and the output signal. Therefore, by adding a certain offset voltage to the input voltage of the VCO (voltage-controlled oscillator) 92, it is possible to limit the frequency range of the output signal of the PLL (phase-locked loop) 39.

[0104] For example, the resonant frequency ω of Loop II ｈ The frequency range of ω is 70 kHz to 90 kHz, and the resonant frequency of loop I is ω ｌ is 50 kHz to 70 kHz, the resonant frequency ω of loop I can be adjusted by setting the frequency range of the output signal of PLL (phase locked loop) 39 to 70 kHz to 90 kHz. ｌ Therefore, the resonant frequency of Loop II, ω ｈThat is, by controlling the input voltage of the VCO (voltage controlled oscillator) 92, the oscillation frequency of the VCO (voltage controlled oscillator) 92 is limited, and the frequency range of the output signal of the PLL (phase locked loop) 39 is limited, thereby improving the operational stability when one of the two resonant loops is selected.

[0105] Here, a signal obtained by inverting an AC signal based on the current detected by the current sensor 4 shown in FIG. 13 using a preamplifier (inverting amplifier circuit) 36 is input to a comparator 31, and the input pulse generated by the comparator 31 is the input signal to a PLL (phase-locked loop) 39. Therefore, when the output signal of the PLL (phase-locked loop) 39 is input to the gate driver 32, the frequency or phase can be controlled by controlling the input voltage of the VCO (voltage-controlled oscillator) 92 (controlling the control value of the VCO 92), compared to when phase control is not performed in a positive feedback mode. Utilizing this, the resonant frequency ω of loop II is set as the resonant frequency of the power supply side resonant circuit 1. ｈ or the resonant frequency ω of loop I ｌ Alternatively, a limit may be imposed on the oscillation frequency of the VCO (Voltage Controlled Oscillator) 92 of the PLL (Phase Locked Loop) 39, and the resonant frequency ω ｈ Only the resonant frequency ω of loop I oscillates. ｌ It is also possible to control the oscillation so that only the

[0106] In this way, a limit is placed on the fluctuation range of the oscillation frequency of the VCO (Voltage Controlled Oscillator) 92 in the PLL (Phase Locked Loop) 39, and the resonant frequency ω ｌ If we make it so that oscillation is absolutely impossible, the resonant frequency ω of Loop II will inevitably be ｈ Therefore, the resonant frequency ω of loop II is forced to ｈIn other words, by using a PLL (phase locked loop) 39 that imposes a frequency limit on a VCO (voltage controlled oscillator) 92, the output frequency of the PLL (phase locked loop) 39 can be limited, so that the resonant loop can be fixed to either one of the frequencies, and unstable switching can be prevented.

[0107] This prevents the problem of random switching of the current loop due to fluctuations in the positions of the power supply coil and power receiving coil during wireless power supply. As described above, adding the PLL (phase-locked loop) 39 not only enables phase control by the PLL, but also has the advantage of improving operational stability, since phase fluctuations and the like can be reduced and stable operation can be expected.

[0108] Furthermore, as shown in Fig. 15, instead of the time delay circuit 37 shown in Fig. 11, the all-pass filter 38 shown in Fig. 12, or the PLL (phase-locked loop) 39 shown in Fig. 13, a method of controlling the phase using both an all-pass filter 38 and a PLL (phase-locked loop) 39 may be considered. Fig. 15 is a circuit diagram showing an example of an inverter and a power supply-side resonant circuit having an all-pass filter and a PLL (phase-locked loop) according to the first embodiment of the present invention. As shown in Fig. 15, an inverter 3 is connected to the power supply-side resonant circuit 1, and the elements constituting the inverter 3 are the same as those shown in Figs. 5, 9, 11, 12, and 13, including a comparator 31, a gate driver 32, a high-side FET (field-effect transistor) 33, and a low-side FET (field-effect transistor) 34.

[0109] FIG. 16 shows the actual device according to the first embodiment of the present invention, in which the AC current i １ With respect to the input voltage V １ When adjusted to be a leading phase, the AC current i １ and input voltage v １ 1 is a graph showing the measured waveform of the AC current i １ , the solid line is the input voltage v １16 shows the phase difference between the AC voltage applied to the power supply-side resonant circuit 1 and the AC current flowing through the power supply coil 11. Fig. 17 is a graph showing the results of measuring the change in resonant frequency when the distance between the power supply coil 11 provided in the power supply-side resonant circuit 1 and the power receiving coil 21 provided in the power receiving-side resonant circuit 2, i.e., the transmission distance d between these two coils, is changed in the phase relationship shown in Fig. 16, where ● marks indicate experimental results and dashed lines indicate calculation results.

[0110] In the experiment, when the phase difference shown in FIG. 16 is 30 to 35 degrees, that is, when the AC voltage v １ AC current i １ When the phase is advanced by 30 to 35 degrees with respect to ｈ 17, it was confirmed that the resonant frequency ω of loop II was always selected and fixed regardless of the transmission distance, i.e., even when the transmission distance d was changed. ｈ It can be seen that the resonant loop is selected and fixed.

[0111] Figure 18 is a graph showing the results of measuring the transmitted power when the transmission distance d between the two coils is changed, with the ● marks indicating the experimental results and the dashed line indicating the calculation results. Looking at Figure 18, it can be seen that power can always be supplied regardless of the transmission distance, that is, even when the transmission distance d is changed.

[0112] As described above, according to the present invention, in a magnetic resonance type wireless power feeder that utilizes PT symmetry and feeds power contactlessly by magnetically resonating a feed coil and a receiver coil, the provision of a phase adjustment circuit for adjusting the phase makes it possible to select and fix a resonance loop without being influenced by the surrounding environment and without imposing constraints on the design of the feed coil and the receiver coil. That is, it is possible to select and fix one of two resonance loops without being influenced by the surrounding environment and without being affected by the selection of a frequency suitable for wireless power feeding or the core shape or material of the coil (no adjustments in coil design are necessary), and without imposing constraints on the design of the feed coil and the receiver coil, and the coils can be made to have any shape and dimensions.

[0113] It should be noted that, within the scope of the present invention, any of the components of the embodiments may be modified or omitted.

[0114] The magnetic resonance type wireless power feeder of this invention can be widely applied to wireless power feed in a wide variety of environments, including not only short-range wireless power feed such as charging a mobile phone, but also environments with water around, such as underwater drones, and environments with a lot of metal around, such as power transmission to equipment in a factory.

[0115] REFERENCE SIGNS LIST 1 Power supply side resonant circuit 2 Power receiving side resonant circuit 3 Inverter 4 Current sensor 11 Power supply coil 21 Power receiving coil 30 Phase adjustment circuit 31 Comparator 32 Gate driver 33 High side FET 34 Low side FET 35 Phase delay circuit 36 ​​Preamplifier (inverting amplifier circuit) 37 Time delay circuit 38 All-pass filter 39 PLL (phase locked loop) 91 PFD (phase detector) 92 VCO (voltage controlled oscillator)

Claims

1. A magnetic resonance type wireless power supply device comprising a power supply side resonant circuit including a power supply coil and a power receiving side resonant circuit including a power receiving coil, which utilizes parity-time symmetry to supply power in a non-contact manner by magnetically resonating the power supply coil and the power receiving coil, wherein when the power supply side resonant circuit and the power receiving side resonant circuit are regarded as a multiple resonant circuit coupled to each other by mutual inductance, there are two resonant loops ("resonant loop I" and "resonant loop II") through which a resonant current flowing in the multiple resonant circuit can circulate, an inverter and a sensor that detects the current or magnetic field of the power supply coil are connected to the power supply side resonant circuit, and the inverter comprises a phase adjustment circuit that adjusts the phase relationship between the AC voltage applied to the power supply side resonant circuit and the AC current flowing in the power supply coil based on a current phase based on the current of the power supply coil detected by the sensor or a magnetic field phase based on the magnetic field of the power supply coil, a switch timing, which is the time to turn on or off a switching element inside the inverter, is determined based on a pulse generated by a signal after the phase relationship has been adjusted by the phase adjustment circuit, thereby selecting and fixing one of the two resonant loops.

2. The magnetic resonance type wireless power feeder according to claim 1, characterized in that a comparator is provided inside the inverter, and a phase delay circuit serving as the phase adjustment circuit is provided before the input to the comparator, and the inverter performs control to adjust the phase relationship by causing the phase delay circuit to delay the AC current based on the current or magnetic field of the power feed coil detected by the sensor by more than 180° and input the AC current to the comparator before the AC current is input to the comparator, for the purpose of selecting "resonance loop II" out of the two resonant loops, so as to make the phase of the AC voltage applied to the power feed side resonant circuit advance relative to the AC current flowing in the power feed coil.

3. The magnetic resonance type wireless power supply device according to claim 1, characterized in that a comparator is provided inside the inverter, and an AC current based on the current or magnetic field of the power supply coil detected by the sensor is input to the comparator, and a time delay circuit serving as the phase adjustment circuit is provided after the output of the comparator, and the inverter controls to adjust the phase relationship by controlling the delay time by the time delay circuit.

4. The magnetic resonance type wireless power supply device according to claim 1, characterized in that a comparator is provided inside the inverter, and an AC current based on the current or magnetic field of the power supply coil detected by the sensor is input to the comparator, and an all-pass filter is provided as the phase adjustment circuit after the output of the comparator, and the inverter controls to adjust the phase relationship by the all-pass filter.

5. The magnetic resonance type wireless power feeder according to claim 1, characterized in that a comparator is provided inside the inverter, and an AC current based on the current or magnetic field of the power feed coil detected by the sensor is input to the comparator, and a PLL (phase-locked loop) is provided as the phase adjustment circuit after the output of the comparator, and the inverter performs control to adjust the phase relationship by using the PLL (phase-locked loop) to advance or delay the phase of the AC voltage applied to the power feed side resonant circuit with respect to the AC current flowing in the power feed coil.

6. The magnetic resonance type wireless power supply device according to claim 5, characterized in that the PLL (phase-locked loop) comprises at least a VCO (voltage-controlled oscillator), and limits the oscillation frequency of the VCO (voltage-controlled oscillator) by controlling the input voltage of the VCO (voltage-controlled oscillator), thereby limiting the frequency range of the output signal of the PLL (phase-locked loop).

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