A reconfigurable bidirectional series-resonant DC-DC power converter

Abstract

A reconfigurable bidirectional series-resonant DC-DC power converter apparatus has a LC resonant tank split on either side of a transformer that prevents DC bias introduced by the input switching circuit and the full bridge output switching circuit from effecting the transformer. The converter operates the full bridge switching circuit to reconfigure the converter topology. When the converter gain is less than a nominal gain, the converter is reconfigured as a half bridge / full bridge topology. When the converter gain is greater than or equal to the nominal gain and the output voltage is less than a maximum voltage, the converter is reconfigured as a half bridge / semi-full bridge topology. When the output voltage is greater than or equal to the maximum voltage the converter is reconfigured as a half bridge / half bridge topology. Converter reconfiguration allows an ultra-wide output voltage rage to be handled efficiently without increased component count.

Description

A RECONFIGURABLE BIDIRECTIONAL SERIES-RESONANT DC-DC POWER CONVERTERTECHNICAL FIELD

[0001] The aspects of the disclosed embodiments relate generally to power conversion apparatus and more particularly to series resonant DC-DC power converters.BACKGROUND

[0002] DC-to-DC power converters are used in many power systems. For example, a DC-to-DC conversion stage is often included in bi-directional AC -DC power converter systems used in electric vehicle (EV) charging / discharging applications. EV power systems are often required to support wide output voltage variations such as output voltage variation from 200 volts to 1000 volts yielding output voltage variation of one to five times the input voltage.

[0003] Various conventional converter solutions have been proposed. Some use extra components to support wide output range and bi-directional power flow. Others incorporate power relays at the converter output or use extra magnetic and switching components to reconfigure the impedance network. Unfortunately, the extra components are undesirable and reduce power density, reduce converter efficiency, and increase cost.

[0004] Some success has been achieved using multi-mode control to operate asymmetric dual-active-bridge series-resonant-converters (DAB SRC). However, the existing converter topologies and their control methods perform best with limited output ranges such as one to two times the input voltage.

[0005] Thus, there is a need for improved reconfigurable DC-DC series-resonant power converters that can efficiently handle ultra-wide range voltage variations and provide improved component utilization without unnecessarily increasing component count. Accordingly, it would be desirable to provide methods and apparatus that addresses at least some of the problems described above.SUMMARY

[0006] The aspects of the disclosed embodiments are directed to a reconfigurable bidirectional series-resonant DC-DC power converter apparatus adapted to provide an ultrawide voltage range capability while improving converter efficiency without increasingcomponent count. The aspects of the disclosed embodiments achieve improved performance and avoid increased component count by operating an existing full bridge switching circuit to reconfigure the converter topology and achieve efficient operation for different output voltage ranges without adding components.

[0007] According to a first aspect, the above and further objectives and advantages are obtained by an apparatus that includes: a transformer having a primary winding magnetically coupled to a secondary winding; a primary series resonant circuit which includes a primary inductor, a primary capacitor, and the primary winding connected in series; a secondary series resonant circuit which includes a secondary inductor, a secondary capacitor, and the secondary winding connected in series; a primary switching circuit coupled between the primary series resonant circuit and an input voltage; a first half bridge switching circuit coupled in parallel with an output voltage; a second half bridge switching circuit coupled in parallel with the output voltage; and a controller. The secondary series resonant circuit is coupled between a central node of the first half bridge switching circuit and a central node of the second half bridge switching circuit. The controller is configured to sense the input voltage and the output voltage, and calculate a converter gain based at least in part on the input voltage and the output voltage. When the converter gain is greater than a predetermined nominal converter gain and the output voltage is less than a predetermined maximum output voltage, the controller is configured to operate the first half bridge switching circuit at a first duty cycle of fifty percent and operate the second half bridge switching circuit at a second duty cycle, wherein the second duty cycle is greater than the first duty cycle.

[0008] In a possible implementation form, the first half bridge switching circuit comprises a first upper switching device and a first lower switching device connected in series to form a first central node, the second half bridge switching circuit comprises a second upper switching device and a second lower switching device connected in series to form a second central node, and the secondary series resonant circuit is coupled between the first central node and the second central node. The second duty cycle corresponds to an on-time of the second lower switching device. Increasing the duty cycle of the lower switching device slices the output voltage and lowers the effective voltage across the impedance thereby avoiding performance degradation caused by increased circulating currents.

[0009] In a possible implementation form, the predetermined nominal converter gain corresponds to one half of the predetermined maximum output voltage. Setting the nominal converter gain to half of the maximum output voltage allows larger output voltage variations to be handled while maintaining acceptable converter efficiency.

[0010] In a possible implementation form, the predetermined nominal converter gain is between zero point nine five (0.95) and zero point nine nine (0.99). Setting the predetermined nominal converter gain close to one, allows the more efficient half bridge / full bridge operating mode to be employed for lower output voltages and changing to a more efficient operating mode when the converter circulating currents begin to impact performance.

[0011] In a possible implementation form, when the converter gain is greater than the predetermined nominal converter gain and the output voltage is less than a predetermined maximum output voltage, the second duty cycle is linearly proportional to the output voltage. Using a linearly proportional relationship maintains the equivalent nominal converter gain across the impedance near one throughout the half bridge / semi-full bridge (HB / S-FB) operation range.

[0012] In a possible implementation form, when the output voltage is greater than a predetermined minimum output voltage and the converter gain is less than or equal to the predetermined nominal converter gain, the controller is configured to operate the first half bridge switching circuit and the second half bridge switching circuit as a full bridge switching circuit. When operating below the nominal converter gain, the half bridge / full bridge (HB / FB) operating mode is the most efficient mode.

[0013] In a possible implementation form, when the output voltage is greater than or equal to the maximum output voltage, the controller is configured to operate the full bridge switching circuit in a half bridge mode, where the half bridge mode is achieved by turning the second upper switching device off, turning the second lower switching device on, and operating the first half bridge switching circuit at a fifty percent (50%) duty cycle. At higher output voltages the half bridge / half bridge (HB / HB) operating mode provides the best performance.

[0014] In a possible implementation form, the transformer comprises a turn ratio configured to produce the predetermined nominal converter gain when the output voltage is one half of the predetermined maximum output voltage. Selecting the nominal converter gain in this fashion provides a suitable trade-off between supporting an ultra- wide range of output voltages and converter performance throughout the operating range.

[0015] According to a second aspect, the above and further objectives and advantages are obtained by a method for operating a power conversion apparatus. The power conversion apparatus includes: a transformer having a primary winding magnetically coupled to a secondary winding; a primary series resonant circuit including a primary inductor, a primary capacitor, and the primary winding connected in series; a secondary series resonant circuit including a secondary inductor, a secondary capacitor, and the secondary winding connected inseries; a primary switching circuit coupled between the primary series resonant circuit and an input voltage; a first half bridge switching circuit coupled between the secondary series resonant circuit and an output voltage; and a second half bridge switching circuit coupled between the secondary series resonant circuit and the output voltage. The method includes: sensing the input voltage and the output voltage; calculating a converter gain based at least in part on the input voltage and the output voltage; and when the converter gain is greater than a predetermined nominal converter gain and the output voltage is less than a predetermined maximum output voltage, operating the first half bridge switching circuit at a first duty cycle of fifty percent (50%) and operating the second half bridge switching circuit at a second duty cycle, wherein the second duty cycle is greater than the first duty cycle.

[0016] In a possible implementation form, the predetermined nominal converter gain corresponds to one half of the predetermined maximum output voltage. Setting the nominal converter gain to half the maximum output voltage allows larger output voltage variations to be handled while maintaining acceptable converter efficiency.

[0017] In a possible implementation form, the predetermined nominal converter gain is between zero point nine five (0.95) and zero point nine nine (0.99). Setting the predetermined nominal converter gain close to one, allows the more efficient HB / FB operating mode to be employed for lower output voltages and changing to a more efficient operating mode when the converter circulating currents begin to impact performance.

[0018] In a possible implementation form, when the converter gain is greater than the predetermined nominal converter gain and the output voltage is less than a predetermined maximum output voltage, the second duty cycle is linearly proportional to the output voltage. Using a linearly proportional relationship maintains the nominal converter gain across the impedance near one throughout the HB / S-FB operation range.

[0019] In a possible implementation form, when the output voltage is greater than a predetermined minimum output voltage and the converter gain is less than or equal to the predetermined nominal converter gain, the controller is configured to operate the first half bridge switching circuit and the second half bridge switching circuit as a full bridge switching circuit. When operating below the nominal converter gain, the full bridge operating mode provides the best performance.

[0020] In a possible implementation form, when the output voltage is greater than or equal to the maximum output voltage, the controller is configured to operate the full bridge switching circuit in a half bridge mode, where the half bridge mode is achieved by turning the second upper switching device off, turning the second lower switching device on, and operatingthe first half bridge switching circuit at a fifty percent (50%) duty cycle. At higher output voltages the HB / HB operating mode provides the best performance.

[0021] In a possible implementation form, the transformer comprises a turn ratio configured to produce the predetermined nominal converter gain when the output voltage is one half of the predetermined maximum output voltage. Selecting the nominal converter gain in this fashion provides a suitable trade-off between supporting an ultra- wide range of output voltages and improving converter performance throughout the operating range.

[0022] These and other aspects, implementation forms, and advantages of the exemplary embodiments will become apparent from the embodiments described herein considered in conjunction with the accompanying drawings. It is to be understood, however, that the description and drawings are designed solely for purposes of illustration and not as a definition of the limits of the disclosed invention, for which reference should be made to the appended claims. Additional aspects and advantages of the invention will be set forth in the description that follows, and in part will be obvious from the description, or may be learned by practice of the invention. Moreover, the aspects and advantages of the invention may be realized and obtained by means of the instrumentalities and combinations particularly pointed out in the appended claims.BRIEF DESCRIPTION OF THE DRAWINGS

[0023] In the following detailed portion of the present disclosure, the invention will be explained in more detail with reference to the example embodiments shown in the drawings, in which like references indicate like elements and:

[0024] Figure 1 illustrates a diagram of an exemplary reconfigurable bidirectional series-resonant DC-DC power converter apparatus incorporating aspects of the disclosed embodiments.

[0025] Figure 2 illustrates graphs showing key operating waveforms of an exemplary series-resonant DC-DC power converter topology during a first operating mode with power flow from the HV-side to the LV-side incorporating aspects of the disclosed embodiments.

[0026] Figure 3 illustrates graphs depicting the key operating waveforms of a series- resonant DAB power converter apparatus configured for HB / S-FB operation incorporating aspects of the disclosed embodiments.

[0027] Figure 4 illustrates graphs depicting duty cycle and converter factor variations of the second half bridge switching circuit incorporating aspects of the disclosed embodiments.

[0028] Figure 5 illustrates graphs depicting the key operating waveforms of a series- resonant DAB power converter apparatus configured for HB / HB operation incorporating aspects of the disclosed embodiments.

[0029] Figure 6 illustrates a flow chart of an exemplary method for operating a reconfigurable bidirectional series-resonant DC-DC power converter incorporating aspects of the disclosed embodiments.DETAILED DESCRIPTION OF THE DISCLOSED EMBODIMENTS

[0030] Figure 1 illustrates a diagram of an exemplary reconfigurable bidirectional series-resonant DC-DC power converter apparatus 100 incorporating aspects of the disclosed embodiments. The exemplary power conversion apparatus 100 of the disclosed embodiments is directed to an improved series-resonant dual active-bridge (SRDAB) power conversion apparatus 100 incorporating a multi-mode control scheme operable to reconfigure the full bridge switching circuit 104 in accordance with variations of the converter gain (M) and output voltage Vo. The apparatus 100 is appropriate for use as a DC-DC conversion stage in power conversion systems such as electric vehicle (EV) charging / discharging systems and AC -DC rectifier systems designed to convert grid power to conditioned DC power for powering telecom equipment, or DC-AC inverter systems used to convert DC power to AC power as used in photovoltaic systems.

[0031] As shown in Figure 1, in one embodiment, the apparatus 100 includes a transformer T having a primary winding 106 that magnetically coupled to a secondary winding 108. A primary series resonant circuit Zpincludes a primary inductor Lpand a primary capacitor Cp, with the primary winding 106 connected in series. A secondary series resonant circuit Zsincludes a secondary inductor Lsand a secondary capacitor G, with the secondary winding connected in series. A primary switching circuit 102 is coupled between the primary series resonant circuit Zpand an input voltage Vi.

[0032] A first half bridge switching circuit 110 is coupled in parallel with an output voltage and a second half bridge switching circuit coupled in parallel with the output voltage Vo. The secondary series resonant circuit Zsis coupled between a central node 120 of the first half bridge switching circuit 110 and a central node 122 of the second half bridge switching circuit.

[0033] A controller 118 is configured to sense the input voltage Vi and the output voltage Vo, and calculate a converter gain M based at least in part on the input voltage Vi and the output voltage Vo. When the converter gain M is greater than a predetermined nominalconverter gain M^ / and the output voltage Vois less than a predetermined maximum output voltage Vo, max. The controller 118 is configured to operate the first half bridge switching circuit 110 at a first duty cycle I)q2 of fifty percent and operate the second half bridge switching circuit 112 at a second duty cycle Dq4. The second duty cycle I)qs greater than the first duty cycle Dq2.

[0034] Generally, for optimal and efficient performance, a DC-to-DC conversion stage should be able to efficiently operate under ultra- wide-range output voltage variations, such as one to five times the input voltage. Low switching losses may be achieved through zero- voltage-switching, and low conduction losses may be achieved by reducing current through high-frequency impedance networks and by maintaining the effective voltage gain across the impedance network near one. Lower magnetic losses may be achieved by reducing effective voltage across the transformer. It is also beneficial to provide these efficiencies while maintaining a low component count and efficient component utilization.

[0035] The primary half bridge switching circuit 102 of the apparatus 100 configured to couple with an input voltage Vi on a primary side 128 of the power converter apparatus 100. The full bridge switching circuit 104 is configured to couple with an output voltage Voon a secondary side 130 of the power converter apparatus 100.

[0036] In alternate embodiments, any suitable switching circuit, such as the half bridge switching circuit illustrated in Figure 1, capable of transferring power between the DC input voltage Vi and the AC primary voltage vpmay be advantageously employed as the primary half bridge switching circuit 102. As will be discussed further below, in certain operating modes it is useful to operate the full bridge switching circuit 104 as two half bridge switching circuits, where a first half bridge switching circuit 110 is coupled in parallel with a second half bridge switching circuit 112.

[0037] Control of the primary half bridge switching circuit 102 and the full bridge switching circuit 104 is provided by a controller 118 configured to generate a first set of control signals 124 to operate the primary half bridge switching circuit 102 and a second set of control signals 126 to operate the full bridge switching circuit 104. Efficient operation throughout an ultra- wide range of output voltages is provided by adapting the controller 118 to reconfigure the full bridge switching circuit 104 as a half bridge (HB) switching circuit, a full bridge (FB) switching circuit, or a semi-full bridge (S-FB) switching circuit based on the desired operating mode. When operating the converter apparatus 100 in a HB / S-FB mode, the first half bridge switching circuit 110 is operated as a half bridge switching circuit and the second half bridgeswitching circuit 112 is used, as will be discussed further below, to improve converter efficiency by reducing the effective secondary AC voltage vsduring certain operating conditions.

[0038] A series resonant tank circuit Z couples the primary half bridge switching circuit 102 with the full bridge switching circuit 104, and includes a primary series resonant circuit Zpon a primary side 128 of a transformer T and a secondary series resonant circuit Zson a secondary side 130 of the transformer T. The primary series resonant circuit Zpincludes a primary inductor Lp, a primary capacitor Cp, and the primary winding 106 of the transformer T connected in series, where the transformer Thas a turns ratio between the primary winding 106 and the secondary winding 108 of N to one, N:l. The secondary series resonant circuit Zsincludes a secondary inductor Ls, a secondary capacitor G, and the secondary winding 108 of the transformer T connected in series. As will be discussed further below, splitting the resonant tank Z into two parts Zp, Zsand distributing the parts on either side 106, 108 of the transformer T, prevents DC bias which may be created by the primary half bridge switching circuit 102 or the full bridge switching circuit 104 when configured to semi-half bridge or half-bridge form from adversely impacting operation of the transformer T.

[0039] It should be understood that the terms input voltage, and output voltage, are used herein as an aid to understanding only and should not be construed as limiting operation of the converter apparatus 100. Those skilled in the art will readily recognize that the power conversion apparatus 100 is capable of bi-directional operation and may be operated to transfer power in either direction (input voltage Vi to output voltage Vo, or output voltage Voto input voltage Vo) by varying the phase relation 0 between the primary half bridge switching circuit 102 and the full bridge switching circuit 104.

[0040] In the exemplary converter apparatus 100, two series connected switching devices Si, S2 form the primary half bridge switching circuit 102, and four switching devices Qi, Q2, 0.3, Q4 connected in an H-bridge configuration, also referred to herein as two parallel half bridge switching circuits, form the full bridge switching circuit 104. In the exemplary apparatus 100 illustrated in Figure 1, the switching devices Si, S2, Qi, Q2, Q3, Qi are illustrated as metal oxide semiconductor field effect transistors (MOSFET) having a parasitic capacitance Cossand substrate diode D. However, any suitable switching device, capable of switching the desired power at the desired frequencies, may be advantageously employed as the switching devices Si, S2, Qi, Q- Q3, Q4 without straying from the spirit and scope of the present disclosure.

[0041] As used herein, a switch or switching device is referred to as ‘on’ or ‘turned on’ when it is conducting current, and is referred to as ‘off or ‘turned off when it is not conducting current.

[0042] When desired, a first link capacitor Q may be coupled in parallel with the input voltage Vi. Similarly, a second link capacitor Comay be coupled in parallel with the output voltage Vo. Including link capacitors may be desirable to provide smoothing, or removal of higher frequency components, of the DC input and DC output power.

[0043] Figure 2 illustrates graphs depicting key operating waveforms 200 of an exemplary SRDAB power converter when operated in a HB / FB configuration incorporating aspects of the disclosed embodiments. The illustrated operating waveforms 200 are representative of a SRDAB power converter such as the exemplary power converter apparatus 100 described above and illustrated in Figure 1. In the illustrated graphs 200, time is depicted along horizontal axes 202, 204, 206 increasing to the right. Graph 210 shows the on or off state of switching devices Si, S2 in the primary half bridge switching circuit 102, and graph 212 shows the on or off state of the switching devices Qi, Q2, Q3, Qi in the full bridge switching circuit 104.

[0044] Graph 214 depicts the primary voltage vp, the secondary voltage vs, and the primary current ipduring one full cycle of the switching circuits where the cycle is depicted along the horizontal time axis 206 between time t=0 and t=T=2n. The primary voltage vpis generated by the primary half bridge switching circuit 102 and varies between zero and the DC input voltage Vi, while the secondary voltage vsis controlled by the full bridge switching circuit 104 and varies between plus and minus the DC output voltage Vo. Current through the primary inductor ipfollows a roughly sinusoidal waveform. Note that magnitude of the current through the secondary inductor istracks the same waveform as the primary current ipand differs in magnitude from the primary current ipby the transformer turn ratio (1 / ).

[0045] While operating as a HB / FB converter, the duty cycle Dp, Dsof both the primary half bridge switching circuit 102 and the full bridge switching circuit 104 are maintained at fifty percent (50%), where each switching device Si, S2, Qi, Q2, Q3, Qi is on for about half the time with a small dead time tai, t i2 included between state transition to avoid shoot through currents.

[0046] When the illustrated cycle 214 begins, at time t=0, the upper switching device Si is on and the lower switching device S2 is off creating a primary voltage vpequal to the input voltage Vi. On the secondary side 130, the first upper switching device Qi and the second lower switching device Q4 are off and the first lower switching device Q2 and the second upperswitching device Q3 are on thereby creating a secondary voltage vsequal to the negative output voltage, -Vo.

[0047] After a phase delay 0, at time t2, the full bridge switching circuit 104 changes state with the first upper switching device Qi and the second lower switching device Q4 turning on and the first lower switching device Q2 and the second upper switching device Q3 turning off causing the secondary voltage vsto rise to the output voltage Vo. In the graphs 200 the phase delay 0 is positive resulting in a power flow from the input voltage Vi to the output voltage Vo. A negative phase delay 0 provides reverse power flow from the output voltage Voto the input voltage Vi.

[0048] The HB / FB configuration of the exemplary converter apparatus 100 is referred to herein as operating mode a, and occurs when both the primary half bridge switching circuit 102 and the full bridge switching circuit 104 are operated at the same switching frequency with a fifty percent (50%) duty cycle. It can be shown that the HB / FB configuration of a series- resonant DC-DC converter, such as the exemplary apparatus 100 described above, is best suited for applications where the input voltage Vi is higher than the output voltage Vo, such as when providing power to a low voltage battery system.

[0049] As an aid to understanding, the following analysis assumes a forward power flow, where the phase shift is positive 0>O, and power is flowing from the input voltage Vi to the output voltage Vo.- It should be understood that the exemplary apparatus 100 is capable of bi-directional power flow and is equally capable of supporting power flowing from the output voltage Vo to the input voltage Vi.

[0050] Output power Poand current Iofor the HB / FB configuration (mode a) of the exemplary converter apparatus 100 are provided by equations 1 and 2:where -n / 2 < 0 < n / 2 represents the phase shift between the primary voltage vpand the secondary voltage vs, Z represents the impedance, and M is the converter gain.

[0051] Magnitude and direction of power / current flow between the primary side 128 and secondary side 130 of the power converter apparatus 100 is determined by the phase shift 0. Positive phase shift +0 provides power flow from the primary side 128 to the secondary side130 while negative phase shift -0 provides power flow form the secondary side 130 to the primary side 128.

[0052] The impedance Z and voltage gain M across the impedance Z are given by equations 3 and 4:where fsis the switching frequency of both the primary half bridge switching circuit 102 and the full bridge switching circuit 104.

[0053] Conditions for minimum root-mean- square currents and soft-switching operation of the switching devices Si, S2, Qi, Q2, Q3, Qi while in mode a are easily derived using standard analyses. A key condition for optimal variation of the phase-shift 0 when operating a SRDAB converter in buck mode (M<1) is show by equation 5:(5)0 = cos-1M for M < 1 .

[0054] Operation of the converter apparatus 100 in mode a begins with the lowest desired output voltage Vo,min , which occurs at the lowest converter gain M, and continues until the output voltage Vorises to a nominal output voltage Vo, M~I which occurs as the converter gain M comes close to one, i.e., M~l. When the converter apparatus 100 enters boost mode, which occurs when the converter gain is greater than or equal to one, M>1. increased circulating currents can adversely affect converter performance. It has therefore been recommended that, when designing a converter apparatus 100 for operation in mode a, the transformer turn ratio N:1 should be selected to maintain a converter gain less than 1, M<1.

[0055] In applications where the output voltage Vovaries between 1 and 5 times the input voltage , selecting a transformer turn ratio N:1 that keeps the converter gain less than one M<1 produces a converter gain M at the minimum output voltage Vo,min that puts the converter into deep buck operation, which leads to lower efficiency.

[0056] The converter gain M at which the nominal output voltage Vo, M~I occurs is referred to herein as the predetermined nominal converter gainand is selected during converter design. The predetermined converter gain M^i is a value close to one and may be in a range of about 0.95 to 0.99. Selection of the desired predetermined converter gain M i maybe based on the switching speed of the switching devices Si, S2, Qi, Q2, Q3, Q4, where faster switching devices allow a predetermined converter gain M i closer to one.

[0057] When the converter apparatus 100 is operated in a HB / FB converter configuration with converter gains greater than one M>1, conduction loss dominates, and converter efficiency may be adversely affected. Converter operation with a converter gain M greater than one are referred to as boot mode operation. To mitigate boost mode performance degradation, the duty cycle of the second lower switching device Q4 is increased above fifty percent (50%). This reduces the effective secondary voltage vsapplied to the impedance network Z and avoids much of the conduction losses created by boost mode operation.

[0058] As used herein the term Half Bridge / Semi-Full Bridge (HB / S-FB) refers to converter operation where the duty cycle Dq2 of the first lower switching device Q2 is fifty percent (50%) and the duty cycle Dq4 of the second lower switching device Q4 is greater than fifty percent (50%), such as between about fifty percent (50%) and one hundred percent (100%). For convenience, HB / S-FB operation is also referred to herein as mode b.

[0059] As used herein, the duty cycle of a half bridge switching circuit, such as the second half bridge switching circuit 112, is represented by the on-time of the lower switching device, such as the on-time of the second lower switching device Q4, For example, when the on-time of the second lower switching device Q4 is greater than the on-time of the second upper switching device Q3, the duty cycle Dq4 of the second half bridge switching circuit 112 is said to be greater than fifty percent (50%).

[0060] Figure 3 illustrates graphs depicting the key operating waveforms 300 of a series-resonant DAB power converter apparatus 100 configured for HB / S-FB operation incorporating aspects of the disclosed embodiments. In the illustrated graphs 300, time is depicted along horizontal axes 310, 312, 314, 316 increasing to the right. Graph 302 shows the on or off state of the switching devices Si, S2 in the primary half bridge switching circuit 102. Graph 304 shows the on or off state of the switching devices Qi, Q2, in the first half bridge switching circuit 110, and Graph 306 shows the on or off state of the switching devices Q3, Q , in the second half bridge switching circuit 112.

[0061] Graph 308 depicts the primary voltage vp, the secondary voltage vs, and the primary current ipduring one full cycle T of the switching circuits 102, 104, where signal magnitude is depicted along a vertical axis 318 increasing upwards. Time for the illustrated cycle is depicted along the horizontal axis 316 between time t=0 and t=T=2n. The primary voltage vpis generated by the primary half bridge switching circuit 102 and varies between zero and the DC input voltage Vi. Waveforms 300 illustrated in Figure 3 show that increasing theduty cycle Dq4 of the second half bridge switching circuit 112 reduces the effective secondary voltage vsapplied to the impedance network Z.

[0062] Because the resonant LC tank Z is split Zp-Zsacross the primary side 128 and secondary side 130 of the transformer, the DC-offset generated by the secondary side half bridge 112 can be removed. When the duty cycle Dq4 is increased, the effective secondary voltage vsis reduced, and the converter gain M shown above in equation 4, may be rewritten as shown in equation 6:where x is a converter factor which varies between 0.5 and 1, i.e., 0.5 <x < 1. When the output voltage is between the nominal output voltage Vo, M~I and the maximum converter output voltage Vo,max, the converter factor x is inversely proportional to the rise in output voltage Vo.

[0063] Variation of the converter factor x and the duty cycle Dq4 of the second haff bridge switching circuit 112 are given by equations 7 and 8:0.5 for Vojnin < o < Vo,M«l (7)Combining equations (7) and (8) gives an equation for the converter factor x in terms of the duty cycle Dq4 of the second haff bridge converter 112 as shown in equation (9): x = (1.5 — £>q4). (9)

[0064] Figure 4 illustrates graphs 402, 404 depicting duty cycle Dq4 and converter factor x variations of the second half bridge switching circuit incorporating aspects of the disclosed embodiments. In the graphs 402, 404, output voltage Vois depicted along horizontal axes 406, 410 increasing to the right while magnitude of the duty cycle is depicted along the vertical axis 408 increasing upwards, and converter factor x is depicted along the vertical axis 412 increasing upwards. As illustrated in graphs 402, 404, the duty cycle and converter factor remain constant while the converter apparatus 100 is configured as a HF / FB topology, mode a, between the minimum output voltage Vo,min and the nominal output voltage Vo, M~I. When the output voltage Vo exceeds the nominal output voltage Vo, M~I, the duty cycle I)q4 increases linearly and the converter factor x decreases linearly until the output voltage Voreaches the maximum output voltage Vo ,max-

[0065] Efficiency is lost in HF / FB operation (mode a) when it goes too deep into buck mode (converter gain much less than one) or into boost mode (converter gain greater than one). When handling large output voltage Vovariations which lead to converter gain greater than one, A£>1, HB / S-FB (mode b) may be employed to slice the secondary voltage vsin order to maintain an effective / equivalent converter gain near one, M~1. This reduction in converter gain is shown above in equations (6) through (9).

[0066] In certain embodiments, such as converter applications where the output voltage Vo varies over an ultra-wide voltage range of one to five times the input voltage , it is beneficial to select a transformer turn ratio Nto produce a converter gain M near one when the output voltage Vois close to half of the maximum output voltage Vo,max. This condition may be expressed in terms of the nominal output voltage VO, M~I as shown in equation (10):

[0067] As an example, consider a power converter apparatus, such as the apparatus 100, where the input voltage is eight hundred twenty volts, Vi = 820 volts, and the output voltage varies over the range Vo= 200 to 800 volts. A transformer turn ratio of one, N=l, will bring the converter gain M=0.48 at the minimum output voltage Vo,min = 200 volts, and will produce a converter gain near one M~1 at a nominal output voltage of Vo, M~I = 400 volts. In this example the converter apparatus 100 will transition from mode a operation to mode b operation when the output voltage Voexceeds the nominal output voltage VO, M~I of 400 volts.

[0068] Once the output voltage Voreaches the maximum output voltage Vo,max, the duty cycle of the second half bridge switching circuit 112 equals 1, Dq4 =1, the converter factor x equals 0.5, =0.5, and the converter apparatus 100 is reconfigured as a half bridge / half bridge (HB / HB) topology. Operation in the HB / HB topology is referred to herein as operating mode c. When operating in mode c, the converter gain M shown in equation (6) can be rewritten as shown in equation (11):

[0069] During operation in mode c, the converter apparatus 100 is reconfigured as a HB / HB topology by turning the second upper switching device Q3 off and turning the second lower switching device Q4 on, thereby effectively removing the second half bridge switching circuit 112 leaving only the first half bridge switching circuit 110 on the secondary side 130 of the transformer T.

[0070] Figure 5 illustrates graphs depicting the key operating waveforms 500 of a series-resonant DAB power converter, such as the exemplary apparatus 100, configured for HB / HB operation incorporating aspects of the disclosed embodiments. In the illustrated graphs 500, time is depicted along horizontal axes 510, 512, 514, 516 increasing to the right. Graph 502 shows the on or off state of the switching devices Si, S2 in the primary half bridge switching circuit 102. Graph 504 shows the on or off state of the switching devices Qi, Q2, in the first half bridge switching circuit 110, and Graph 506 shows the on or off state of the switching devices Q3, Qi, in the second half bridge switching circuit 112. Note that in mode c, the second half bridge switching circuit 112 is effectively removed by turning the second upper switching device Q3 off and the second lower switching device Q on.

[0071] Graph 508 depicts the primary voltage vp, the secondary voltage vs, and the primary current ipduring one full cycle T of the switching circuit 102, 104, where signal magnitude is depicted along a vertical axis 518 increasing upwards. Time for the illustrated cycle is depicted along the horizontal axis 516 between time t=0 and t=T=2n. The primary voltage vpis generated by the primary half bridge switching circuit 102 and varies between zero and the input voltage Vi, and the secondary voltage vsis generated by the first half bridge switching circuit 110 and varies between zero and the output voltage .

[0072] Referring now to Figure 6 there can be seen a flow chart of an exemplary method600 for operating a reconfigurable bidirectional series-resonant DC-DC power converter incorporating aspects of the disclosed embodiments. The exemplary method 600 may be beneficially employed to operate any appropriate reconfigurable bidirectional series-resonant DC-DC power converter such as the exemplary converter apparatus 100 described above and with reference to Figure 1. During operation, the exemplary method 600 reconfigures the converter topology to provide efficient power conversion over a wide output voltage range, such as an output voltage that varies from one to five time the input voltage, without increasing converter component count.

[0073] The exemplary method 600 configures 618 an appropriate turn ratio N for the transformer T where an appropriate turn ratio N produces the predetermined nominal converter gain Mi / when the output voltage Vois one half of the predetermined maximum output voltage Vo, max. In one embodiment the predetermined nominal converter gain Mi / is a value near unity gain such as a converter gain M within the range of about 0.95 to 0.99.

[0074] During converter operation, measurements of the input voltage and output voltage are made 602. These measurements may be acquired through any appropriate sensingmeans such as analog to digital conversion or other appropriately accurate means of measuring an analog voltage.

[0075] Measurements of the input voltage Vi and output voltage Voare used to calculate604 a converter gain M. In addition to the input voltage Vi and output voltage Vomeasurements, it may be beneficial, in certain embodiments, to include additional values when calculating 604 the converter gain A / , such as the transformer turn ratio N or converter factor , as described above and with reference to equations (4), (6), and (11). The converter gain M may then be used as a basis for reconfiguring the converter topology and selecting an appropriate converter operating mode.

[0076] When 606 the output voltage Vois greater than a pre-determined minimum output voltage Vo,mm and the converter gain M is less than or equal to the predetermined nominal converter gain M=j, the method 600 reconfigures the converter apparatus 100 as a HB / FB converter topology where the first half bridge switching circuit 110 and the second half bridge switching circuit 112 are operated 608 as a full bridge switching circuit 104.

[0077] When 610 the converter gain M is greater than a predetermined nominal converter gain M^i and the output voltage Vois less than a predetermined maximum output voltage Vo, max, the converter is operated 612 in a HB / S-FB mode where the first half bridge switching circuit 110 is operated at a first duty cycle I)q2 of fifty percent (50%) and the second half bridge switching circuit 112 is operated at a second duty cycle Dq, wherein the second duty cycle Dqis greater than the first duty cycle Dq2. When operating 610 the converter apparatus 100 in a HB / S-FB mode, it is beneficial, in certain embodiment, to adapt the second duty cycle Dqto be linearly proportional to the output voltage Vo, as described above and illustrated in equations (6) through (9). Operating 612 the second half bridge switching 112 circuit at a duty cycle greater than fifty percent (50%) places the converter in a HB / S-FB configuration where the second lower switching device Q4 slices the output voltage Vo, thereby effectively reducing the secondary voltage vsseen by the resonant tank Z and mitigating efficiency issues often seen when a HB / FB converter topology is operated with higher converter gains.

[0078] As discussed above, in one embodiment it may be beneficial to configure the converter apparatus 100, being operated in accordance with the exemplary method 600, to generate one half of the predetermined maximum output voltage Vo,max when the converter apparatus 100 is operated at the predetermined nominal converter gain M i. Setting the nominal converter gain to half the maximum output voltage helps the converter apparatus handle larger output voltage variations while maintaining acceptable converter efficiency.

[0079] When 614 the output voltage Vois greater than or equal to the predetermined maximum output voltage Vo,max, the method 600 configures the converter apparatus as a HB / HB topology where the full bridge switching circuit 104 is operated 616 in a half bridge mode. When operating a full bridge switching circuit in a half bridge mode, the second upper switching device Q3 is turned off and the second lower switching device Q4 is turned on, effectively removing the second half bridge switching circuit from the converter topology. The first half bridge switching circuit 110 may then be operated at a fifty percent (50%) duty cycle to complete the HB / HB topology.

[0080] Thus, while there have been shown, described, and pointed out, fundamental novel features of the invention as applied to the exemplary embodiments thereof, it will be understood that various omissions, substitutions and changes in the form and details of devices and methods illustrated, and in their operation, may be made by those skilled in the art without departing from the spirit and scope of the presently disclosed invention. Further, it is expressly intended that all combinations of those elements, which perform substantially the same function in substantially the same way to achieve the same results, are within the scope of the invention. Moreover, it should be recognized that structures and / or elements shown and / or described in connection with any disclosed form or embodiment of the invention may be incorporated in any other disclosed or described or suggested form or embodiment as a general matter of design choice. It is the intention, therefore, to be limited only as indicated by the scope of the claims appended hereto.

Claims

CLAIMSWhat is claimed is:

1. An apparatus (100) comprising: a transformer (I) comprising a primary winding (106) magnetically coupled to a secondary winding (108); a primary series resonant circuit (Zp) comprising a primary inductor (Lp), a primary capacitor ( ), and the primary winding (106) connected in series; a secondary series resonant circuit (Zs) comprising a secondary inductor (Zs), a secondary capacitor (G), and the secondary winding (108) connected in series; a primary half bridge switching circuit (102) coupled between an input voltage (Vi) and the primary series resonant circuit (Zp) a first half bridge switching circuit (110) coupled in parallel with an output voltage ( >); a second half bridge switching circuit (112) coupled in parallel with the output voltage (Vo) and wherein the secondary series resonant circuit (Zs) is coupled between a central node (120) of the first half bridge switching circuit (110) and a central node (122) of the second half bridge switching circuit (112), and a controller (118), wherein the controller (118) is configured to: sense the input voltage (Vi) and the output voltage (Vo); calculate a converter gain (M) based at least in part on the input voltage (Vi) and the output voltage (Vo); and, when the converter gain (M) is greater than a predetermined nominal converter gain (M=j) and the output voltage (Vo) is less than a predetermined maximum output voltage (Vo, max), the controller (118) is configured to operate the first half bridge switching circuit (110) at a first duty cycle (Dq2) of fifty percent (50%) and operate the second half bridge switching circuit (112) at a second duty cycle (Dqi), wherein the second duty cycle (Dqi) is greater than the first duty cycle (Dq2).

2. The apparatus (100) according to claim 1 wherein the first half bridge switching circuit (110) comprises a first upper switching device (Qi) and a first lower switching device (Q2) connected in series to form a first central node (120), the second half bridge switching circuit (112) comprises a second upper switching device (Q3) and a second lower switching device (Qi) connected in series to form a second central node (122), and the secondaryseries resonant circuit (Zs) is coupled between the first central node (120) and the second central node (122), and wherein the second duty cycle Dq4 corresponds to an on-time of the second lower switching device ( ).

3. The apparatus (100) according to any one of claims 1 or 2 wherein the predetermined nominal converter gain (M=j) corresponds to one half of the predetermined maximum output voltage Vo, max)-4. The apparatus (100) according to any one of the preceding claims wherein the predetermined nominal converter gain (M^ / ) is between 0.95 and 0.99.

5. The apparatus (100) according to any one of the preceding claims wherein, when the converter gain (M) is greater than the predetermined nominal converter gain (Mxi and the output voltage (Vo) is less than a predetermined maximum output voltage Vo,max), the second duty cycle Dq4 is linearly proportional to the output voltage (Vo).

6. The apparatus (100) according to any one of the preceding claims wherein, when the output voltage Vo) is greater than a pre-determined minimum output voltage (Vo,mm) and the converter gain (M) is less than or equal to the predetermined nominal converter gain (Mxi), the controller (118) is configured to operate the first half bridge switching circuit (110) and the second half bridge switching circuit (112) as a full bridge switching circuit (104).

7. The apparatus (100) according to any one of the preceding claims wherein, when the output voltage (Vo) is greater than or equal to the maximum output voltage (Vo,max), the controller (118) is configured to operate the full bridge switching circuit (104) in a half bridge mode, wherein the half bridge mode comprises turning the second upper switching device (Q3) off, turning the second lower switching device (Q4) on, and operating the first half bridge switching circuit (110) at a fifty percent (50%) duty cycle.

8. The apparatus (100) according to any one of the preceding claims wherein the transformer(T) comprises a turn ratio (N) configured to produce the predetermined nominal converter gain (M^) when the output voltage (Vo) is one half of the predetermined maximum output voltage (Vo ,max)-9. A method (600) for operating a power conversion apparatus, wherein the power conversion apparatus comprises: a transformer comprising a primary winding magnetically coupled to a secondary winding; a primary series resonant circuit comprising a primary inductor, a primary capacitor, and the primary winding connected in series;a secondary series resonant circuit comprising a secondary inductor, a secondary capacitor, and the secondary winding connected in series; a primary half bridge switching circuit coupled between an input voltage and the primary series resonant circuit; a first half bridge switching circuit coupled between the secondary series resonant circuit and an output voltage; and a second half bridge switching circuit coupled between the secondary series resonant circuit and the output voltage, the method (600) comprising: sensing (602) the input voltage and the output voltage; calculating (604) a converter gain based at least in part on the input voltage and the output voltage; and when (610) the converter gain is greater than a predetermined nominal converter gain and the output voltage is less than a predetermined maximum output voltage, operating (612) the first half bridge switching circuit (110) at a first duty cycle of fifty percent (50%) and operating the second half bridge switching circuit at a second duty cycle, wherein the second duty cycle is greater than the first duty cycle.

10. The method (600) according to claim 9 wherein the predetermined nominal converter gain corresponds to one half of the predetermined maximum output voltage.

11. The method (600) according to any one of claims 9 or 10 wherein the predetermined nominal converter gain is between 0.95 and 0.99.

12. The method (600) according to any one claims 9 through 11 wherein, when the converter gain is greater than the predetermined nominal converter gain, the output voltage is less than a predetermined maximum output voltage, and the second duty cycle is linearly proportional to the output voltage.

13. The method (600) according to any one of claims 9 through 12 wherein, when the output voltage is greater (606) than a pre-determined minimum output voltage and the converter gain is less than or equal to the predetermined nominal converter gain, the method (600) further comprises operating (608) the first half bridge switching circuit and the second half bridge switching circuit as a full bridge switching circuit.

14. The method (600) according to any one of claims 9 through 13 wherein, when the output voltage is greater than or equal to the predetermined maximum output voltage, the method (600) further comprises operating (616) the full bridge switching circuit in a half bridge mode, wherein the half bridge mode comprises turning the second upper switching deviceoff, turning the second lower switching device on, and operating the first half bridge switching circuit at a fifty percent (50%) duty cycle.

15. The method (600) according to any one of claims 9 through 14 wherein the transformer comprises a turn ratio configured to produce the predetermined nominal converter gain when the output voltage is one half of the predetermined maximum output voltage.

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