Dual three-phase asymmetric stator-type double-permanent-magnet vernier electric motor and improved model predictive current control method therefor

Abstract

Disclosed in the present invention are a dual three-phase asymmetric stator-type double-permanent-magnet vernier electric motor and an improved model predictive current control method therefor. The electric motor comprises a stator core, stator permanent magnets, a rotor core and a rotor ferrite, wherein the stator core is formed by alternately arranging umbrella-shaped armature teeth and quasi-rectangular armature teeth in the circumferential direction, the stator permanent magnet, which is magnetized radially outwards, is adhered to each of openings on two sides of the umbrella-shaped armature teeth, and a fan-shaped auxiliary slot is formed between the umbrella-shaped armature teeth and the stator permanent magnet; and the shape of the rotor core is similar to a gear, the rotor ferrite is embedded in the rotor core using a Halbach-based alternating-pole array, and the length of the rotor ferrite is greater than the length of the rotor core and the length of the stator core. In addition, the present invention uses a model predictive current control system based on a virtual voltage vector to construct a cost function to select the voltage vector having the minimum switching frequency, and make same operate preferentially. The present invention not only improves the torque density of the electric motor, but also significantly reduces switching losses in model predictive control.

Description

A dual three-phase asymmetric stator type dual permanent magnet vernier motor and its improved model predictive current control method Technical Field

[0001] The present invention relates to a dual permanent magnet motor and a control strategy thereof, in particular to a dual three-phase asymmetric stator dual permanent magnet vernier motor and an improved model prediction current control method thereof, belonging to the technical field of permanent magnet motors. Background Art

[0002] With the rapid development of rare earth permanent magnet materials and power electronics technology, permanent magnet synchronous motors with high power density and high efficiency have broad application prospects in many fields such as electric vehicles, rail transportation, agricultural machinery and equipment.

[0003] A dual permanent magnet vernier motor is a motor with permanent magnets distributed on both the stator and rotor and high torque density. Its unique bilateral excitation structure has maximized the use of the excitation source distribution. Therefore, how to further improve the motor torque density has become a research hotspot for many scholars. The paper "Design and analysis of novel asymmetric-stator-pole flux reversal PM machine (IEEE Transactions on Industrial Electronics, 67(1):101-114, 2020)" proposes a flux reversal motor with asymmetric stator poles. Compared with traditional flux reversal motors, although the torque density of this motor is improved, it still does not fully utilize the internal space of the motor and has a large amount of inter-pole leakage. The paper "A novel dual-permanent-magnet-excited machine with non-uniformly distributed permanent-magnets and flux modulation poles on the stator (IEEE Transactions on Vehicular Technology, 69(7): 7104-7115, 2020)" proposes a dual permanent magnet vernier motor with non-uniformly distributed stator poles. Due to the large number of stator and rotor permanent magnet pole pairs of the motor, eddy current loss increases, which reduces its efficiency. At the same time, the model predictive current control for the permanent magnet synchronous motor converts each available voltage vector into a corresponding current value and substitutes it into the current cost function. For a dual three-phase permanent magnet synchronous motor, there are 64 available voltage vectors. The computational complexity of voltage vector screening is greatly increased compared with vector control and direct torque control, which has high hardware requirements and poor steady-state performance. Therefore, some scholars have proposed to reduce the number of alternative voltage vectors by the sector of the stator flux and the torque error, and introduce duty cycle control to improve the steady-state performance of the algorithm. Although this method can effectively reduce the computational complexity and improve the control performance, it still has a high computational complexity. Therefore, the model-predicted current control strategy needs to be further optimized.

[0004] In order to reduce the amount of calculation, the paper "Dual-vector model predictive torque control of dual three-phase permanent magnet synchronous motors (Journal of Electric Machines and Control, 26(9):97-107, 2022)" synthesizes 24 large and medium vectors out of 64 voltage vectors into 12 virtual voltage vectors, and then reduces the iterative voltage vectors from 12 to 4 according to the sector where the stator flux is located and the torque error, which greatly reduces the amount of calculation. However, this method still cannot get rid of the disadvantage of the large computational load caused by iterating the voltage vector into the cost function. In addition, the traditional model predictive current control directly outputs the voltage vector to the inverter, without considering the loss of power switching devices caused by the number of switching times. Therefore, further research on model predictive current control is needed to further reduce the amount of calculation and improve the steady-state performance of the system while ensuring that the dynamic performance remains unchanged, and to optimize the switching times of the voltage source inverter.

[0005] The purpose of this patent is to design a new type of permanent magnet vernier motor with dual permanent magnet, dual winding structure and good dynamic and steady-state performance and its control strategy.

[0006] Summary of the Invention

[0007] In response to the shortcomings and defects of the prior art, the present invention proposes a dual three-phase asymmetric stator dual permanent magnet vernier motor and an improved model predictive current control method thereof. The motor uses dual permanent magnets in combination with a dual winding structure to improve torque density. On this basis, the concept of alternating poles and an asymmetric design are introduced. While reducing the amount of permanent magnets and improving the utilization rate of permanent magnets, ferrites with Halbach arrays are arranged between adjacent rotor teeth to enhance the operating harmonics in the air gap flux density, thereby further improving the torque density of the motor. At the same time, the improved model predictive current control strategy proposed in the present invention is adopted to reduce the model prediction calculation time, simplify the duty cycle calculation, optimize the PWM average switching frequency, and ensure that the model predictive control achieves excellent dynamic and steady-state performance, as well as lower computational complexity and lower average switching frequency.

[0008] The technical solution of the present invention is as follows: a dual three-phase asymmetric stator type dual permanent magnet vernier motor, comprising a stator and a rotor, both of which are salient pole structures; the stator comprises a stator core (1), a dual three-phase winding (2) and a stator permanent magnet (3); the stator core (1) is composed of a plurality of umbrella-shaped armature teeth (11) and rectangular armature teeth (12) arranged alternately along the circumferential direction; the rectangular armature teeth (12) are composed of a piece of iron core; the umbrella-shaped armature teeth (11) are composed of a sandwich array of "permanent magnet + iron core + permanent magnet", and a stator permanent magnet is respectively pasted on the openings on both sides; the stator permanent magnet (3) is pasted on The openings on both sides of the umbrella-shaped armature teeth are magnetized radially outward with the same polarity; the dual three-phase winding (2) is wound on the umbrella-shaped armature teeth (11) and the rectangular armature teeth (12) respectively, and the dual three-phase winding is a centralized winding; the rotor includes a rotor core (5) and a rotor ferrite (6); the rotor core (5) is shaped like a gear; the rotor ferrite (6) is embedded in the rotor core using a Halbach-based alternating pole array; the rotor ferrite adopts an unequal length structure, and its length is greater than the length of the rotor core and the length of the stator core; the stator core (1) and the rotor core (5) are both laminated silicon steel sheets.

[0009] Furthermore, the umbrella-shaped armature tooth (11) is in an inverted "T" shape in the circumferential cross section, and both sides of the tooth end are further processed with fan-shaped groove openings.

[0010] Furthermore, the stator permanent magnet (3) is tile-shaped and pasted on the openings at both ends of the umbrella-shaped armature tooth. A fan-shaped auxiliary groove is opened between the umbrella-shaped armature tooth (11) and the stator permanent magnet (3). The auxiliary groove (4) is filled with non-magnetic material such as epoxy resin or is not filled with any filler.

[0011] Furthermore, the stator permanent magnet (3) is made of a high coercive force rare earth permanent magnet material such as a neodymium iron boron magnet, and is magnetized radially outward with the same polarity, and an alternating pole permanent magnet array is attached to the openings on both ends of the umbrella-shaped armature teeth; the rotor ferrite (6) is made of a low coercive force permanent magnet material such as a ferrite magnet, and is embedded in the rotor core using a Halbach array composed of radial (61) and parallel (62) magnetization. The magnetic concentration direction of the rotor ferrite is radially pointing to the stator, and the length of the rotor ferrite is greater than the length of the rotor core and the length of the stator core.

[0012] Furthermore, the maximum lengths of the two ferrites magnetized in the radial direction (61) and the parallel direction (62) along the circumferential direction are tr0=(0.25-0.5)*τ and tr1=(0-0.25)*τ, respectively, where τ is the pole pitch.

[0013] Furthermore, the dual three-phase winding (2) adopts a centralized winding structure and is wound on the umbrella-shaped armature teeth (11) and the quasi-rectangular armature teeth (12), respectively. The windings wound on the umbrella-shaped armature teeth are A1 phase, B1 phase and C1 phase, respectively, and the phase difference between A1 phase and B1 phase, A1 phase and C1 phase, and B1 phase and C1 phase is 120 degrees; the windings wound on the quasi-rectangular armature teeth are A2 phase, B2 phase and C2 phase, respectively, and the phase difference between A2 phase and B2 phase, A2 phase and C2 phase, and B2 phase and C2 phase is 120 degrees; the phase difference between A1 phase and A2 phase, B1 phase and B2 phase, and C1 phase and C2 phase is 30 degrees.

[0014] Furthermore, based on the bilateral excitation structure and the bidirectional magnetic field modulation effect, the equivalent armature winding pole pair number P a , number of stator teeth N s , number of stator permanent magnet pole pairs P s , number of rotor teeth N r , rotor ferrite pole pair number P r , the five satisfy the following relationship:

[0015] Where: p is the number of pole pairs in the unit stator symmetry period, and s is the number of slots in the unit stator symmetry period.

[0016] The improved model prediction current control method of a dual three-phase asymmetric stator dual permanent magnet vernier motor of the present invention comprises the following steps:

[0017] Step 1: Establish a structural model of a dual three-phase asymmetric stator dual permanent magnet vernier motor;

[0018] Step 2: Use the actual speed of the detected dual three-phase asymmetric stator dual permanent magnet vernier motor as the motor's feedback speed n, so that the motor's given speed n ref The motor speed error is obtained by subtracting n, and the q-axis reference current i of the motor is obtained through the PI controller. q ref , set the d-axis reference current i d ref =0, the stator current vector in the dq coordinate system is expressed as I s ref =i d ref +ji q ref , I s ref The projection on the α-β subplane is I s_αβ ref ;

[0019] Step 3: Use Clarke transformation matrix to transform the dual three-phase current i A1 、i B1、i C1 、i A2 、i B2 、i C2 The current i transformed into the α-β subplane α 、i β and the current i in the xy subplane x 、i y , and then use the Park transformation matrix to transform i α 、i β The current i transformed into the dq coordinate system d k ,i q k ;

[0020] Step 4: Based on the principle of suppressing the current in the xy subspace, the base voltage vector is used to synthesize the virtual voltage vector VV i (i=1, 2, ..., 12), so that it is 0 in the xy subspace and its amplitude in the α-β subplane is 0.596U dc , where U dc is the DC bus voltage; the two base voltage vectors participating in the synthesis of the virtual voltage vector are large voltage vectors V with the same direction in the α-β sub-plane max and the medium and large voltage vector V midL , the virtual voltage vector is divided into sectors on the α-β sub-plane, the angle bisector of two adjacent virtual voltage vectors is the dividing line of each sector, and each virtual voltage vector is on the center line of its sector;

[0021] Step 5: Establish the current equation of the dual three-phase asymmetric stator dual permanent magnet vernier motor in the dq coordinate system and discretize it to obtain

[0022] Where: d k and i q k kT s The components of the current on the d-axis and q-axis at the moment i d k+1 、i q k+1 、u d k+1 and u q k+1 They are (k+1)T s The components of the current and voltage on the d and q axes at the moment; R is the stator resistance; L is the stator inductance; ω is the electrical angular velocity; Ψ f is the permanent magnet flux; T s is the control period; k represents kT s moment; i d 、iq 、u d 、u q are the components of current and voltage on the d and q axes; thus, the current-related term i is separated id k+1 、i iq k+1 and voltage-related terms i ud k+1 、i uq k+1

[0023] It is expressed in vector form as I in the dq coordinate system si =i id k+1 +ji iq k+1 , I su =i ud k+1 +ji uq k+1 , I si and I su The projection on the α-β subplane is I si_αβ and I su_αβ , where I su_αβ In phase with the virtual voltage vector, with an amplitude equal to the virtual voltage vector T s / L times;

[0024] Step 6: Calculate the current increment vector I based on the reference current and feedback current sDB And its projection in the α-β coordinate system I sDB_αβ Phase angle θ αβ

[0025] Where: ∠I sDB For I sDB Angle in the dq coordinate system; θ e is the rotor electrical angle; therefore, in order to track the reference current faster, it is necessary to make I su_αβ with I sDB_αβ The angle is the smallest; due to I su_αβ The phase angle is the same as the virtual voltage vector, then according to θ αβ Judgment I sDB_αβ The sector where the optimal voltage vector is located is the virtual voltage vector at the center line of the sector, thereby directly obtaining the optimal voltage vector and avoiding the continuous iteration of the voltage vector in the cost function.

[0026] Step 7: Calculate the base voltage vector V corresponding to the optimal voltage vector max 、V midL and the action time T of the zero voltage vector max、 T midL and T0 is

[0027] Step 8: In order to reduce the number of switching times of the power switch device in the voltage source inverter, the current voltage vector of the system and the voltage vector to be applied are substituted into the switching cost function J = (a1 i -a1) 2 +(b1 i -b1) 2 +(c1 i -c1) 2 +(a2 i -a2) 2 +(b2 i -b2) 2 +(c2 i -c2) 2

[0028] Where: a1, b1, c1, a2, b2, c2 represent the power switching devices of the upper bridge arm of the voltage source inverter A1, B1, C1, A2, B2, C2 corresponding to the current base voltage vector in kT s The switch state at the moment, where 0 is off and 1 is on; a1 i 、b1 i 、c1 i 、a2 i 、b2 i 、c2 i The six-phase upper-arm power switch devices of the voltage source inverter corresponding to the i-th base voltage vector are at (k+1)T s The switch state at the moment, thus obtaining J i The smallest voltage vector gives priority.

[0029] The dual three-phase asymmetric stator dual permanent magnet vernier motor and its improved model predictive current control strategy of the present invention have the following beneficial effects:

[0030] 1) The dual three-phase asymmetric stator type dual permanent magnet vernier motor of the present invention adopts a dual permanent magnet, dual salient pole structure. Its unique bidirectional flux modulation effect combined with the dual three-phase winding structure can maximize the torque density of the motor.

[0031] 2) The dual three-phase asymmetric stator dual permanent magnet vernier motor of the present invention combines the alternating pole concept with an asymmetric design to generate additional operating harmonics to increase torque, and introduces a Halbach array in the rotor ferrite to enhance the effective air gap harmonics, thereby further improving the torque density while also improving the utilization rate of the permanent magnets.

[0032] 3) The stator poles of the dual three-phase asymmetric stator type dual permanent magnet vernier motor of the present invention are processed with auxiliary slots, which can suppress the leakage flux between the stator poles, thereby improving the torque output capacity of the motor and reducing torque pulsation.

[0033] 4) The dual three-phase asymmetric stator dual permanent magnet vernier motor of the present invention adopts a dual permanent magnet structure and introduces the concept of alternating poles, asymmetric design and Halbach array, which greatly expands the operating range of the high-efficiency area and reduces material costs.

[0034] 5) The rotor ferrite of the dual three-phase asymmetric stator dual permanent magnet vernier motor of the present invention adopts an unequal length structure, so that the length of the rotor ferrite is greater than the length of the rotor core and the length of the stator core, thereby maximizing the use of the internal space of the motor, further enhancing the effective magnetic flux of the air gap, and thus enhancing the torque output capability of the motor.

[0035] 6) The improved model predictive current control strategy of the motor of the present invention is based on the current tracking concept. The optimal voltage vector is directly obtained according to the angle of the current increment vector in the α-β coordinate system, which effectively avoids the continuous iteration of the voltage vector in the cost function and greatly reduces the computational load and complexity.

[0036] 7) The improved model-predictive current control strategy for the motor in this invention calculates the magnitude of the current increment vector based on the current and voltage vectors to determine the optimal duty cycle for the voltage vector. This simplifies the traditional model-predictive duty cycle calculation method based on the derivation of the current cost function, significantly reducing the computational complexity. The combination of the zero vector and the current increment vector effectively improves the accuracy and response speed of current tracking.

[0037] 8) The improved model-predictive current control strategy for this motor designs a cost function for the switching states of the power devices in each bridge arm of the voltage source inverter. By adjusting the base voltage vector action sequence within a control cycle, the base voltage vector action sequence with the lowest switching frequency is obtained, significantly reducing the average PWM switching frequency. Combined with the optimal voltage vector acquisition method, this strategy not only reduces the computational effort but also lowers the average switching frequency of the power devices, thereby reducing switching losses. BRIEF DESCRIPTION OF THE DRAWINGS

[0038] FIG1 shows the topological structures of a conventional flux reversal motor and a dual three-phase asymmetric stator dual permanent magnet vernier motor proposed in the present invention: (a) conventional flux reversal motor; (b) motor of the present invention;

[0039] FIG2 is a diagram showing the changing process of the motor of the present invention;

[0040] FIG3 is a Fourier decomposition comparison diagram of the magnetomotive force generated by the stator permanent magnets of a conventional flux reversal motor and the motor of the present invention;

[0041] FIG4 is a comparison diagram of the air gap flux density Fourier decomposition of a conventional flux reversal motor and the motor of the present invention;

[0042] FIG5 is a comparison of the contribution of harmonics to torque of a conventional flux reversal motor and a motor of the present invention with and without rotor ferrite;

[0043] FIG6 is a torque comparison diagram of a conventional flux reversal motor and a motor of the present invention with and without rotor ferrite;

[0044] FIG7 is an efficiency map of a conventional flux reversal motor and a motor of the present invention: (a) conventional flux reversal motor; (b) motor of the present invention;

[0045] FIG8 is a cross-sectional view of the unequal length structure of the motor rotor ferrite according to the present invention;

[0046] FIG9 is a block diagram of the improved model prediction current control of the motor of the present invention;

[0047] Figure 10 shows the distribution of large and medium voltage vectors in the α-β and xy subspaces;

[0048] Figure 11 shows the virtual voltage vector distribution and sector division;

[0049] Figure 12 is a schematic diagram of the current vector control principle;

[0050] FIG13 is a flow chart of the PWM switching times optimization algorithm;

[0051] FIG14 is a schematic diagram showing the effect of optimizing the number of PWM switching times;

[0052] Figure 15 is a comparison of the optimization effects of PWM switching times under various working conditions;

[0053] FIG16 is a Fourier decomposition waveform of the A1 phase current of the motor of the present invention;

[0054] FIG17 is a waveform of torque and speed under normal operation of the motor of the present invention;

[0055] FIG18 is a torque step waveform of the motor of the present invention under normal operation;

[0056] FIG19 is a speed step waveform of the motor of the present invention under normal operation.

[0057] In the figure: 1 is the stator core, 11 is the umbrella-shaped armature tooth, 12 is the quasi-rectangular armature tooth; 2 is the double three-phase winding; 3 is the stator permanent magnet; 4 is the auxiliary slot; 5 is the rotor core; 6 is the rotor ferrite, 61 is the radially charged ferrite, and 62 is the parallelly charged ferrite. DETAILED DESCRIPTION

[0058] The technical solutions in the embodiments of the present invention will be described clearly and completely below with reference to the accompanying drawings in the embodiments of the present invention.

[0059] In order to more simply and clearly illustrate the characteristics and beneficial effects of the dual three-phase asymmetric stator dual permanent magnet vernier motor and its improved model predictive current control strategy of the present invention, a detailed description is given below in conjunction with a specific dual three-phase asymmetric stator dual permanent magnet vernier motor.

[0060] A dual three-phase asymmetric stator type dual permanent magnet vernier motor, a conventional flux reversal motor and an embodiment of the motor of the present invention are shown in FIG1 . Both motors include a stator and a rotor, and both adopt a 12-slot / 34-pole slot-pole combination. The conventional flux reversal motor adopts a conventional NS-pole surface-mounted permanent magnet array to be pasted on the stator surface. Unlike the conventional flux reversal motor, the umbrella-shaped armature teeth (11) and the quasi-rectangular armature teeth (12) of the embodiment of the present invention are different due to the difference in the installation of the permanent magnets. The two stator teeth are respectively composed of an iron core and a sandwich array of "permanent magnet + iron core + permanent magnet"; the two stator permanent magnets (3) on the umbrella-shaped armature teeth (11) are magnetized radially outward with the same polarity, and a fan-shaped auxiliary slot is opened between the two permanent magnets and the middle iron core. The width of the auxiliary slot (4) is 2°, and the auxiliary slot (4) is filled with non-magnetic conductive material such as epoxy resin or without any filler; the stator is composed of 6 umbrella-shaped armature teeth (11) and 6 quasi-rectangular armature teeth (12) arranged alternately along the circumferential direction. The rotor core (5) is similar to a gear with 17 teeth. Ferrites using a Halbach array are embedded between adjacent rotor teeth. The rotor ferrite (6) has 17 pole pairs and is composed of ferrites magnetized radially (61) and parallelly (62). The magnetic concentration direction of the rotor ferrite (6) is radially pointing toward the stator. The maximum lengths of the two ferrites magnetized radially (61) and parallelly (62) along the circumferential direction are tr0=5.7mm and tr1=3.1mm respectively. The length of the rotor ferrite is greater than the length of the stator core and the length of the rotor core by 10mm. The dual three-phase winding (2) adopts a centralized winding structure and is wound on the umbrella-shaped armature teeth (11) and the rectangular armature teeth (12). The windings wound on the umbrella-shaped armature teeth (11) are A1 phase, B1 phase and C1 phase respectively, and the phase differences between A1 phase and B1 phase, A1 phase and C1 phase, and B1 phase and C1 phase are 120 degrees; the windings wound on the quasi-rectangular armature teeth (12) are A2 phase, B2 phase and C2 phase respectively, and the phase differences between A2 phase and B2 phase, A2 phase and C2 phase, and B2 phase and C2 phase are 120 degrees; the phase differences between A1 phase and A2 phase, B1 phase and B2 phase, and C1 phase and C2 phase are 30 degrees.

[0061] Figure 2 shows the changing process of the motor structure of the present invention. Unlike the traditional flux reversal motor, the stator permanent magnet of the present invention adopts an alternating pole permanent magnet array and an asymmetric design, and introduces auxiliary slots. The alternating pole ferrite based on the Halbach array is embedded in the rotor core.

[0062] According to the bilateral excitation structure and bidirectional magnetic field modulation effect, the equivalent armature winding pole pair number Pa, the stator teeth number Ns, the stator permanent magnet pole pair number Ps, the rotor teeth number Nr, and the rotor ferrite pole pair number Pr satisfy the following relationship:

[0063] Where: p is the number of pole pairs in the unit stator symmetry period, and s is the number of slots in the unit stator symmetry period.

[0064] Figure 3 compares the Fourier decomposition of the magnetomotive force generated by the stator permanent magnets of a conventional flux reversal motor and the motor of the present invention. The figure shows that due to the differences in adjacent stator teeth, the magnetomotive force of the motor of the present invention adds harmonics of odd multiples of 6, such as the 6th, 18th, and 30th orders. These harmonics, modulated by the rotor teeth, generate additional harmonics, as shown in Figure 4. Figure 5 compares the contribution of harmonics to torque in a conventional flux reversal motor and the motor of the present invention, with and without rotor ferrite. The figure shows that the asymmetric design of the stator poles adds additional harmonics, such as the 6th, 11th, and 18th orders, to generate torque, while in the motor of the present invention, the 17th order harmonic also contributes to torque due to the presence of rotor ferrite.

[0065] Figure 6 compares the torque of a conventional flux reversal motor and the motor of the present invention with and without rotor ferrites. The figure shows that the torque of the conventional flux reversal motor is 5.59 Nm with a torque ripple of 0.8%; the torque of the motor of the present invention based on rotorless ferrites is 7.53 Nm with a torque ripple of 1.6%; and the torque of the motor of the present invention is 10.05 Nm with a torque ripple of 1.26%. While reducing the amount of permanent magnets by half, the motor of the present invention based on rotorless ferrites utilizes an asymmetric stator pole design combined with an alternating pole concept, resulting in a 34.7% increase in torque. Furthermore, the motor of the present invention incorporates an alternating pole rotor ferrite based on a Halbach array, further increasing torque by 33.47%. Furthermore, due to the presence of dual three-phase windings, the torque ripple of all three motors is low. The optimal slot width for the auxiliary slots in the motor of the present invention is 2°.

[0066] Figure 7 is an efficiency map of a conventional flux reversal motor and a motor of the present invention. As can be seen from the figure, under the operating conditions of a rated speed of 300 rpm and a rated peak current of 10 A, the efficiency of the conventional flux reversal motor is 81.11%, and the efficiency of the motor of the present invention is 87.88%. In addition, the motor of the present invention improves the torque performance in the constant torque area; in the weak magnetic area, the operating range of the high-efficiency area is greatly expanded. Figure 8 is a cross-sectional view of the unequal length structure of the rotor ferrite of the motor of the present invention, and the length of the rotor ferrite is greater than the length of the rotor core and the stator core. Considering the amount of ferrite material used, the optimal length of the unequal length of one end of the rotor ferrite is 5 mm.

[0067] The control strategy of the motor of the present invention is shown in Figure 9. It adopts a dual closed-loop control system consisting of a current loop based on model predictive control and a PI control speed loop. First, the given speed is subtracted from the actual speed to obtain the speed error. The speed error is input into the speed loop PI controller to obtain the q-axis reference current i q ref , set the d-axis reference current i d ref = 0. The stator current vector in the dq coordinate system can be expressed as I s ref =i d ref +ji q ref , I s ref The projection on the α-β subplane is I s_αβ ref .

[0068] Using Clarke transformation matrix T clarke The dual three-phase current i collected by the current sensor A1 、i B1 、i C1 、i A2 、i B2 、i C2 Transform i into α-β and xy coordinate systems α 、i β 、i x 、i y , where i α 、i β is the fundamental current, i x 、i y is the harmonic current. Use Park transformation matrix to transform i α 、i β Transformed into the current i in the dq coordinate system d k ,i q k .

[0069] As shown in Figure 10, the large voltage vector and the medium-large voltage vector with smaller amplitude in the α-β subspace have the same direction but are in opposite directions in the xy subspace. In order to suppress the current in the xy subspace, a set of large voltage vectors and medium-large voltage vectors can be output in one control cycle to satisfy

[0070] Let |V xy |=0, we can get

[0071] It can be seen that when the ratio of the action time of the large voltage vector and the medium-large voltage vector is 0.731:0.269, the effect is equivalent to the amplitude of 0.596U in the α-β sub-plane. dc And the resultant vector with an amplitude of 0 in the xy sub-plane. For convenience, the resultant vector is represented as a virtual voltage vector VV i (i=1, 2, ..., 12). Then, the virtual voltage vector is divided into sectors in the α-β coordinate system, and the angle bisectors of adjacent virtual voltage vectors are used as sector demarcation lines. The synthesized virtual voltage vector diagram is shown in FIG11 .

[0072] The current equation of the motor of the present invention in the dq coordinate system is established and discretized to obtain the current prediction equation

[0073] Where: R is the stator resistance; L is the stator inductance; ω is the electrical angular velocity; Ψ f is the permanent magnet flux; T s is the control period; k represents kT s moment; i d 、i q 、u d 、u q are the components of the current and voltage vectors on the d and q axes, respectively.

[0074] On this basis, the current related term i is separated according to formula (5): id k+1 、i iq k+1 and voltage-related terms i ud k+1 、i uq k+1 . And express it in vector form in dq coordinate system as I si =i id k+1 +ji iq k+1 , I su =i ud k+1 +ji uq k+1 , Isi and I su The projection on the α-β subplane is I si_αβ and I su_αβ Therefore, we can know that I su_αβ The phase angle is the same as the virtual voltage vector phase angle, and the amplitude is the virtual voltage vector amplitude T s / L times.

[0075] According to formula (6), we can get tracking I s ref The required current increment vector I sDB and I sDB Projection in α-β coordinate system I sDB_αβ The phase angle θ αβ

[0076] Where: ∠I sDB For I sDB Phase angle in the dq coordinate system; θ e is the motor electrical angle.

[0077] As shown in Figure 12, in order to make I si_αβ Fast Track I s_αβ ref , need to make I su_αβ with I sDB_αβ The angle between the two is the smallest. su_αβ The phase angle is the same as that of the virtual voltage vector, so it is necessary to make it sDB_αβ The virtual voltage vector with the smallest angle between θ and θ is taken as the optimal voltage vector. αβ Judgment I sDB_αβ The virtual voltage vector sector is located, and then the virtual voltage vector at the center line of the sector is selected as the optimal voltage vector to realize current tracking control.

[0078] Will I su The range of action is approximately a radius of |I su |circle. Because I su The magnitude of the virtual voltage vector T s / L times, we can get

[0079] The basic voltage vector V that constitutes the optimal voltage vector max 、V midL Calculate V by the following formula max 、V midL and the action time T of the zero voltage vector max 、T midL and T0.

[0080] Available, when IsDB Beyond I su The scope of action, that is, |I sDB |Greater than|I su |, the zero voltage vector is not used. sDB Not exceeding I su The scope of action, that is, |I sDB |Less than|I su |When the zero voltage vector is added, the duty cycle is optimized to improve the accuracy of current tracking control.

[0081] In each control cycle, the controller must output a large voltage vector, a medium-large voltage vector, and a zero voltage vector, or a large voltage vector and a medium-large voltage vector to the PWM module. In order to reduce the number of switching times of the power devices in the voltage source inverter, the base voltage vector is substituted into the switching cost function of formula (9). It is obtained that J i The smallest base voltage vector is selected and given priority. The specific flow chart is shown in Figure 13. i -a1) 2 +(b1 i -b1) 2 +(c1 i -c1) 2 +(a2 i -a2) 2 +(b2 i -b2) 2 +(c2 i -c2) 2 (9)

[0082] Where: a1, b1, c1, a2, b2, c2 are 0 and 1 to represent the upper arm power switch devices of the voltage source inverter A1, B1, C1, A2, B2, C2 in kT s The switch state at the moment, where 0 is off and 1 is on; a1 i 、b1 i 、c1 i 、a2 i 、b2 i 、c2 i Indicates the power devices on the six-phase upper bridge arm corresponding to the i-th base voltage vector at (k+1)T s The switch status at the moment.

[0083] Assume that the current voltage vector is V 24 , and will input VV to the PWM module 10 According to the traditional voltage vector action sequence, the base voltage vector action sequence is V 24 -V 51 -V 15If the base voltage vector to be applied is substituted into equation (9), we can get J i The smallest base voltage vector is selected and given priority. The final base voltage vector action order will become V 24 -V 15 -V 51 ,The effect is shown in Figure 14. The number of switches is reduced from 9 to 7.

[0084] When the motor is running, the optimization effect of the voltage source inverter switch is shown in Figure 15. It can be seen that the number of switches is significantly reduced under different working conditions. The current harmonics, torque, and speed waveforms of the motor in steady state are shown in Figures 16 and 17. It can be seen that the speed is stable at 300rpm, and the harmonic content of the A1 phase current is 1.44%. It proves that the invented control algorithm has good steady-state performance and also successfully suppresses current harmonics. The motor load and speed step waveforms are shown in Figures 18 and 19. It can be seen that when the load steps from 5Nm to 10Nm, the torque tracking is timely and accurate; when the speed steps from 200rpm to 300rpm, there is no obvious overshoot in the feedback speed. The above dynamic simulation fully proves that the motor of the present invention adopts the improved model prediction current control strategy with excellent steady-state and dynamic performance.

[0085] Although the present invention has been disclosed above with preferred embodiments, the embodiments are not intended to limit the present invention. Any equivalent changes or modifications made without departing from the spirit and scope of the present invention shall fall within the scope of protection limited by the appended claims of this application.

[0086] Throughout this specification, reference to terms such as "one embodiment," "some embodiments," "illustrative embodiments," "examples," "specific examples," or "some examples" means that a specific feature, structure, material, or characteristic described in conjunction with that embodiment or example is included in at least one embodiment or example of the present invention. In this specification, illustrative uses of the above terms do not necessarily refer to the same embodiment or example. Furthermore, the specific features, structures, materials, or characteristics described may be combined in any suitable manner in any one or more embodiments or examples.

[0087] While embodiments of the present invention have been shown and described, it will be appreciated by those skilled in the art that various changes, modifications, substitutions, and variations may be made to the embodiments without departing from the principles and spirit of the invention, and that the scope of the invention is defined by the claims and their equivalents.

Claims

1. A dual three-phase asymmetric stator type dual permanent magnet vernier motor, characterized by: The invention comprises a stator and a rotor, both of which are salient pole structures; the stator comprises a stator iron core (1), a double three-phase winding (2) and a stator permanent magnet (3); the stator iron core (1) is composed of a plurality of umbrella-shaped armature teeth (11) and rectangular armature teeth (12) arranged alternately along the circumferential direction; the rectangular armature teeth (12) are composed of a piece of iron core; the umbrella-shaped armature teeth (11) are composed of a sandwich array of "permanent magnet + iron core + permanent magnet", with a stator permanent magnet respectively attached to the openings on both sides; the stator permanent magnet (3) is attached to the openings on both sides of the umbrella-shaped armature teeth and adopts the same method. The polarity is magnetized radially outward; the dual three-phase winding (2) is respectively wound on the umbrella-shaped armature teeth (11) and the rectangular armature teeth (12), and the dual three-phase winding is a centralized winding; the rotor includes a rotor core (5) and a rotor ferrite (6); the rotor core (5) is shaped like a gear; the rotor ferrite (6) is embedded in the rotor core using a Halbach-based alternating pole array; the rotor ferrite adopts an unequal length structure, and its length is greater than the length of the rotor core and the length of the stator core; the stator core (1) and the rotor core (5) are both laminated silicon steel sheets.

2. The dual three-phase asymmetric stator dual permanent magnet vernier motor according to claim 1, characterized in that: The umbrella-shaped armature tooth (11) is in an inverted "T" shape in the circumferential cross section, and both sides of the tooth end are grooved in a fan-shaped manner.

3. The dual three-phase asymmetric stator dual permanent magnet vernier motor according to claim 1, characterized in that: The stator permanent magnet (3) is tile-shaped and pasted on the openings at both ends of the umbrella-shaped armature tooth. A fan-shaped auxiliary groove is opened between the umbrella-shaped armature tooth (11) and the stator permanent magnet (3). The auxiliary groove (4) is filled with non-magnetic material such as epoxy resin or without any filler.

4. The dual three-phase asymmetric stator dual permanent magnet vernier motor according to claim 1, characterized in that: The stator permanent magnet (3) is made of a high coercive force rare earth permanent magnet material such as a neodymium iron boron magnet, and is magnetized radially outward with the same polarity, and an alternating pole permanent magnet array is attached to the openings on both sides of the umbrella-shaped armature teeth; the rotor ferrite (6) is made of a low coercive force permanent magnet material such as a ferrite magnet, and is embedded in the rotor core using a Halbach array composed of radial (61) and parallel (62) magnetization. The magnetic focusing direction of the rotor ferrite is radially pointing to the stator, and the length of the rotor ferrite is greater than the length of the rotor core and the length of the stator core.

5. The dual three-phase asymmetric stator dual permanent magnet vernier motor according to claim 1, characterized in that: The maximum lengths of the two ferrites magnetized in the radial direction (61) and the parallel direction (62) along the circumferential direction are tr0=(0.25-0.5)*τ and tr1=(0-0.25)*τ, respectively, where τ is the pole pitch.

6. The dual three-phase asymmetric stator dual permanent magnet vernier motor according to claim 1, characterized in that: The dual three-phase winding (2) adopts a centralized winding structure and is wound on the umbrella-shaped armature teeth (11) and the quasi-rectangular armature teeth (12). The windings wound on the umbrella-shaped armature teeth are A1 phase, B1 phase and C1 phase, and the phase difference between A1 phase and B1 phase, A1 phase and C1 phase, and B1 phase and C1 phase is 120 degrees; the windings wound on the quasi-rectangular armature teeth are A2 phase, B2 phase and C2 phase, and the phase difference between A2 phase and B2 phase, A2 phase and C2 phase, and B2 phase and C2 phase is 120 degrees; the phase difference between A1 phase and A2 phase, B1 phase and B2 phase, and C1 phase and C2 phase is 30 degrees.

7. The dual three-phase asymmetric stator dual permanent magnet vernier motor according to claim 1, characterized in that: According to the bilateral excitation structure and bidirectional magnetic field modulation effect, the equivalent armature winding pole pair number P a , number of stator teeth N s , number of stator permanent magnet pole pairs P s , number of rotor teeth N r , rotor ferrite pole pair number P r , the five satisfy the following relationship: Where: p is the number of pole pairs in the unit stator symmetry period, and s is the number of slots in the unit stator symmetry period.

8. An improved model predictive current control method for a dual three-phase asymmetric stator dual permanent magnet vernier motor according to any one of claims 1 to 7, characterized in that: The steps include: Step 1: Establish a structural model of a dual three-phase asymmetric stator dual permanent magnet vernier motor; Step 2: Use the actual speed of the detected dual three-phase asymmetric stator dual permanent magnet vernier motor as the motor's feedback speed n, so that the motor's given speed n ref The motor speed error is obtained by subtracting n, and the q-axis reference current i of the motor is obtained through the PI controller. q ref , set the d-axis reference current i d ref =0, the stator current vector in the dq coordinate system is expressed as I s ref =i d ref +ji q ref , I s ref The projection on the α-β subplane is I s_αβ ref ; Step 3: Use Clarke transformation matrix to transform the dual three-phase current i A1 、i B1 、i C1 、i A2 、i B2 、i C2 The current i transformed into the α-β subplane α 、i β and the current i in the xy subplane x 、i y , and then use the Park transformation matrix to transform i α 、i β The current i transformed into the dq coordinate system d k ,i q k ; Step 4: Based on the principle of suppressing the current in the xy subspace, the base voltage vector is used to synthesize the virtual voltage vector VV i (i=1, 2, ..., 12), so that it is 0 in the xy subspace and its amplitude in the α-β subplane is 0.596U dc , where U dc is the DC bus voltage; the two base voltage vectors participating in the synthesis of the virtual voltage vector are large voltage vectors V with the same direction in the α-β sub-plane max and the medium and large voltage vector V midL , the virtual voltage vector is divided into sectors on the α-β sub-plane, the angle bisector of two adjacent virtual voltage vectors is the dividing line of each sector, and each virtual voltage vector is on the center line of its sector; Step 5: Establish the current equation of the dual three-phase asymmetric stator dual permanent magnet vernier motor in the dq coordinate system and discretize it to obtain Where: d k and i q k kT s The components of the current on the d-axis and q-axis at the moment i d k+1 、i q k+1 、u d k+1 and u q k+1 They are (k+1)T s The components of the current and voltage on the d and q axes at the moment; R is the stator resistance; L is the stator inductance; ω is the electrical angular velocity; Ψ f is the permanent magnet flux; T s is the control period; k represents kT s moment; i d 、i q 、u d 、u q are the components of current and voltage on the d and q axes; thus, the current-related term i is separated id k+1 、i iq k+1 and voltage-related terms i ud k+1 、i uq k+1 It is expressed in vector form as I in the dq coordinate system si =i id k+1 +ji iq k+1 , I su =i ud k+1 +ji uq k+1 , I si and I su The projection on the α-β subplane is I si_αβ and I su_αβ , where I su_αβ In phase with the virtual voltage vector, with an amplitude equal to the virtual voltage vector T s / L times; Step 6: Calculate the current increment vector I based on the reference current and feedback current sDB And its projection in the α-β coordinate system I sDB_αβ Phase angle θ αβ Where: ∠I sDB For I sDB Angle in the dq coordinate system; θ e is the rotor electrical angle; therefore, in order to track the reference current faster, it is necessary to make I su_αβ with I sDB_αβ The angle is the smallest; due to I su_αβ The phase angle is the same as the virtual voltage vector, then according to θ αβ Judgment I sDB_αβ The sector where the optimal voltage vector is located is the virtual voltage vector at the center line of the sector, thereby directly obtaining the optimal voltage vector and avoiding the continuous iteration of the voltage vector in the cost function. Step 7: Calculate the base voltage vector V corresponding to the optimal voltage vector max 、V midL and the action time T of the zero voltage vector max 、T midL and T0 is Step 8: In order to reduce the number of switching times of the power switching device in the voltage source inverter, the current voltage vector of the system and the voltage vector to be applied are substituted into the switching cost function <h2 style=";text-align:left;direction:ltr">J=(a1<h2 style=";text-align:left;direction:ltr"> i <h2 style=";text-align:left;direction:ltr"> -a1)<h2 style=";text-align:left;direction:ltr"> 2 <h2 style=";text-align:left;direction:ltr"> +(b1<h2 style=";text-align:left;direction:ltr"> i <h2 style=";text-align:left;direction:ltr"> -b1)<h2 style=";text-align:left;direction:ltr"> 2 <h2 style=";text-align:left;direction:ltr"> +(c1<h2 style=";text-align:left;direction:ltr"> i <h2 style=";text-align:left;direction:ltr"> -c1)<h2 style=";text-align:left;direction:ltr"> 2 <h2 style=";text-align:left;direction:ltr"> +(a2<h2 style=";text-align:left;direction:ltr"> i <h2 style=";text-align:left;direction:ltr"> -a2)<h2 style=";text-align:left;direction:ltr"> 2 <h2 style=";text-align:left;direction:ltr"> +(b2<h2 style=";text-align:left;direction:ltr"> i <h2 style=";text-align:left;direction:ltr"> -b2)<h2 style=";text-align:left;direction:ltr"> 2 <h2 style=";text-align:left;direction:ltr"> +(c2<h2 style=";text-align:left;direction:ltr"> i <h2 style=";text-align:left;direction:ltr"> -c2)<h2 style=";text-align:left;direction:ltr"> 2 Where: a1, b1, c1, a2, b2, c2 represent the power switching devices of the upper bridge arm of the voltage source inverter A1, B1, C1, A2, B2, C2 corresponding to the current base voltage vector in kT s The switch state at the moment, where 0 is off and 1 is on; a1 i 、b1 i 、c1 i 、a2 i 、b2 i 、c2 i The six-phase upper-arm power switch devices of the voltage source inverter corresponding to the i-th base voltage vector are at (k+1)T s The switch state at the moment, thus obtaining J i The smallest voltage vector gives priority to it.

9. The improved model predictive current control method based on a dual three-phase asymmetric stator dual permanent magnet vernier motor according to claim 8, characterized in that: The specific process of step 4 is as follows: Step 4.1, the switching state of the upper arm power device in the dual three-phase asymmetric stator dual permanent magnet vernier motor inverter corresponds to the phase voltage, which can be expressed as Step 4.2: Use the Clarke transformation matrix to transform the six-phase voltage to the two-phase stationary coordinate system, thereby obtaining the base voltage vectors of the α-β and xy sub-planes: The large voltage vector V in the α-β subplane 44 、V 64 、V 66 、V 26 、V 22 、V 32 、V 33 、V 13 、V 11 、V 51 、V 55 、V 45 The amplitude is 0.644U dc , medium and large voltage vector V 65 、V 46 、V 24 、V 62 、V 36 、V 23 、V 12 、V 31 、V 53 、V 15 、V 41 、V 54 The amplitude is 0.471U dc , small and medium voltage vector V 04 、V 74 、V 60 、V 67 、V 06 、V 76 、V 20 、V 27 、V 02 、V 72 、V 30 、V 37 、V 03 、V 73 、V 10 、V 17 、V 01 、V 71 、V 50 、V 57 、V 05 、V 75 、V 40 、V 47 The amplitude is 0.333U dc , small voltage vector V 56 、V 25 、V 42 、V 34 、V 63 、V 16 、V 21 、V 52 、V 35 、V 43 、V 14 、V 61 The amplitude is 0.173U dc ; Step 4.3, due to the large voltage vector V max and the medium and large voltage vector V midL The directions are consistent on the α-β sub-plane, but opposite on the xy sub-plane. In one control cycle, these two base voltage vectors are selected and then assigned a suitable action time, which is equivalent to a virtual voltage vector. The amplitude of the virtual voltage vector in the α-β sub-plane and the xy sub-plane must satisfy Let |V xy |=0, that is Where: t is V max Action time in one control cycle; |V αβ | is the composite voltage vector amplitude in the α-β sub-plane; |V xy | is the magnitude of the composite voltage vector in the xy subplane; therefore, when V max and V midL When the ratio of action time is 0.731:0.269, the amplitude of the resultant voltage vector in the α-β sub-plane is 0.596U dc , the amplitude of the xy subplane is 0; for convenience of representation, the synthetic voltage vector is represented as a virtual voltage vector VV i (i=1, 2, ..., 12); Step 4.4: divide the virtual voltage vector into sectors in the α-β coordinate system. The angle bisector of two adjacent virtual voltage vectors is the boundary line of each sector, and each virtual voltage vector is located at the center line of each sector.

10. The improved model predictive current control method based on a dual three-phase asymmetric stator dual permanent magnet vernier motor according to claim 8, characterized in that: In step 7, the range of action of the virtual voltage vector on the current is approximated as a radius of |I su |Circle, when I sDB When the virtual voltage vector is out of range, in order to track I s ref , zero voltage vector does not work; when I sDB When the virtual voltage vector does not exceed its action range, the zero voltage vector is introduced and the duty cycle is optimized, and the action time ratio of the zero voltage vector is 1-|I su | / |I sDB |.

Images