Power conversion device, motor drive device, and refrigeration cycle application device

Abstract

A power conversion device (1) comprises a converter (100), a capacitor (200), an inverter (300), and a control device (3) for driving and controlling each of the converter (100) and the inverter (300). The converter (100) and the inverter (300) are each provided with at least one current detection unit (301a, 301b) in order to detect a converter current flowing through the converter (100) and an inverter current flowing through the inverter (300). The control device (3) is provided with a capacitor current estimation unit (700) that uses the detection values of the converter current and the inverter current detected by the current detection units (301a, 301b), sets the capacitor current flowing through the capacitor (200) as a calculation target, switches calculation processing according to the operation state of the converter (100), and performs an estimation calculation of the capacitor current.

Description

Power conversion devices, motor drive devices, and refrigeration cycle application equipment

[0001] The present disclosure relates to a power conversion device that converts AC power into desired power, a motor drive device, and a refrigeration cycle device.

[0002] Conventionally, a power conversion device that converts AC voltage from an AC power source into DC voltage and then converts the DC voltage into a desired AC power has been provided with a smoothing capacitor to reduce fluctuations in the ripple voltage of the DC voltage. A ripple current corresponding to the ripple voltage of the DC voltage flows through this capacitor.

[0003] The ripple current flowing through a smoothing capacitor causes the capacitor to heat up. Furthermore, if excessive ripple current continues to flow through a smoothing capacitor, the capacitor may fail or its lifespan may be shortened. One way to reduce the ripple current in a capacitor is to increase its capacitance. However, increasing the capacitance of the capacitor increases the cost and weight of the device. Preventing increases in the cost and weight of the device is a common challenge that is always required of power conversion devices, regardless of the application.

[0004] Various control techniques for reducing capacitor ripple current have been proposed. For example, there is a technique for appropriately controlling the motor current output from an inverter so as to suppress pulsation in the current flowing into and out of the capacitor. However, in any of these control techniques, if the control for reducing the capacitor ripple current does not function properly, excessive ripple current continues to flow through the capacitor, making it difficult to avoid capacitor failure or deterioration of its lifespan. Therefore, it is important to understand the ripple current flowing through the capacitor during operation of a power conversion device. Patent Document 1 listed below, for example, discloses a technique for estimating the capacitor current flowing through a capacitor. Patent Document 1 discloses a technique for estimating the capacitor current using a detected DC voltage value and a capacitor capacitance value.

[0005] Japanese Patent Application Laid-Open No. 2005-295712

[0006] However, in the technology of Patent Document 1, the capacitor capacitance value used to estimate the capacitor current is subject to errors due to individual differences and changes over time relative to the nominal value, which results in errors in the estimated capacitor current.

[0007] The present disclosure has been made in consideration of the above, and aims to provide a power conversion device that can accurately estimate capacitor current even when the capacitor capacitance value has individual differences or changes over time from the nominal value.

[0008] In order to solve the above-mentioned problems and achieve the object, a power conversion apparatus according to the present disclosure includes a converter, a capacitor, and an inverter, as well as a control device that drives and controls the converter and the inverter, respectively. The converter includes at least one switch element and converts a first AC voltage applied from an AC power source into a DC voltage. The capacitor smoothes the voltage output by the converter. The inverter converts the DC voltage smoothed by the capacitor into a second AC voltage and applies the converted second AC voltage to a motor to rotate it. The converter and the inverter each include at least one current detection unit for detecting a converter current flowing through the converter and an inverter current flowing through the inverter. The control device drives and controls the converter and the inverter, respectively. The control device also includes a capacitor current estimation unit that uses the detected values ​​of the converter current and the inverter current detected by the current detection unit, and switches calculation processing depending on the operating state of the converter to estimate the capacitor current, with the capacitor current flowing through the capacitor as the calculation target.

[0009] The power conversion device according to the present disclosure has the advantage of being able to accurately estimate the capacitor current even when the capacitor capacitance value varies from the nominal value depending on the individual capacitor or changes over time.

[0010] FIG. 1 is a diagram showing an example of the configuration of a power conversion device and a motor drive device according to the first embodiment; FIG. 2 is a diagram explaining the operation of a converter provided in the power conversion device according to the first embodiment; FIG. 3 is a diagram explaining the operation of a converter control unit and a converter according to the first embodiment; FIG. 4 is a block diagram showing an example of the configuration of an inverter control unit provided in the control device according to the first embodiment; FIG. 5 is a diagram showing an example of the configuration of an inverter according to the first embodiment; FIG. 6 is a diagram showing eight switching patterns in the inverter according to the first embodiment; FIG. 1 is a block diagram showing an example of the configuration of a power conversion device and a motor drive device according to a second embodiment; FIG. 2 is a diagram explaining the operation of a converter provided in the power conversion device according to the second embodiment; FIG. 3 is a block diagram showing an example of the configuration of an inverter control unit and a capacitor current estimating unit provided in a control device according to the third embodiment;

[0011] Hereinafter, a power conversion device, a motor drive device, and a refrigeration cycle applied device according to embodiments of the present disclosure will be described in detail with reference to the accompanying drawings.

[0012] First Embodiment Fig. 1 is a diagram showing an example of the configuration of a power conversion device 1 and a motor drive device 2 according to a first embodiment. The power conversion device 1 is connected to a commercial power source 110 and a compressor 400. The power conversion device 1 converts first AC power supplied from the commercial power source 110 into second AC power having a desired amplitude and phase, and supplies the second AC power to the compressor 400. The commercial power source 110 may be single-phase or three-phase. In this paper, a case where the commercial power source 110 is single-phase will be described.

[0013] The compressor 400 shown in Fig. 1 is an example of a mechanical device to be driven. The compressor 400 includes a motor 401 driven by the power conversion device 1 and a mechanical load 402. The motor 401 drives the mechanical load 402, and the mechanical load 402 performs mechanical work. In this paper, the compressor 400 will be described as an example, but the present invention is also applicable to mechanical devices other than the compressor 400. The motor drive device 2 according to the first embodiment is configured by the power conversion device 1 and the motor 401 provided in the compressor 400.

[0014] In the compressor 400, a periodic load torque changes when the refrigerant is drawn in, compressed, and discharged. When the speed of the motor 401 is approximately constant, the load torque in the compressor 400 has a periodic pulsating waveform. Note that the mechanical structure of the compressor 400 for compressing the refrigerant is not particularly limited. The compressor 400 may be a rotary compressor, a reciprocating compressor, a scroll compressor, or a screw compressor. Alternatively, the compressor 400 may be a compressor other than these.

[0015] 1, the power conversion device 1 includes a converter 100, a capacitor 200, an inverter 300, current detection units 301a and 301b, voltage detection units 201a and 201b, and a control device 3. The control device 3 controls the driving of the converter 100 and the inverter 300 based on at least one of the detection values ​​detected by the current detection units 301a and 301b and the voltage detection units 201a and 201b.

[0016] Converter 100 is connected between commercial power supply 110 and capacitor 200. Converter 100 is a power converter that includes at least one switch element and converts a first AC voltage applied from commercial power supply 110, which is an AC power supply, into a DC voltage and outputs the DC voltage. Capacitor 200, voltage detection unit 201b, and inverter 300 are connected to the output terminal of converter 100.

[0017] When obtaining DC voltage from an AC power source, it is common to use a power factor correction circuit, which has the functions of controlling the AC current so as to comply with harmonic standards and suppressing fluctuations in the output DC bus voltage.

[0018] The converter 100 shown in FIG. 1 is configured with a power factor correction circuit called a simplified switching circuit. Specifically, the converter 100 includes a reactor 120 installed on the input side of a commercial power supply 110, a diode rectifier 130, and a simplified switching cell 160 in which a switch element 150 is connected to two diodes 140a and 140b. Current and voltage detection in the converter 100 can be performed using a current detection unit 301a and voltage detection units 201a and 201b, as shown in FIG. 1 . When open-loop control is applied, the current detection unit 301a and voltage detection units 201a and 201b shown in FIG. 1 do not need to be used. Although the switch element 150 in FIG. 1 is represented by the symbol IGBT (Insulated Gate Bipolar Transistor), a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) may also be used; the switch element 150 is not limited to an IGBT. 1 illustrates a circuit configuration including a simple switching cell 160 as an example of the converter 100 using a power factor correction circuit, but other circuit configurations may also be used. Examples of other power factor correction circuits will be described in the following embodiments.

[0019] As shown in FIG. 1 , the current detection unit 301a is installed at the lower output terminal of the diode rectifier 130. The lower output terminal of the diode rectifier 130 is the side where the current resulting from the power rectified by the diode rectifier 130 is fed back to the commercial power supply 110. The current detection unit 301a detects the converter current Iconv flowing through this portion and outputs the detected current value to the control device 3. The installation position of the current detection unit 301a is not limited to the position shown in FIG. 1 , and the current detection unit 301a may be installed at any position where the converter current Iconv can be detected or estimated. For example, instead of installing the current detection unit 301a at the lower output terminal of the diode rectifier 130, the current detection unit 301a may be installed at a position where the input AC current Iac flowing through the reactor 120 is detected. In this case, by using the relationship Iconv = |Iac|, the converter current Iconv can be estimated and used for control calculations in the control device 3.

[0020] Capacitor 200 is a smoothing capacitor that smoothes the voltage output by converter 100. Examples of capacitor 200 include an electrolytic capacitor and a film capacitor. Capacitor 200 only needs to have a capacity that can smooth the power rectified by converter 100 to some extent.

[0021] The inverter 300 is connected to the capacitor 200 and the motor 401. The inverter 300 converts the DC voltage smoothed by the capacitor 200 into a second AC voltage by switching an internal switch element, and applies the converted second AC voltage to the motor 401 to drive and rotate the motor 401. The second AC voltage has a desired amplitude, frequency, and phase. The motor 401 rotates at a desired rotational speed by the second AC voltage.

[0022] The current detection unit 301b detects the inverter current Iinv fed back from the inverter 300 at least twice per control cycle of the inverter 300 and outputs the detected current value to the control device 3. The inverter current Iinv is a current flowing in and out of the inverter 300. When the motor 401 is a three-phase Y-connection, a technique for reconstructing the three-phase currents Iu, Iv, and Iw flowing through the motor 401 from the inverter current Iinv is known, and this makes it possible to reconstruct the three-phase currents Iu, Iv, and Iw flowing through the motor 401. Of course, a current detector may be installed to directly detect the three-phase currents Iu, Iv, and Iw output from the inverter 300 and output the detected values ​​to the control device 3.

[0023] The control device 3 receives the detected value of the converter current Iconv detected by the current detection unit 301a, the detected value of the inverter current Iinv detected by the current detection unit 301b, and the detected value of the DC bus voltage Vdc detected by the voltage detection unit 201b, and performs various control calculations using a converter control unit 500, an inverter control unit 600, and a capacitor current estimation unit 700, which will be described later, to generate various switching signals Q ０ , Q １ and performs an estimation calculation to estimate the capacitor current Ic.

[0024] Although not shown, the control device 3 is configured to include a CPU (Central Processing Unit), ROM (Read Only Memory), RAM (Random Access Memory), and various interface electronic circuits such as an A / D (Analog-Digital) converter. The control device 3 is also configured to read out a program stored in the ROM, load it into the RAM, and have the CPU execute various processes. Note that, in FIG. 1, one control device 3 is configured to realize the functions of the converter control unit 500, the inverter control unit 600, and the capacitor current estimating unit 700, but each function may be implemented as a separate control device, i.e., a control device including various interface electronic circuits such as a CPU, ROM, RAM, A / D converter, etc.

[0025] In a typical power conversion device in a refrigeration cycle device, the converter 100 and the inverter 300 have conventionally been provided with at least one current detection unit each to realize the functions of the converter control unit 500 and the inverter control unit 600 described below or to protect the switch elements of the converter 100 and the inverter 300 from overcurrent. Therefore, if the capacitor current Ic can be estimated using the current detection units provided in the converter 100 and the inverter 300, there is no need to provide an additional current detection unit in the capacitor 200. Therefore, in the power conversion device 1 according to the first embodiment, a capacitor current estimation unit 700 is provided inside the control device 3 to estimate the capacitor current Ic flowing through the capacitor 200 based on the detected values ​​of the converter current Iconv and the inverter current Iinv detected by the current detection units 301a and 301b provided in the power conversion device 1. In addition, the method of estimating the capacitor current Ic using the detection values ​​of the current detection units 301a and 301b has the advantage that errors due to differential calculations or errors due to variations in the capacitor capacitance value do not occur, compared to the conventional method of estimating it from the voltage detection value.

[0026] Next, the control calculations performed in the converter control unit 500, the inverter control unit 600, and the capacitor current estimating unit 700 configured in the control device 3 according to the first embodiment will be described in detail.

[0027] The converter control unit 500 references at least the detected value of the AC voltage Vac or its synchronous signal to generate a switching signal Q which is an ON / OFF command value for the switch element 150 of the converter 100. ０ As shown in FIG. 1, the converter control unit 500 performs a calculation by inputting the detected values ​​of the DC bus voltage Vdc and the converter current Iconv, and by feedback control, outputs a switching signal Q ０A typical feedback control method is PID (Proportional Integral Differential) control. Alternatively, the switching signal Q is determined by open loop control without using the detected voltage value and the detected current value, by referring to a switching pattern stored in advance in the control device 3. ０ A control method for determining the switching signal Q is also known, and this known method may be used. ０ The switching method may be a high-frequency switching method or a switching method called a simple switching method, which is also called a partial switching method, in which switching control is performed on the switch element 150 of the converter 100 only once or a few times per half cycle of the AC power supply.

[0028] Next, the operation of the converter control unit 500 and the converter 100 will be described with reference to several drawings. Fig. 2 is a diagram illustrating the operation of the converter 100 provided in the power conversion device 1 according to the first embodiment.

[0029] 2 , the path of the current flowing due to the power supplied from the commercial power supply 110 is switched by switching control of the switch element 150 provided in the converter 100, i.e., by on / off (hereinafter referred to as “ON / OFF”) control of the switch element 150. Specifically, when the switch element 150 is controlled to be OFF, the input AC current Iac flowing from the commercial power supply 110 is rectified by the diode rectifier 130 via the reactor 120 as shown by the thick dashed line in the figure, and becomes a charging current flowing to the capacitor 200. At this time, no current flows through the simple switching cell 160. When the switch element 150 is controlled to be ON, as shown by the thick solid line in the figure, a short-circuit current flows through the path that passes through the simple switching cell 160, and magnetic energy is stored in the reactor 120 via the diode rectifier 130. At this time, the capacitor 200 is not charged by the power from the commercial power supply 110. When the switching element 150 is switched from ON to OFF by the switching control of the switching element 150, the magnetic energy stored in the reactor 120 is simultaneously transferred to the capacitor 200 via the diode rectifier 130. In the converter 100 equipped with the power factor correction circuit of the first embodiment, this series of operations makes it possible to widen the conduction angle of the input AC current Iac compared to when no switching control is performed, i.e., when the switching element 150 is always OFF, and therefore it is possible to improve the power factor to a certain extent. Note that, regardless of the ON / OFF control of the switching element 150, a current due to the discharge of the capacitor 200 flows to the inverter 300.

[0030] 3 is a diagram illustrating the operation of the converter control unit 500 and the converter 100 according to the first embodiment. The left side of FIG. 3 shows the AC voltage Vac of the commercial power supply 110, the input AC current Iac, and the switching signal Q when the converter control unit 500 and the converter 100 are operated in the "(a) simple switching method." ０ 3 shows waveforms of the AC voltage Vac, the input AC current Iac, and the switching signal Q when the converter control unit 500 and the converter 100 are operated in the "(b) high frequency switching method." ０The waveforms of the switching signal Q are shown at the bottom of each of Figs. 3(a) and 3(b), where a "High" level is "ON" and a "Low" level is "OFF". ０ is shown as an example.

[0031] In the simple switching method, the switch element 150 is controlled to be turned on and off one or more times per half power supply cycle using open-loop control. This causes the input AC current Iac to flow as shown in FIG. 3(a). While FIG. 3 illustrates a case in which the switch element 150 is controlled to be switched on only once per half power supply cycle, the number of times the switch element 150 is controlled may be any number of times. The simple switching method has a lower power factor correction function than the high-frequency switching method, but the reduced number of switching times allows the switching loss of the switch element 150 to be kept low. The simple switching method can control the energy stored in the reactor 120 by controlling the short circuit start time Tdl, short circuit duration Ton, and switching frequency Nsw. The short circuit start time Tdl is the time from the zero-crossing point of the AC voltage Vac to when the switch element 150 is first turned on. The short circuit duration Ton is the time during which the switch element 150 remains on. The number of switching operations Nsw is the number of times per half power supply cycle that the switch element 150 is turned on / off. The DC bus voltage Vdc can be steplessly boosted to any higher voltage value compared to when the switch element 150 is not subjected to switching control.

[0032] The high-frequency switching method is a method of feedback-controlling the DC bus voltage Vdc and the input current Iin. The input current Iin is a current flowing from the converter 100 to the capacitor 200 and the inverter 300. In the high-frequency switching method, as shown in FIG. 3(b), the switching frequency of the simple switching method is set to several kHz or more, and the ON / OFF times of the switch element 150 are controlled so that the DC bus voltage Vdc is at a desired value and the input AC current Iac approaches a sine wave synchronized with the AC voltage Vac. Furthermore, as shown in FIG. 3(b), the high-frequency switching method can make the waveform of the input AC current Iac approach a sine wave synchronized with the AC voltage Vac, thereby improving the power factor to approximately 1. Furthermore, the high-frequency switching method has a higher boost capability than the simple switching method. Therefore, the DC bus voltage Vdc in the high-frequency switching method can be steplessly boosted to any desired voltage value, higher than when switching control of the switch element 150 is not performed.

[0033] The control method in converter control unit 500 of the first embodiment is not limited to the simple switching method and high-frequency switching method described above, and various known control methods may be used.

[0034] FIG. 4 is a block diagram showing an example of the configuration of the inverter control unit 600 provided in the control device 3 of the first embodiment. The inverter control unit 600 includes a speed control unit 601, a d-axis current command determination unit 602, a current restoration unit 603, a current control unit 604, a position estimation unit 605, and a PWM (Pulse Width Modulation) signal generator 606. The operation of the inverter control unit 600 will be described below. Here, the case where control is performed on a dq rotating coordinate system synchronized with the rotor position of the motor 401 is described. However, the dq rotating coordinate system is merely an example, and control may be performed on other coordinate systems. Furthermore, control on the dq rotating coordinate system requires information about the rotor position of the motor 401. The rotor position of the motor 401 may be detected by a position sensor (not shown). Alternatively, the position estimation unit 605 may calculate a speed electromotive force using the output voltage of the inverter 300 and the current flowing through the motor 401, and estimate the rotor position using the calculated speed electromotive force.

[0035] The speed control unit 601 outputs the q-axis current command Iq* generated by performing speed control so that the speed command ω* coincides with the estimated speed ωest estimated by the position estimation unit 605. PID control is a well-known speed control technique.

[0036] The d-axis current command determiner 602 determines a d-axis current command Id* and outputs it to the current controller 604. Although not shown, the d-axis current command Id* is determined by referring to the three-phase voltage command Vuvw*, the q-axis current command Iq*, the speed command ω*, etc. Typical methods for determining the d-axis current command Id* include a method for determining the d-axis current command Id* so that the motor drive device 2 achieves maximum efficiency driving, and a method for determining the d-axis current command Id* by flux-weakening control so as to suppress voltage saturation of the motor 401.

[0037] The current restoration unit 603 uses the detected value of the inverter current Iinv detected at least twice per control period of the inverter 300 to calculate a three-phase current restoration value Iuvw, which is a restoration value of the three-phase currents Iu, Iv, and Iw of the motor 401. The operation of the current restoration unit 603 will be further described with reference to several drawings. In the following description, the portion between the capacitor 200 and the inverter 300 through which the inverter current Iinv flows will be referred to as the "DC bus portion."

[0038] Fig. 5 is a diagram showing a configuration example of an inverter 300 according to the first embodiment. In Fig. 5, the inverter 300 includes six switch elements Sup, Sun, Svp, Svn, Swp, and Swn. In this document, the switch elements Sup, Svp, and Swp are referred to as "upper arm switch elements" or simply "upper arms" as appropriate, and the switch elements Sun, Svn, and Swn are referred to as "lower arm switch elements" or simply "lower arms" as appropriate.

[0039] As described above, the current restoration unit 603 calculates the three-phase current restoration value Iuvw using the detected value of the inverter current Iinv, and in doing so, uses information about the switching states of the six switch elements Sup, Sun, Svp, Svn, Swp, and Swn of the inverter 300.

[0040] FIG. 6 is a diagram showing eight switching patterns in the inverter 300 of the first embodiment. The switching patterns are patterns of switching states in the inverter 300. The six switch elements Sup, Sun, Svp, Svn, Swp, and Swn of the inverter 300 are switched by the inverter control unit 600 using different combinations of ON / OFF control switching patterns. These switching patterns can be classified into eight switching patterns (a) to (h) as shown in FIG. 6. In this case, the relationship between the three-phase currents Iu, Iv, and Iw and the inverter current Iinv is as shown in (a) to (h) in FIG. 6, respectively.

[0041] First, let us consider "(a) All upper arms OFF" and "(h) All upper arms ON." In either of these cases, no power is supplied from the DC bus, and the three-phase currents Iu, Iv, and Iw flowing through the motor 401 circulate through the upper arms or lower arms of the motor 401 and the inverter 300. Therefore, no current flows through the DC bus, and Iinv=0.

[0042] Next, let us consider "(b) Only the U-phase upper arm is ON." In this case, power supplied from the DC bus is supplied to the motor 401 via the U-phase upper arm and is returned to the DC bus via the V-phase lower arm and the W-phase lower arm. Therefore, Iinv = -Iv - Iw, and from the relationship Iu + Iv + Iw = 0 according to Kirchhoff's law, Iinv = Iu.

[0043] For the other switching patterns (c) to (g), the relationship between the three-phase currents Iu, Iv, Iw and the inverter current Iinv can be derived in the same way as for (b). The current restoration unit 603 restores the three-phase currents Iu, Iv, Iw from the inverter current Iinv by using the relationship between the three-phase currents Iu, Iv, Iw and the inverter current Iinv shown in FIG.

[0044] FIG. 6 shows that the three-phase currents Iu, Iv, and Iw can be reconstructed from the inverter current Iinv. However, only one of the three-phase currents Iu, Iv, and Iw can be reconstructed from a single detection value of the inverter current Iinv. To achieve control by the inverter control unit 600, all values ​​of the three-phase currents Iu, Iv, and Iw are required for each control period. Even if one phase is calculated using Kirchhoff's law, the currents for at least the remaining two phases must be detected. That is, to achieve control by the inverter control unit 600, the inverter current Iinv must be detected at least twice for each control period of the inverter control unit 600. Furthermore, current detection is not possible when the switching state of the inverter 300 is "(a) all upper arms OFF" or "(h) all upper arms ON." Therefore, current detection must be performed at least twice when the switching state is one of (b) to (g), and the currents that can be detected at each detection must be different phase currents.

[0045] Fig. 7 is a time chart illustrating the operation of the inverter 300 and the inverter control unit 600 according to embodiment 1. Fig. 7 shows the phase voltage commands Vu*, Vv*, Vw* applied to the respective phases of the inverter 300, the switching signals applied to the switch elements Sup, Svp, Swp of the upper arm of the inverter 300, the inverter current Iinv, the timing of detection of the inverter current Iinv, and the like.

[0046] 7 shows an example in which the magnitude relationship between the phase voltage commands Vu*, Vv*, and Vw* of the respective phases is Vu* > Vv* > Vw*. The switching signals of the upper arm switch elements are determined by triangular wave comparison PMW with the respective phase voltage commands Vu*, Vv*, and Vw*, and are as shown in the second to fourth rows of FIG. 7. As a result, the switching pattern of the inverter 300 is as shown in the bottom row.

[0047] The waveform of the inverter current Iinv in Figure 7 is indicated by black dots, which indicate the timing at which the inverter current Iinv is detected. As shown in Figure 7, by detecting the inverter current Iinv when the switching patterns during the carrier cycle are (b) and (c), it is possible to detect the U-phase current Iu and the W-phase current Iw. Furthermore, if the U-phase current Iu and the W-phase current Iw can be detected, then Kirchhoff's law also allows the V-phase current Iv to be detected.

[0048] 7, the detection timings of the switching patterns (b) and (c) for detecting the inverter current Iinv are included in the period in which the carrier wave is monotonically decreasing. That is, in the example of FIG. 7, the inverter current Iinv is detected once each when the switching patterns (b) and (c) are included in the half carrier cycle in which the carrier wave is monotonically decreasing. This makes it possible to restore all of the three-phase currents Iu, Iv, and Iw for each control cycle of the inverter control unit 600 based on the detected value of the inverter current Iinv.

[0049] In FIG. 7 , the switching patterns (b) and (c) can be rephrased as the times before and after the switching state of the V-phase switching element Svp changes, as can be seen from the waveform of the V-phase switching element Svp in the third row. Therefore, the detection timing of the inverter current Iinv can be rephrased as simply detecting the inverter current Iinv once before and once after the switching state of the switching element Svp changes. Here, FIG. 7 illustrates a case where the relationship between the phase voltage commands Vu*, Vv*, and Vw* of each phase is Vu* > Vv* > Vw*, and the phase voltage command Vv* is the middle phase when the phase voltage commands Vu*, Vv*, and Vw* of each phase are arranged in descending order. Therefore, it can be generalized that the detection of the inverter current Iinv can be rephrased as simply detecting the inverter current Iinv once before and once after the switching state of the upper arm switching element of the middle phase changes.

[0050] As explained above, the current reconstruction unit 603 detects the inverter current Iinv once before and once after the switching state of the intermediate phase changes for each carrier down half cycle in which the carrier wave monotonically decreases, thereby reconstructing the current values ​​for two phases of the three-phase currents Iu, Iv, and Iw. The current value for the remaining phase is then calculated using Kirchhoff's law to reconstruct the three-phase currents Iu, Iv, and Iw from the detected value of the inverter current Iinv. Note that the example shown here is merely an example, and values ​​detected three or more times per carrier cycle may also be used. For example, while FIG. 7 illustrates two detections per carrier down half cycle, it is also possible to detect two detections per carrier down half cycle and two detections per carrier up half cycle and use the average values ​​of the detection values ​​for each half cycle. Furthermore, current detection may be performed multiple times for each of the switching patterns (b) to (g), and the average values ​​may be used for current reconstruction. In these cases, although the amount of calculation increases, it is possible to average the changes in the phase currents during one carrier cycle to some extent and reconstruct the current.

[0051] The current control unit 604 performs current control based on the d-axis current command Id*, the q-axis current command Iq*, the three-phase current restored value Iuvw, and the estimated position θest, and outputs a three-phase voltage command Vuvw*. As with speed control, PID control is a well-known current control method.

[0052] The position estimator 605 calculates a speed electromotive force based on the three-phase voltage command Vuvw* and the three-phase current restored value Iuvw, and estimates the rotor position and speed based on the calculated speed electromotive force. Various methods have been proposed for position sensorless control of the motor 401, and for example, sensorless vector control using an adaptive observer is well known.

[0053] The PWM signal generator 606 generates a PWM signal based on the three-phase voltage command Vuvw* and the DC bus voltage Vdc. The generated PWM signal is used as a switching signal Q to the inverter 300. １ The signal is output to the inverter 300 as a PWM signal. Carrier comparison modulation is well known as a method for generating a PWM signal, but other methods may also be used.

[0054] As described above, the inverter control unit 600 outputs the switching signal Q １ The inverter 300 performs a switching operation in accordance with the PWM signal, thereby enabling the motor 401 to operate in accordance with the desired speed command ω*.

[0055] 8 is a block diagram showing an example of the configuration of the capacitor current estimating unit 700 provided in the control device 3 of embodiment 1. The capacitor current estimating unit 700 is configured to include an estimated current calculating unit 710 and a current effective value calculating unit 720.

[0056] The estimated current calculation unit 710 calculates the detected value of the converter current Iconv, the detected value of the inverter current Iinv, and the switching signal Q ０ The estimated capacitor current Ic_est is calculated based on the above.

[0057] In the circuit configuration of the power conversion device 1 shown in FIG. 1 , the capacitor current Ic flowing through the capacitor 200 satisfies the relationship Ic = Iin - Iinv according to Kirchhoff's law. Therefore, by detecting the input current Iin in addition to the inverter current Iinv, it is possible to estimate the capacitor current Ic without installing a current detector in the electrical wiring connected to the capacitor 200. Here, reference is made to the current path when the switch element 150 of the converter 100 is OFF and the current path when it is ON, as shown in FIG. 2 . When the switch element 150 of the converter 100 is ON, the input current Iin is zero. In this case, the capacitor current Ic flowing through the capacitor 200 is Ic = -Iinv. Furthermore, when the switch element 150 of the converter 100 is OFF, the input current Iin is equal to the converter current Iconv, so the capacitor current Ic flowing through the capacitor 200 is Ic = Iconv - Iinv.

[0058] Taking the above relationship into consideration, the estimated current calculation unit 710 calculates the detected value of the converter current Iconv, the detected value of the inverter current Iinv, and the switching signal Q of the converter 100. ０ Based on this, the estimated capacitor current Ic_est is calculated according to the following equation (1).

[0059]

[0060] At least in the power conversion device 1 assumed in embodiment 1, the method of using the detection values ​​of the current detection units 301a, 301b installed in the converter 100 and the inverter 300, respectively, and switching the calculation formula depending on the switching state of the switch element 150 of the converter 100 to calculate the estimated capacitor current Ic_est, which is an estimated value of the capacitor current Ic, is considered to be a new method that, as far as the inventors know, has not been mentioned in other literature.

[0061] As described above, the new method according to the first embodiment makes it possible to estimate the capacitor current Ic by calculating the estimated capacitor current Ic_est, without installing an additional new current detection unit at the portion of the electrical wiring that connects the capacitor 200. Furthermore, the new method according to the first embodiment directly uses the detection values ​​detected by the current detection units 301a and 301b, as shown in equation (1), and therefore has the advantage of being unaffected by errors in the differential operation and the capacitor capacitance value, compared to the conventional method of making an estimation by referring to the DC bus voltage Vdc and the capacitor capacitance value.

[0062] The effective current value calculation unit 720 receives the estimated capacitor current Ic_est as an input and calculates the effective value of the estimated capacitor current Ic_est. Here, the effective value of the estimated capacitor current Ic_est is the effective current value obtained by normalizing the estimated capacitor current Ic_est by a reference frequency according to the frequency characteristics of the temperature rise caused by the ripple current of the capacitor 200.

[0063] After a sufficient amount of time has passed since the start of power supply from the commercial power source 110, and in a steady state where the motor 401 is operating at a constant load and a constant speed, the capacitor current Ic becomes an AC current with a DC component, which is the average value of the capacitor current Ic, of 0. Therefore, the magnitude of the capacitor current Ic is determined by the peak value or effective value of the capacitor current Ic.

[0064] When a current flows through capacitor 200, capacitor 200 generates heat due to the influence of its internal resistance. Because the internal resistance of capacitor 200 has a frequency characteristic, the lower the frequency, the greater the internal resistance. Therefore, the lower the frequency of capacitor current Ic, the greater the temperature rise in capacitor 200. The AC component of capacitor current Ic includes multiple AC components, such as an AC component synchronized with the frequency of commercial power supply 110 and an AC component synchronized with fluctuations in the power consumption of mechanical load 402. Therefore, determining the magnitude of capacitor current Ic is preferably based on its effective value rather than its peak value. Furthermore, to determine the magnitude of the influence on the temperature rise of capacitor 200, it is preferable to calculate the effective current value normalized by a reference frequency according to the frequency characteristic of capacitor 200's internal resistance.

[0065] Based on the above, the current effective value calculation unit 720 performs frequency analysis using the estimated capacitor current Ic_est as an input, and calculates the amplitude value of each frequency component contained in the estimated capacitor current Ic_est. Although the fast Fourier transform (FFT) is a well-known frequency analysis method, a frequency analysis method other than FFT may also be used. After performing the frequency analysis, the current effective value calculation unit 720 calculates the amplitude value Ic_f of each frequency component. ０ , Ic_f １ , Ic_f ２ , ..., Ic_f Ｎ and frequency correction coefficient K(f ０ ), K(f １ ), K(f ２ ), ..., K(f Ｎ ) and calculates the frequency-corrected estimated capacitor current effective value Ic_est_rms according to the following equation (2).

[0066]

[0067] In the above equation (2), the frequency correction coefficient K(f ０ ), K(f １ ), K(f ２ ), ..., K(f Ｎ) may be defined as a function of the frequency f. Alternatively, the frequency correction coefficient K(f ０ ), K(f １ ), K(f ２ ), ..., K(f Ｎ ) may be referred to by the current effective value calculation unit 720.

[0068] In the above equation (2), when calculating the effective value of the estimated capacitor current Ic_est by adding up the frequency components, the amplitude values ​​Ic_f of all the frequency components obtained by frequency analysis are used. ０ , Ic_f １ , Ic_f ２ , ..., Ic_f Ｎ and frequency correction coefficient K(f ０ ), K(f １ ), K(f ２ ), ..., K(f Ｎ ) to calculate the effective value, and the amplitude value Ic_f ０ , Ic_f １ , Ic_f ２ , ..., Ic_f Ｎ In addition, in the frequency analysis described above, the amplitude values ​​Ic_f of all frequency components may be used and summed. ０ , Ic_f １ , Ic_f ２ , ..., Ic_f Ｎ It is not necessary to calculate Ic_est_rms, and only the amplitude value of a specific frequency component may be calculated. By taking these points into consideration, it becomes possible to calculate the estimated capacitor current effective value Ic_est_rms with a small calculation load. A well-known method for calculating the amplitude value of a specific frequency component is calculation using Fourier series expansion.

[0069] As described above, the AC component of the capacitor current Ic includes various AC components, such as an AC component synchronized with the frequency of the commercial power supply 110 and an AC component synchronized with fluctuations in the power consumption of the compressor 400. The AC voltage Vac applied to the diode rectifier 130 is full-wave rectified by the diode rectifier 130. As a result, the AC components synchronized with the frequency of the commercial power supply 110 have large current amplitudes, including a component twice the frequency of the AC voltage Vac applied from the commercial power supply 110 and its harmonic components. Therefore, when the effective current value calculator 720 calculates the amplitude value of a specific frequency component, the estimated current calculator 710 calculates the amplitude value of a component contained in the capacitor current Ic that is an integer multiple of the frequency of the AC voltage Vac. In this way, the effective current value calculator 720 can calculate the effective value of the AC component contained in the capacitor current Ic that is synchronized with the frequency of the AC voltage Vac.

[0070] Next, an example of an AC component synchronized with fluctuations in power consumption of the compressor 400 will be described. As described above, in the compressor 400, periodic load torque pulsation occurs when the refrigerant is drawn in, compressed, and discharged. This load torque pulsation is generally synchronized with the rotation speed of the compressor 400. When the torque generated by the motor 401 is constant, the load torque pulsation causes speed pulsation in the rotation speed of the compressor 400.

[0071] In the prior art, to suppress this type of speed pulsation, pulsation suppression control is performed in which the motor torque pulsates in synchronization with the load torque pulsation. In this pulsation suppression control, the motor torque pulsates. Therefore, the inverter current Iinv and the motor current, which are the DC loads from the capacitor 200, fluctuate in synchronization with the load torque pulsation. Furthermore, the capacitor current Ic also experiences current pulsation synchronized with the load torque pulsation, which is expected to increase the current amplitude of its frequency component. Therefore, it is desirable for the RMS current calculator 720 to calculate the amplitude value of a component synchronized with the load torque pulsation, i.e., a frequency component that is an integer multiple of the rotational speed of the motor 401, as the amplitude value of a specific frequency component. In this way, the RMS current calculator 720 can calculate the RMS value of the AC component synchronized with the fluctuations in the current flowing through the inverter 300 and the motor 401, which may be included in the capacitor current Ic.

[0072] Furthermore, due to interference between an AC component synchronized with the frequency of commercial power supply 110 and an AC component synchronized with fluctuations in the power consumption of compressor 400, a ripple current due to the sum and difference of the respective frequency components may be generated in capacitor current Ic. As a specific example of this case, consider a case where commercial power supply 110 has a frequency of 50 Hz and compressor 400 rotates at 30 revolutions per second. In this case, it is expected that capacitor current Ic will generate a ripple current of 130 Hz (= 100 Hz + 30 Hz) due to the sum of the double frequency component of commercial power supply 110 and the frequency of the rotation speed, and a ripple current of 70 Hz (= 100 Hz - 30 Hz) due to the difference between the double frequency component of commercial power supply 110 and the frequency of the rotation speed. Therefore, when calculating the amplitude value of a specific frequency component, current effective value calculation unit 720 desirably performs the calculation including these frequency components. In this way, the current effective value calculation unit 720 can calculate the effective value of the AC component synchronized with the sum and difference components of the double component of the frequency of the commercial power supply 110 and the frequency component of the rotation speed, which may be included in the capacitor current Ic. Note that although the sum and difference components of the double component of the frequency of the commercial power supply 110 and the frequency component of the rotation speed are given as examples here, frequency components that are combinations of these natural number multiple components may also be targeted.

[0073] In addition to the frequency components described above, the capacitor current Ic is expected to contain ripple currents of frequency components synchronized with the switching frequency of the converter 100 and the inverter 300. Therefore, when calculating the amplitude value of a specific frequency component, the effective current calculator 720 desirably includes these frequency components. Furthermore, if there is sufficient processing capacity, the calculation may further include frequency components that are combinations of natural number multiples of these frequencies. In this way, the effective current calculator 720 can calculate the effective values ​​of frequency components synchronized with the switching frequency of the converter 100 and the inverter 300, which may be included in the capacitor current Ic.

[0074] 9 is a block diagram showing an example of the configuration of the current effective value calculation unit 720 provided in the capacitor current estimator 700 according to embodiment 1. The current effective value calculation unit 720 is configured to include Fourier series expansion calculation units 721a to 721h and a frequency component summing unit 722.

[0075] The Fourier series expansion calculation units 721a to 721h input the estimated capacitor current Ic_est and perform Fourier series expansion to calculate the amplitude value of a specific frequency component contained in the estimated capacitor current Ic_est. The Fourier series expansion calculation units 721a and 721b in Fig. 6 calculate and output, as the specific frequency component, an amplitude value Ic_2fac of a component that is twice the frequency of the commercial power supply 110 and an amplitude value Ic_4fac of a component that is four times the frequency of the commercial power supply 110. Furthermore, the Fourier series expansion calculation units 721c and 721d calculate and output, as the specific frequency component, an amplitude value Ic_fω of a component that is once the rotational speed of the motor 401 and an amplitude value Ic_2fω of a component that is twice the rotational speed of the motor 401. Furthermore, the Fourier series expansion calculation units 721e and 721f calculate and output, as specific frequency components, an amplitude value Ic_2fac+fω of the frequency sum component and an amplitude value Ic_2fac-fω of the frequency difference component between the double component of the frequency of the commercial power supply 110 and the single component of the rotational speed of the motor 401. Furthermore, the Fourier series expansion calculation units 721g and 721h calculate and output, as specific frequency components, an amplitude value Ic_2fconv of the double component of the switching frequency of the converter 100 and an amplitude value Ic_2finv of the double component of the switching frequency of the inverter 300.

[0076] Hereinafter, a procedure will be described representatively in which the Fourier series expansion calculation unit 721a generates, as the specific frequency component, the amplitude value Ic_2fac of the component twice the frequency of the commercial power supply 110.

[0077] First, the Fourier series expansion calculation unit 721a multiplies the estimated capacitor current Ic_est by a sine wave component sin(2fac×t) and a cosine wave component cos(2fac×t), each having a frequency twice that of the commercial power supply 110, to detect the component twice the frequency of the commercial power supply 110. The amplitudes of the sine wave component and cosine wave component corresponding to twice the frequency of the commercial power supply 110 are doubled by twice the average values ​​of the sine wave term and cosine wave term obtained by the multiplication. The Fourier series expansion calculation unit 721a calculates the square root of the sum of the squares of the amplitudes of the sine wave component and cosine wave component corresponding to twice the frequency of the commercial power supply 110, thereby calculating the amplitude value Ic_2fac of the component twice the frequency of the commercial power supply 110, which is included in the estimated capacitor current Ic_est. This completes the Fourier series expansion calculation performed by the Fourier series expansion calculation unit 721a. The calculations of the Fourier series expansion calculation units 721b to 721h are the same as those of the Fourier series expansion calculation unit 721a except for the frequency to be detected. Furthermore, if the estimated capacitor current Ic_est is a periodic waveform, the output signals of the Fourier series expansion calculation units 721a to 721h will be approximately constant.

[0078] The frequency component summing unit 722 refers to each amplitude value calculated in the Fourier series expansion calculation units 721a to 721h and the frequency correction coefficient K(f) corresponding to each frequency, and substitutes these values ​​into the above equation (2) to calculate, thereby summing all frequency components and calculating the estimated capacitor current effective value Ic_est_rms.

[0079] 9, it is possible to extract only frequency components with a specific high degree of influence from among the frequency components contained in the capacitor current Ic, thereby enabling the estimated capacitor current effective value Ic_est_rms to be calculated with a small calculation load and high accuracy.

[0080] As described above, the power conversion apparatus according to the first embodiment includes a converter, an inverter, and a control device that drives and controls the converter and the inverter. The converter and the inverter each include at least one current detection unit for detecting the converter current flowing through the converter and the inverter current flowing through the inverter. The control device includes a capacitor current estimation unit that uses the detected values ​​of the converter current and the inverter current detected by the current detection unit, and estimates the capacitor current by switching the calculation process depending on the operating state of the converter, with the capacitor current flowing through the capacitor being the calculation target. The power conversion apparatus according to the first embodiment has the advantage of being able to estimate the capacitor current without installing a current detection unit in the electrical wiring to which the capacitor is connected. Furthermore, the power conversion apparatus according to the first embodiment estimates the capacitor current using the detected value detected by the current detection unit, which has the advantage of eliminating errors due to differential calculations and errors due to the capacitor capacitance value, compared to conventional techniques that estimate the capacitor current using a detected DC voltage value and a capacitor capacitance value. Furthermore, the power conversion apparatus according to the first embodiment has the advantage of being able to accurately estimate the capacitor current even when the capacitor capacitance value varies with individual capacitors or changes over time from the nominal value.

[0081] Furthermore, in the power conversion device according to the first embodiment, the capacitor current estimator provided in the control device performs frequency analysis on the estimated capacitor current to calculate the current amplitudes of a plurality of specific frequency components contained in the capacitor current. The specific frequency components are frequency components with a high degree of influence among the frequency components contained in the capacitor current. The capacitor current estimator normalizes each of the calculated current amplitudes of the plurality of specific frequency components in accordance with the frequency characteristics of the capacitor, and calculates the effective value of the estimated capacitor current using the normalized current amplitudes of the plurality of specific frequency components. According to the power conversion device configured in this manner, when calculating the estimated capacitor current, which is an estimate of the capacitor current, only the frequency components with a high degree of influence among the frequency components contained in the capacitor current are extracted to calculate the capacitor current, thereby achieving the effects of enabling highly real-time calculation processing with a small calculation load and obtaining a highly accurate estimated value of the capacitor current.

[0082] In the power conversion device according to the first embodiment, the plurality of specific frequencies analyzed by the capacitor current estimation unit may be configured to include at least one of a frequency N times the frequency of the AC voltage applied from the AC power supply, a frequency M times the rotational frequency of the mechanical device to be driven, a frequency sum that is the sum of the N-fold frequency and the M-fold frequency, and a frequency difference that is the difference between the N-fold frequency and the M-fold frequency, where N and M are any natural numbers. The components of the capacitor current corresponding to these specific frequencies are frequency components with a high degree of influence. Therefore, by extracting the components of the capacitor current corresponding to these specific frequencies, it is possible to obtain an accurate estimated value of the capacitor current.

[0083] Furthermore, in the power conversion device according to the first embodiment, the plurality of specific frequencies analyzed by the capacitor current estimator may be configured to include at least one of a frequency K times the converter switching frequency and a frequency L times the inverter switching frequency, where K and L are any natural numbers. The components of the capacitor current corresponding to these specific frequencies are frequency components with high influence. Therefore, by extracting the components of the capacitor current corresponding to these specific frequencies, it is possible to obtain an accurate estimated value of the capacitor current.

[0084] Embodiment 2 In Embodiment 2, a power conversion device 1a and a motor drive device 2a having a different configuration of the converter 100 from those in Embodiment 1 will be described. FIG. 10 is a diagram showing an example of the configuration of the power conversion device 1a and the motor drive device 2a according to Embodiment 2. Compared with the configuration shown in FIG. 1, in FIG. 10, the converter 100 is replaced with a converter 100a. Also, compared with the configuration shown in FIG. 1, in FIG. 10, the capacitor current estimator 700 provided in the control device 3 is replaced with a capacitor current estimator 700a. The motor drive device 2a according to Embodiment 2 is composed of the power conversion device 1a and a motor 401 provided in the compressor 400. The other configuration is the same or equivalent to that in FIG. 1, and the same or equivalent components are designated by the same reference numerals, and redundant description will be omitted.

[0085] The converter 100a is configured to include a diode rectifier 130, a reactor 120a installed on the output side of the diode rectifier 130, a switch element 150a, and a diode 140c. To detect the current and voltage in the converter 100a, a current detection unit 301a and voltage detection units 201a and 201b can be used as shown in FIG. 10. When open-loop control is applied, the current detection unit 301a and voltage detection units 201a and 201b shown in FIG. 10 do not need to be used. Note that although the switch element 150a is represented by the symbol IGBT in FIG. 10, a MOSFET may also be used, and the switch element 150a is not limited to an IGBT. The ON / OFF of the switch element 150a is controlled by a switching signal Q output from the converter control unit 500. ０ can be switched by

[0086] The basic function of the converter 100a of the second embodiment is the same as that of the converter 100 of the first embodiment. That is, the converter 100a of the second embodiment is connected between the commercial power supply 110 and the capacitor 200, and converts a first AC voltage applied from the commercial power supply 110 into a DC voltage and outputs it to the capacitor 200 and the inverter 300. However, unlike the first embodiment, in the converter 100a, the reactor 120a is installed on the DC side rather than the AC side. The converter 100a differs from the converter 100 of the first embodiment in that a short-circuit operation caused by an ON operation of the switch element 150a is also performed on the DC side.

[0087] The control device 3 receives as input the detected value of the converter current Iconv detected by the current detection unit 301a, the detected value of the inverter current Iinv detected by the current detection unit 301b, and the detected value of the DC bus voltage Vdc detected by the voltage detection unit 201b, and similarly to the contents described in embodiment 1, performs control calculations by the converter control unit 500, the inverter control unit 600, and the capacitor current estimating unit 700 to output switching signals for the converter 100 and the inverter 300, and also performs estimation calculations to estimate the capacitor current Ic.

[0088] The converter control unit 500 outputs a switching signal Q which is an ON / OFF command value for the switch element 150 of the converter 100.０ In the power conversion device 1a according to the second embodiment, the switching signal Q ０ The calculation of may be performed using either the simple switching method or the high-frequency switching method described in the first embodiment, or may be performed using a known switching method different from these methods.

[0089] FIG. 11 is a diagram illustrating the operation of a converter 100a included in a power conversion device 1a according to the second embodiment. The path of a current flowing due to power supplied from the commercial power supply 110 is switched as shown in FIG. 11 by ON / OFF control of a switch element 150a included in the converter 100a. Specifically, when the switch element 150a is controlled to be OFF, the input AC current Iac flowing from the commercial power supply 110 is rectified by the diode rectifier 130 as shown by the bold dashed line in the figure, and becomes a charging current flowing to the capacitor 200 via the reactor 120. When the switch element 150a is controlled to be ON, a short-circuit current flows via the path passing through the switch element 150a as shown by the bold solid line in the figure, and magnetic energy is stored in the reactor 120 by the rectified current via the diode rectifier 130. At this time, the capacitor 200 is not charged with power from the commercial power supply 110. When the switching element 150a is switched from ON to OFF by the switching control of the switching element 150a, the magnetic energy stored in the reactor 120 is simultaneously transferred to the capacitor 200 via the diode 140c. In the converter 100a equipped with the power factor correction circuit of the second embodiment, this series of operations makes it possible to widen the conduction angle of the input AC current Iac compared to when no switching control is performed, i.e., when the switching element 150a is always OFF, and therefore it is possible to improve the power factor to a certain extent. Note that, regardless of the ON / OFF control of the switching element 150a, a current due to the discharge of the capacitor 200 flows to the inverter 300.

[0090] Next, the operation of the capacitor current estimator 700 will be described. As in the first embodiment, the capacitor current Ic flowing through the capacitor 200 satisfies the relationship Ic = Iin - Iinv according to Kirchhoff's law. Therefore, by detecting the input current Iin in addition to the inverter current Iinv, it is possible to estimate the capacitor current Ic without installing a current detector in the electrical wiring to which the capacitor 200 is connected. Here, reference is made to the current path when the switch element 150a of the converter 100a is OFF and the current path when it is ON, as shown in FIG. 11 . When the switch element 150 of the converter 100a is ON, the input current Iin is zero. In this case, the capacitor current Ic flowing through the capacitor 200 is Ic = -Iinv. Furthermore, when the switch element 150 of the converter 100a is OFF, the input current Iin is equal to the converter current Iconv, so the capacitor current Ic flowing through the capacitor 200 is Ic = Iconv - Iinv.

[0091] As explained above, the capacitor current Ic corresponding to the ON / OFF of the switch element 150a of the converter 100a of the second embodiment is ultimately the same as the capacitor current Ic corresponding to the ON / OFF of the switch element 150 of the converter 100 of the first embodiment, and can be expressed by the above-mentioned formula (1). Therefore, the calculation process in the capacitor current estimator 700 may be the same as that of the first embodiment. Therefore, the same effects as those of the first embodiment can be obtained even when the power conversion device according to the second embodiment is used.

[0092] Converter 100a shown in FIG. 10 is an example of a converter having the same function as converter 100 shown in FIG. 1, and the control described in embodiment 1 can be applied to any converter having the same function, even if the converter has a configuration different from that of FIG. 10.

[0093] Third Embodiment Fig. 12 is a diagram showing an example of the configuration of a power conversion device 1b and a motor drive device 2b according to a third embodiment. Compared to the configuration shown in Fig. 1, in Fig. 12, the control device 3 is replaced with a control device 3a, and in the control device 3a, the capacitor current estimation unit 700 is replaced with a capacitor current estimation unit 700a. Also, compared to the configuration shown in Fig. 1, in Fig. 12, the switching signal Q output from the inverter control unit 600 to the inverter 300 is １ is also input to capacitor current estimation unit 700a. Motor drive device 2b according to the third embodiment is configured by power conversion device 1b and motor 401 provided in compressor 400. Other configurations are the same as or equivalent to those in Fig. 1, and the same or equivalent components are denoted by the same reference numerals, and redundant explanations will be omitted.

[0094] Consider the waveforms in the time chart shown in FIG. 7 . When the inverter 300 is not operating in overmodulation mode, i.e., when the phase voltage commands Vu*, Vv*, and Vw* of each phase are all equal to or less than Vdc / 2, the switching pattern of the inverter 300 during one carrier cycle of the carrier wave changes between four states: (a) and (h) switching patterns in which the inverter current Iinv is zero, and (b) through (g) switching patterns in which the inverter current Iinv matches the absolute value of one of the three-phase currents Iu, Iv, and Iw flowing through the motor 401. In this case, the inverter current Iinv has a pulse-like waveform corresponding to the ON / OFF switching operation of each phase, as shown in FIG. 7 . Therefore, the inverter current Iinv includes a ripple current consisting of a double component of the carrier frequency of the inverter 300 and its harmonic components. Furthermore, since the capacitor current Ic is expressed by the above formula (1), a ripple current similar to the ripple current of the inverter current Iinv flows in the capacitor current Ic.

[0095] To detect the influence of harmonic components of the ripple current, it is necessary to detect the inverter current Iinv at a cycle that is sufficiently fast relative to the carrier cycle of the inverter 300 while simultaneously performing calculations using equation (1). A guideline for a cycle that is sufficiently fast is, for example, 1 / 10 or less of the carrier cycle. To perform processing to speed up the detection cycle in this way, the control device 3a requires an A / D converter, a CPU, and other components capable of high-speed calculations, which leads to increased costs and weight for the control device 3a and the power conversion device 1b.

[0096] Another method uses an integrator circuit and an AD conversion port for average value calculation to detect the average value of the inverter current Iinv at certain cycles and then uses the detected average value to estimate the capacitor current Ic. This method does not require a faster detection cycle and can detect the influence of harmonic components due to the carrier frequency. However, this method also requires the installation of an additional integrator circuit, which increases costs and increases the weight of the device. Furthermore, current detection errors can occur due to variations in the constants of the resistors and capacitors that make up the integrator circuit, which is undesirable.

[0097] Therefore, the power conversion device 1b according to the third embodiment proposes the following method. First, the capacitor current estimation unit 700a estimates the inverter current Iinv detected at least twice per carrier cycle of the inverter 300 and the switching signal Q １ and calculates an average value Iinv_m of the inverter current Iinv for each carrier cycle of the inverter 300. Then, the capacitor current estimator 700a uses the calculated average value Iinv_m of the inverter current Iinv and the converter current Iconv to calculate an estimated capacitor current Ic_est for each carrier cycle of the inverter 300. This method makes it possible to calculate the average value Iinv_m of the inverter current Iinv and estimate the capacitor current Ic without performing current detection at high speed or requiring the provision of an additional circuit for current detection.

[0098] Next, a specific description will be given of the control calculation performed by the capacitor current estimator 700a according to the third embodiment. Fig. 13 is a block diagram showing an example of the configuration of the inverter control unit 600 and the capacitor current estimator 700a provided in the control device 3a according to the third embodiment.

[0099] The configuration of the inverter control unit 600 is the same as that shown in FIG. 4 in the first embodiment, and the calculation processing of each unit is also the same, so a description thereof will be omitted. The difference from the first embodiment is that the switching signal Q output from the inverter control unit 600 to the inverter 300 is １ is also input to the capacitor current estimation unit 700a.

[0100] The capacitor current estimator 700a includes an estimated current calculator 710a, an effective current value calculator 720, and an inverter current average value calculator 730. Compared to the configuration shown in Fig. 8, the capacitor current estimator 700a shown in Fig. 13 differs from the configuration shown in Fig. 8 in that the estimated current calculator 710 is replaced with an estimated current calculator 710a and an inverter current average value calculator 730 is added before the estimated current calculator 710a. Another difference is that, whereas the calculation cycle of the capacitor current estimator 700 in the first embodiment is not limited, the capacitor current estimator 700a in the third embodiment performs calculation processing synchronized with the carrier cycle of the inverter 300.

[0101] The inverter current average value calculation unit 730 calculates the inverter current Iinv detected at least twice per carrier cycle of the inverter 300 and the switching signal Q １ , and calculates an average value Iinv_m of the inverter current Iinv for each carrier period of the inverter 300.

[0102] The operation of the inverter current average value calculation unit 730 will be further described with reference to several drawings. Fig. 14 is a time chart illustrating the operation of the inverter 300, inverter control unit 600, and capacitor current estimator 700a according to the third embodiment. Fig. 14 shows the phase voltage commands Vu*, Vv*, and Vw* applied to each phase of the inverter 300, the switching signals applied to the upper arm switch elements Sup, Svp, and Swp of the inverter 300, the inverter current Iinv, and the timing of detecting the inverter current Iinv. In Fig. 14, the parts related to the operation of the inverter 300 and inverter control unit 600 are the same as those in Fig. 7.

[0103] In the first embodiment, with reference to Fig. 6 , it has been shown that, depending on the switching state of the inverter 300, the inverter current Iinv may be 0 or may match the absolute value of any one of the three-phase currents Iu, Iv, and Iw of the motor 401. Also, in the first embodiment, with reference to Fig. 7 , it has been shown that the inverter current Iinv matches the absolute value of any one of the three-phase currents Iu, Iv, and Iw of the motor 401 around the time when the switching state of the upper arm switch element of the intermediate phase, which is the middle one when the phase voltage commands Vu*, Vv*, and Vw* of each phase are arranged in order of magnitude, is changed. This also applies to the operating state in Fig. 14 . Therefore, as an operation of the inverter current average value calculation unit 730, to calculate the average value Iinv_m of the inverter current Iinv for each carrier period of the inverter 300, in FIG. 14 , the average value Iinv_m can be calculated using the following equation (3) using the switching state durations Ti, Tj before and after the switching state of the upper arm switch element of the middle phase is switched, the detection values ​​Iinv1, Iinv2 of the inverter current Iinv detected during each period, and one carrier period Tc of the carrier wave.

[0104]

[0105] In this paper, of the switching state durations Ti and Tj, the switching state duration Ti may be referred to as the "first switching state duration," and the switching state duration Tj may be referred to as the "second switching state duration." When the carrier frequency of the inverter 300 is not changed, one carrier period Tc is a fixed value. Therefore, the elements that vary in equation (3) are the switching state durations Ti and Tj and the detected values ​​Iinv1 and Iinv2 of the inverter current Iinv. Therefore, it can be said that the average value Iinv_m of the inverter current Iinv can be calculated based on the product of the switching state duration Ti and the detected value Iinv1 of the inverter current Iinv, and the product of the switching state duration Tj and the detected value Iinv2 of the inverter current Iinv.

[0106] 14, the switching state durations Ti and Tj can be calculated using the ON times Tup, Tvp, and Twp of the upper arm switch elements of each phase during one carrier period. １ In Fig. 14, since the V phase is the intermediate phase, Ti = (Tup - Tvp) / 2 and Tj = (Tvp - Twp) / 2. In the case of other combinations where the intermediate phase is the U phase or W phase, calculations can be made in the same way.

[0107] Furthermore, the detected values ​​Iinv1 and Iinv2 of the inverter current Iinv detected before and after the upper arm switch element of the middle phase used in equation (3) is switched may be the same as the detected values ​​used by the current restoration unit 603 of the inverter control unit 600 to restore the three-phase currents Iu, Iv, and Iw of the motor 401. Therefore, the capacitor current estimator 700a does not need to perform additional current detection at different timings to estimate the capacitor current Ic, and it is sufficient to commonly use the detected values ​​detected to restore the three-phase currents Iu, Iv, and Iw. This makes it possible to calculate the average value Iinv_m of the inverter current Iinv for each carrier period of the inverter 300 and calculate the estimated capacitor current Ic_est without increasing the calculation load of A / D conversion.

[0108] Alternatively, to improve the accuracy of current detection, the inverter current Iinv may be detected multiple times before and after the upper arm switch element of the intermediate phase switches, and the average values ​​Iinv1 and Iinv2 may be used to calculate the average value Iinv_m of the inverter current Iinv. For simplicity of explanation, FIG. 14 shows the inverter current Iinv as constant during each of the switching state durations Ti and Tj. However, in reality, the current changes during each of the switching state durations Ti and Tj in response to changes in the phase current of the motor 401. On the other hand, by detecting the current multiple times before and after switching, the current changes during the switching state durations Ti and Tj can be taken into account, thereby improving the accuracy of current detection. Because multiple current detections increase the computational load of A / D conversion, there is a trade-off between improved current detection accuracy and increased computational load. Whether to perform multiple current detections can be determined based on which is prioritized.

[0109] 14 shows an example in which current detection is performed during a carrier down cycle, which is a period in which the carrier wave monotonically decreases, to calculate the average value Iinv_m of the inverter current Iinv, but this example is not limiting. Current detection may also be performed during a carrier up cycle, which is a period in which the carrier wave monotonically increases, to calculate the average value Iinv_m of the inverter current Iinv. Alternatively, current detection may be performed during both the carrier down cycle and the carrier up cycle, and the average value of the current during the period before and after switching in the carrier down cycle and the average value of the current during the period before and after switching in the carrier up cycle may be further averaged to calculate the average value Iinv_m of the inverter current Iinv.

[0110] As explained above, in the method of the third embodiment, the inverter current average value calculation unit 730 calculates the inverter current average value Iinv detected at least twice per carrier cycle of the inverter 300 and the switching signal Q １ is referred to, and an average value Iinv_m of the inverter current Iinv is calculated for each carrier period.

[0111] The estimated current calculation unit 710a calculates the detected value of the converter current Iconv detected in synchronization with the carrier cycle of the inverter 300, the average value Iinv_m of the inverter current Iinv calculated by the inverter current average value calculation unit 730, and the switching signal Q ０ The estimated capacitor current Ic_est is calculated by referring to the switching signal Q of the converter 100. Note that the calculation method of the estimated capacitor current Ic_est is different from that of the first embodiment in that the inverter current Iinv in the equation (1) shown in the first embodiment is replaced with the average value Iinv_m of the inverter current Iinv. ０ The same applies to the point that the calculation formula is switched depending on the

[0112] The current effective value calculation unit 720 receives the estimated capacitor current Ic_est as input and calculates the estimated capacitor current effective value Ic_est_rms for each carrier period of the inverter 300. The calculation content of the current effective value calculation unit 720 is similar to the calculation by the current effective value calculation unit 720 described in the first embodiment.

[0113] As described above, in the power conversion device according to the third embodiment, the inverter includes a plurality of switch elements whose switching is controlled by phase voltage commands generated by the control device and assigned to each phase of the inverter. When a phase having an intermediate relationship between the magnitudes of the phase voltage commands for the plurality of switch elements is defined as an intermediate phase, the capacitor current estimator included in the control device calculates an average value of the inverter current for each switching period of the inverter based on the product of the first switching state duration before the switching state of the switch element of the intermediate phase is switched and the detected value of the inverter current detected during the first switching state duration, and the product of the second switching state duration after the switching state of the switch element of the intermediate phase is switched and the detected value of the inverter current detected during the second switching state duration. The power conversion device according to the third embodiment can achieve the same effects as the power conversion device according to the first embodiment. Furthermore, the technique of the third embodiment can achieve accurate current detection without providing a new detector or additional circuit for current detection.

[0114] Fourth Embodiment Fig. 15 is a diagram showing an example of the configuration of a power conversion device 1c and a motor drive device 2c according to a fourth embodiment. Compared to the configuration shown in Fig. 1, in Fig. 15, the control device 3 is replaced with a control device 3b, and an alarm signal generating unit 750 is added to the control device 3b. The motor drive device 2c according to the fourth embodiment is made up of the power conversion device 1c and a motor 401 provided in a compressor 400. The other configuration is the same as or equivalent to that in Fig. 1, and the same or equivalent components are designated by the same reference numerals, and redundant description will be omitted.

[0115] The alarm signal generating unit 750 outputs an alarm signal ALM based on the estimated capacitor current effective value Ic_est_rms when a specific condition occurs, such as when the estimated capacitor current effective value Ic_est_rms becomes greater than a capacitor current limit value Ic_lim stored in advance in the control device 3b.

[0116] Even in a typical power conversion device, an allowable current is set for each capacitor. If a capacitor is operated for a long period of time while a current greater than the allowable current continues to flow through it, the capacitor may fail or its lifespan may be rapidly shortened. Therefore, in the power conversion device 1c according to the fourth embodiment, a capacitor current limit value Ic_lim, which is a preset limit value for the capacitor current, is stored in the control device 3b. The control device 3b constantly monitors the estimated capacitor current effective value Ic_est_rms calculated during operation of the power conversion device 1c. When the estimated capacitor current effective value Ic_est_rms exceeds the capacitor current limit value Ic_lim, the alarm signal generating unit 750 of the control device 3b outputs an alarm signal ALM indicating that the estimated capacitor current effective value Ic_est_rms has exceeded the capacitor current limit value Ic_lim.

[0117] The alarm signal ALM allows the user of the device or a higher-level control device (not shown) to know that there is a possibility of a failure or a deterioration in the life of the capacitor 200. If the alarm signal ALM is not released for a certain period of time, the power conversion device 1c outputs a switching signal Q１ In this way, various measures such as warnings and protection can be taken before the device breaks down.

[0118] As described above, the control device provided in the power conversion device according to embodiment 4 includes an alarm signal generating unit that outputs an alarm signal when the estimated value of the capacitor current exceeds a preset limit value of the capacitor current. The power conversion device according to embodiment 4 has the advantage that it is possible to monitor the capacitor current during operation of the power conversion device, to determine whether there is a possibility of a failure or deterioration of the capacitor's life, and, if there is a possibility of a failure or deterioration of the capacitor's life, to appropriately notify the user of the device, a higher-level control device, etc.

[0119] Fifth Embodiment Fig. 16 is a diagram showing an example configuration of a power conversion device 1d and a motor drive device 2d according to a fifth embodiment. Compared to the configuration shown in Fig. 15, in Fig. 16, the control device 3b is replaced with a control device 3c, and a speed command drooping unit 760 is added to the control device 3c. The motor drive device 2d according to the fifth embodiment is composed of the power conversion device 1d and a motor 401 provided in a compressor 400. The other configuration is the same as or equivalent to that in Fig. 15, and the same or equivalent components are designated by the same reference numerals, and redundant description will be omitted.

[0120] Although not limited to the power conversion device 1d according to the fifth embodiment, the greater the power consumption of the inverter 300 and the motor 401, which are located on the load side as viewed from the capacitor 200, the greater the capacitor current Ic flowing through the capacitor 200. Therefore, when the estimated capacitor current effective value Ic_est_rms exceeds the capacitor current limit value Ic_lim and an alarm signal ALM is output, the power consumption on the load side can be reduced and the capacitor current Ic can also be reduced by reducing the rotational speed of the motor 401.

[0121] 16 , when speed command drooping unit 760 receives alarm signal ALM, it generates signal Δω for reducing the rotation speed of motor 401. Speed ​​command drooping unit 760 generates corrected speed command ω** (=ω*−Δω) by subtracting signal Δω from speed command ω*, and outputs the generated corrected speed command ω** to inverter control unit 600.

[0122] The inverter control unit 600 performs the calculations shown in the first embodiment based on the corrected speed command ω** instead of the speed command ω*, and controls the speed so that the rotation speed of the motor 401 coincides with the corrected speed command ω**.

[0123] Although the output power to the motor 401 is reduced by the operation of the speed command drooping unit 760, the operation of the power conversion device 1d can be continued while preventing breakdown or deterioration of the capacitor 200.

[0124] As described above, the control device provided in the power conversion device according to the fifth embodiment includes a speed command drooping unit that receives an alarm signal output from the alarm signal generating unit and reduces the rotation speed command of the motor. According to the power conversion device according to the fifth embodiment, when an alarm signal indicating the possibility of a capacitor failure or life degradation is received, the motor rotation speed command is reduced to reduce the output power to the motor, thereby achieving the effect of enabling the operation of the power conversion device to continue while preventing the capacitor from failing or life degradation.

[0125] Sixth Embodiment. Fig. 17 is a diagram showing a configuration example of a refrigeration cycle-applied apparatus 900 according to a sixth embodiment. The refrigeration cycle-applied apparatus 900 according to the sixth embodiment includes the power conversion device 1 described in the first embodiment. Note that the power conversion device 1 may be replaced with any of the power conversion devices 1a to 1d described in the second to fifth embodiments. The refrigeration cycle-applied apparatus 900 according to the sixth embodiment can be applied to products equipped with a refrigeration cycle, such as air conditioners, refrigerators, freezers, and heat pump water heaters. Note that in Fig. 17, components having the same functions as those in the first embodiment are denoted by the same reference numerals as those in the first embodiment.

[0126] The refrigeration cycle application equipment 900 includes a compressor 400 incorporating the motor 401 in embodiment 1, a four-way valve 902, an indoor heat exchanger 906, an expansion valve 908, and an outdoor heat exchanger 910, which are attached via refrigerant piping 912.

[0127] Inside the compressor 400, a compression mechanism 904 that compresses the refrigerant and a motor 401 that operates the compression mechanism 904 are provided.

[0128] The refrigeration cycle device 900 can perform heating or cooling operation by switching the four-way valve 902. The compression mechanism 904 is driven by a motor 401 that is variably controlled in speed.

[0129] During heating operation, as shown by the solid arrow, the refrigerant is pressurized by the compression mechanism 904 and sent out, passes through the four-way valve 902, the indoor heat exchanger 906, the expansion valve 908, the outdoor heat exchanger 910 and the four-way valve 902 and returns to the compression mechanism 904.

[0130] During cooling operation, as shown by the dashed arrow, the refrigerant is pressurized by the compression mechanism 904 and sent out, and returns to the compression mechanism 904 through the four-way valve 902, the outdoor heat exchanger 910, the expansion valve 908, the indoor heat exchanger 906 and the four-way valve 902.

[0131] During heating operation, the indoor heat exchanger 906 acts as a condenser to release heat, and the outdoor heat exchanger 910 acts as an evaporator to absorb heat. During cooling operation, the outdoor heat exchanger 910 acts as a condenser to release heat, and the indoor heat exchanger 906 acts as an evaporator to absorb heat. The expansion valve 908 reduces the pressure of the refrigerant to expand it.

[0132] The power conversion device 1 can estimate the capacitor current Ic flowing through the capacitor 200 using the current detection units 301a, 301b installed in the converter 100 and the inverter 300, respectively. Therefore, the capacitor current Ic can be estimated without installing an additional current detection unit in the electrical wiring connecting the capacitor 200. As a result, by monitoring the capacitor current Ic during operation of the refrigeration cycle applied device 900, it is possible to take various measures, such as appropriate warnings and protection, before the device breaks down or the capacitor reaches the end of its life due to an overcurrent in the capacitor 200.

[0133] The configurations shown in the above embodiments are merely examples, and may be combined with other known technologies, or different embodiments may be combined with each other. It is also possible to omit or modify parts of the configurations as long as they do not deviate from the gist of the invention.

[0134] 1, 1a to 1d power conversion device, 2, 2a to 2d motor drive device, 3, 3a to 3c control device, 100, 100a converter, 110 commercial power supply, 120, 120a reactor, 130 diode rectifier, 140a to 140c diode, 150, 150a, Sup, Sun, Svp, Svn, Swp, Swn switch element, 160 simple switching cell, 200 capacitor, 201a, 201b voltage detection unit, 300 inverter, 301a, 301b current detection unit, 400 compressor, 401 motor, 402 mechanical load, 500 converter control unit, 600 inverter control unit, 601 speed control unit, 602 d-axis current command determination unit, 603 current restoration unit, 604 current control unit, 605 Position estimation unit, 606 PWM signal generator, 700, 700a capacitor current estimation unit, 710, 710a estimated current calculation unit, 720 current effective value calculation unit, 721a to 721h Fourier series expansion calculation unit, 722 frequency component summation unit, 730 inverter current average value calculation unit, 750 alarm signal generation unit, 760 speed command drooping unit, 900 refrigeration cycle applied equipment, 902 four-way valve, 904 compression mechanism, 906 indoor heat exchanger, 908 expansion valve, 910 outdoor heat exchanger, 912 refrigerant piping.

Claims

1. A power conversion device comprising: a converter having at least one switch element and converting a first AC voltage applied from an AC power source into a DC voltage; a capacitor that smooths the voltage output by said converter; an inverter that converts the DC voltage smoothed by said capacitor into a second AC voltage and applies said second AC voltage to a motor to drive it in rotation; and a control device that drives and controls said converter and said inverter, respectively, wherein said converter and said inverter each have at least one current detection unit for detecting a converter current flowing in said converter and an inverter current flowing in said inverter, and said control device comprises a capacitor current estimation unit that uses the detected values ​​of the converter current and the inverter current detected by the current detection unit, calculates the capacitor current flowing in the capacitor, switches calculation processing depending on the operating state of the converter, and performs an estimation calculation of the capacitor current.

2. The power conversion device according to claim 1, wherein the inverter comprises a plurality of switch elements whose switching is controlled by phase voltage commands generated by the control device and given to each phase of the inverter, and the capacitor current estimating unit, when an intermediate phase is defined as a phase in which the magnitude relationship of the phase voltage commands for the plurality of switch elements is intermediate, calculates an average value of the inverter current for each switching period of the inverter based on the product of a first switching state duration before the switching state of the switch element of the intermediate phase is switched and a detected value of the inverter current detected during the first switching state duration, and the product of a second switching state duration after the switching state of the switch element of the intermediate phase is switched and a detected value of the inverter current detected during the second switching state duration.

3. The power conversion device according to claim 1 or 2, characterized in that the capacitor current estimation unit calculates the current amplitudes of a plurality of specific frequency components contained in the capacitor current by frequency analysis of the estimated capacitor current, normalizes each of the calculated current amplitudes of the plurality of specific frequency components in accordance with the frequency characteristics of the capacitor, and calculates the effective value of the estimated capacitor current using the normalized current amplitudes of the plurality of specific frequency components.

4. The power conversion device according to claim 3, wherein the plurality of specific frequencies analyzed in the capacitor current estimation unit include at least one of a frequency N times the frequency of the AC voltage applied from the AC power supply, a frequency M times the rotational frequency of the mechanical device to be driven, a frequency sum that is the sum of the N-times frequency and the M-times frequency, and a frequency difference that is the difference between the N-times frequency and the M-times frequency, where N and M are any natural numbers.

5. The power conversion device according to claim 3 or 4, characterized in that the plurality of specific frequencies analyzed in the capacitor current estimation unit include at least one of a frequency that is K times the switching frequency of the converter and a frequency that is L times the switching frequency of the inverter, where K and L are any natural numbers.

6. A power conversion device according to any one of claims 1 to 5, characterized in that the control device is provided with an alarm signal generating unit that outputs an alarm signal when the estimated value of the capacitor current exceeds a preset limit value of the capacitor current.

7. The power conversion device according to claim 6, wherein the control device comprises a speed command drooping section that receives the alarm signal and reduces the rotation speed command of the motor.

8. A motor drive device comprising the power conversion device according to any one of claims 1 to 7.

9. A refrigeration cycle device comprising the power conversion device according to any one of claims 1 to 7.

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