Resonant converter and parameter calculation method therefor

Abstract

A resonant converter and a parameter calculation method therefor. The parameter calculation method for a resonant converter comprises the following steps: (A) setting a preset operating condition of a resonant converter, the preset operating condition comprising a specific frequency range of an operating frequency of the resonant converter; (B) calculating a plurality of sets of first resonance parameters within the specific frequency range; (C) obtaining, from the first resonance parameters, a first resonance parameter at which the current ripple of a resonant inductor of the resonant converter is the minimum value; and (D) on the basis of the turns ratio of the first resonance parameter and the preset operating condition, calculating a plurality of sets of second resonance parameters, and selecting one of the second resonance parameters as a resonance parameter.

Description

Resonant converter and its parameter calculation method Technical Field

[0001] This invention relates to a method for calculating the parameters of a resonant converter, and more particularly to a method for calculating the resonant parameters of a resonant converter. Background Technology

[0002] Dual Active Bridge (DAB) converters are typically used in high-power power conversion applications and are suitable as battery charging converters due to their wide adjustable voltage range. Traditional DAB converters operate in a fixed-frequency mode, resulting in relatively simple circuit designs. However, while DAB converters can perform zero-voltage switching (ZVS) when the pulse width control switch is on, the switch-off operation is a hard switch, which leads to significant current stress and higher switching losses during turn-off.

[0003] On the other hand, the circuit design of dual active bridge converters typically uses existing formula derivations for parameter design. However, this parameter design method cannot optimize the obtained parameters, resulting in excessive current ripple and consequently, an overly large resonant circuit. Furthermore, the unoptimized parameter design prevents the dual active bridge converter from achieving a good zero-voltage switching (ZVS) range, thus hindering effective efficiency improvement.

[0004] Therefore, how to design a resonant converter and its parameter calculation method, and how to obtain better resonant parameters through optimized processes and precise analysis, is a major research topic that the creators of this project intend to undertake. Summary of the Invention

[0005] To address the aforementioned problems, this disclosure provides a method for calculating the parameters of a resonant converter, overcoming the limitations of existing technologies. Therefore, the method for calculating the parameters of a resonant converter disclosed herein is used to obtain the resonant parameters of the resonant converter. The method for calculating the resonant parameters of a resonant converter includes the following steps: (A) setting preset operating conditions for the resonant converter, wherein the preset operating conditions include a specific frequency range of the operating frequency of the resonant converter. (B) calculating multiple sets of first resonant parameters within the specific frequency range, wherein the first resonant parameters include the turns ratio of the transformer of the resonant converter. (C) among the first resonant parameters, obtaining the first resonant parameter where the current ripple of the resonant inductor of the resonant converter is minimized. (D) calculating multiple sets of second resonant parameters based on the turns ratio of the first resonant parameters and the preset operating conditions, and selecting one of the second resonant parameters as the resonant parameter.

[0006] To address the aforementioned problems, this disclosure provides a resonant converter to overcome the limitations of the prior art. Therefore, the resonant converter of this disclosure includes a primary-side circuit, a resonant circuit, and a secondary-side circuit. The primary-side circuit is coupled to a DC power supply, and the resonant circuit includes multiple transformers. The primary winding of the transformer is coupled to the primary-side circuit. The secondary-side circuit includes multiple secondary-side switching circuits, each comprising a first secondary-side bridge arm and a second secondary-side bridge arm connected in parallel. The first and second secondary-side bridge arms of the secondary-side switching circuits are respectively coupled to the two ends of the secondary winding of the transformer, and the secondary-side switching circuits are connected in parallel to the device.

[0007] The main purpose and effect of this disclosure is to provide an optimization process for resonant converters, thereby obtaining better resonant parameters through precise analysis to achieve low current ripple and improve the efficiency of the resonant converter.

[0008] To gain a deeper understanding of the techniques, means, and effects employed by this invention to achieve its intended purpose, please refer to the following detailed description and accompanying drawings. It is believed that the purpose, features, and characteristics of this invention can be understood in a thorough and specific manner from these drawings. However, the accompanying drawings are provided for reference and illustration only and are not intended to limit this invention. Attached Figure Description

[0009] Figure 1A is a circuit block diagram of the first embodiment of the resonant converter of this disclosure;

[0010] Figure 1B is a circuit block diagram of the second embodiment of the resonant converter of this disclosure;

[0011] Figure 1C is a circuit block diagram of the third embodiment of the resonant converter of this disclosure;

[0012] Figure 1D is a circuit block diagram of the fourth embodiment of the resonant converter of this disclosure;

[0013] Figure 1E is a circuit block diagram of the fifth embodiment of the resonant converter of this disclosure;

[0014] Figure 2A is a circuit block diagram of the first embodiment of the resonant conversion module of this disclosure;

[0015] Figure 2B is a simplified equivalent circuit diagram of the resonant conversion module according to the first embodiment of this disclosure;

[0016] Figure 2C shows the power and voltage curves of the resonant conversion module of this disclosure for battery charging and discharging;

[0017] Figure 3A shows the waveform of the resonant conversion module of this disclosure operating in three-phase shift mode;

[0018] Figure 3B is a timing diagram of the first waveform of the resonant conversion module of this disclosure operating in three-phase shift mode;

[0019] Figure 3C is a second waveform timing diagram of the resonant conversion module of this disclosure operating in three-phase shift mode;

[0020] Figure 3D is the third waveform timing diagram of the resonant conversion module of this disclosure operating in three-phase shift mode;

[0021] Figure 3E is a waveform diagram of the resonant conversion module of this disclosure operating in single-phase shift mode;

[0022] Figure 3F is a waveform diagram of the resonant conversion module of this disclosure operating in dual phase-shift mode;

[0023] Figure 4A is a flowchart of the parameter calculation method for obtaining resonance parameters in the resonance conversion module of this disclosure;

[0024] Figure 4B is a schematic diagram of the numerical distribution of the second resonance parameter of this disclosure;

[0025] Figure 5A is a first simulated waveform diagram of the second resonant parameter of a resonant converter using an embodiment of the present disclosure.

[0026] Figure 5B is a second simulated waveform diagram of the second resonant parameter of a resonant converter using an embodiment of the present disclosure;

[0027] Figure 5C is a third analog waveform diagram of the second resonant parameter of the resonant converter of this disclosure using an embodiment;

[0028] Figure 6A is a first analog waveform diagram of the resonant converter of this disclosure using the second resonant parameter of another embodiment;

[0029] Figure 6B is a second simulated waveform diagram of the resonant converter of this disclosure using the second resonant parameter of another embodiment;

[0030] Figure 6C is a third simulated waveform diagram of the resonant converter of this disclosure using the second resonant parameter of another embodiment;

[0031] Figure 7A is a system control block diagram of the resonant converter of this disclosure;

[0032] Figure 7B is a block diagram of the operation frequency lookup table control of this disclosure;

[0033] Figure 7C is a block diagram of the numerical / time conversion of this disclosure;

[0034] Figure 7D is a control block diagram of the dead time lookup table of this disclosure;

[0035] Figure 8 is a flowchart of the closed-loop control of the resonant converter of this disclosure;

[0036] Figure 9A shows the distribution of the zero-voltage switching range obtained by using the simulation program to execute the resonant converter and substituting the first implementation parameters;

[0037] Figure 9B shows the zero-voltage switching range distribution obtained by using the simulation program to execute the resonant converter and substituting the second implementation parameters;

[0038] Figure 9C shows the zero-voltage switching range distribution obtained by using the simulation program to execute the resonant converter and substituting the third implementation parameters;

[0039] Figure 10A is a circuit block diagram of a second embodiment of the resonant conversion module of this disclosure;

[0040] Figure 10B is a simplified equivalent circuit diagram of the resonant conversion module according to the second embodiment of this disclosure;

[0041] Figure 11A is a control block diagram of the closed-loop system for battery charging by the resonant conversion mode according to the second embodiment of this disclosure; and

[0042] Figure 11B is a control block diagram of the closed-loop system for battery discharge by the resonant conversion mode according to the second embodiment of this disclosure.

[0043] Figure Label Explanation: 100: Resonant Converter; A: Primary Side Circuit; 1, 1A, 1B: Resonant Converter Modules; C1, Co: Capacitors; 4: Switching Circuit; 42: First Switching Arm; Q1: First Switch; Q2: Second Switch; 44: Second Switching Arm; Q3: Third Switch; Q4: Fourth Switch; 46: Third Switching Arm; Q5: Third Switch; Q6: Fourth Switch; M: Common Core Module; Pgnd: Primary Side Ground Terminal; B: Resonant Circuit; Lr1, Lr2: Resonant Inductors; Cr1, Cr2: Resonant Capacitors; T Transformer Wp: Primary winding Wp1: First primary winding Wp2: Second primary winding Pc: Center tap Ws: Secondary winding C: Secondary circuit 5: Secondary switching circuit 52: First secondary bridge arm Qs1: First secondary switch Qs2: Second secondary switch 54: Second secondary bridge arm Qs3: Third secondary switch Qs4: Fourth secondary switch Sgnd: Secondary ground terminal Pp1, Pp2, Pp3, Ps1, Ps2: Node 6: Controller 200: Device Vdc: DC Voltage Vo, V1, V2, V3, Vpp, Vss, Eb: Voltage PWM: Pulse Width Modulation Signal Le, Lb: Equivalent Inductance Ce, Ce1, Ce2: Equivalent Capacitance Rb: Equivalent Impedance Vl: Inductor Voltage Vc: Capacitor Voltage Vdc*: Voltage Command Ip(avg), Is(avg), Ib: Current Ib*: Current Command Il, I0, I1, -I0, -I1: Inductor Current Q: Current Ripple CC: Constant Current Charging Mode CP: Constant Power Charging Mode CV: Constant Voltage Charging Mode ψ, D1, D2: Phase shift 1 / R*: Compensation n, N: Turns ratio Lv, Lv1, Lv2: Inductance value Cv, Cv1, Cv2: Capacitance value Fs: Operating frequency Cf: Frequency command Fv: Frequency value T: Period DTp, DTs: Dead time value Cpwm: Pulse width modulation command Cdt: Dead time command Po, Pmax, Pmin: Power S100~S680: Step BI: Bandwidth interval 7, 7A, 7B: Parameter reference table PF: Filter block 8: ψ conversion block 9: Dead time control block Detailed Implementation

[0044] The technical content and detailed description of the present invention are explained below with reference to the accompanying drawings:

[0045] The converters disclosed herein, and their parameter calculation methods, are all described using a dual active bridge (DAB) converter (hereinafter referred to as a DAB converter) or a series resonant dual active bridge (SR DAB) converter (hereinafter referred to as an SR DAB converter) circuit structure. However, while the technical content of this disclosure is more applicable to DAB converters and SR DAB converters, it is not inapplicable to resonant converters such as LLC and LC. Therefore, unless a specific converter is specified herein, the resonant converter 100 described later in this disclosure may represent a DAB converter, an SR DAB converter, and resonant converters such as LLC and LC.

[0046] Please refer to Figure 1A, which is a circuit block diagram of the first embodiment of the resonant converter of this disclosure. The resonant converter 100 can be mainly composed of one or more resonant conversion modules (1A, 1B), and the resonant conversion modules (1A, 1B) include a primary side circuit A, a resonant circuit B, and a secondary side circuit C. The primary side circuit A is coupled to a DC voltage Vdc, and the resonant circuit B includes multiple sets of resonant slots (here illustrated by a single resonant inductor Lr1) and a transformer T. Each transformer T includes a primary side winding Wp and a secondary side winding Ws, and the primary side winding Wp is coupled to the primary side circuit A. The secondary side circuit C includes multiple capacitors Co and multiple sets of secondary side switching circuits 5, and the secondary side switching circuits 5 respectively include a first secondary side bridge arm 52 and a second secondary side bridge arm 54 connected in parallel with capacitors Co. The primary side bridge arm 52 and the secondary side bridge arm 54 of the secondary side switching circuit 5 are respectively coupled to the two ends of the secondary side winding Ws, and the secondary side switch 5 and the capacitor Co circuit are connected in parallel and coupled to a device 200. Because DAB converters and SR DAB converters are characterized by their ability to perform large-power conversion and the ability to adjust the output voltage Vo over a wide range, they are particularly suitable for systems used for charging and discharging electric vehicles. Therefore, device 200 is preferably used for batteries for electric vehicles, but is not limited thereto.

[0047] Furthermore, in Figure 1A, the primary-side circuit A includes a capacitor C1 and multiple switching circuits 4, with capacitor C1 and switching circuits 4 connected in parallel and coupled to a DC voltage Vdc. Capacitor C1 is primarily used for voltage regulation or filtering; therefore, it can be replaced with any device capable of voltage regulation and filtering (e.g., but not limited to, a filter). Switching circuits 4 include a first switching bridge arm 42 and a second switching bridge arm 44, with the first switching bridge arm 42 connected in parallel with the second switching bridge arm 44. The first switching bridge arm 42 may include a first switching switch Q1 and a second switching switch Q2 connected in series, and the second switching bridge arm 44 may include a third switching switch Q3 and a fourth switching switch Q4 connected in series. Node Pp1 between the first switching switch Q1 and the second switching switch Q2 is coupled to one end of the primary-side winding Wp, and node Pp2 between the third switching switch Q3 and the fourth switching switch Q4 is coupled to the other end of the primary-side winding Wp. The resonant slot (here illustrated by a single resonant inductor Lr1) can be coupled between one end of the primary winding Wp and the first switching bridge arm 42, or between the other end of the primary winding Wp and the second switching bridge arm 44.

[0048] Similarly, the first-stage side bridge arm 52 includes a first-stage side switch Qs1 and a second-stage side switch Qs2, and the second-stage side bridge arm 54 includes a third-stage side switch Qs3 and a fourth-stage side switch Qs4. Node Ps1 between the first-stage side switch Qs1 and the second-stage side switch Qs2 is coupled to one end of the secondary-side winding Ws, and node Ps2 between the third-stage side switch Qs3 and the fourth-stage side switch Qs4 is coupled to the other end of the secondary-side winding Ws. Figure 1A only shows the architecture of the DAB converter, therefore the resonant slot only includes the resonant inductor Lr1. However, it is not limited to a single resonant inductor Lr1, which can be adjusted according to the type of converter, and will not be elaborated here. Furthermore, in one embodiment, the turns ratio of the primary-side winding Wp to the secondary-side winding Ws is indicated as n:1, but it is not limited to this and can be any ratio implementable in the resonant converter 100.

[0049] Furthermore, the controller 6 provides a pulse width modulation (PWM) signal to control the switching switches Q1-Q4 and the secondary-side switches Qs1-Qs4 to convert the DC voltage Vdc into voltage Vo. Since the DAB converter and the SR DAB converter have bidirectional conversion capabilities, the controller 6 can also control the resonant converter 100 to convert voltage Vo into DC voltage Vdc by providing a PWM signal. Because the resonant conversion modules (1A, 1B) in Figure 1A are connected in parallel, the resonant converter 100 can perform high-power conversion. In one embodiment, the controller 6 can control the resonant converter 100 to operate in single-phase shift (SPS) mode, dual-phase shift (DPS) mode, and triple-phase shift (TPS) mode. That is, the controller 6 can control the switching switches Q1-Q4 or the secondary-side switches Qs1-Qs4 to perform phase shifts, or even control both the switching switches Q1-Q4 and the secondary-side switches Qs1-Qs4 to perform phase shifts. In addition, controller 6 can also have frequency conversion function (i.e., adjustment period T), which will be described in more detail later, and will not be repeated here.

[0050] On the other hand, in one embodiment, the controller 6 can be a digital signal processor (DSP), but is not limited thereto; any type of controller capable of controlling the resonant converter 100 should be included in the scope of this embodiment. Furthermore, the resonant conversion modules (1A, 1B) of FIG1A can also be controlled using interleaved control technology to reduce the current stress on the input filter of the switching power supply (SPS) modulation. This is because interleaved control has the effect of ripple current cancellation, which can reduce ripple current and thus reduce the size of the filter components. On another note, in the topology of FIG1A, the resonant converter 100 is mainly a structure of two resonant conversion modules (1A, 1B) connected in parallel (i.e., the primary side is connected in parallel, and the secondary side is also connected in parallel). Therefore, it can convert high input voltages, and even medium-voltage (MV, generally exceeding 1000V) DC voltages Vdc, and provide high power output.

[0051] Please refer to Figure 1B, which is a circuit block diagram of the second embodiment of the resonant converter of this disclosure, and refer in conjunction with Figure 1A, and repeatedly refer to Figures 1A and 1B. The difference between Figure 1B and Figure 1A is that the switching circuit 4 is connected in series with the DC voltage Vdc to form a circuit structure in which the primary side of the resonant converter 100 is connected in series and the secondary side is connected in parallel. It is worth mentioning that, in one embodiment, the secondary-side switching circuit 5 of Figure 1B can be similar to the structure of the primary side, that is, it can also be connected in series, in parallel, or even have a single output (the secondary side can be coupled to a 400 / 800V battery). The secondary-side switching circuit 5 of Figure 1A is also like this, and will not be described again here.

[0052] Furthermore, since the DC voltage Vdc (e.g., but not limited to 800V) received by the resonant converter 100 is affected by the series connection on the primary side, the capacitor C1 at the two input terminals divides the DC voltage Vdc in half (Vdc / 2, 400V). Therefore, the voltage rating of the switching switches Q1 to Q4 can be reduced, i.e., for example, but not limited to, using GaN components with a voltage rating of approximately 650V to replace the switching switches Q1 to Q4 with a voltage rating of 800V or higher. Other coupling relationships and their achievable effects can be seen in Figure 1A, and will not be elaborated further here.

[0053] Please refer to Figure 1C, which is a circuit block diagram of the third embodiment of the resonant converter of this disclosure, and refer in conjunction with Figures 1A to 1B, and repeatedly refer to Figures 1A and 1C. The difference between Figure 1C and Figure 1A is that the primary side circuit A includes a set of switching circuits 4, and the switching circuit 4 includes a first switching bridge arm 42 and a second switching bridge arm 44 connected in parallel. Furthermore, the primary side winding Wp of the transformer T is coupled in series to form a primary side winding string. The node Pp1 between the first switching switch Q1 and the second switching switch Q2 is coupled to one end of the primary side winding string, and the node Pp2 between the third switching switch Q3 and the fourth switching switch Q4 is coupled to the other end of the primary side winding string.

[0054] Furthermore, the resonant converter 100 in Figure 1C is mainly a structure of two resonant conversion modules (1A, 1B) combined into a single H-bridge resonant conversion module 1, with the two primary windings Wp connected in series. Similarly, the secondary windings can also be connected in series, in parallel, or even in a single-output structure. Since the voltage across the transformer T in Figure 1C is evenly distributed across the two primary windings Wp, the voltage across a single primary winding Wp is only half, thus the turns ratio of the primary winding Wp to the secondary winding Ws can be reduced (indicated by m:1, where m can be any number greater than 0). Therefore, the architecture in Figure 1C is particularly suitable for applications converting from high voltage to low voltage (e.g., but not limited to, converting from 400V to 12V). Other coupling relationships and their achievable effects can be found in Figure 1A, and will not be elaborated further here.

[0055] Please refer to Figure 1D, which is a circuit block diagram of the fourth embodiment of the resonant converter of this disclosure, and also refer to Figures 1A to 1C, and repeatedly refer to Figures 1A and 1D. The difference between Figure 1D and Figure 1A is that the resonant inductor Lr1 of the resonant circuit B forms a common iron core structure M. Furthermore, the dotted terminals (same-name terminals) of the resonant inductor Lr1 are in the same position. Since the two resonant inductors Lr1 form a common iron core structure M, the two resonant inductors Lr1 are connected in parallel, which reduces current ripple. Furthermore, since the current ripple is reduced, the size of the iron core can be reduced (compared to two separate iron cores). Other than this, the remaining coupling relationships and the effects they achieve can be referred to in conjunction with Figure 1A, and will not be elaborated further here.

[0056] Please refer to Figure 1E, which is a circuit block diagram of the fifth embodiment of the resonant converter of this disclosure, and refer in conjunction with Figures 1A to 1C, and repeatedly refer to Figures 1C and 1E. The difference between Figure 1E and Figure 1C is that the switching circuit 4 further includes a third switching bridge arm 46 connected in parallel with the first switching bridge arm 42, and the third switching bridge arm 46 includes a fifth switching switch Q5 and a sixth switching switch Q6 connected in series. In addition to forming a primary winding string structure, the transformer T also forms a center tap Pc between the two primary windings Wp. Furthermore, the node Pp3 between the fifth switching switch Q5 and the sixth switching switch Q6 is coupled to the center tap Pc. Since the primary winding Wp of the transformer T has a center tap structure, the cross voltage of the two primary windings Wp is Vdc, making this structure suitable for applications with small differences in the transformer turns ratio (e.g., but not limited to 1:1). Apart from this, the other coupling relationships and the effects they can achieve can be referred to in conjunction with Figure 1C, and will not be described in detail here.

[0057] Please refer to Figure 2A, which is a circuit block diagram of the first embodiment of the resonant conversion module of this disclosure, and also refer to Figures 1A to 1E. In Figure 2A, the resonant conversion module 1 mainly shows the circuit structure of a series resonant dual active bridge, which mainly includes a primary-side resonant slot and a secondary-side resonant slot. The primary-side resonant slot includes a resonant inductor Lr1 and a resonant capacitor Cr1, and the secondary-side resonant slot includes a resonant inductor Lr2 and a resonant capacitor Cr2. Its detailed circuit architecture and coupling relationship are similar to the resonant conversion modules (1A, 1B) in Figure 1A, and will not be described again here. Since resonant capacitors Cr1 and Cr2 are added to the resonant circuit of the resonant conversion module 1, the resonant capacitors Cr1 and Cr2 will resonate with the resonant inductors Lr1 and Lr2, respectively. Due to the resonance of the resonant capacitors Cr1 and Cr2 and the resonant inductors Lr1 and Lr2, the inductor current Il rises or falls linearly (the frequency is equal to the resonant frequency of the resonant capacitors Cr1 and Cr2 and the resonant inductors Lr1 and Lr2). It is worth mentioning that, in one embodiment, since the circuit architecture of FIG2A has only a single set of resonant conversion modules 1, this single set of resonant conversion modules 1 can be regarded as resonant converter 100, and the resonant conversion module 1 described below is also like that.

[0058] Please refer to Figure 2B, which is a simplified equivalent circuit diagram of the resonant conversion module according to the first embodiment of this disclosure, and also refer to Figures 1A to 2A. In Figure 2B, the voltages of nodes Pp1 and Pp2 in Figure 2A are equivalent to voltage sources (voltages) Vpp, and the voltages of nodes Ps1 and Ps2 are equivalent to voltage sources (voltages) Vss. Furthermore, the equivalent inductance of the resonant conversion module 1 is Le, and the equivalent capacitance is Ce. In the phase shift mode of the above three components, the voltages Vpp and Vss will exhibit waveform lead / lag due to phase shift, thereby improving the dynamic performance of the circuit in response to load changes and shortening the adjustment time.

[0059] Please refer to Figure 2C, which is a power and voltage curve of the resonant conversion module of this disclosure for charging and discharging a battery, and also refer to Figures 1A to 2B. Although the controller 6 of this disclosure can control the resonant conversion module 1 to provide a fixed voltage Vo to the device 200 coupled to the back end, the resonant conversion module 1 is particularly suitable for coupling a battery to charge and discharge it because it has the characteristic of adjusting the voltage Vo. Furthermore, the following description of this disclosure mainly focuses on the concept when the device 200 is a battery. Specifically, Figure 2C is mainly a power Po and voltage Vo curve of Figure 2A. When the voltage Vo is above V1 (e.g., but not limited to 200V), the controller 6 can set the mode of the resonant conversion module 1 to charge and discharge the battery in a constant current charging mode CC, and the power Po starts to increase from Pmin (e.g., but not limited to 3.3kW). Once the voltage Vo rises to V2 (e.g., but not limited to 280V), the controller 6 can set the resonant conversion module 1 to a constant power charging mode (CP) for charging and discharging the battery, and the power Po is maintained at Pmax (e.g., but not limited to 6.6kW). Finally, once the voltage Vo rises to V3 (e.g., but not limited to 460V), the controller 6 can set the resonant conversion module 1 to a constant voltage charging mode (CV) for charging and discharging the battery. Since the battery is nearly fully charged, the power begins to gradually decrease.

[0060] Please refer to Figure 3A, which shows the waveform of the resonant conversion module of this disclosure operating in three-phase shift mode, and also refer to Figures 1A to 2C. In Figure 3A, Vpp is the voltage waveform between nodes Pp1 and Pp2 of switching circuit 4 (refer to Figure 2A) when switching switches Q1 to Q4 are switched, and Vss is the voltage waveform between nodes Ps1 and Ps2 of secondary-side switching circuit 5 (refer to Figure 2A) when secondary-side switches Qs1 to Qs4 are switched. Specifically, Figure 3A mainly shows the waveform of the resonant conversion module 1 of Figure 2A operating in three-phase shift (TPS) mode. Therefore, the controller 6 in Figure 2A can control the switching circuit 4 to perform phase shifting, and its phase shift amount D1 (first phase shift amount) can be 0-0.5 (the phase shift of the switching switches Q1 to Q4 will cause the waveform of voltage Vpp to produce a situation similar to the increase or decrease of the duty cycle). Similarly, the controller 6 in Figure 2A can control the secondary-side switching circuit 5 to perform phase shifting, and its phase shift amount D2 (second phase shift amount) can also be 0-0.5. In this way, voltages Vpp and Vss will increase or decrease with the change of phase shift amount. In addition, the controller 6 in Figure 2A can also control the switching circuit 4 and the secondary-side switching circuit 5 to perform phase shifting, and its phase shift amount ψ (third phase shift amount) can be -T / 2 to T / 2 (where T is the period, which corresponds to the operating frequency of the resonant conversion module 1). Therefore, voltages Vpp and Vss will move relative to each other. Furthermore, in addition to performing three-phase shift mode, this invention also performs frequency variation compared to conventional fixed-frequency technology (the period is fixed at 0 to T), which will be explained in more detail later.

[0061] Specifically, based on the operation of the three-phase shift (TPS) mode described above, the following three waveform timing diagrams in Figures 3B to 3D can be summarized. In region I of Figure 3B, the phase shift D1 is greater than the phase shift D2 plus a certain amount, thus forming the corresponding voltage waveforms Vpp and Vss, as well as the corresponding inductor voltage Vl and inductor current Il (refer to Figure 2A). Furthermore, the switching switches Q1-Q4 and the secondary-side switches Qs1-Qs4 can operate in zero-voltage switching (ZVS). Region III of Figure 3C is similar to Figure 3B, thus yielding waveforms similar to those in Figure 3B. Switches Q1-Q4 and the secondary-side switches Qs1-Qs4 can also operate in ZVS. In region VI of Figure 3D, the phase shift D2 is greater than the phase shift D1 plus a certain amount, thus yielding waveforms similar to those in Figure 3B. Switches Q1-Q4 and the secondary-side switches Qs1-Qs4 can also operate in ZVS. Therefore, through the three-phase shift (TPS) mode and frequency variation technology disclosed in this application, the resonant conversion module 1 of this application can provide better efficiency at low power. In one embodiment, for ease of presentation, the inductor current Il waveforms in Figures 3B to 3D are shown only as simplified point-to-point linear waveforms.

[0062] Please refer to Figure 3E, which shows the waveform diagram of the resonant conversion module of this disclosure operating in single-phase-shift mode, and Figure 3F, which shows the waveform diagram of the resonant conversion module of this disclosure operating in dual-phase-shift mode. Also refer to Figures 1A to 3D. In Figure 3E, the controller 6 mainly controls the resonant conversion module 1 to operate in single-phase-shift (SPS) mode. Therefore, the controller 6 only phase-shifts the control signals of the switching switches Q1-Q4 or the secondary-side switches Qs1-Qs4. Conversely, in Figure 3E, the controller 6 mainly controls the resonant conversion module 1 to operate in single-phase-shift (SPS) mode. Therefore, the controller 6 phase-shifts the control signals of the switching switches Q1-Q4 and the secondary-side switches Qs1-Qs4, causing the waveform of the inductor current Il to exhibit an exponential increase / decrease from I0, I1, -I0 to -I1.

[0063] In Figure 3F, controller 6 primarily controls the resonant conversion module 1 to operate in dual phase-shift (DPS) mode. This causes the resonant inductors Lr1 and Lr2 to resonate with the resonant capacitors Cr1 and Cr2, respectively. Consequently, the waveform of the inductor current Il changes non-exponentially from I0, I1, -I0 to -I1. Therefore, the inductor current Il becomes more distorted due to resonance, making it impossible to design parameters such as the resonant slot using simple formulas, and to calculate the value of the inductor current Il during operation to obtain the inductor current waveform.

[0064] Therefore, please refer to Figure 4A, which is a flowchart of the parameter calculation method for obtaining resonant parameters in the resonant conversion module of this disclosure, and also refer to Figures 1A to 3F. The parameter calculation method in Figure 4A is mainly applicable to the resonant conversion module 1 of the SR DAB architecture. Although the resonant conversion module 1 of the DAB architecture generally only has a resonant inductor Lr1 and the resonant parameters can be obtained through a simple calculation formula, the parameter calculation method in Figure 4A can also be applied to the resonant conversion module 1 of the DAB architecture to obtain more accurate resonant parameters, and even to resonant converters such as LLC and LC for calculating resonant parameters. To avoid obscuring the technical characteristics of this disclosure, the parameter calculation method in Figure 4A is still mainly applied to the resonant conversion module 1 of the SR DAB architecture.

[0065] Specifically, the parameter calculation method of this disclosure first includes setting preset operating conditions for the resonant converter, and these preset operating conditions include a specific frequency range of the operating frequency of the resonant converter (S100). Before calculating the resonant parameters, the preset operating conditions of the resonant converter 100 (resonant conversion module 1) must first be determined. These preset operating conditions include, but are not limited to, the upper power limit Po = Pmax at the device end of the resonant converter 100 (resonant conversion module 1) (which can be set to 6.6kW with reference to Figure 2C), the voltage Vo variation range Vo = V2~V3 (which can be set to 280V~460V with reference to Figure 2C), and may also include, but are not limited to, the upper and lower limits of the values ​​of the resonant inductors Lr1 and Lr2, the upper and lower limits of the values ​​of the resonant capacitors Cr1 and Cr2, and the DC voltage Vdc. The preset operating conditions need to include a specific frequency range of the operating frequency of the resonant converter 100 (resonant conversion module 1) (for example, but not limited to, the frequency variation range of the operating frequency is 400kHz~600kHz).

[0066] The frequency variation in this disclosure refers to a function similar to frequency conversion. Specifically, in constant power charging mode CP, the output power of the resonant conversion module 1 remains constant, but the current varies with the voltage. Therefore, as the voltage of the device 200 (e.g., but not limited to, a battery) increases during charging, the current decreases, causing the output conditions of the resonant conversion module 1 to constantly change. Consequently, the operating frequency calculated by the controller 6 will vary (in the above example, it will vary between 400kHz and 600kHz). Furthermore, the parameter calculation method of this disclosure can be performed by a device with a processor (e.g., but not limited to, a computer). The operator can input preset operating conditions into the pre-designed calculation software in the device, and then obtain the optimal value through selection and operation.

[0067] Then, multiple sets of first resonant parameters are calculated within a specific frequency range, and these multiple sets of first resonant parameters include the turns ratio of the resonant converter transformer (S200). After the operating frequency (400kHz~600kHz) is set, numerous combinations of values ​​for the resonant circuit B can be calculated based on the operating frequency. These combinations may include, but are not limited to, multiple inductance values ​​of resonant inductors Lr1 and Lr2, multiple capacitance values ​​of resonant capacitors Cr1 and Cr2, and multiple turns ratios n of the transformer T. These combinations of resonant parameters (including inductance values, capacitance values, and turns ratio n) are all resonant parameters that satisfy preset operating conditions when the controller 6 controls the resonant converter 100 (resonant conversion module 1).

[0068] Then, among the multiple sets of first resonant parameters, the first resonant parameter with the minimum resonant inductor current ripple of the resonant converter is obtained, and multiple sets of second resonant parameters are calculated based on the turns ratio of the first resonant parameter and preset operating conditions (S300). Among the combinations of multiple resonant parameters, the first resonant parameter with the minimum current ripple (min_Irms) among the effective values ​​Il(rms) of the inductor current Il of the resonant inductor Lr1 of the resonant converter 100 (resonant conversion module 1) is found and selected. The current ripple Q can be calculated using the formula Q=min 1 / N∑Il_rms(i)^2*(1+n^2), but is not limited to this, and has various existing calculation methods. Then, the turns ratio n of the selected first resonant parameter is used as the selected turns ratio N=n, and multiple sets of second resonant parameters are calculated based on the selected turns ratio N and the same preset operating conditions. These second resonant parameters, besides having a fixed value for the selected turns ratio N=n, can include combinations of multiple inductor values ​​and multiple capacitor values.

[0069] Please refer to Figure 4B, which is a schematic diagram of the numerical distribution of the second resonant parameters of this disclosure, and also refer to Figure 4A. The aforementioned multiple second resonant parameters, after being graphically represented according to the inductance value Lv and capacitance value Cv, can be arranged as shown in Figure 4B. Each point represents one second resonant parameter, and each second resonant parameter includes at least one inductance value Lv, capacitance value Cv, and a selected turns ratio N; therefore, the turns ratio parameters of the second resonant parameters are all the same. In Figure 4B, the strip interval BI represents the preferred second resonant parameters, which mainly include smaller inductance values ​​Lv or capacitance values ​​Cv, primarily to reduce the size of the inductor and capacitor.

[0070] Referring again to Figure 4A, the parameter calculation method of this disclosure finally includes selecting one of the multiple sets of second resonant parameters as the resonant parameter (S400). The second resonant parameters in the strip segment BI of Figure 4B are all preferred second resonant parameters, and these second resonant parameters can all be used to set the resonant circuit B of the resonant converter 100 (resonant conversion module 1). Furthermore, among these second resonant parameters, the uppermost second resonant parameter has a larger capacitance value Cv (lower voltage stress) and a smaller inductance value Lv, but the total harmonic distortion (THD) of the inductor current Il will be higher. Conversely, the lowermost second resonant parameter has a smaller capacitance value Cv (higher voltage stress), a larger inductance value Lv, and the total harmonic distortion (THD) of the inductor current Il will be lower. Moreover, a larger value means that the component occupies a larger volume, especially the capacitance value Cv.

[0071] Therefore, the designer can select any one of the multiple second resonant parameters in the strip interval BI to set the resonant circuit B of the resonant converter 100 (resonant conversion module 1) according to specific conditions, such as, but not limited to, manufacturer and customer requirements, circuit size specifications, etc. In this way, the designer can select more suitable parameters in the strip interval BI according to the application and actual needs of the resonant converter 100 (resonant conversion module 1). Therefore, in summary, this disclosure provides an optimization process for the resonant converter 100 (resonant conversion module 1), thereby obtaining better values ​​for the resonant parameters (i.e., inductance value Lv, capacitance value Cv, and turns ratio n) through precise analysis.

[0072] Please refer to Figures 5A-5C, which are simulated waveforms of the second resonant parameters using one embodiment of the resonant converter of this disclosure, and Figures 6A-6C, which are simulated waveforms of the second resonant parameters using another embodiment of the resonant converter of this disclosure, in conjunction with Figures 1A-4B. The second resonant parameters introduced in Figures 5A-5C are simulated waveforms of a low inductance value Lv, a high capacitance value Cv, and an operating frequency Fs varying from 400kHz to 600kHz. Furthermore, as can be seen from the power / voltage distribution diagram in Figure 5A, at any point, the resonant converter 100 (resonant conversion module 1) can achieve zero-voltage switching (ZVS) under the control of the controller 6. In addition, Figures 5B and 5C are adjusted to three-dimensional perspective views. Besides the power / voltage distribution, Figure 5B also includes the parameter of the operating frequency Fs, and Figure 5C also includes the parameter of the inductor current Il (rms). Therefore, as can be seen from Figures 5A to 5C, the second resonant parameter of low inductance value Lv and high capacitance value Cv can achieve the function of zero voltage switching (ZVS) in a large range of power / voltage.

[0073] Figures 6A to 6C are similar to Figures 5A to 5C, except that the second resonant parameter introduced in Figures 6A to 6C is a high inductance value Lv and a low capacitance value Cv. Furthermore, it can be seen that the region where the zero-voltage switching (ZVS) function can be achieved differs slightly from Figures 5A to 5C, but the overall range is roughly equivalent to 5A to 5B. Therefore, through the parameter (optimization) calculation method of the resonant converter disclosed herein, selective matching can be performed based on the respective costs of the resonant inductors Lr1 and Lr2, the resonant capacitors Cr1 and Cr2, and the transformer T, thereby reducing the cost of the resonant circuit B and achieving the zero-voltage switching (ZVS) function, thus improving the efficiency of the resonant converter 100 (resonant conversion module 1).

[0074] On the other hand, after the resonant parameters are calculated and designed, the closed-loop design and operation of the resonant converter 100 (resonant conversion module 1) must be performed to confirm parameters such as phase shift and dead time of the resonant converter 100 (resonant conversion module 1), and to adjust the pulse width modulation signal (PWM) provided by the controller 6 to increase the stability and efficiency of the resonant converter 100 (resonant conversion module 1) during operation. Specifically, please refer to Figure 7A, which is the system control block diagram of the resonant converter of this disclosure, and refer to Figures 1A to 6C in conjunction with Figures 2A and 7A repeatedly. Since the resonant converter 100 (resonant conversion module 1) uses the inductor current of the SR DAB conversion module to resonate, its current change is more complex than that of the DAB architecture, making it more difficult to calculate its current parameters using simple formulas. Therefore, in addition to using the parameter calculation methods shown in Figures 6A to 6B to calculate the resonant parameters, the resonant converter 100 (resonant conversion module 1) also needs to obtain the voltage Vo and power Po changes at the device end through program simulation, and obtain appropriate parameters through the closed-loop system control block shown in Figure 7A, so that the controller 6 can adjust the pulse width modulation signal PWM accordingly.

[0075] Specifically, in the battery charging process of the resonant converter 100 (resonant conversion module 1), the main control is the current Ib at the control device end, and the sum of the current Ib and the current supplied to the capacitor Co is the average current Is(avg) after the secondary-side switching of the secondary-side switching circuit 5. Therefore, the closed-loop system control block of the resonant converter 100 (resonant conversion module 1) for battery charging shown in FIG7A can be obtained. In FIG7A, the controller 6 can set the current command Ib*, and the controller 6 can adjust the closed loop according to the error between the current command Ib* and the current Ib. Furthermore, the adjustment is mainly made by adjusting the operating frequency Fs, phase shifts D1, D2, and ψ, so that the current Ib can be adjusted according to the current command Ib*.

[0076] Furthermore, since the processing speed and throughput of controller 6 are related to its cost, and since the voltage Vo and current Ib are time-varying in the constant power charging mode CP, if the controller 6 were to perform real-time calculations, it would require a faster processing speed or a larger throughput, making it impossible to reduce the cost of controller 6. Therefore, in addition to increasing the stability and efficiency of the resonant converter 100 (resonant conversion module 1) by obtaining appropriate parameters through a closed-loop system control block, this disclosure further reduces the cost of controller 6 by using a lookup table, allowing it to use a lower-order digital signal processor (DSP).

[0077] Specifically, please refer to Figure 7B, which is the control block diagram for the operating frequency lookup table of this disclosure, and refer in conjunction with Figures 1A to 7A, and repeatedly refer to Figures 2A, 7A to 7B. In the closed-loop block of Figure 7A, the controller 6 can read the voltage Vo and power Po at the device end, and the voltage Vo and power Po are used to obtain the corresponding operating frequency Fs by substituting them into the pre-set first parameter lookup table 7 shown in Figure 7B. Specifically, the controller 6 can obtain the frequency value Fv corresponding to the voltage Vo and power Po through the first parameter lookup table 7. Furthermore, since the controller 6 generally reads the voltage Vo and power Po at a relatively fast speed, it is more prone to transient noise, which causes the frequency value Fv to fluctuate accordingly. Therefore, the frequency value Fv can be filtered by the filter block PF to obtain a more stable frequency command Cf, and the frequency command Cf corresponds to the operating frequency Fs of the control switches (switching switches Q1 to Q4 and secondary side switches Qs1 to Qs4). In this context, the filter block PF is preferably a low-pass filter block to remove high-frequency noise, but it is not limited to this. Furthermore, besides directly using the value obtained from the filter block PF as the frequency command Cf, a difference representation can also be used. That is, the frequency command Cf can be the difference between the frequency value Fv and the value obtained from the filter block PF, and adjustments made using this difference are generally more accurate. Therefore, the frequency command Cf can be represented in various ways, not limited to the examples above.

[0078] Referring to Figure 7A, the frequency command Cf corresponding to the operating frequency Fs is input into the block f^-1 to obtain the phase shifts D1, D2, and ψ, and the current Ib is adjusted through the closed-loop control block. Since the phase shifts D1, D2, and ψ are numerical values, and the controller 6 can achieve the phase shift effect by converting the numerical values ​​into time differences or other delays, referring to Figure 7C, the phase shifts D1, D2, and ψ can be converted from numerical values ​​to time values ​​through the ψ conversion block to convert the numerical values ​​into corresponding delay times, and a pulse width modulation command Cpwm is provided based on the delay time. Furthermore, the controller 6 modulates the pulse width modulation signal PWM according to the pulse width modulation command Cpwm, and controls the switching of switches (switching switches Q1~Q4 and secondary side switches Qs1~Qs4) through the pulse width modulation signal PWM. However, in addition to the delay time, the controller 6 should have various other numerical conversion methods, and is not limited to the above-described implementation.

[0079] On the other hand, please refer to Figure 7D, which is a block diagram of the dead-time lookup control of this disclosure, and refer in conjunction with Figures 1A to 7A, and repeatedly refer to Figures 2A, 7A, and 7D. Since dead time is required when switching the upper and lower arms of the switching circuits Q1 to Q4 and the secondary-side switches Qs1 to Qs4, and control is also required based on zero-voltage switching (ZVS), the appropriate dead time can also be obtained by looking up a table using the control strategy of the resonant converter 100 (resonant conversion module 1) disclosed in this disclosure. That is, the dead time during the switching of the upper and lower arms of the switching circuit 4 and the secondary-side switching circuit 5 of the resonant converter 100 (resonant conversion module 1) can be obtained by looking up a table using the voltage Vo and power Po at the device end. In this way, the controller 6 can control the switching circuits Q1 to Q4 and the secondary-side switches Qs1 to Qs4 according to the dead time to achieve the effect of zero-voltage switching (ZVS).

[0080] Specifically, in the closed-loop block of Figure 7A, the controller 6 can read the voltage Vo and power Po at the device end, and input the voltage Vo and power Po into the pre-set second parameter lookup tables 7A and 7B shown in Figure 7D to obtain the dead time during the switching of the switching circuit 4 and the secondary-side switching circuit 5. Specifically, the controller 6 can obtain the dead time values ​​DTp and DTs through the second parameter lookup tables 7A and 7B, respectively. Among them, the dead time value DTp corresponds to the primary-side dead time when the primary-side switching switches Q1 to Q4 are switched, and the dead time value DTs corresponds to the secondary-side dead time value DTs when the secondary-side switches Qs1 to Qs4 are switched. Then, the dead time command Cdt is generated by the control of the dead time control block 9, and the dead time command Cdt corresponds to the dead time during the switching of the switching circuit 4 and the secondary-side switching circuit 5.

[0081] Finally, integrating Figures 2A and 7A-7D, we can obtain the closed-loop control flowchart of the resonant converter shown in Figure 8. Specifically, in Figure 8, the closed-loop control flowchart of the resonant converter includes a first process (S500-S560) and a second process (S600-S680). The first process (S500-S560) is a fast-response process, mainly aimed at improving the response speed of the resonant converter 100 (resonant conversion module 1) to increase its stability and provide better control. Therefore, the controller 6 typically executes this process at a relatively fast frequency (e.g., but not limited to 50µs), and the steps of the first process (S500-S560) are executed regardless of whether the resonant converter 100 is operating in a steady state. The second process (S600-S680) is a slow-response process, mainly related to the dead time, and is used to increase the operating efficiency of the resonant converter 100. Since the response speed of this process does not easily affect the stability of the resonant converter 100 (resonant conversion module 1), and it does not require constant table checks and value changes. Therefore, the controller 6 typically executes the process at a slower frequency (e.g., but not limited to 1ms) and needs to wait for the resonant converter 100 to enter a steady state before executing the second process (S600 to S680).

[0082] The first process (S500-S560) first includes confirming the current at the device terminal of the resonant converter (S500). Then, the phase shift of the resonant converter is calculated based on the current (S520). Specifically, as shown in Figure 7A, the controller 6 can set the current command Ib*, and calculate the secondary side average current Is(avg) based on the first error between the current Ib and the current command Ib*, and calculate the phase shifts D1, D2, and ψ based on the secondary side average current Is(avg). Then, the operating frequency of the voltage and power at the corresponding device terminal is confirmed through the first parameter lookup table (S540, this can be referred to Figure 7B, and will not be elaborated here). Finally, the pulse width signal used to control the resonant converter is adjusted by the phase shift and the operating frequency (S560, this can be referred to Figure 7C, and will not be elaborated here).

[0083] The second process (S600-S680) follows the first process (S500-S560), and can be initiated at any point during the execution of steps (S500) to (S520), as long as the timing is appropriate (step (S500) is used as an illustrative example here). Furthermore, after the resonant converter 100 enters a steady state, the second process (S600-S680) is executed. The controller 6 can determine whether the resonant converter 100 is operating in a steady state by detecting, for example but not limited to, whether parameters such as the operating frequency Fs, current Ib, phase shift ψ, D1, and D2 of the resonant converter 100 are within a steady-state range. The second process (S600-S680) first includes confirming the phase shift and operating frequency of the resonant converter (S600). After the first process (S500-S560) has been executed at least once, the phase shifts D1, D2, ψ and the operating frequency Fs of the resonant converter 100 (resonant conversion module 1) can be obtained. Then, it is confirmed whether the phase shifts and the operating frequency are within a predetermined range (S620). When both the phase shifts D1, D2, ψ and the operating frequency are within the predetermined range, it means that the efficiency of the resonant converter 100 (resonant conversion module 1) can be further improved by adjusting the dead time. Therefore, the dead time of the voltage and power at the device terminal of the corresponding resonant converter is confirmed by the second parameter lookup table (S640, this can be referred to Figure 7D, and will not be elaborated here).

[0084] Then, the pulse width modulation signal used to control the resonant converter is adjusted by the dead time (S660). The controller 6 can adjust the pulse width modulation signal PWM by the dead time command Cdt corresponding to the dead time, and the specific adjustment method is similar to that in Figure 7C, that is, the value of the dead time command Cdt is converted into the pulse width modulation command Cpwm corresponding to the delay time, which will not be described in detail here. On the other hand, when the judgment in step (S620) is negative, it means that at least one of the phase shifts D1, D2, ψ and the operating frequency Fs is not within the predetermined range. Therefore, the dead time is set to a preset value (S680). When at least one of the phase shifts D1, D2, ψ and the operating frequency Fs is not within the predetermined range, it means that the resonant converter 100 (resonant conversion module 1) may not have been adjusted to a better state. Therefore, the dead time is set to a preset value without adjustment, and the process continues with step (S660).

[0085] It is worth mentioning that, in one embodiment, the closed-loop control flow shown in Figure 8 is preferably applied to the resonant converter 100 (resonant conversion module 1) operating in constant power charging mode CP (i.e., the battery charging and discharging mode is constant power charging mode CP), and after confirming that the operation is in constant power charging mode CP, the process proceeds to step (S500). The main reason for this is that in constant power charging mode CP, the voltage Vo and current Ib are not constant values, so parameters such as phase shift D1, D2, ψ and operating frequency Fs will be adjusted at any time to achieve the requirement of high efficiency. However, constant current mode CC and constant voltage mode CV can also be implemented using the flow in Figure 8, but since the voltage Vo or current Ib is constant, some parameters such as phase shift D1, D2, ψ and operating frequency Fs will not be adjusted at any time, and the overall efficiency is not significantly improved.

[0086] Please refer to Figure 9A, which shows the zero-voltage switching range distribution obtained by executing the resonant converter using an analog program and substituting the first implementation parameters; Figure 9B, which shows the zero-voltage switching range distribution obtained by executing the resonant converter using an analog program and substituting the second implementation parameters; and Figure 9C, which shows the zero-voltage switching range distribution obtained by executing the resonant converter using an analog program and substituting the third implementation parameters. Also refer to Figures 1A to 8. Figures 9A to 9C mainly show the circuit architecture of the resonant converter 100 (resonant converter module 1) executed using analog programs such as MATLAB / Simulink, and the zero-voltage switching (ZVS) range distribution plots drawn using the parameters mentioned in Figures 1A to 8. The only difference between Figures 9A and 9C is the parameter. Specifically, in Figure 9A, the inductance value Lv of the resonant inductor Lr1 is set to 4uH, and the operating frequency Fs is set to a frequency variation range of 400kHz to 600kHz. Figure 9B is similar to Figure 9A, except that the inductance value Lv of the resonant inductor Lr1 is set to 5uH, and Figure 9C is similar to Figure 9B, except that the operating frequency Fs is fixed at 400kHz.

[0087] As can be seen from the range in Figure 9A, in the range where the resonant converter 100 (resonant converter module 1) outputs low power Po and low voltage Vo, the resonant converter 100 (resonant converter module 1) can still operate under zero-voltage switching (ZVS) conditions. On the other hand, comparing Figures 9A and 9B, different inductance values ​​Lv have their own advantages. When the inductance value Lv is 4uH, although the range of zero-voltage switching (ZVS) is smaller under high power Po (or high voltage Vo) conditions, the range of zero-voltage switching (ZVS) is larger under low power Po (or low voltage Vo) conditions. The opposite is true when the inductance value is 5uH, so the circuit components can be configured according to the actual needs of the circuit. On the other hand, in Figure 9C, the operating frequency Fs is fixed at 400kHz. Therefore, comparing Figures 9B and 9C, in the range of low power Po and low voltage Vo in Figure 9B, the resonant converter 100 (resonant converter module 1) can still have a larger range of zero-voltage switching (ZVS). Therefore, with the same inductance value Lv, a larger zero-voltage switching (ZVS) range can be obtained by using the frequency variation operation mode, which makes the efficiency of the resonant converter 100 (resonant converter module 1) better.

[0088] Please refer to Figure 10A, which is a circuit block diagram of the second embodiment of the resonant conversion module of this disclosure, and also refer to Figures 1A to 9C. The difference between Figure 10A and Figure 2A is that the resonant conversion module 1 is a dual active bridge circuit architecture. The biggest difference is that the resonant circuit B includes a transformer T and a resonant inductor Lr1. The rest of the circuit structure and coupling method are the same as those in Figure 2A, and will not be described again here. The resonant circuit B of the resonant conversion module 1 in Figure 10A may also include capacitors Cr1 and Cr2 as shown in Figure 2A, but their capacitance values ​​are smaller (e.g., but not limited to single-digit nF or pF). Therefore, if it includes capacitors Cr1 and Cr2, then capacitors Cr1 and Cr2 do not participate in resonance, but are used as DC blocking capacitors. Refer to Figure 10B, which is a simplified equivalent circuit diagram of the resonant conversion module of the second embodiment of this disclosure. In Figure 10B, Le is the equivalent inductance on the primary side, and Ce1 and Ce2 are the equivalent capacitances on the primary and secondary sides, respectively. Furthermore, Lb is the equivalent inductance of the battery, Rb is the equivalent impedance of the battery, and Eb is the battery voltage. Referring to Figure 11A, which is a block diagram of the closed-loop system control for battery charging using the resonant switching mode according to the second embodiment of this disclosure, in the closed-loop system control block, since the charging mode primarily involves charging the battery, the battery voltage will increase or decrease with changes in battery capacity rather than remaining constant. Therefore, in the charging mode, the focus is on controlling the current Ib.

[0089] Furthermore, under the closed-loop control of the charging mode, the resonant converter 100 (resonant conversion module 1) enters the closed-loop control shown in Figures 7A and 11A, and can also enter the closed-loop control process shown in Figure 8. Specifically, the current command Ib* is compared with the actual current Ib fed back to the battery to obtain the current Is(avg) for charging the capacitor Co. On the other hand, the phase shifts D1, D2, and ψ can be calculated using conversion equations, and the capacitor voltage Vc can be calculated using conversion equations from the currents Is(avg) and Ib. After calculating the error between the capacitor voltage Vc and the battery voltage Eb, the current Ib for charging the battery is obtained. In the above-mentioned closed-loop control operation, the stability of the resonant converter 100 is mainly increased by executing the first process (S500~S560). Furthermore, after confirming that the resonant converter 100 (resonant conversion module 1) is operating in a steady state, the operating efficiency of the resonant converter 100 is improved by executing the second process (S600~S680). The reason for this is that when the resonant converter 100 (resonant conversion module 1) operates in a transient state, it is usually either newly started or experiencing an abnormal state. At this time, the detected values ​​will be abnormal (for example, but not limited to, large fluctuations in operating frequency Fs and current Ib). Therefore, the first process (S500-S560) is needed to stabilize the control of the resonant converter 100 as quickly as possible. Furthermore, once it is confirmed that the resonant converter 100 (resonant conversion module 1) is operating in a steady state, the second process (S600-S680) is initiated.

[0090] Please refer to Figure 11B, which is a block diagram of the closed-loop system control for battery discharge by the resonant converter module according to the second embodiment of this disclosure, and also refer to Figures 1A to 11A. In the closed-loop control, since the discharge mode is that the battery discharges to the power supply terminal (i.e., the DC voltage Vdc terminal) of the resonant converter 100 (resonant converter module 1), the DC voltage Vdc at the power supply terminal is regarded as the load (which represents the equivalent resistance fed into the power supply terminal by the analog inverter, and the change of the load will have a significant impact on the settling time). Therefore, the DC voltage Vdc at the power supply terminal is a constant value, and the difference lies in the primary side current Ip(avg) it draws. Therefore, in the discharge mode, the focus is on the control of the DC voltage Vdc.

[0091] Specifically, in the closed-loop control of the resonant converter 100 (resonant conversion module 1) in discharge mode, the voltage command Vdc* is compared with the actual DC voltage Vdc at the power supply terminal. The primary side current Ip(avg) drawn from the power supply terminal, as well as the phase shifts D1, D2, and ψ, can be calculated using the conversion equation. Thus, the DC voltage Vdc can be obtained to form the closed-loop control shown in Figure 11B. Since load changes can significantly affect the settling time, this disclosure further adds a feedforward loop to increase the stability of the closed-loop control. Referring to Figure 11B, 1 / R* is the compensation amount, R* = (2Vdc^2 / Vm*Im), and Vm and Im are the excitation voltage and excitation current of the transformer T, respectively.

[0092] Specifically, the compensation amount is the value estimated by the feedforward loop, which helps to offset load changes. Furthermore, the main purpose of providing the compensation amount in advance is to avoid needing to compensate too much at once when the controller 6 adjusts other parameters. This makes the closed-loop control of the discharge mode more stable. Therefore, the controller 6 can first confirm whether the battery charging and discharging mode of the resonant converter 100 (resonant conversion module 1) is a discharge mode, and if it is confirmed to be a discharge mode, it provides a compensation amount 1 / R* based on the DC voltage. Then, it sets the voltage command Vdc* and calculates the error between the DC voltage Vdc and the voltage command Vdc*, and calculates the primary-side current Ip(avg) based on the error between the two. Finally, the sum of the primary-side current Ip(avg) and the pre-provided compensation amount 1 / R* is the second error amount, and the DC voltage Vdc is adjusted based on the second error amount.

[0093] In summary, the resonant converter 100 of this disclosure can be mainly composed of one or more resonant conversion modules 1, which enables the resonant converter 100 to handle high power through the parallel connection of the resonant conversion modules 1. Furthermore, this disclosure mainly employs interleaved control technology to control the resonant conversion modules 1 to reduce the current stress on the input filter of the switching power supply (SPS) modulation. This is because interleaved control has the effect of ripple current cancellation, which can reduce ripple current and thus reduce the size of the filter components.

[0094] However, the above description is only a detailed description and accompanying drawings of preferred embodiments of the present invention. The features of the present invention are not limited thereto and are not intended to limit the present invention. The full scope of the present invention should be determined by the following claims. All embodiments that conform to the spirit of the claims of the present invention and similar variations thereof should be included in the scope of the present invention. Any variations or modifications that can be easily conceived by those skilled in the art within the field of the present invention can be covered by the following claims.

Claims

1. A parameter calculation method of a resonant converter for obtaining resonant parameters of the resonant converter, the parameter calculation method comprising the steps of: setting a predetermined operating condition of the resonant converter, the predetermined operating condition including a specific frequency range of an operating frequency of the resonant converter; calculating a plurality of sets of first resonant parameters in the specific frequency range, the plurality of sets of first resonant parameters including a turns ratio of a transformer of the resonant converter; obtaining a first resonant parameter of the plurality of sets of first resonant parameters in which a current ripple of a resonant inductor of the resonant converter is a minimum value; calculating a plurality of sets of second resonant parameters from the turns ratio of the first resonant parameter and the predetermined operating condition, and selecting one of the plurality of sets of second resonant parameters as the resonant parameters.

2. The parameter calculation method of claim 1, wherein the second resonant parameters include an inductance value of the resonant inductor and a capacitance value of a resonant capacitor.

3. The parameter calculation method of claim 1, further comprising the steps of: (a) determining a current of a device terminal of the resonant converter, and calculating a phase shift amount of the resonant converter from the current; (b) determining an operating frequency corresponding to a voltage and a power of the device terminal by a first parameter reference table; and (c) adjusting a pulse width modulation signal for controlling the resonant converter by the phase shift amount and the operating frequency.

4. The parameter calculation method of claim 3, further comprising the steps of: setting a current command, and calculating a secondary side average current from a first error amount of the current and the current command; and calculating the phase shift amount from the secondary side average current.

5. The parameter calculation method of claim 3, further comprising the steps of: determining a frequency value corresponding to the voltage and the power by the first parameter reference table; and filtering the frequency value to obtain the operating frequency.

6. The parameter calculation method of claim 3, wherein the resonant converter includes a switching circuit of a primary side and a secondary side switching circuit of a secondary side, and the phase shift amount includes a first phase shift amount of the switching circuit, a second phase shift amount of the secondary side switching circuit, and a third phase shift amount between the switching circuit and the secondary side switching circuit.

7. The parameter calculation method of claim 3, further comprising the steps of: converting the phase shift amount to a corresponding delay time by a value / time conversion, and adjusting the pulse width modulation signal from the delay time.

8. The parameter calculation method of claim 3, further comprising the steps of: (d) determining a phase shift amount and an operating frequency of the resonant converter, and determining whether the phase shift amount and the operating frequency are within a predetermined range; (e) when both the phase shift amount and the operating frequency are within the predetermined range, determining a dead time corresponding to a voltage and a power of a device terminal of the resonant converter by a second parameter reference table; and (f) adjusting a pulse width modulation signal for controlling the resonant converter by the dead time.

9. The parameter calculation method of claim 8, wherein the resonant converter comprises a switching circuit on a primary side and a switching circuit on a secondary side, and the parameter calculation method further comprises the steps of: when both the phase shift and the operating frequency are within the predetermined range, confirming a primary side dead time of the switching circuit on the primary side and a secondary side dead time of the switching circuit on the secondary side by the second parameter reference table.

10. The parameter calculation method of claim 8, further comprising the step of: when one of the phase shift and the operating frequency is not within the predetermined range, setting the dead time as a preset value.

11. The parameter calculation method of claim 8, wherein a device end of the resonant converter is coupled to a battery to charge or discharge the battery, and the parameter calculation method further comprises the steps of: confirming whether the resonant converter operates in a steady state; and when the resonant converter operates in the steady state, proceeding to step (d).

12. The parameter calculation method of claim 3, wherein a device end of the resonant converter is coupled to a battery to charge or discharge the battery, and the parameter calculation method further comprises the steps of: confirming that a mode of charging or discharging the battery is a constant power charging mode; and proceeding to step (a).

13. The parameter calculation method of claim 12, further comprising the steps of: confirming that the mode of charging or discharging the battery is a discharging mode; providing a compensation amount according to a direct current voltage of a power end of the resonant converter; setting a voltage command, and calculating a primary side current according to the direct current voltage and the voltage command; and summing the primary side current and the compensation amount as a second error amount, and adjusting the direct current voltage according to the second error amount.

14. A resonant converter applying the parameter calculation method of claim 1, and the resonant converter comprises: a primary side circuit coupled to a direct current power supply; a resonant circuit comprising a plurality of transformer sets, primary side windings of the plurality of transformer sets being coupled to the primary side circuit; and a secondary side circuit comprising a plurality of secondary side switching circuit sets, and the plurality of secondary side switching circuit sets each comprising a first secondary side bridge arm and a second secondary side bridge arm in parallel wherein the first secondary side bridge arm and the second secondary side bridge arm of the plurality of secondary side switching circuit sets are coupled to two ends of a secondary side winding of the plurality of transformer sets, respectively, and the plurality of secondary side switching circuit sets are coupled in parallel to a device.

15. The resonant converter of claim 14, wherein the primary side circuit comprises: a plurality of switching circuit sets each comprising a first switching bridge arm and a second switching bridge arm in parallel, and the first switching bridge arm and the second switching bridge arm of the plurality of switching circuit sets are coupled to two ends of a primary side winding of the plurality of transformer sets, respectively.

16. The resonant converter of claim 15, wherein the plurality of switching circuit sets are coupled in parallel to the direct current power supply or coupled in series to the direct current power supply.

17. The resonant converter of claim 15, wherein the plurality of switching circuit sets are coupled in parallel to the direct current power supply, and a resonant inductor of the resonant circuit forms a common core structure.

18. The resonant converter of claim 15, wherein the primary side circuit comprises: a switching circuit comprising a first switching leg and a second switching leg in parallel, and a plurality of primary side winding sets of transformers coupled in series to form a primary side winding string; wherein the first switching leg and the second switching leg are coupled to two ends of the primary side winding string, respectively.

19. The resonant converter of claim 18, wherein the switching circuit further comprises a third switching leg in parallel, and a center tap terminal formed between two primary side winding sets of the plurality of transformers, the third switching leg coupled to the center tap terminal.

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