A fault-tolerant control method for six-phase permanent magnet synchronous motor based on position sensorless

Abstract

The present invention discloses a fault-tolerant control method for a six-phase permanent magnet synchronous motor based on a position sensorless motor, and belongs to the technical field of motor control. The present invention aims to solve the problem that when a single-phase fault occurs in the motor, conventional vector control causes the motor current to be distorted, the harmonic current to increase, and the torque to fluctuate. When any phase of the six-phase permanent magnet synchronous motor fails, the present invention establishes a general single-phase fault reduction-order transformation matrix C 5s , and establish a mathematical model after the motor open circuit fault, so that the motor transitions to a normal state. The present invention can also make the motor run normally when a single-phase fault occurs in the motor.

Description

Technical Field

[0001] The present invention relates to the field of motor fault-tolerant control, and in particular to a position sensor-free six-phase permanent magnet synchronous motor fault-tolerant control method. Background Art

[0002] Compared with three-phase motors, multi-phase motors have the characteristics of small torque pulsation and strong fault tolerance, which makes multi-phase motors widely used in many occasions with high reliability requirements, especially in rail transportation, vehicles, wind power generation, aerospace and other fields. Considering the comprehensive processing difficulty, fault tolerance and system complexity, symmetrical six-phase permanent magnet synchronous motors are attracting widespread attention. In order to detect the rotor position information, the multi-phase PMSM control system needs to install a position sensor, but it will increase the system cost and reduce the system reliability, limiting its application occasions. Therefore, in order to improve the system reliability, position sensorless detection technology is a feasible technical solution.

[0003] At present, there are two main fault-tolerant methods: hardware redundancy and software fault-tolerant control. Hardware redundancy is to increase system redundancy by repeatedly configuring hardware resources. This method improves the reliability of the system, but with the increase in the number of motor phases, the system cost will increase significantly, and it is easy to increase the system volume and weight. Therefore, software fault-tolerant control is the current development trend to improve the reliability of the drive system. When a multi-phase motor fails, the system switches to the fault state through a fault-tolerant algorithm and can meet certain performance indicators; multi-phase PMSM position-free detection technology is generally divided into two categories: one is a high-frequency injection method suitable for low-speed and zero-speed areas, which uses the salient pole effect of the motor rotor to estimate the rotor position and speed. The estimation accuracy is independent of the speed and is insensitive to changes in motor parameters, but this method requires the PMSM to have a certain salient pole. In addition, the amplitude of the injected high-frequency signal must be grasped, otherwise it will bring electromagnetic noise. The other is the estimation of the rotor position and speed suitable for medium and high speeds, which depends on the motor back electromotive force, such as the extended Kalman filter method, the model reference method, and the sliding mode observer method. Summary of the invention

[0004] In order to solve the above problems, the present invention provides a fault-tolerant control method for a six-phase permanent magnet synchronous motor based on a position sensorless system, which realizes the transition of the permanent magnet synchronous motor to a normal state when any phase of the motor fails according to a single-phase fault reduced-order transformation matrix.

[0005] The present invention discloses a position sensor-free six-phase permanent magnet synchronous motor fault-tolerant control method, comprising:

[0006] When any phase of the six-phase permanent magnet synchronous motor fails, according to the general single-phase fault reduction-order transformation matrix C 5s To make the motor transition to the normal state, the general single-phase fault reduction-order transformation matrix C5s for:

[0007]

[0008] Among them, [] indicates rounding, k1 = 0, 1, 2, 3, 4, respectively indicating that the E phase, B phase, D phase, F phase (A phase), and C phase winding phase are broken;

[0009] According to the general single-phase fault reduced-order transformation matrix C 5s Establish a mathematical model of the motor after an open circuit fault;

[0010] Under the mathematical model after the startup open circuit fault, the rotor position angle is estimated.

[0011] Preferably, the method of the mathematical model after the motor open circuit fault includes:

[0012] Obtain the transformation matrix T5 from the natural coordinate system to the dq coordinate system according to the following formula:

[0013] T5=P 5s C 5s ;

[0014] The mathematical model in the dq coordinate system when one phase is open is:

[0015]

[0016] Among them, R dq =T5R s T5 -1 =R S I2,I dq =T5I s =[i d i q ] T ψ dq =L dq I dq +ψ f F dq ;Ω is the velocity matrix,

[0017] L dq Inductance matrix, L dq =T5L S T5 -1 ;

[0018] The stator voltage equation in the dq coordinate system is obtained as follows:

[0019]

[0020] Preferably, when any phase of the six-phase permanent magnet synchronous motor fails, the rotor angle is estimated by:

[0021] The stator voltage equation in the dq coordinate system is transformed to obtain a voltage equation in the αβ coordinate system;

[0022] Establish the current state space equation of the system:

[0023]

[0024] In the formula, z α 、z β To reconstruct the control law required for the system; is the current observation value in the coordinate system;

[0025] An improved sliding mode observer is established, and the back electromotive force is estimated by the improved sliding mode observer as follows:

[0026]

[0027] in,

[0028] The angle of the rotor is obtained by the inverse tangent function method and speed

[0029]

[0030]

[0031] Preferably, the improved sliding mode observer is:

[0032]

[0033] Wherein, F() represents the FVT function.

[0034] Preferably, the FVT function is:

[0035]

[0036] in, K r is the resonant gain, ω c is the cut-off frequency, f is the frequency of the stator current, K p is the proportional gain.

[0037] Preferably, the frequency f of the stator current is:

[0038]

[0039] in,

[0040] As described above, the present invention provides a position sensor-free six-phase permanent magnet synchronous motor fault-tolerant control method, which has the following effects:

[0041] 1. The present invention provides a transformation matrix for decoupling the αβ and z1z2 subspaces of a common neutral point symmetrical six-phase permanent magnet synchronous motor, which effectively reduces the torque pulsation after phase loss.

[0042] 2. Taking into account the complexity of motor faults, the present invention obtains a general single-phase fault reduction-order transformation matrix expression. The general transformation matrix and the mathematical model after the motor open circuit fault are established to convert the motor voltage equation under the dq axis into the voltage equation under the αβ axis; so that the u required in the subsequent improvement of the sliding film observer algorithm α and u β Estimation of the motor back EMF ensures smooth transition to normal state when any phase of the motor has an open circuit fault, and plays a key role in reconstructing the motor model after an open circuit fault.

[0043] 3. The present invention extends the concept of traditional three-phase position sensorless control technology to the motor mathematical model in multi-dimensional space, and realizes the stable operation of the six-phase permanent magnet synchronous motor based on position sensorless control.

[0044] 4. The present invention adopts an improved sliding mode observer based on the FVT function, and adopts the FVT function to replace the switching function of the traditional sliding mode, so as to resist the influence of torque pulsation and harmonic components on the back electromotive force value estimated by the traditional SMO, and improve the estimation accuracy of the back electromotive force.

[0045] 5. The present invention is suitable for applications such as aerospace, electric vehicles, etc. that have high requirements on the reliability and dynamic performance of motors. BRIEF DESCRIPTION OF THE DRAWINGS

[0046] Figure 1 This is a topological structure diagram of a symmetrical six-phase permanent magnet synchronous motor system according to a specific embodiment of the present invention;

[0047] Figure 2 is the conversion between the natural coordinate system and the αβ coordinate system of a specific embodiment of the present invention;

[0048] Figure 3 This is a schematic diagram of the value of k1 when any phase is disconnected in a specific embodiment of the present invention;

[0049] Figure 4 is a block diagram of stator current frequency detection based on PLL according to a specific embodiment of the present invention;

[0050] Figure 5 is a block diagram of the FVT structure of a specific embodiment of the present invention;

[0051] Figure 6This is a comparison diagram of the 800r / min speed waveform of a specific embodiment of the present invention;

[0052] Figure 7 This is a waveform diagram of the 800r / min speed error of a specific embodiment of the present invention;

[0053] Figure 8 This is a comparative waveform diagram of the rotor position angle at 800 r / min according to a specific embodiment of the present invention;

[0054] Fig. 9 This is a waveform diagram of the rotor position angle error at 800 r / min according to a specific embodiment of the present invention;

[0055] Fig.10 It is a main flow chart of a system of a specific embodiment of the present invention;

[0056] Fig.11 It is a main program flow chart of a specific embodiment of the present invention;

[0057] Fig.12 is a flow chart of an interruption procedure of a specific embodiment of the present invention;

[0058] Fig.13 is a DSP power supply circuit diagram of a specific embodiment of the present invention;

[0059] Fig.14 is a voltage sampling circuit diagram of a specific embodiment of the present invention;

[0060] Fig.15 is an AC current sampling circuit diagram of a specific embodiment of the present invention;

[0061] Fig.16 is a DC bias circuit diagram of a specific embodiment of the present invention;

[0062] Fig.17 is an overcurrent protection circuit diagram of a specific embodiment of the present invention;

[0063] Fig.18 It is a driving circuit diagram of 2SD315AI according to a specific embodiment of the present invention. DETAILED DESCRIPTION

[0064] The following describes the embodiments of the present invention by specific examples, and those skilled in the art can easily understand other advantages and effects of the present invention from the contents disclosed in this specification. The present invention can also be implemented or applied through other different specific embodiments, and the details in this specification can also be modified or changed in various ways based on different viewpoints and applications without departing from the spirit of the present invention. It should be noted that the following embodiments and features in the embodiments can be combined with each other without conflict.

[0065] It should be noted that the illustrations provided in the following embodiments are only used to illustrate the basic concept of the present invention in a schematic manner, and thus the illustrations only show components related to the present invention rather than being drawn according to the number, shape and size of components in actual implementation. In actual implementation, the type, quantity and proportion of each component may be changed arbitrarily, and the component layout may also be more complicated.

[0066] In a specific embodiment, a position sensorless six-phase permanent magnet synchronous motor fault-tolerant control method is provided, which specifically includes:

[0067] In this embodiment, when the E phase of the permanent magnet synchronous motor fails, the ClARKE transformation matrix C 6s The row vectors in are no longer orthogonal, C 6s It is expressed as follows:

[0068]

[0069] Therefore, the transformation matrix needs to be reconstructed as follows:

[0070]

[0071] According to the spatial distribution of windings after the fault, α and β can be expressed as:

[0072]

[0073]

[0074] Since the motor has a common neutral point, the sum of the remaining five-phase stator currents is still zero, and we can get:

[0075] z3=[1 1 1 1 1] T (5)

[0076] To make C 5s The row components in are mutually orthogonal, and z1, z2, z3, α and β need to satisfy the following constraints, namely:

[0077]

[0078] In order to make α in equation (3) and equation (5) orthogonal to z3, α in equation (3) is modified as follows:

[0079] α=[0.8 0.3 -0.7 -0.7 0.3] T (7)

[0080] At this time, α, β and z3 are orthogonal to each other and do not affect the electromechanical energy conversion.

[0081] According to formula (6) and formula (7), we can get:

[0082] z1=[-1.3331 -0.333 -0.333 1] T (8)

[0083] z2=[0 0.866 -0.866 0.866 -0.866] T (9)

[0084] We can get:

[0085]

[0086] When other phases of the motor are disconnected, it is similar to the disconnection of phase E. It is only necessary to change the motor rotor position θ.

[0087] In this embodiment, let k1=0, 1, 2, 3, 4 respectively represent the phase failure of the windings of phase E, phase B, phase D, phase F (phase A), and phase C. Figure 3 shown.

[0088] When any phase is disconnected, the corrected electrical angle after the phase disconnection is:

[0089] θ=θ * -k1·60° (11)

[0090] In the formula, θ * is the motor feedback angle with the α axis as the reference, where the electrical angle θ is obtained by estimating the angle without a position sensor.

[0091] When a phase open circuit fault occurs, the transformation matrix C 5s The remaining five-dimensional vectors in are corrected, that is, the α, β, z1, z2, and z3 planes are corrected, where the fundamental wave vectors α and β can be obtained by the CLARK transformation matrix C of the normal case of the symmetrical six-phase motor. 6s It is obtained that the harmonic plane vectors z1 and z2 can be calculated from the fundamental wave vectors α and β; the zero-sequence plane vector z3 is obtained based on the condition that the sum of the phase currents is zero.

[0092] This embodiment is described by taking the E phase fault as an example. When a phase failure occurs in other phases, the order of the column vectors in the CLARKE transformation matrix can be changed.

[0093] The third and fourth row vectors in formula (10) represent the harmonic planes z1 and z2, respectively. In order to facilitate the subsequent arrangement of the general transformation matrix, according to the principle of equal amplitude transformation, one third of the z1 and z2 vectors are extracted as a whole. After extraction, the vector z1 is:

[0094]

[0095] The fourth row vector z2 is:

[0096]

[0097] Since the motor windings have a common neutral point, the plane vector z3 needs to satisfy [1 1 1 1 1]. At the same time, in order to meet the requirements that z1, z2, z3, α and β are mutually orthogonal, the α vector needs to be adjusted accordingly. According to the matrix operation rules, a coefficient m should be subtracted from α to obtain the corrected α, which can be expressed as follows:

[0098]

[0099] After calculation, it is found that αT.z3=1, which means 5m=1, m is 0.2, and finally the corrected α needs to be subtracted from 0.5 on the five elements of this row;

[0100] In order to further obtain the expression result of the general transformation matrix, this embodiment introduces sin and cos functions to represent the α and β vectors. When the remaining phases of the motor have an open circuit fault, it is only necessary to use the broken E phase as a reference and rotate the coordinate system about the origin in a specified direction by a certain angle to obtain the transformation matrix expression result when the remaining phases have an open circuit fault. There are two methods to achieve the above transformation in the transformation matrix: using formula (11) to achieve a corresponding change in the motor feedback angle, or adjusting the order of the column elements of the transformation matrix. Here, by introducing the α, β, z1 and z2 vectors in the transformation matrix Figure 3 The value of k1 in the transformation matrix is ​​obtained by bringing in the corresponding k1 values ​​of different phase open circuit faults to adjust the order of the transformation matrix column elements, and finally obtaining the universal transformation matrix C 5s :

[0101]

[0102] In the formula, [x] represents the integer of x, and mod5 represents the modulus of the element with respect to 5. From formula (14), we can get the transformation matrix C when any phase is disconnected: 5s , that is, formula (10) represents the transformation matrix when the motor has an E-phase open circuit fault, which is a special case of formula (14).

[0103] Rotation transformation matrix P 5s for:

[0104]

[0105] Where I3 represents the three-dimensional unit matrix.

[0106] The transformation matrix T5 from the natural coordinate system to the dq coordinate system is:

[0107] T5=P 5s C 5s (16)

[0108] Substituting T5 in equation (15) into the voltage equation, flux equation and torque equation of the motor, we can obtain the mathematical model in the dq coordinate system when one phase is open:

[0109]

[0110] Consider only the dq subplane and define: U dq =T5U s =[u d u q ] T ; R dq =T5R s T5 -1 =R S I2;I dq =T5I s =[i d i q ] T ψ dq =T5ψ s

[0111] ψ dq =L dq I dq +ψ f F dq (18)

[0112] U s The voltage equation representing the normal state of the motor is:

[0113]

[0114] ψ s The flux equation representing the normal state of the motor is:

[0115] ψ s =L s I s +ψ f F s (θ) (20)

[0116] in:

[0117] U s =[U A U D U B U E U C U F ] T ,

[0118] ψ s =[ψ A ψ D ψB ψ E ψ C ψ F ] T ,

[0119] I s =[I A I D I B I E I C I F ] T ,

[0120] F s (θ)=[cosθ cos(θ-60°) cos(θ-120°) cos(θ-180°) cos(θ-240°) cos(θ-300°)] T .

[0121] R s is the stator winding resistance; F dq is the flux coefficient matrix under the dq axis of the motor unidirectional open circuit fault, which is represented by F s (θ) is obtained by removing the corresponding phase-out elements in the matrix and performing T5 transformation.

[0122] Ω is the velocity matrix:

[0123]

[0124] Inductor Matrix:

[0125] L dq =T5L S T5 -1 (twenty two)

[0126] Finally, the stator voltage equation in the dq coordinate system is obtained as:

[0127]

[0128] When a symmetrical six-phase motor full-order model is used to observe the state of the motor system, many parameters need to be determined; the increase in the order of the system state observer brings a huge computational burden to the DSP, and more uncertain parameters will also affect the performance of the state observer. The electromagnetic torque of the six-phase motor is mainly determined by the fundamental stator magnetomotive force. Since the dq-axis inductance under the third harmonic is relatively small compared to the fundamental component, when establishing the system state observer, the fundamental and harmonic space mutual inductance electromotive force can be ignored, or the influence of the dq-axis inductance under the third harmonic can be regarded as a disturbance, and only the fundamental space voltage equation is retained; since the six-phase motor magnetic field oriented control requires the voltage and current to be converted into DC quantities, it is more suitable to observe the internal state of the motor in the αβ coordinate system without a position sensor. In order to make the αβ coordinate system inductance matrix no longer coupled with the rotor position information, the motor model is transformed as follows:

[0129]

[0130] The extended back EMF is defined as follows:

[0131] E αβ =(L d -L q )(ωi d -i q )+ωψ m (25)

[0132] Where E αβ To expand the back EMF, ψ m is the effective rotor flux of the six-phase motor when considering the fault mode.

[0133] Then, the voltage equation shown in equation (21) is transformed into the αβ coordinate system, and the sliding mode stator current variable frequency tracker designed for the permanent magnet synchronous motor is used to independently estimate the electromotive force. The specific process is as follows:

[0134]

[0135] Where: e α 、e β is the component of the back electromotive force on the αβ axis, which can be expressed as:

[0136]

[0137] Formula (23) is expressed as the current state space equation:

[0138]

[0139] Reconstruct the current state space equation of the system:

[0140]

[0141] Where z α 、z β To reconstruct the control law required for the system; is the observed current value in the coordinate system.

[0142] In z α and z β The motion trajectory of the system to be observed under the control law tends to the actual current motion trajectory. When the two trajectories coincide, the control law will include the back electromotive force e α 、e β As well as the modeling uncertainty part, filtering out the harmonic interference caused by the uncertainty component can achieve the estimation of the fundamental back EMF.

[0143] Subtracting equation (25) from equation (26) yields the current error state space equation:

[0144]

[0145] In the formula is the current observation error value in the coordinate system.

[0146] According to sliding mode variable structure control theory, the sliding surface is defined as

[0147] When the sliding surface function s α 、s β When it is 0, it means that the motion trajectory of the system to be observed coincides with the motion trajectory of the actual system, that is, the back electromotive force observation is realized. At this time, the control law z α 、z β It can be determined by the Lyapunov function method.

[0148] Define the Lyapunov function:

[0149]

[0150] Derivative of the Lyapunov function:

[0151]

[0152] In order to make the derivative of V less than 0, the control law is designed as follows:

[0153]

[0154] In the formula, K s is the sliding mode gain, and its value range must satisfy the stability, existence and reachability conditions of the sliding mode observer.

[0155] After the system disturbance occurs, the current error can converge to zero in a finite time, that is, the observed system can return to the motion trajectory of the actual system.

[0156] The traditional sliding mode switching function is the switching function, and the current sliding mode observer in the αβ coordinate system is:

[0157]

[0158] From equations (33) and (34), the current deviation equation can be obtained as follows:

[0159]

[0160] Since the discrete switching characteristics of the sign function in the traditional sliding mode observer easily lead to high-frequency jitter in the estimated back electromotive force, this embodiment introduces the FVT function to replace the original switching function sign() to form an improved sliding mode observer to track the fundamental wave of the stator current.

[0161]

[0162] Proportional resonant control (PR) can accurately track a specific frequency, and the stator current error can converge at a specific frequency. However, during actual motor operation, the frequency of the stator current varies with speed. When the frequency changes, the traditional PR controller cannot accurately track the AC signal. Therefore, the FVT function can be implemented by modifying the PR as follows to adapt to the changing frequency, Figure 4 FIG. 1 is a block diagram of a stator current frequency detection based on a PLL, where f is the frequency of the stator current obtained by a phase-locked loop (PLL).

[0163]

[0164] ω e =2πf (38)

[0165]

[0166] Where, f is the frequency of the stator current, K p is the proportional gain, ω e is the angular frequency.

[0167] make The FVT function can be expressed as:

[0168]

[0169] Among them, K r is the resonant gain, ω c is the cutoff frequency.

[0170] To accommodate the changing frequency, the transfer function (40) is transformed into the Z domain:

[0171]

[0172] have to:

[0173]

[0174] Define the state variables:

[0175]

[0176] Deriving the derivative of (43) yields the following state-space equation:

[0177]

[0178] FVT can be obtained from formula (44):

[0179]

[0180] Figure 5 The structural diagram of FVT is given.

[0181] When the moving point converges to the sliding surface, the back electromotive force estimated by the sliding mode observer is equivalent to:

[0182]

[0183] The angle and speed of the rotor can be obtained by rearranging equation (46) using the inverse tangent function:

[0184]

[0185]

[0186] From this, we can get the prediction results of the sliding film observer structure under the motor open circuit fault. It can be seen that the dq axis inductance has a great influence on the position sensorless control algorithm based on the back electromotive force method under the condition of motor fault. For the symmetrical six-phase permanent magnet synchronous motor, the influence of inductance on magnetic saturation enhancement is small, and the dq axis inductance is less affected by the motor open circuit fault. Therefore, the symmetrical six-phase permanent magnet synchronous motor has excellent sensorless performance and high-speed operation capability under fault conditions.

[0187] In order to verify this embodiment, a test bench is built in this embodiment. Fig.10As shown in the main flow chart of the system, the hardware circuit of the control system of this embodiment includes a control circuit, a voltage and current sampling circuit, an overcurrent protection circuit, an IGBT drive circuit and a six-phase PMSM. The power side topology of the six-phase PMSM control system is an AC-DC-AC topology, that is, the 220V industrial frequency AC power is first passed through a rectifier bridge to obtain a DC bus voltage, and then the DC bus voltage output by the rectifier bridge is filtered and stabilized by a non-polarity filter capacitor and sent to a six-phase voltage inverter. This embodiment uses a six-phase voltage inverter with an anti-parallel diode structure to achieve inverter processing. The phase voltage and phase current of the motor are processed by the voltage and current sampling circuit and transmitted to the DSP control circuit. The voltage and current analog signals obtained by the sampling circuit are digitized by the ADC conversion unit of the DSP. Then, the motor rotor position information is estimated by the SMO algorithm on the DSP chip to achieve a double closed loop of speed and current. The current closed loop is output to the SVPWM modulation algorithm to obtain a drive signal for the drive circuit; SVPWM modulation is achieved by controlling the on and off of the power devices in the inverter, thereby controlling the operation of the six-phase PMSM.

[0188] The main program of this embodiment mainly completes system initialization and interruption, and the interruption program includes AD sampling, fault diagnosis, FVT and ESO calculation, speed loop and current loop, etc. Fig.11 As shown in the figure, in the main program flow of the system, when the system starts running, all interrupts are turned off and the system is initialized, completing the initial settings of each unit used in the program. After the initialization is completed, the interrupt is turned on, the timer is started, and the interrupt is waited. The flowchart of the interrupt subroutine is as follows Fig.12 It is used to complete the sampling of phase voltage and current, estimate the rotor position and speed, speed loop PI regulation, current loop PI regulation, coordinate transformation, and output the control signal to the power module through the SVPWM algorithm to control the motor operation.

[0189] The hardware circuit of the control system of this embodiment is mainly composed of control circuit, voltage and current sampling circuit, overcurrent protection circuit, IGBT driving circuit, etc. DSP architecture is adopted. During operation, the DSP chip uses TMS320F28335 of TI company. The DSP chip is mainly responsible for the operation of processed sampling signals, command current extraction, current tracking control algorithm, and inductive control algorithm. The phase voltage and phase current of the motor sampled by the voltage and current sampling circuit are processed and transmitted to the DSP control circuit. The voltage and current analog signals obtained by the sampling circuit are processed by digital signal through the ADC conversion unit of the DSP. Then, the motor rotor position information is estimated by the SMO algorithm on the DSP chip to realize the double closed loop of speed and current. The received modulation signal data is compared with the modulation wave and the carrier to obtain a PWM signal with a dead zone. The PWM is amplified by the driving circuit to drive the power switch tubes in each phase inverter to work.

[0190] like Fig.13 The figure shows the DSP power supply circuit, which uses the TPS767D301 chip to power the DSP and outputs two stable DC voltages, 1.9V to supply the DSP core and 3.3V DC to supply the I / O port.

[0191] like Fig.14 The figure shows a voltage sampling circuit. The phase voltage and DC bus voltage are collected through an isolation amplifier. The principle is to use a voltage divider resistor on the high-voltage side to obtain a small voltage, and isolate the voltage in a differential manner through an isolation amplifier. The output side of the isolation amplifier is also output in a differential manner, and then passes through an operational amplifier to convert the differential signal into a single-ended signal and send it to the DSP.

[0192] like Fig.15 The figure shows the AC current sampling circuit, which collects the AC current of the motor through the current Hall sensor. The model of the current Hall sensor is HA2020 produced by YHDC. The maximum sampling current value is 100A, the power supply is ±15V, and the transformation ratio is 2000:1.

[0193] like Fig.16 The figure shows a DC bias circuit, which limits the voltage amplitude of the current sampling signal to between 0-3V through the bias circuit. The bias voltage of 1.65V is generated by the voltage divider of resistors R37 and R39. Resistor R39 and capacitor C10 form a first-order RC filter circuit. The Schottky diode D2 constitutes a clamping circuit for the sampling voltage to prevent the voltage entering the DSP from being too large, which may damage the chip.

[0194] The overcurrent protection circuit in this embodiment is as follows: Fig.17As shown, the main function is to prevent the phase current of the motor from exceeding the rated value of the IGBT switch tube, causing the IGBT switch tube to burn out. The circuit is composed of a comparator to form a voltage comparator. The sampled current signal passes through the bias circuit and is compared with the limit value. If the voltage value of the biased current signal is higher than 4.5V or lower than 0.6V, the control system blocks the PWM output. The selection of the limit value is related to the motor rating and the gain of the sampling circuit.

[0195] The driving circuit is used to amplify the low-level and low-power control signal output by the DSP so that it can drive the power switch tube. Fig.18 As shown, the driving circuit of this embodiment uses the driving module model 2SD315AI launched by the Swiss CONCEPT company, which has two working modes: direct mode and half-bridge mode. The 8-pin MOD of the driver is short-circuited with VDD, and it works in direct mode. At this time, channels A and B have no relationship, and the two channels work independently. RC1 and RC2 are short-circuited with GND, and the state output SO1 / SO2 also works independently. The 8-pin MOD of the driver is short-circuited with GND, and it works in half-bridge mode. A dead time is generated between the two channels. The dead time is adjusted by the RC network between pins 5 and 7. At this time, INB is connected to a high level to enable, and INA is the total input terminal of the two signals.

[0196] The simulation analysis of the symmetrical six-phase PMSM non-sensing operation under a single-phase fault in this embodiment obtains the following results:

[0197] The motor is started with a load torque of 5 N·m and the given speed is set to 800 r / min. Figure 6 and Figure 8 The speed and rotor position angle waveforms are shown respectively. The red solid line in the figure represents the estimated speed and position of the rotor, and the blue solid line represents the actual speed and position measured by the motor. Figure 7 and Fig. 9 The waveforms of the speed error and rotor position angle error are shown respectively. It can be seen that the speed error is within ±1r / min, and the rotor position angle error is very small, indicating that the observer can accurately estimate the speed and track the rotor position angle.

[0198] The above embodiments are merely illustrative of the principles and effects of the present invention, and are not intended to limit the present invention. Anyone familiar with the art may modify or alter the above embodiments without departing from the spirit and scope of the present invention. Therefore, all equivalent modifications or alterations made by a person of ordinary skill in the art without departing from the spirit and technical concept disclosed by the present invention shall still be covered by the claims of the present invention.

Claims

1. A fault-tolerant control method for a six-phase permanent magnet synchronous motor based on a position sensorless system, characterized in that: include: When any phase of the six-phase permanent magnet synchronous motor fails, a general single-phase fault reduced-order transformation matrix C is established. 5s ,for: ； Wherein, [ ] indicates rounding, k1=0, 1, 2, 3, 4, respectively indicating that the E phase, B phase, D phase, F phase or A phase, C phase winding is broken; mod5 means the remainder of 5; According to the general single-phase fault reduced-order transformation matrix C 5s After that, a mathematical model after the motor open circuit fault is established, and the method for establishing the mathematical model after the motor open circuit fault includes: Obtain the transformation matrix T5 from the natural coordinate system to the dq coordinate system according to the following formula: ； in, is the rotation transformation matrix; The mathematical model in the dq coordinate system when one phase is open is: ； in, , R s is the stator winding resistance; ; ; is the velocity matrix, which is: ; Inductor matrix, ; F dq It is the flux coefficient matrix under the dq axis of the motor unidirectional open circuit fault; The stator voltage equation in the dq coordinate system is obtained as follows: ； Among them, F dq It is the flux coefficient matrix under the dq axis of the motor unidirectional open circuit fault; Under the mathematical model after the open circuit fault of the motor, the angle of the rotor is estimated, and the method for estimating the rotor angle includes: The stator voltage equation in the dq coordinate system is transformed to obtain a voltage equation in the αβ coordinate system; Establish the current state space equation of the system: ； In the formula, z α 、z β To reconstruct the control law required for the system; is the current observation value in the coordinate system; An improved sliding mode observer is established, and the back electromotive force is estimated by the improved sliding mode observer as follows: ； in, , is the current observation error value in the coordinate system, , ; F( ) represents FVT function; The angle of the rotor is obtained by the inverse tangent function method and speed : ； ； Among them, ψ m is the effective rotor flux of the six-phase motor when considering the fault mode.

2. According to the position sensorless six-phase permanent magnet synchronous motor fault-tolerant control method of claim 1, it is characterized in that: The improved sliding mode observer is: ； Where F( ) represents the FVT function, K s is the sliding mode gain.

3. According to the position sensorless six-phase permanent magnet synchronous motor fault-tolerant control method of claim 2, it is characterized in that: The FVT function is: ； in, , , K r is the resonant gain, ω c is the cut-off frequency, f is the frequency of the stator current, K p is the proportional gain.

4. According to the position sensorless six-phase permanent magnet synchronous motor fault-tolerant control method of claim 3, it is characterized in that: The frequency f of the stator current is: ； in, .

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