Current controller design method, current controller and permanent magnet synchronous motor

Abstract

The invention provides a permanent magnet synchronous motor current controller design method, a current controller and a permanent magnet synchronous motor. The permanent magnet synchronous motor current controller design method comprises the steps of 1, designing a model-free nonsingular terminal sliding mode current controller; 2, designing a quasi-resonance regulator; and step 3, a quasi-resonance compensation step: respectively connecting two ends of the d-axis model-free nonsingular terminal sliding mode current controller and the q-axis model-free nonsingular terminal sliding mode current controller in parallel with a quasi-resonance regulator. According to the current controller design method, a current tracking controller of a permanent magnet synchronous motor with a current tracking function is designed, and the two ends of a d-axis model-free nonsingular terminal sliding mode current controller and the two ends of a q-axis model-free nonsingular terminal sliding mode current controller are respectively connected with a quasi-resonance regulator in parallel. Therefore, the motor not only has good dynamic performance, but also can effectively reduce the current fluctuation, suppress the torque ripple and improve the steady-state performance.

Description

Technical Field

[0001] This invention belongs to the field of motor control technology, specifically, it relates to a design method for a current controller of a permanent magnet synchronous motor, a current controller, and a permanent magnet synchronous motor. Background Technology

[0002] Permanent magnet synchronous motors (PMSMs) achieve speed regulation through variable voltage and variable frequency technology. They are characterized by small size, low loss, high efficiency, and wide speed range, and have been widely used in recent decades. As the application fields continue to expand and the performance requirements become increasingly stringent, while the design of the PMSM's main structure is continuously optimized and innovated, its motor control technology has also developed rapidly.

[0003] In applications requiring high control performance of permanent magnet synchronous motors (PMSMs), smooth torque output capability is a crucial indicator of motor performance. However, harmonics generated by factors such as imperfect motor design, inverter nonlinearity, and current measurement errors can cause torque ripple, leading to decreased speed control accuracy. Currently, in industrial control, PI control is still the primary method for controlling PMSM drive systems. However, with the widespread application of PMSMs in various high-performance applications such as electric vehicles, robotic systems, and CNC machine tools in recent years, the traditional PI method is no longer sufficient to meet the requirements of high-performance control due to the nonlinear motion dynamics of PMSMs and their susceptibility to various sources of disturbance and uncertainty, including external disturbances such as load torque changes and friction torque, parameter uncertainties including changes in mechanical and electrical parameters, and unmodeled dynamics influenced by motor design, inverter nonlinearity, and current measurement errors.

[0004] In permanent magnet synchronous motor speed control systems, sliding mode control has become a research hotspot due to its good dynamic performance and strong robustness. Sliding mode control mainly includes the design of the sliding mode reaching law and the sliding surface. The sliding mode reaching law brings the system state to the designed sliding surface, but this state is difficult to maintain at zero error on the sliding surface, leading to unavoidable chattering. To alleviate this phenomenon, a popular method is to improve the design of the reaching law by replacing discontinuous sign functions with continuous functions, such as saturation functions, sigmoid functions, and hyperbolic tangent functions, to construct a continuous control law that suppresses chattering. Another method is to combine sliding mode control with a disturbance observer to reduce chattering. The disturbance observer estimates the total disturbance of the system in real time and compensates for it in the sliding mode control to mitigate its impact. Although sliding mode control has good current response dynamic performance and is robust to system parameter changes and external disturbances, it still cannot eliminate periodic disturbances. In vector control, if the dead time, cogging torque, magnetic flux harmonics, and current measurement are inaccurate, the dq-axis current obtained through coordinate transformation will contain harmonics and cannot be completely controlled as DC, thus causing current fluctuations and torque pulsations, which are unacceptable in high-performance motor control applications.

[0005] Commonly used linear control algorithms, such as proportional-integral (PI) control, are often used for speed control of permanent magnet synchronous motors (PMSMs) due to their simple design and implementation. However, PMSM drive systems are nonlinear and multivariable, with various internal disturbances (parameter variations and unmodeled dynamics) and external disturbances (load torque variations and friction torque, etc.). Although system disturbances can be suppressed through closed-loop control, these disturbances inevitably cause fluctuations in the PMSM speed response, thus affecting the motor's static and dynamic performance. Summary of the Invention

[0006] In order to solve the technical problems of current fluctuations, torque pulsation, and poor steady-state performance in existing current controllers, this invention proposes a current controller that can solve the above problems.

[0007] To solve the above-mentioned technical problems, the present invention adopts the following technical solution:

[0008] A current controller design method, comprising: Step 1: Design a model-free, non-singular terminal sliding mode current controller; Step 2, design the quasi-resonant modulator: ; Where, ω c ω is the cutoff frequency, k is the resonance coefficient, ω is the resonant frequency, and s is the complex frequency domain. Step 3, Quasi-resonant compensation step: Connect one of the aforementioned quasi-resonant regulators in parallel across the two ends of the d-axis modelless non-singular terminal sliding mode current controller and the q-axis modelless non-singular terminal sliding mode current controller, and output the compensation voltage to the control variable of the corresponding current controller.

[0009] This invention also proposes a current controller for a permanent magnet synchronous motor, which is designed using the aforementioned design method.

[0010] This invention also proposes a permanent magnet synchronous motor, which includes a current controller designed according to the aforementioned design method; The current controller is used to obtain the control variable based on the desired current and the control law of the control variable, and is used to control the power supply of the permanent magnet synchronous motor. The extended state observer is used to observe unknown disturbances to obtain an estimate of the lumped disturbance, and transmits the estimate of the lumped disturbance to the current controller for compensation of the control variables.

[0011] Compared with the prior art, the advantages and positive effects of the present invention are as follows: The current controller design method of the present invention designs a current tracking controller for a permanent magnet synchronous motor with current tracking function. By connecting a quasi-resonant regulator in parallel at both ends of the d-axis modelless non-singular terminal sliding mode current controller and the q-axis modelless non-singular terminal sliding mode current controller, the quasi-resonant regulator is used to amplify the current error at the resonant frequency and output a compensation voltage to the control variable of the corresponding current controller, so that the motor not only has good dynamic performance, but also effectively reduces current fluctuations, suppresses torque ripple, and improves steady-state performance.

[0012] By designing the current controller as a model-free, non-singular terminal sliding mode current controller, and by optimizing the sliding surface design and control law, the improvement lies in overcoming the singularity problem that may occur in traditional terminal sliding mode control, while also improving the system's convergence speed and anti-interference capability.

[0013] Other features and advantages of the present invention will become clearer after reading the detailed description of the embodiments of the present invention in conjunction with the accompanying drawings. Attached Figure Description

[0014] Figure 1 This is a system block diagram of an embodiment of a permanent magnet synchronous motor using the current controller proposed in this invention; Figure 2 These are simulation diagrams of the speed change of a permanent magnet synchronous motor under disturbance applied using existing PI control and the present scheme, respectively. Figure 3a This is a simulation diagram of the a-phase current when a disturbance is applied to a permanent magnet synchronous motor using existing PI control. Figure 3bThis is a simulation diagram of the a-phase current of a permanent magnet synchronous motor subjected to disturbance using this scheme; Figure 4a This is a simulation diagram of the fundamental frequency content when a disturbance is applied to a permanent magnet synchronous motor using existing PI control. Figure 4b This is a simulation diagram of the fundamental wave content of a permanent magnet synchronous motor subjected to disturbance using this scheme. Figure 5 This is a simulation diagram of torque change when a disturbance is applied to a permanent magnet synchronous motor using existing PI control. Detailed Implementation

[0015] The specific embodiments of the present invention will be described in further detail below with reference to the accompanying drawings.

[0016] To make the objectives, technical solutions, and advantages of the embodiments of the present invention clearer, the technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0017] Example 1, as Figure 1 The figure shown is a system block diagram of the permanent magnet synchronous motor in this embodiment.

[0018] Existing permanent magnet synchronous motor drive systems suffer from unmodeled dynamics caused by imperfect motor design (magnetic flux harmonics, cogging torque), inverter nonlinearity, and current measurement errors. These factors generate periodic harmonic disturbances. In the dq coordinate system, a 6th harmonic exists. This invention introduces a quasi-resonant regulator to compensate for the current harmonics generated by these periodic disturbances, suppressing the resulting torque ripple and thus improving the system's steady-state performance. The following detailed description uses specific embodiments to illustrate this solution.

[0019] This embodiment proposes a current controller design method, including:

[0020] Step 1: Design a model-free, non-singular terminal sliding mode current controller.

[0021] Step 2, design the quasi-resonant modulator: ; Where, ω c ω is the cutoff frequency, k is the resonance coefficient, ω is the resonant frequency, and s is the complex frequency domain.

[0022] Step 3, Quasi-resonant compensation step: Connect the two ends of the d-axis modelless non-singular terminal sliding mode current controller and the q-axis modelless non-singular terminal sliding mode current controller in parallel with one of the aforementioned quasi-resonant regulators. The quasi-resonant regulator is used to amplify the current error at the resonant frequency and output the compensation voltage to the control variable of the corresponding current controller.

[0023] Quasi-resonant regulators enhance the ability to compensate for interference at specific frequencies by introducing resonant characteristics, and are particularly suitable for handling periodic disturbances introduced by motor structure, driver or load.

[0024] The transfer function of an ideal resonant controller is: .

[0025] Obviously, the denominator is zero when s = jω, meaning the gain is infinite. However, in practical systems, due to limitations in the accuracy of component parameters and the accuracy of digital control systems, ideal resonant controllers are difficult to implement in both analog and digital systems. Furthermore, because the gain of an ideal resonant controller drops sharply outside the resonant frequency, it cannot effectively suppress higher harmonics when fluctuations in motor speed cause fluctuations in the actual motor frequency near the resonant frequency. Therefore, to improve the ideal resonant controller, a quasi-resonant controller structure as shown below is adopted, with the following transfer function: .

[0026] The current controller design method of this embodiment designs a current tracking controller for a permanent magnet synchronous motor with current tracking function. By connecting a quasi-resonant regulator in parallel at both ends of the d-axis modelless non-singular terminal sliding mode current controller and the q-axis modelless non-singular terminal sliding mode current controller, the quasi-resonant regulator is used to amplify the current error at the resonant frequency and output a compensation voltage to the control variable of the corresponding current controller. This enables the motor to not only have good dynamic performance, but also suppress the 6th harmonic. It can effectively suppress current fluctuations caused by imperfect motor design (magnetic flux harmonics, cogging torque), inverter nonlinearity factors, and current measurement errors, suppress torque pulsation, and improve the steady-state performance of the motor.

[0027] By designing the current controller as a model-free, non-singular terminal sliding mode current controller, and by optimizing the sliding surface design and control law, the improvement lies in overcoming the singularity problem that may occur in traditional terminal sliding mode control, while also improving the system's convergence speed and anti-interference capability.

[0028] In some embodiments, the non-singular terminal integral sliding surface of the model-free non-singular terminal sliding mode current controller is: ; Where e is the state variable, which is the difference between the desired stator current and the stator current, c is the sliding diaphragm gain, p and q are positive odd numbers, and 1 < p / q < 2.

[0029] The non-singular terminal integral sliding surface designed in this embodiment firstly solves the singularity problem in traditional terminal sliding mode control by designing nonlinear terms to ensure that the control law does not become infinite or unrealizable when the system state approaches the equilibrium point, thus improving the physical realizability of the control. Combining the characteristics of terminal sliding mode, the system state can quickly converge to the sliding surface and eventually reach the equilibrium point within a finite time. Compared with the asymptotic convergence of traditional linear sliding mode control, this significantly improves the response speed and steady-state accuracy, making it particularly suitable for high-speed, high-precision control scenarios. It has strong anti-interference capabilities against parameter changes and external disturbances. Through the design of nonlinear terms, the system can maintain stability and control accuracy even under uncertain environments. Secondly, by introducing an integral sliding surface, steady-state errors can be further eliminated, enabling the system to converge the error to zero within a finite time, making it particularly suitable for high-precision applications such as sensorless control.

[0030] Differentiating the above sliding surface, we get: .

[0031] In some embodiments, the reaching law of the model-free nonsingular terminal sliding mode current controller is: ; in, , The parameters are all positive real numbers, and the parameter α satisfies 0 < α < 1.

[0032] In some embodiments, the permanent magnet synchronous motor establishes a hyperlocal model of the permanent magnet synchronous motor current loop, and the hyperlocal model is as follows: ; Where i is the stator current, which is the output variable. Let u represent the stator voltage, and F represent the lumped disturbance of the permanent magnet synchronous motor, which is the control variable.

[0033] In some specific embodiments, the output variables of the hyperlocal model include the stator current on the d-axis and the stator current on the q-axis; the control variables include the stator voltage on the d-axis and the stator voltage on the q-axis; and the unknown disturbance is a lumped disturbance that includes the influence of motor parameters, the uncertainty of the hyperlocal model, and external unknown disturbances.

[0034] The mathematical model of a permanent magnet synchronous motor in the dq coordinate system is as follows: .

[0035] Among them, Rs L is the stator resistance. d and L q These are the stator inductances along the d-axis and q-axis, respectively, n p For extreme logarithms, T e Where J is the electromagnetic torque, J is the moment of inertia, ω is the mechanical angular velocity, and B is the coefficient of friction; φ is the magnetic flux of the permanent magnet, and f is the magnetic flux of the permanent magnet. d f q f ω Disturbances caused by changes in model parameters and external loads.

[0036] f d f q f ω It can be defined as: .

[0037] Wherein, △R s =R st -R s , △L d =L dt -L d , △L q =L qt -L q △B=B t -B, △φ=φ t -φ,△J= J t -J represents the parameter variation value. R st L dt L qt B t φ t J t τ represents the actual parameters during motor operation. L This represents the external load torque.

[0038] According to the hyperlocal model theory, the first and second equations above can be expressed as: .

[0039] According to the model-free control principle, based on the system's input and output, a hyperlocal model can be used to replace some nonlinear, complex, and variable systems. A first-order hyperlocal model of a single-input, single-output system is represented as: .

[0040] In the formula, y and u are the output variable and control variable of the system, respectively; α is a constant; and F represents the unknown quantity of the system.

[0041] To reduce the dependence of the speed controller and current controller on the permanent magnet synchronous motor system model, the hyperlocal models of the speed loop and current loop of the permanent magnet synchronous motor are designed as follows: .

[0042] in, , Fd, Fq, and Fω are the voltage gains of the stator on the q-axis and d-axis, respectively, and the lumped disturbances on the d-axis, q-axis, and speed caused by motor parameters, model uncertainties, and external unknown disturbances, respectively. and These represent the q-axis and d-axis stator voltages, respectively, and are the control variables for the output.

[0043] This scheme eliminates the dependence on the system's motor parameters by establishing a hyperlocal model of the permanent magnet synchronous motor system that does not consider the motor parameters.

[0044] In some embodiments, the state variable of the q-axis and the state variables of the d-axis They are respectively: .

[0045] and These are the desired currents along the q-axis and the d-axis, respectively. and These are the q-axis stator current and the d-axis stator current, respectively.

[0046] By taking the derivatives of each state variable and combining them with the hyperlocal model, a mathematical relationship is established between the derivatives of the state variables and the control variables and lumped disturbances of the permanent magnet synchronous motor.

[0047] Differentiating the state variable equations yields: .

[0048] By combining the derivatives of each state variable with the hyperlocal model, a model-free nonsingular terminal sliding mode control law can be obtained, which serves as the control law for a model-free nonsingular terminal sliding mode current controller: ; .

[0049] , These are the estimated values ​​of the q-axis lumped disturbance and the d-axis lumped disturbance, respectively. , These represent the voltage gains along the q-axis and d-axis of the stator to be designed, respectively. , These are the state variables of the stator's q-axis and d-axis, respectively, to be designed.

[0050] The control performance of existing model-free control theories is related to the estimation accuracy of the unknown parts in the hyperlocal model of the system, and the estimation accuracy of the observer will directly affect the control performance of the system.

[0051] To address the issue of estimation accuracy, in some embodiments, the permanent magnet synchronous motor is also equipped with an extended state observer to output an estimate of the q-axis lumped disturbance. and the estimated value of the d-axis lumped disturbance To the modelless non-singular terminal sliding mode current controller.

[0052] In some embodiments, the extended state observer is: .

[0053] The observed value of the stator current. This is an estimate of the lumped disturbance. It is a nonlinear factor. β1 and β2 are the gain of the observer parameters to be designed, and ε is the observation error.

[0054] .

[0055] This scheme designs an extended state observer to estimate unknown disturbances in the system. This observer features high estimation accuracy and fast response speed, effectively improving the system's anti-interference capability. Simultaneously, the estimated disturbance is used for feedforward compensation, further enhancing the system's control accuracy.

[0056] In some embodiments, the design method of the model-free non-singular terminal sliding mode current controller also includes setting a PI control module. The real-time angular velocity is obtained; the desired current along the q-axis is obtained by the PI control module based on the reference angular velocity and the real-time angular velocity; the reference angular velocity is obtained by setting.

[0057] In some specific embodiments, the real-time angular velocity is obtained by an encoder located within the permanent magnet synchronous motor.

[0058] The simulation verification steps in this embodiment include: This step verifies the superiority of the PMSM model-free current controller based on the quasi-resonant sliding mode control strategy using MATLAB / Simulink. To better verify the speed tracking performance and anti-interference capability of the control strategy designed in this invention, it is compared with a PI-based controller. Figure 2As shown, a 2μs dead time was added to the simulation. A step load of 10 was suddenly applied to the motor at 0.2s. Due to harmonics generated by factors such as imperfect motor design, inverter nonlinearity, and current measurement errors, torque pulsation occurred, leading to a decrease in speed control accuracy. From... Figure 2 As can be seen, after the motor is in steady-state operation, the speed fluctuation of the motor using PI control is relatively large, while the speed fluctuation of the motor using quasi-resonant regulator and model-free sliding mode current control strategy is significantly smaller, and it has better anti-interference performance.

[0059] By comparison Figure 3a , Figure 3b The figures show the a-phase current waveforms of the PI controller control system and the sliding diaphragm control (SMC) + quasi-resonant regulation (PR) scheme after the application of a step load. Clearly, the a-phase current deformation of this scheme is smaller, smoother, and more stable. This is because the designed current controller is model-free and unaffected by unmodeled dynamics and model parameters, thus ensuring better steady-state performance of the motor.

[0060] On the other hand, through comparison Figure 4a , Figure 4b The figures show the percentage of harmonic content in the fundamental current of the PI controller after a step load is applied, and the percentage of harmonic content in the fundamental current of this scheme. Clearly, the total harmonic content of this scheme is a lower percentage of the fundamental current, indicating superior harmonic rejection performance.

[0061] The addition of a quasi-resonant term results in significantly lower harmonics in the motor current, leading to higher steady-state accuracy.

[0062] Figure 5 The diagram shows the control system of the PI controller after a step load is applied, and the torque change waveform of this scheme. It can be seen that the torque output of this scheme is more stable.

[0063] Example 2 presents a current controller for a permanent magnet synchronous motor, designed using the method described in Example 1. The design method is described in Example 1 and will not be repeated here.

[0064] Example 3: This example proposes a permanent magnet synchronous motor, including a current controller designed according to the design method described in Example 1.

[0065] The current controller is used to obtain the control variable based on the desired current and the control law of the control variable, and is used to control the power supply of the permanent magnet synchronous motor.

[0066] The extended state observer is used to observe unknown disturbances to obtain an estimate of the lumped disturbance, and transmits the estimate of the lumped disturbance to the current controller for compensation of the control variables.

[0067] The design method can be found in Example 1, and will not be repeated here.

[0068] Of course, the above description is not intended to limit the present invention, and the present invention is not limited to the examples given above. Any changes, modifications, additions or substitutions made by those skilled in the art within the scope of the present invention should also fall within the protection scope of the present invention.

Claims

1. A current controller design method, characterized in that, include: Step 1: Design a model-free, non-singular terminal sliding mode current controller; Step 2, design the quasi-resonant modulator: ； Where, ω c ω is the cutoff frequency, k is the resonance coefficient, ω is the resonant frequency, and s is the complex frequency domain. Step 3, Quasi-resonant compensation step: Connect one of the aforementioned quasi-resonant regulators in parallel across the two ends of the d-axis modelless non-singular terminal sliding mode current controller and the q-axis modelless non-singular terminal sliding mode current controller, and output the compensation voltage to the control variable of the corresponding current controller.

2. The current controller design method according to claim 1, characterized in that, The non-singular terminal integral sliding surface of the model-free non-singular terminal sliding mode current controller is: ； Where e is the state variable, which is the difference between the desired stator current and the stator current, c is the sliding diaphragm gain, p and q are positive odd numbers, and 1 < p / q < 2.

3. The current controller design method according to claim 2, characterized in that, The reaching law of the model-free non-singular terminal sliding mode current controller is: ； in, , The design parameters are all positive real numbers, and the parameter α satisfies 0 < α < 1; Solving for the model-free nonsingular terminal sliding mode control law yields the control law for the model-free nonsingular terminal sliding mode current controller: ； ； , These are the estimated values ​​of the q-axis lumped disturbance and the d-axis lumped disturbance, respectively. , These represent the voltage gains along the q-axis and d-axis of the stator to be designed, respectively. , These are the state variables of the stator's q-axis and d-axis, respectively, to be designed.

4. The current controller design method according to claim 3, characterized in that, A hyperlocal model of the permanent magnet synchronous motor current loop is established, and the hyperlocal model is as follows: ； Where i is the stator current, which is the output variable. Let u represent the stator voltage, and F represent the lumped disturbance of the permanent magnet synchronous motor, which is the control variable.

5. The current controller design method according to any one of claims 1-4, characterized in that, The permanent magnet synchronous motor is also equipped with an extended state observer, which is used to output an estimate of the q-axis lumped disturbance. and the estimated value of the d-axis lumped disturbance To the modelless non-singular terminal sliding mode current controller.

6. The current controller design method according to claim 5, characterized in that, The extended state observer is: ； The observed value of the stator current. This is an estimate of the lumped disturbance. It is a nonlinear factor. ε is the filter factor, β1 and β2 are the gain of the observer parameters to be designed, and ε is the observation error; 。 7. The current controller design method according to claim 4, characterized in that, q-axis state variables and the state variables of the d-axis They are respectively: ； and These are the desired currents along the q-axis and the d-axis, respectively. and These are the q-axis stator current and the d-axis stator current, respectively. By taking the derivatives of each state variable and combining them with the hyperlocal model, a mathematical relationship is established between the derivatives of the state variables and the control variables and lumped disturbances of the permanent magnet synchronous motor.

8. The current controller design method according to any one of claims 1-4, characterized in that, The design method of the model-free non-singular terminal sliding mode current controller also includes setting a PI control module; Obtain the real-time angular velocity; the desired current along the q-axis. The PI control module obtains the angular velocity based on the reference angular velocity and the real-time angular velocity. The reference angular velocity is obtained by setting it.

9. A current controller for a permanent magnet synchronous motor, characterized in that, It is designed by the design method described in any one of claims 1-8.

10. A permanent magnet synchronous motor, characterized in that, Including a current controller designed according to the design method of any one of claims 1-8; The current controller is used to obtain the control variable based on the desired current and the control law of the control variable, and is used to control the power supply of the permanent magnet synchronous motor. The extended state observer is used to observe unknown disturbances to obtain an estimate of the lumped disturbance, and transmits the estimate of the lumped disturbance to the current controller for compensation of the control variables.

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