Hybrid DC / DC converter having a push-pull full bridge topology and secondary-side resonance circuit

Abstract

The invention relates to a converter device comprising: a primary unit with two DC voltage inputs for connecting to a fuel cell and a secondary unit with two DC voltage outputs for connecting to a consumer; wherein the primary unit and the secondary device are coupled via a transformer; wherein the transformer has a series circuit of a first primary coil (Lp1) and a second primary partial coil (Lp2) and a secondary coil (Ls), wherein a first of the DC voltage inputs is connected via a first inductance (L1) to a connection point of the first and second primary partial coil (Lp1, Lp2); wherein the primary unit comprises an inverter, the bridge points of which are connected in parallel to the series circuit of the first and second primary coil (Lp1, Lp2); wherein the secondary unit has a rectifier, the AC inputs of which are connected in parallel to a series circuit of the secondary coil (Ls) and a second inductance (L2); and wherein the AC input of the rectifier connected to the second inductance (L2) and one of the two DC outputs is connected via a resonance capacitor Cr, wherein the second inductance (L2) and the resonance capacitor Cr form a series resonance circuit.

Description

[0001] HYBRID DC / DC CONVERTER WITH PUSH-PULL FULL BRIDGE TOPOLOGY AND SECONDARY SIDE RESONANCE CIRCUIT

[0002] The invention relates to a converter device, in particular a high-performance voltage converter, for using a fuel cell as an energy source for vehicles, aircraft, rail vehicles, watercraft, etc., a method for operating such a converter device, a drive system for a vehicle having such a converter device, and a vehicle having such a drive system.

[0003] Vehicles with multiple fuel cells (i.e., a fuel cell stack) as their energy source are constantly being developed further, and it is foreseeable that they will play a significant role in the near future. Their main application is in mobile transportation, where the required energy quantities are so large that carrying correspondingly large batteries becomes either uneconomical or too heavy for the vehicle. Although fuel cell vehicles also have a traction battery, this is comparatively small. It enables energy recovery (recuperation) during braking and the dynamic compensation of load fluctuations that the rather sluggish fuel cell cannot immediately follow; fuel cell vehicles are therefore, strictly speaking, series hybrid vehicles.

[0004] The fuel cell as an energy source also has the advantage that the hydrogen required for operation can be refueled in a relatively short time, whereas charging a large battery at a similarly rapid rate would require a very powerful and expensive infrastructure. This applies particularly to buses, trucks, and ships, but aircraft are also conceivable in the near future.

[0005] What all of these applications as energy sources in vehicles have in common is that they generally require high continuous electrical power outputs of well over 100 kW to drive them. To ensure that the operating currents and required cable cross-sections are not excessive despite this high power, the operating voltages are selected accordingly. A frequently used DC voltage range is between 600 V and 900 V, but higher voltages up to 1500 V are also conceivable in the future. On the other hand, a certain technologically driven "standardization" is emerging in the development of fuel cell stacks; the nominal power of a stack is around 150 kW, the maximum current up to 900 A, resulting in a nominal operating voltage of around 167 V. To increase this voltage to the aforementioned operating voltage range, a powerful voltage converter is required. To date, a simple boost converter has proven to be a suitable basic circuit (seeFig. 1 (state of the art)), whereby the power is usually distributed over several staggered “phases” in order to achieve the most continuous power flow possible (comparable to an internal combustion engine, where the smoothness of running increases with the number of cylinders).

[0006] As the power and operating voltage of the drive increase, this simple configuration also has its disadvantages:

[0007] • The electronic power switches of the boost converter (IGBTs or SiC MOSFETs) must be dimensioned for both the maximum voltage of the drive intermediate circuit (e.g. 900V) and the high fuel cell current (e.g. 900A), so that the so-called "corner power", ie the product of maximum current and maximum voltage, is 900V x 900A = 810kVA, which is more than five times the usual nominal power of the fuel cell of 150kW; this represents a significant over-dimensioning.

[0008] • The dielectric strength of the insulation between the fuel cell and earth (vehicle chassis) must meet more demanding standards of the drive network (max. 900V), although the fuel cell voltage itself hardly reaches 300V.

[0009] • In the event of a boost converter failure (e.g., if the freewheeling diode fails), complex safety measures must be taken to prevent current from flowing back from the drive circuit into the fuel cell, as otherwise the fuel cell stack would be immediately destroyed, which poses certain risks in a hydrogen environment; this applies particularly to systems with multiple fuel cell stacks and thus multiple boost converters.

[0010] • Both the boost converter and the drive's inverter undergo periodic, rapid switching operations, which cause electromagnetic interference that must be filtered out in accordance with prescribed EMC (electromagnetic compatibility) standards. The filters provided for this purpose use, among other things, Y capacitors to divert the interference from the traction circuit to electrical ground. For safety reasons, the maximum total energy of these capacitors—and thus their total capacitance—must not exceed a certain value.Since every component in the traction circuit – including the fuel cell – already has a certain capacitance to ground, in systems where multiple fuel cell stacks are connected to the same traction circuit, the technical possibilities for dissipating electromagnetic interference using additional Y capacitors become increasingly limited with the increasing number of fuel cell stacks, even though more systems also generate more interference. This means that, above a certain system size, the maximum total energy of the Y capacitors must exceed the prescribed standard value in order to comply with the EMC standard.

[0011] One known solution to this problem is the introduction of a transformer between each fuel cell circuit and the traction circuit. Figure 2 (state of the art) shows such a circuit, which is known in the prior art. A transformer stage is connected downstream of the booster. The transformer stage comprises a DC-AC converter (inverter) connected upstream of the transformer and a rectifier connected downstream of the transformer. The circuit allows the connection of a fuel cell stack to the input of the booster (boost converter) with input voltages in the range of approximately 150 to 300 V. D C for currents up to 900A.

[0012] This solution works well, but also has the following disadvantages:

[0013] • It is very complex: an additional transformer and two additional electronic switching stages (inverter and rectifier) ​​are required, which entails additional costs and weight.

[0014] • The additional processing stages generate more power loss and thus reduce the efficiency of the system.

[0015] The object of the invention is to provide a converter device which overcomes the aforementioned disadvantages and, in particular, reduces the number of switching stages.

[0016] The invention is based on the features of the independent claims. Advantageous developments and refinements are the subject of the dependent claims. Further features, possible applications, and advantages of the invention will become apparent from the following description and the explanation of exemplary embodiments of the invention illustrated in the figures.

[0017] A first aspect of the invention relates to a converter device comprising: a primary device with two DC voltage inputs for connecting a fuel cell and a secondary device with two DC voltage outputs for connecting a consumer; wherein the primary device and the secondary device are coupled via a transformer; wherein the transformer has a series connection of a first primary coil (Lp1) and a second primary coil section (Lp2) and a secondary coil (Ls), wherein a first of the DC voltage inputs is connected via a first inductance (L1) to an electrical connection point of the first and second primary coil sections (Lp1, Lp2); wherein the primary device has an inverter whose bridge points are connected in parallel to the series connection of the first and second primary coil sections (Lp1, Lp2);wherein the secondary device comprises a rectifier whose AC inputs are connected in parallel to a series circuit of a secondary coil (Ls) and a second inductance (L2); and wherein the AC input of the rectifier connected to the second inductance (L2) and one of the two DC voltage outputs are connected via a resonant capacitor C; r is connected, the second inductance (L2) and the resonance capacitor C r form a series resonant circuit.

[0018] The inverter advantageously has the controllable switches S1, ..., S4, which can be designed, for example, as IGBT diode combinations or as MOSFETs.

[0019] The proposed converter device realizes a powerful, potential-isolated or galvanically isolated DC-DC converter device that has a smaller number of switching stages and combines the transformer with the choke of the boost converter.

[0020] The proposed converter device is advantageously designed for bidirectional operation, i.e. power is fed back from the secondary side of the transformer to the primary side of the transformer. For this purpose, the alternating current input of the rectifier connected to the secondary coil (Ls) is advantageously connected to a bridge point of an inverter, the inputs of which are each connected in parallel to the two direct voltage outputs of the rectifier. This circuit acts as a resonant buck converter from the secondary side to the primary side of the transformer. To expand the secondary voltage range to smaller values, it is also advantageous to also connect the alternating current input of the rectifier connected to the second inductance (L2) to a bridge point of an inverter, the inputs of which are each connected in parallel to the two direct voltage outputs of the rectifier.

[0021] In fuel cell applications of the converter, the feedback from the primary-side intermediate circuit from the battery is used, for example, to "start up" the fuel cell system. However, this advantageously only operates various auxiliary units (pumps, compressors, heaters) whose power requirements are significantly lower than the rated power of the fuel cell stack itself. The reduced feedback power therefore hardly represents a limitation for this application.

[0022] In an advantageous development of the proposed converter device, the AC input of the rectifier connected to the second inductance (L2) and the two DC outputs of the rectifier are each connected via a resonance capacitor C r1 , C r2 The resonance capacitors C r1 , C r2 identically dimensioned.

[0023] An advantageous further development of the proposed converter device is characterized in that the number of coil windings of the first and second primary coil section (Lp1, Lp2) is identical.

[0024] An advantageous further development of the proposed converter device is characterized in that the inverter has an intermediate circuit capacitor C z The inverter is advantageously a full-bridge inverter. The full-bridge inverter advantageously comprises four controllable switches S1, S4. The four controllable switches S1, S4 each advantageously comprise an IGBT diode combination or MOSFETs.

[0025] An advantageous further development of the proposed converter device is characterized in that a capacitor C inAn advantageous development of the proposed converter device is characterized in that a capacitor C a is switched on.

[0026] The resonance capacitor C is advantageous r dimensioned as follows:

[0027] Cro / 2 2 C r S 2 Cro with

[0028] P max : maximum electrical power of the secondary device, max : maximum electrical voltage of the secondary device f0: nominal switching frequency for controlling the inverter.

[0029] The second inductance (L2) is advantageously dimensioned as follows:

[0030] L20 / 2 ​​< L2 < 2 L2 with

[0031] L2: Inductance in [H]

[0032] C r : Capacitance of the resonance capacitor f0: Nominal clock frequency for controlling the inverter.

[0033] An advantageous further development of the proposed converter device is characterized in that the primary device is designed for an input voltage in the range of 100 to 400 V DC or 150 to 350 V DC or 150 to 200 V DC or preferably 167 V to 333 V.

[0034] An advantageous further development of the proposed converter device is characterized in that the primary device is designed for a continuous power in the range of 100 to 300 kW, in particular in the range of 100 to 150 kW.

[0035] An advantageous further development of the proposed converter device is characterized in that the secondary device is designed for an output voltage in the range of 600 to 1,500 V DC or 600 to 900 V DC.

[0036] A second aspect of the invention relates to a method for operating a converter device as described above, which is characterized in that two half-bridges of the inverter are operated with identical PWM signals (PWM: pulse width modulation) with a clock frequency 0, but which are offset by half a period (180°).

[0037] Advantageously, the clock frequency f0 is selected in the range from 10 kHz to 150 kHz or is in particular 50 kHz or 49 kHz or 48 kHz or 47 kHz or 46 kHz or 45 kHz.

[0038] An advantageous development of the proposed method is characterized in that, in order to regulate an input voltage between the DC voltage inputs of the primary device, and / or to regulate an output voltage between the DC voltage outputs of the secondary device and / or to regulate a transmission power of the converter device, a duty cycle of PWM signals used to control the inverter and / or a clock frequency f0 of the PWM signals are varied.

[0039] An advantageous development of the proposed method is characterized in that for regulating an input voltage between the DC voltage inputs of the primary device and / or for regulating an output voltage between the DC voltage outputs of the secondary device and / or for regulating a transmission power of the converter device, a duration of a primary transmission current pulse l Dc jn greater than or equal to the duration of the secondary current pulse l puis-S ek is selected, where the secondary current pulse l puis-S ek is determined by the series resonant circuit consisting of second inductance L2 and resonance capacitor Cr and an operating point of the converter device.

[0040] A further aspect of the invention relates to a drive system for a vehicle, aircraft, rail vehicle, watercraft, underwater vehicle, etc., wherein the drive system comprises a converter device as described above.

[0041] Such a drive system is in particular a drive system based on one or more electric motors or on a combination of at least one electric motor and an internal combustion engine.

[0042] A final aspect of the invention relates to a vehicle, in particular a road vehicle, an aircraft, a rail vehicle, a watercraft or an underwater vehicle with such a propulsion system.

[0043] Further advantages, features, and details will become apparent from the following description, which – where appropriate with reference to the drawings – describes at least one embodiment in detail. Identical, similar, and / or functionally equivalent parts are provided with the same reference numerals.

[0044] They show:

[0045] FIG. 1 Simple boost converter (state of the art)

[0046] FIG. 2 Boost converter with transformer assembly and rectifier (state of the art)

[0047] FIG. 3 shows a converter device according to the invention

[0048] FIG. 4a, b representations to explain preferred operating phases of the converter device, here: resonant, continuous operation with x = Vout / Vz < 2

[0049] FIG. 5a, b representations to explain preferred operating phases of the converter device, here: resonant, discontinuous operation with x = Vout / Vz < 2

[0050] FIG. 6a, b representations to explain preferred operating phases of the converter device, here: resonant voltage doubler with x = Vout / Vz = 2

[0051] FIG. 7a, b representations to explain preferred operating phases of the converter device, here: hard-switching operation with x = Vout / Vz < 2 FIG. 8a, b, c representations to explain a converter device according to the invention with a modified secondary device for feeding back the primary-side intermediate circuit from the battery (bidirectional operation) and its preferred operating phases.

[0052] FIG. 3 shows an embodiment of the proposed converter device, comprising the following components:

[0053] A transformer assembly, which may consist entirely of a leakage field transformer, but is shown with three components for a better understanding of its functionality:

[0054] ■ A first choke L1

[0055] ■ A transformer with two identical primary windings Lp1, Lp2 connected in series, with the first choke L1 being connected at the junction point of the two primary windings, and a secondary winding Ls, with the coupling between Lp1, Lp2 and Ls being considered ideal in the first consideration.

[0056] ■ A second choke L2 connected in series with the secondary winding Ls.

[0057] A full-bridge inverter, consisting of four switches S1,..., S4 (e.g. realized as an IGBT diode combination, but MOSFETs are also conceivable) and the intermediate circuit capacitor Cz, whose bridge points are connected to the series connection of the primary windings Lp1, Lp2 of the transformer assembly

[0058] A full-bridge rectifier consisting of the output capacitor Ca and four diodes D1,..., D4, whose input is connected to the series circuit of transformer secondary winding Ls and choke L2.

[0059] A capacitor C in, which is connected between the still free terminal of L1 and one of the DC voltage poles of the inverter, preferably the negative pole. A resonant capacitor Cr, which is connected between one of the two bridge points (i.e., AC inputs) of the rectifier and one of the two DC voltage outputs of the rectifier (the negative pole). The input of the converter is parallel to the capacitor C in switched, ie the fuel cell is connected there

[0060] The output of the converter is connected in parallel to the capacitor Ca, where, for example, the traction circuit of the vehicle is connected, here represented by the connection of a traction battery.

[0061] The nominal value of the clock frequency is f=f0, with for example f o=50kHz. All components of the system are advantageously dimensioned for operation at f=f0, but the system can then be operated at any other clock frequency f, provided that its proper function is not impaired.

[0062] A maximum output voltage Va,max at the DC output of the rectifier is determined by the maximum traction circuit voltage and is, for example, Va,max=900V.

[0063] The primary intermediate circuit voltage is designated Vz. A nominal intermediate circuit voltage VzO is determined by the transformation ratio ü = ^Ls / Lpl = ■ Ls I Lpl and is VzO = Va,max / ü.

[0064] Depending on requirements, it may be advantageous to divide the power to be processed by the converter device between several converter devices, each of which is connected in parallel at the input and output. If, for example, a typical fuel cell power of 150 kW is divided equally between three converters, then the maximum power of a converter device is, for example, Pmax = 150 kW / 3 = 50 kW. It may be advantageous to operate the converters synchronously at the same clock frequency, with the clock signals for the individual converters having a defined phase shift cp relative to one another, for example cp = 1807n, where n is the number of parallel converters. The above also applies to the maximum input current U max of the converter device. If a typical maximum fuel cell current of 900 A is divided equally between three parallel converter devices, the individual maximum converter input current is I max = 300 A.

[0065] The input choke L1 is preferably dimensioned so that the resulting ripple current in the choke 11 pk-pk = AI does not exceed a defined value Almax. For example, let Almax = 60% of U max = 300A, so Almax = 0.6 300A = 180A, the result is:

[0066] VzO 450V

[0067] L1 = - — - - = - — - = 3.125 pH

[0068] 16 • fn • Va,max 2 16 • 50kHz • 180A

[0069] The nominal dimensioning of the capacitor Cr is advantageously determined according to the following relationship: j) max

[0070] Cr0 / 2 < Cr < 2Cr0, where: Cr0= - - f0• Va, max

[0071] Thus, if a capacitor with capacitance Cr0 is charged to the voltage Va,max, the product of its energy and twice the nominal clock frequency f0 equals the maximum power Pmax. For example, with Pmax=50kW, Va,max=900V, and f o =50kHz: Cr0= 1.234pF

[0072] The capacitor Cr can also be divided into two capacitors, preferably of identical capacitance Cr / 2, with the second capacitor being connected in parallel to D1 instead of D2.

[0073] The nominal dimensioning of the choke L2 is advantageously determined according to the following relationship:

[0074] L20 / 2 ​​< L2 < 2L20, where:

[0075] If a coil of inductance L20 is connected to a capacitor of capacitance Cr, the result is an oscillating circuit with a resonant frequency that is exactly twice the nominal clock frequency f0. For example, with Cr=Cr0=1.234pF, and f o =50kHz: L20= 2.053pH

[0076] FIGS. 4a and b show diagrams illustrating preferred operating phases of the converter device, here: resonant, continuous operation with x = Vout / Vz<2. FIG. 4a shows a circuit diagram of the converter device. FIG. 4b shows the time behavior of the PWM voltages V1 and V2, the difference V7-V2, the secondary voltage Vsec, and the currents In and Isec. Where the respective voltages and currents are measured can be seen in FIG. 4a.

[0077] The inverter's two half-bridges (S1 / S2 and S3 / S4) are operated with identical PWM signals with a clock frequency f, but offset by half a period (180°). The duty cycle is calculated by dividing the pulse width by the period duration, corresponding to a range between 0 and 100%, and is proportional to the load current. The typical PWM duty cycle range is between 25% and 75%, but can also be higher or lower.

[0078] Basically, two operating phases can be distinguished:

[0079] 1. The PWM voltages V1 and V2 are different, i.e., either V1=Vz and V2=0, or V1=0 and V2=Vz. Power is then transferred to the secondary side of the transformer, while simultaneously energy is drawn from the primary side, because the voltage at the primary transformer center point is Vz / 2. This energy transfer occurs as a resonant secondary current pulse, which decays before V1 or V2 changes.

[0080] 2. The PWM voltages VI and V2 are identical; either Vl=V2=0, in which case energy from the input voltage Vin is stored in the inductor LI. Or Vl=V2=Vz, in which case energy from the input and the inductor LI is stored in the primary DC link capacitor Cz. In both cases, no energy is transferred to the secondary side.

[0081] In the present case of a resonant, non-gap operation, with x =Vout / Vz<2, the "resonant" pulses are transmitted completely, and the duration Ton of the transmitted PWM voltage pulse is - except for a "dead time" required for voltage commutation - equal to the current pulse duration T2, ie the voltage pulse is without a "currentless gap" within the primary voltage pulse: with the resonant circuit time constant:

[0082] T = ->jLs Cr and x = Vout / Vz.

[0083] The impulse energy E p is determined by the intermediate circuit voltage Vz and the ratio x between:

[0084] This operating mode is preferable because it results in the lowest crest factor and thus the lowest possible power loss and output current ripple. This results in a clear relationship between the current pulse duration T2=Ton, frequency / , and the PWM duty cycle y: y=f-T2 (y< V2) or y=lf-T2 (y> V2).

[0085] FIGS. 5a, b show illustrations for explaining operating phases of a converter device according to the invention, analogous to the illustrations in FIGS. 4a, b, here for a resonant, intermittent operation of the converter device with x = Vout / Vz < 2.

[0086] Here, the "resonant" pulses are transmitted completely, and the duration T o n of the transmitted PWM voltage pulse is longer than the current pulse duration T 2 . This means that after the end of the current pulse (i.e., the energy transfer to the secondary side), the voltage pulse persists for a while to achieve the desired primary duty cycle; a "current-free gap" is created within the primary voltage pulse: with the resonant circuit time constant: T = -jLs - Cr and x = Vout / Vz. The pulse energy is determined by the intermediate circuit voltage Vz and the ratio x between the output voltage Vout and the intermediate circuit voltage Vz.

[0087] This operating mode is particularly useful when, within the given boundary conditions, a required input voltage at the desired power cannot be achieved in "continuous operation." By extending the duration Tn of the PWM voltage pulse beyond the current pulse duration T2, a degree of freedom is gained for selecting the duty cycle while maintaining the frequency and the intermediate circuit voltage Vz, and thus maintaining the power. The relationship between voltage pulse duration Tn, frequency, and PWM duty cycle y is: y=fTn (y< V2) or y=lf-Tn (y> V2).

[0088] FIGS. 6a, b show illustrations for explaining operating phases of a converter device according to the invention, analogous to the illustrations in FIGS. 4a, b and 5a, b, here for a resonant voltage doubler with x = Vout / Vz = 2.

[0089] The "resonant" current pulses are transmitted completely and have - within the resonance condition - the shortest possible duration T2 = T - with the resonant circuit time constant T = ^Ls Cr . A doubling of the voltage occurs; thus, the resonant capacitor Cr does not necessarily have to be completely recharged, which means that any power P < 2f - ü 1 • Vz 2• Cr can be transmitted. The power is determined solely by the current regulator in the booster. The duty cycle can thus be varied within a wide range, independent of the power, by changing the frequency f while maintaining the voltage pulse duration Ton. This operating mode is preferably used for low power levels, which would otherwise only be achievable with an impermissibly low clock frequency or through "hard switching operation" (see the following operating mode, FIG.7a, b). In practice, "intermittent operation" usually results due to the shortest possible current pulse when using the voltage doubler.

[0090] FIGS. 7a, b show illustrations for explaining operating phases of a converter device according to the invention, analogous to the illustrations in FIGS. 4a, b and 5a, b and FIGS. 6a, b, here for a hard-switching operation with .v = Vout / Vz < 2

[0091] The "resonant" current pulses are transmitted incompletely, i.e., the current pulses are interrupted by "premature" switching on the primary side. Thus, the voltage pulse duration Ton is shorter than the theoretical resonant pulse duration T2. with the resonant circuit time constant: T = 4LS ~CT and x = Vout / Vz.

[0092] This operating mode is preferably used at low power levels to prevent the PWM frequency f from becoming too low. The transmitted pulse energy E Pis thus reduced and is primarily determined by the specified voltage pulse duration Ton. Depending on the duration of Ton, the resonant capacitor Cr may or may not be fully recharged. Either way, the energy of the stray inductance flows into the secondary circuit. Due to the low power, the primary input current is relatively small; therefore, the turn-off current of the primary switch during "hard" turn-off is not necessarily higher than at full load in resonant operation. The decay of the secondary currents occurs more quickly than in resonant operation, which requires special attention to the switching processes in the secondary-side rectifier diodes (power loss, EMC).

[0093] FIGS. 8a, b show illustrations to explain a converter device according to the invention with a modified secondary device for feeding back the primary-side intermediate circuit from the battery (bi-directional operation) and its preferred operating phases.

[0094] The diode half-bridge D3-D4 is advantageously supplemented by parallel active switches S7-S8; thus, the secondary circuit can operate as a resonant converter.

[0095] So that even at low secondary voltages V out a high primary voltage (ie a high intermediate circuit voltage V z ) is possible, the capacitor diodes D1, D2 are advantageously supplemented with parallel switches S5-S6. The current and voltage pulse shapes in the resonant circuit run "backwards" in time compared to the "normal" operating mode, ie, there is first a linear current increase, which changes into a resonant oscillation after S5 or S6 is switched off. The switching off of S5 or S6 should preferably occur exactly when the energy in the coil Ls is just large enough for the capacitor voltage V c can completely swing around (from zero to V out or from V out to zero).

[0096] The current flow in the primary coil L in looks the same as in "forward operation", only the average current is negative.

[0097] The example in FIG. 8b illustrates "continuous operation." Of course, "intermittent operation" can also be achieved by keeping S1 and S4 or S2 and S3 switched on for a longer period. S7 or S8 can be switched off as soon as the secondary current l sek has decayed to zero.

[0098] A "voltage halving" (as a counterpart to the voltage doubling in forward operation) and a "hard turn-off" are also conceivable, but if the resonant capacitor Cr is incompletely recharged, the switches S5 and S6 must not be activated, since otherwise the remaining capacitor energy would be absorbed by the short-circuiting switch, which could exceed its permissible power dissipation.

[0099] FIG. 8c shows the voltage and current waveform after switching on S6 and S7 in the circuit of FIG. 8a.

[0100] Switch S6 is activated simultaneously with switch S7 and remains switched on until the secondary current l sek has reached the required value so that when switched off the capacitor voltage can just completely swing around, although V z > V out / 2.

[0101] Compared to "forward operation," in regenerative operation the signal curve in the resonant circuit is reversed in time, meaning that first there is a linear current increase followed by a resonant oscillation. The transition is triggered by the switching off of switch S5 or switch S6. Although this occurs under load, the switching-off losses are low, since the voltage across capacitor C r builds up comparatively slowly. Is V z < V out, the current continues to rise after switching off as shown and reaches the peak value l0 when V c the output voltage minus the transformed input voltage, for a 1:1 transformer: V c = V out - V z ie the voltage across the leakage inductance is then exactly zero. Now the capacitor voltage V c nor by the value of the transformed input voltage V z increase to the value V out ideally to be achieved exactly when l se k=0, ie a sinusoidal oscillation process with current amplitude l0 and voltage amplitude V occurs z .

[0102] Although the invention has been illustrated and explained in detail by means of preferred embodiments, the invention is not limited by the disclosed examples, and other variations may be derived therefrom by those skilled in the art without departing from the scope of the invention. It is therefore clear that a multitude of possible variations exist. It is also clear that the embodiments mentioned by way of example are truly only examples and should not be construed as limiting the scope, possible applications, or configuration of the invention in any way.Rather, the preceding description and the description of the figures enable the person skilled in the art to implement the exemplary embodiments in concrete terms, whereby the person skilled in the art, with knowledge of the disclosed inventive concept, can make various changes, for example with regard to the function or the arrangement of individual elements mentioned in an exemplary embodiment, without departing from the scope of protection defined by the claims and their legal equivalents, such as a further explanation in the description.

Claims

Patent claims 1. Converter device comprising: a primary device with two DC voltage inputs for connecting a fuel cell and a secondary device with two DC voltage outputs for connecting a consumer; - wherein the primary device and the secondary device are connected via a transformer coupled; - wherein the transformer comprises a series circuit of a first primary coil (Lp1) and a second primary coil (Lp2) and a secondary coil (Ls), - wherein a first of the DC voltage inputs is connected via a first inductance (L1) to a connection point of the first and second primary coil sections (Lp1, Lp2); - wherein the primary device comprises an inverter whose bridge points are connected in parallel to the series connection of the first and second primary coils (Lp 1 , Lp2); - wherein the secondary device comprises a rectifier whose AC inputs are connected in parallel to a series circuit of a secondary coil (LS) and a second inductance (L2); and - wherein the AC input of the device connected to the second inductance (L2) Rectifier and one of the two DC outputs via a resonance capacitor C r is connected, the second inductance (L2) and the resonance capacitor C r form a series resonant circuit.

2. Converter device according to claim 1, wherein the AC input of the rectifier connected to the secondary coil (Ls) is connected to a bridge point of an inverter, the inputs of which are each connected in parallel to the two DC outputs of the rectifier.

3. Converter device according to one of claims 1 or 2, wherein the AC input of the rectifier connected to the second inductance (L2) is connected to a bridge point of an inverter whose inputs are each connected in parallel to the two DC outputs of the rectifier.

4. Converter device according to one of claims 1 to 3, wherein the AC input of the rectifier connected to the second inductance (L2) and the two DC voltage outputs are each connected via a resonance capacitor C r1 , C r2 are connected.

5. Converter device according to one of claims 1 to 4, wherein the inverter comprises an intermediate circuit capacitor C z has.

6. Converter device according to one of claims 1 to 5, wherein the inverter is a full-bridge inverter.

7. Converter device according to one of claims 1 to 6, wherein between the two DC voltage inputs a capacitor C in is switched on.

8. Converter device according to one of claims 1 to 7, wherein between the two DC voltage outputs a capacitor C a is switched on.

9. A method for operating a converter device according to one of claims 1 to 8, wherein two half-bridges of the inverter are operated with identical PWM signals having a clock frequency of 0, but offset by half a period (180°).

10. A method for operating a converter device according to claim 9, wherein the clock frequency f0 is selected in the range from 10 kHz to 150 kHz, in particular 50 kHz or 49 kHz or 48 kHz or 47 kHz or 46 kHz or 45 kHz.

11. A method for operating a converter device according to claim 10, in which, in order to regulate an input voltage between the DC voltage inputs and / or to regulate an output voltage between the DC voltage outputs and / or to regulate a transmission power of the converter device, both a duty cycle of PWM signals for controlling the inverter and / or a clock frequency f0 of the PWM signals are varied.

12. Drive system for a vehicle comprising a converter device according to one of claims 1 to 8.

13. Vehicle, in particular road vehicle, aircraft, rail vehicle, watercraft or underwater vehicle with a propulsion system according to claim 12.

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