Derivative voltage and current sensing devices

Abstract

A device for detecting and measuring at least one of a pulse voltage source, the device comprising: a transformer having a primary winding electromagnetically coupled to a secondary winding, the primary winding having a first inductance; and a differentiator comprising at least one resistive element coupled to the primary winding, thereby forming a closed circuit with the pulse voltage source, the differentiator being configured to electromagnetically induce an induced pulse voltage on the secondary winding; wherein the induced pulse voltage indicates the pulse voltage source.

Description

Technical Field

[0001] The disclosed technology relates generally to voltage and current sensing, and more particularly to an apparatus for derivative voltage and current detection and measurement in power electronic systems. Background Art

[0002] In general, various methods and devices for sensing voltage and current are known in the art. The range of measurement and detection techniques is wide, some of which include: sensing (shunt) resistors; techniques based on Faraday's law of induction, such as current transformers (CTs) and Rogowski methods; techniques based on magnetic field sensors (e.g., Hall effect sensors and fluxgate sensors); techniques based on the magnetoresistive effect using magnetoresistive current sensors; techniques based on the Faraday magneto-optical effect in optical fibers located around current-carrying conductors; application-specific integrated circuits (ICs); conductor trace resistance sensing in printed circuit boards (PCBs); direct methods using conventional ammeters and voltmeters, etc. Each of these methods has advantages and disadvantages. For example, shunt resistor technology is relatively simple, but exhibits an increase in power loss as the current flowing through the shunt resistor increases. CT technology can be used to measure high currents, but may typically exhibit hysteresis and typically exhibit an undesirable direct current (DC) component, which may cause the core material (e.g., ferrite) of the CT to saturate. Rogowski coil technology exhibits low inductance and is not affected by saturation due to the absence of a magnetic core, however, this method exhibits relatively low sensitivity and may typically require the use of an amplifier and integrator circuit, which in turn requires power. Technology based on magnetic field sensors such as Hall effect sensors can achieve high accuracy measurements, but typically produce low level outputs, requiring signal conditioning electronics including low noise amplifiers and temperature compensation circuits. Technology based on the magnetoresistive effect is typically capable of high sensitivity measurements, but may exhibit non-linear behavior and may be susceptible to damage from external magnetic fields. Trace resistance sensing technology is low cost, but is typically affected by thermal drift of the conductor trace, effectively changing the current measurement, and requiring the use of an amplifier to achieve a useful measurement.

[0003] Reference is now made to FIG. 1 , which is a schematic diagram of a series-connected sensing circuit for measuring current flowing through a transistor-implemented switch in the prior art, generally designated 10. Circuit 10 includes transistor 12, sensing resistor 14, and voltage sensing subcircuit 16 (i.e., implemented via a low-pass filter / integrator circuit). Voltage sensing subcircuit 16 includes resistor 18 and capacitor 20, thereby forming a series RC circuit. Transistor 12 is used as a power switch and is typically implemented by a metal oxide semiconductor field effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), etc. Transistor 12 typically includes terminals represented as gate (G), drain (D), source (S), and body (B), wherein the drain terminal is connected to a high voltage power supply 22, the gate terminal is connected to an input (gate drive) signal source (not shown), and the body terminal and the source terminal are interconnected to each other and further connected to the sensing resistor 14 and the low-pass filter / integrator 16. The high voltage is connected between the drain terminal of transistor 12 and ground.

[0004] FIG1 shows an input signal 24 (varying over time) driving the gate terminal of transistor 12. Input signal 24 is used to control the switching state of transistor 12, which in turn acts as a switch between a conducting ('ON') and a non-conducting ('OFF') state. For simplicity, transients are not discussed in this prior art example. <t 1 The input signal 24 at t is zero, and transistor 12 is off (ie, not conducting current between high voltage source 22 and sense resistor 14). 1 ≤t≤t 2 The input signal 24 between 2 and 1 drives the transistor 12 to switch to the ON state, thus enabling current to flow through the transistor 12 (i.e., between the drain terminal and the source terminal) and through the sense resistor 14. The current flowing through the sense resistor 14 (usually low ohmic) produces a voltage drop V across the sense resistor 14. Rs The voltage sensing subcircuit 16 (which is a series RC low pass filter / integrator circuit) senses the voltage drop V Rs and generates an output signal 26. By measuring the output signal 26 and knowing the resistance of the sense resistor 14, the current flowing through the switch (ie, transistor 12) and the sense resistor 14 is determined.

[0005] Several disadvantages of current sensing techniques employing series-connected sense resistors as shown in FIG. 1 may include time-accumulated energy loss (e.g., as heat) due to current flowing through sense resistor 14 (i.e., depending on the current value and resistance value); unwanted inductance between transistor 12 and sense resistor 14, which may cause inductive spikes; typically low voltage output signals (on the order of millivolts); minimal or nonexistent high voltage protection; and RsThe V of transistor 12 is caused gs The reduction in gate-source voltage leads to conduction losses.

[0006] Another known prior art for sensing current uses a current sensing circuit or application specific integrated circuit (IC) connected in parallel with a power switch (e.g., a switching transistor) through which the current intended for sensing flows. Reference is now made to FIGS. 2A and 2B. FIG. 2A is a schematic diagram of a parallel-connected sensing circuit for measuring the current flowing through a transistor-implemented switch in the prior art, generally labeled 30. FIG. 2B is a timing diagram corresponding to the operation of an integrated circuit (IC) in the parallel-connected sensing circuit of FIG. 2A, generally labeled 50. Circuit 30 (FIG. 2A) includes a transistor 32, an integrated circuit (IC) 34, and a gate resistor 36. Transistor 32 is shown as a MOSFET in FIG. 2A; however, the same principles also apply to IGBTs. IC 34 is an IR25750 current sensing IC manufactured by International Rectifier (acquired by Infineon Technologies AG). IC 34 includes a CS pin (interchangeably referred to herein as a "pin," "connector," and "terminal"), a GATE pin, a VS pin, and a COM pin. The GATE terminal of IC 34 is connected to the gate terminal of transistor 32 and one end of gate resistor 36. The other end of gate resistor 36 is connected to gate drive input signal terminal 40. COM terminal is connected to the source terminal of transistor 32, which in turn is connected to ground. VS terminal is connected to high voltage switch node / terminal 42. CS terminal constitutes output signal terminal 44.

[0007] Transistor 32 acts as a power switch for turning an external device and voltage (not shown) 'on' and 'off'. The configuration described in conjunction with FIG. 2A shows that IC 34 is connected in parallel with the drain terminal and source terminal of transistor 32. Therefore, IC 34 acts as a parallel-connected sensing circuit for measuring the voltage between the drain terminal and the source terminal of transistor 32 (in the case where transistor 32 is a MOSFET), that is, VDS(on), or the voltage between the collector terminal and the emitter terminal (in the case where transistor 32 is an IBGT), that is, VCE(on). The internal circuit of IC 34 (not shown, but available online) includes a MOSFET (HVFET), an RC delay circuit, and a p-type metal oxide semiconductor (PMOS) suppression transistor. Detailed information on IR25750 (including internal circuit diagrams (not shown) and operating principles) can be found in "Application Note AN-1199" published by International Rectifier via www.ifr.com. IC 34 uses gate drive input signal 40 to power the internal circuit and turn the internal circuit on and off.

[0008] With further reference to the timing diagram 50 of FIG. 2B , during the 'OFF' time of transistor 32, the gate drive input signal 40 is 'low' (i.e., at COM) and at the CS terminal. When the gate drive input signal 40 becomes 'high' (rectangular waveform), transistor 32 is turned on and the drain voltage decreases from the high voltage to V DS (on) decreases, which depends on the current flowing through transistor 32 and its characteristic R DS The gate resistor 36 enables the IC 34 to be turned on for a short time delay after the on time of the transistor 32. The output signal 44 at the CS output terminal is a target current sensing signal that can be fed to an auxiliary circuit (not shown). Conversely, when the gate drive input signal 40 becomes 'low', the transistor 32 is turned off, then the HVFET is turned off, and the CS terminal is at COM.

[0009] The parallel-connected sensing circuit 30 (FIG. 2A) is considered to be an improvement over the conventional series-connected resistance sensing circuit 10 (FIG. 1), however, both circuits may exhibit noise spikes due to switching. In addition, in the circuit 30, given that the output voltage V DS (on) depends on the current flowing through transistor 32 and R DS (on), thus requiring a "high" current (e.g., 2 amps) or a relatively "high" R DS (on) (e.g., 0.25Ω) to generate an output voltage V (approximately volts). Therefore, circuit 30 may require the use of additional gain blocks (e.g., amplifiers) as well as DS (on) varies with temperature (i.e., ambient and self-heating effects), so temperature compensation and management techniques need to be utilized.

[0010] Another prior art technique for sensing current employs a current transformer connected to a sensing circuit that includes a sensing resistor through which the current intended to be sensed flows. Reference is now made to FIGS. 3A , 3B , 3C , and 3D . FIG. 3A is a schematic diagram of a current sensing circuit in the prior art that employs a current transformer to measure current, generally designated 60 . FIG. 3B is a schematic diagram of a simplified equivalent circuit of a portion of the current sensing circuit of FIG. 3A , generally designated 80 . FIG. 3C is a schematic diagram of an intermediate output voltage waveform corresponding to the operation of the current sensing circuit of FIG. 3A , generally designated 90 . FIG. 3D is a schematic diagram of an output voltage waveform corresponding to the operation of the current sensing circuit of FIG. 3A , generally designated 96 . The current sensing circuit 60 ( FIG. 3A ) includes a current transformer 62 , a diode 72 , a reset resistor 70 , and a sensing resistor 74 (R s ). The current transformer 62 includes a primary winding 64 (e.g., having one winding) electromagnetically coupled to a secondary winding 66 (e.g., typically a plurality of windings). The inductance of the secondary winding is represented by an inductor 68 (while its parasitic capacitance (i.e., parasitic capacitance between its windings)) is not shown in FIG. 3A . The anode of a diode 72 is connected to one terminal of the secondary winding 66, while the cathode of the diode 72 is connected to a sense resistor 74, which in turn forms a closed circuit with a second terminal of the secondary winding 66.

[0011] Marked as I in The time-varying input current 76 (eg, pulses) flows through the primary winding 64, thereby generating a voltage, which in turn electromagnetically induces an induced voltage in the secondary winding 66, that is, the voltage V 2 3C and represented by the intermediate output voltage waveform 90. The reset resistor 70 is used to allow time for the intermediate output voltage V 2 (following each input pulse) so that the transformer 62 is not saturated. FIG3B shows an equivalent circuit 80 of a portion of the current sensing circuit 60 of FIG3A. The equivalent circuit 80 represents certain non-ideal aspects of the non-ideal transformer 62, whose secondary winding 66 (and primary winding 64) exhibits a voltage drop of 0.1 V. The voltage drop of 0.1 V is 0.1 V. T The resistance represented by the equivalent capacitor C T FIG3B also shows the parasitic capacitance represented by the inductor L Ts The inductance 68 of the secondary winding 66 represented by the capacitor C D The capacitance of the diode 72 represented by the resistor R D (and reset resistor R r and the sensing resistor R s ) represents the resistance of diode 72. 1 to2 The 'ON' time between (Fig. 3C) (corresponding to the input current I in The on-time (FIG. 3A)) is characterized by the current I through the reset resistor 70. m and the voltage V of the intermediate output voltage waveform 90 (FIG. 3C) 2 The positive component of 92. Input current I in The 'OFF' time in the equivalent circuit 80 I m Component L Ts , R T , C T and R r The subcircuit formed is characterized by a current flowing in the opposite direction to the current and a voltage V of the intermediate output voltage waveform 90. 2 The diode 72 rectifies the negative component 94 of the intermediate output voltage waveform 90. The output voltage V Out (Figure 3A, Figure 3D) is the voltage V 2 and the forward voltage drop of diode 72 (given by V D represents) and (that is, V Out =V 2 +V D ), as represented by output voltage waveform 96. The value of reset resistor 70 is usually selected so that it is not much larger than the value of RT, otherwise if R r >>R T , then most of the current I m will effectively flow through the secondary winding 66. In addition, the value of the sense resistor 74 is typically selected so that it is not much larger than the value of the reset resistor 70, otherwise if R s >>R r , then most of the current will pass through the reset resistor 70. Therefore, the product of the low value resistance of the sense resistor 70 and the low value current flowing therethrough produces a low value voltage drop across the sense resistor 70, so it is usually necessary to amplify the output voltage V Out . Summary of the invention

[0012] The purpose of the disclosed technology is to provide a novel device for detecting and measuring at least one of a pulse voltage source. The device includes a transformer and a differentiator. The transformer has a primary winding electromagnetically coupled to a secondary winding. The primary winding has a first inductance. The differentiator includes at least one resistive element (e.g., a resistor) coupled to the primary winding, thereby forming a closed circuit with the pulse voltage source. The differentiator is configured to electromagnetically induce an induced pulse voltage on the secondary winding, so that the induced pulse voltage indicates the pulse voltage source.

[0013] According to another aspect of the disclosed technology, a device for detecting and measuring at least one of a pulse voltage source is provided. The device includes a differentiator and a transformer. The differentiator includes at least one resistive element (e.g., a resistor) coupled to at least one capacitive element (e.g., a capacitor), thereby forming a closed circuit with the pulse voltage source. The transformer has a primary winding electromagnetically coupled to a secondary winding. The primary winding is connected in parallel with the at least one resistive element. The primary winding is configured to electromagnetically induce an induced pulse voltage on the secondary winding, so that the induced pulse voltage indicates a pulse voltage source.

[0014] According to another aspect of the disclosed technology, a device for detecting and measuring at least one of a pulse voltage source is provided. The device includes a transformer having a primary winding electromagnetically coupled to a secondary winding. The primary winding has a first inductance. The resonant circuit includes at least one capacitive element (e.g., a capacitor) forming an LC circuit with the primary winding. The resonant circuit forms a closed circuit with the pulse voltage source. The primary winding is configured to electromagnetically induce an induced pulse voltage on the secondary winding, so that the induced pulse voltage indicates a pulse voltage source.

[0015] According to another aspect of the disclosed technology, a method for performing at least one of detecting and measuring a pulse voltage source is provided. The method comprises: differentiating a pulse signal of the pulse voltage source via a differentiator; up-converting the differential signal; and performing at least one of detecting and measuring the up-converted differential signal. The inductive component of the differentiator is a winding of a transformer so that the differentiation generates a differential signal. The up-converting of the differential signal via the transformer generates an up-converted differential signal. The up-converted differential signal indicates a pulse voltage source. BRIEF DESCRIPTION OF THE DRAWINGS

[0016] The disclosed technology will be more fully understood and appreciated from the following detailed description taken in conjunction with the accompanying drawings, in which:

[0017] FIG. 1 is a schematic diagram of a series sensing circuit for measuring current through a switch implemented by a transistor in the prior art;

[0018] FIG. 2A is a schematic diagram of a prior art parallel sensing circuit for measuring current through a switch implemented by a transistor;

[0019] FIG. 2B is a timing diagram corresponding to the operation of an integrated circuit (IC) in the parallel-connected sensing circuit of FIG. 2A ;

[0020] FIG3A is a schematic diagram of a current sensing circuit that uses a current transformer to measure current in the prior art;

[0021] FIG3B is a schematic diagram of a simplified equivalent circuit of a portion of the current sensing circuit of FIG3A ;

[0022] FIG3C is a schematic diagram of an intermediate output voltage waveform corresponding to the operation of the current sensing circuit of FIG3A;

[0023] FIG3D is a schematic diagram of an output voltage waveform corresponding to the operation of the current sensing circuit of FIG3A ;

[0024] Figure 4A is a schematic diagram of an apparatus for derivative voltage and current sensing of a pulsed voltage source, constructed and operative in accordance with an embodiment of the disclosed technology;

[0025] Figure 4B According to the disclosed technology Figure 4A A schematic diagram of the basic equivalent circuit of the device;

[0026] Figure 4C is a schematic diagram of a characteristic curve graph of an output voltage varying with time according to the disclosed technology together with a curve graph of an input time-dependent pulse signal of a voltage source;

[0027] Figure 5 is a schematic diagram of an exemplary implementation of an apparatus 100 for performing derivative voltage and current sensing of and through a transistor-implemented switch for switching voltage and current according to the disclosed technology;

[0028] Fig. 6A is a schematic diagram of an apparatus for derivative voltage and current sensing of a pulsed voltage source constructed and operative in accordance with another embodiment of the disclosed technology;

[0029] Figure 6B is constructed and operated according to embodiments of the disclosed technology Fig. 6A A schematic diagram of a simulation circuit of the device shown;

[0030] Figure 6C Is Figure 6B A graph of the output voltage generated by the simulation circuit;

[0031] Fig.6D yes Figure 6B A curve diagram of the output voltage of the simulation circuit varying with the rise time of the input voltage of the pulse voltage source;

[0032] Fig. 6E yes Figure 6B The simulation circuit is Fig. 6A A graph of the output voltage of a device as the value of the capacitor varies;

[0033] Figure 7is a schematic diagram of an apparatus for derivative voltage and current sensing of a pulsed voltage source constructed and operative in accordance with further embodiments of the disclosed technology;

[0034] Fig. 8A is a schematic diagram of an apparatus for performing derivative voltage and current sensing of a pulsed voltage source using a resonant circuit, constructed and operative in accordance with another embodiment of the disclosed technology;

[0035] Figure 8B yes Fig. 8A A schematic diagram of a basic model of an equivalent circuit of a resonant circuit of a device in FIG.

[0036] Figure 8C It is a schematic diagram of a characteristic curve diagram of an input time-dependent pulse signal of a voltage source;

[0037] Fig.8D is based on Fig. 8A a characteristic graph of a time-dependent voltage drop across a primary winding of a transformer for a first model configuration of a resonant circuit of the device;

[0038] Fig. 8E is based on Fig. 8A a characteristic graph of a time-dependent voltage drop across a primary winding of a transformer for a second model configuration of a resonant circuit of the device;

[0039] Fig.8F According to the embodiments of the disclosed technology Fig. 8A A characteristic curve diagram of the output voltage of the device varying with time;

[0040] Fig.9A is a schematic diagram of an exemplary implementation of an auxiliary output detection and measurement comparator circuit configured for use in conjunction with an apparatus of the disclosed technology;

[0041] Fig. 9B is a schematic diagram of another exemplary implementation of an auxiliary output peak / envelope detector circuit configured for use in conjunction with an apparatus of the disclosed technology;

[0042] Fig. 9C is a schematic diagram of another exemplary implementation of an auxiliary output comparator peak detector configured for use in conjunction with an apparatus of the disclosed technology; and

[0043] Fig.10 is a schematic diagram of a method for performing at least one of detecting and measuring a pulse voltage source according to the principles of the disclosed technology. DETAILED DESCRIPTION

[0044] The disclosed technology overcomes the shortcomings of the prior art by providing a device for detecting and measuring at least one of the following in a derivative manner through the "unconventional" use of a transformer as will be explained in detail herein: (1) the voltage of a voltage source; and (2) the current flowing through the voltage source. A transformer is generally a device configured and operated to transfer electrical energy between at least two circuits via electromagnetic induction. Several common uses of transformers include stepping down and stepping up of voltage and current, electrical isolation between circuits, impedance matching, signal filtering, alternating current (AC) phase angle adjustment, etc. The disclosed technology uses the transformer both in a conventional manner (i.e., for transferring electrical energy between circuits) and more importantly in an unconventional manner. In view of a transformer including a primary winding and a secondary winding that are electromagnetically coupled to each other, the transformer is configured to transfer electrical energy between the primary winding and the secondary winding (i.e., depending on the characteristics of the energy source applied to at least one of the windings). According to one embodiment of the disclosed technology, the unconventional use of the transformer includes utilizing the primary winding as an inductive element in a differentiator circuit. Thus, the primary winding is a dual common main component of both the transformer and the differentiator circuit. According to another embodiment of the disclosed technology, the primary winding is an inductive element in the resonant circuit.

[0045] Specifically, according to an embodiment of the disclosed technology, a device for detecting and measuring at least one of a pulsed voltage source (i.e., also interchangeably labeled as a "pulsed voltage source" herein) characterized as having a constant voltage that does not vary with time is provided. The device includes a transformer and a differentiator. The transformer includes a primary winding (having a first inductance) electromagnetically coupled to a secondary winding. The differentiator includes at least one resistive element (e.g., a resistor, a plurality of resistors having an effective resistance) coupled to the primary winding, thereby forming a closed circuit with the pulsed voltage source. Therefore, the main inductive component of the differentiator is common to a portion of the transformer (i.e., its primary winding). The differentiator is configured to electromagnetically induce an induced pulse voltage (i.e., a pulse voltage that has been electromagnetically induced via the principle of electromagnetic induction) on the secondary winding, wherein the induced pulse voltage indicates the pulsed voltage source.

[0046] The disclosed technology generally addresses the need to provide voltage and current sensing in power electronic systems for the purposes of measurement (e.g., deriving data), control (e.g., power regulation, feedback, etc.), monitoring (e.g., self-monitoring of power supply circuits), and protection and safety (e.g., for protection against overload conditions, short circuits, etc.). For example, there is a need to obtain information related to current through a particular high voltage switching device (e.g., a high voltage transistor or a series of interconnected high voltage switching transistors).

[0047] To further disclose this embodiment of the disclosed technology in more detail, reference is now made to Figure 4A , Figure 4B and Figure 4C . Figure 4A is a schematic diagram of an apparatus for derivative voltage and current sensing of a pulsed voltage source, generally designated 100, constructed and operative in accordance with an embodiment of the disclosed technology. Figure 4B According to the disclosed technology Figure 4A A schematic diagram of a basic equivalent circuit of the device is generally designated 130 . Figure 4C 1 is a diagram of a characteristic curve of an output voltage varying with time according to the disclosed technique, plotted together with a curve of an input time-dependent pulse signal of a voltage source, generally designated 140. The apparatus 100 includes a transformer 104 and a differentiator 106. The transformer 104 includes a primary winding 108 and a secondary winding 110, wherein the primary winding and the secondary winding are electromagnetically coupled to each other. The primary winding 108 includes an N P windings, and having a first inductance L 1 (non-zero). The secondary winding 110 includes N S winding, and has a second inductance L 2 (can be zero). (N P and N S is a digital number). The differentiator 106 includes at least one (electrical) resistive element represented by a resistor 112 coupled to the primary winding 108. Therefore, the main inductive component in the differentiator 106 (i.e., the primary winding 108) is also part of the transformer 104. The device 100 also includes an input terminal 114 1 and 114 2 And output terminal 116 1 and 116 2 ,like Figure 4A The device 100 is configured and operated to receive a signal through the input terminal 114. 1 and 114 2 is coupled to an external time-varying voltage source 120, which is pulsed ("pulsed voltage source"), represented by Vs(t) and characterized as having an internal resistance during the 'ON' state, represented by R s (on). Figure 4A As shown, the differentiator 106, and in particular the series-coupled pair of the resistor 112 and the primary winding 108, is configured to form a closed circuit with the pulse voltage source 120. The differentiator 106 is configured and operated to electromagnetically induce an induced pulse voltage on the secondary winding 110, so that the induced pulse voltage indicates the pulse voltage source 120, as will be described in detail below. In addition, the device 100, and in particular the secondary inductor 110, is configured and operated to transmit the pulse voltage to the secondary winding 110 via the output terminal 116. 1 and 116 2 With external load 122(R L) (i.e., a "sense load") (e.g., a shunt resistor ("sense resistor" or "resistive load")) coupled, current can flow through the external load and can be detected and measured. The device 100 including peripheral components such as a voltage source 120 and an external load 122 is Figure 4A 102. It should be noted that the at least one resistive element effectively represented by resistor 112 may be embodied as a single resistor, a plurality of resistors (in series and / or in parallel), an impedance (i.e., ohmic resistance components and reactive components), or an equivalent device exhibiting resistive properties that is compatible with the principles of the disclosed technology.

[0048] Figure 4B The basic equivalent circuit 130 of the device 100 is shown. Figure 4A ), wherein the voltage source 120 is configured to provide a voltage to the differentiator 106 ( Figure 4A , Figure 4B ) of the series RL circuit applying a pulse signal 142 ( Figure 4C ), thus forming a current I P The closed circuit ( Figure 4B ). The pulse signal 142 is a time-varying signal, the waveform of which includes a rising edge and a falling edge. Figure 4C 142 is shown as a rectangular shape, but other pulse shapes (e.g., with rising and falling edges) are possible. It should be further noted that, in order to illustrate the main principles of the disclosed technology, the basic equivalent circuit 130 shows a simplified model that does not include the equivalent subcircuit of a "real" (i.e., non-ideal) transformer that exhibits leakage reactance, core losses (or magnetizing current losses, such as hysteresis), flux leakage, parasitic capacitance, etc. The resistance of the resistor 112 is represented as R, and the non-zero inductance of the inductor 108 is represented as L. 1 The (time-dependent) voltage drop across resistor 112 is denoted as V R (t)(abbreviated as V R ), and the voltage drop across the inductor 108 is represented as V L1 (t)(abbreviated as V L1 ), so that the following equation holds:

[0049]

[0050] The RL coupling point is denoted as V d (t)(abbreviated as V d ) voltage (which is also equal to V L1 ) is determined by solving the following differential equation:

[0051]

[0052] The time-varying current I in the inductor 108 P A changing magnetic field (not shown) is generated, the flux of which also propagates through the inductor 110, thereby generating a changing voltage (i.e., electromotive force (emf)) at the inductor 110. Therefore, an output voltage ("induced voltage") V is generated between the terminals of the inductor 110 by electromagnetic induction. O (t), where the maximum output voltage (i.e., ) is ideally given by:

[0053]

[0054] Where n represents the turns ratio of transformer 104, which has a conversion factor proportional to n (e.g., step-up ("up-conversion" n>1), step-down ("down-conversion": n<1), 1:1 conversion: (n=1)). Maximum output voltage Therefore, it is n times the maximum voltage on the inductor 108, that is, V L1 =V d , (taking into account the voltage drop V on resistor 112 R , whose minimum value is determined by V Rmin express). Therefore, the voltage of the pulse voltage source 120 (ie, the output signal V O (t) is used as a function of the input signal V s (t) indication or gauge). Figure 4C shows the output voltage V varying with time O (t) (denoted by 144), which is plotted together with a graph of the time-dependent input pulse signal 142 of the voltage source 120. In order to measure the current I S , the secondary inductor 110 is configured and operated to sense a load R , for example in the form of a sense resistor 122 L The current I can be detected and measured via a current measuring device (not shown) (eg, a current detection circuit, an ammeter, etc.). S Alternatively, by measuring the output voltage V O (t) and knowing the value of the sense resistor 122, the current I is determined (eg, via Ohm's law) S (ie, the current through the sense resistor 112). The device 100 does not need to use an amplifier to amplify the output voltage V O (t) signal (i.e., as is usually required in prior art devices), because Typically in the order of volts. For example, given n = 100 and a peak voltage V d ≈5mV, ideal (i.e., no energy loss) and due to the voltage drop across resistor 112 (V R) can be ignored, so the maximum output voltage (ie, on the order of volts).

[0055] The device 100 is implemented for derivative voltage and current sensing of a pulsed voltage source as a switching device (or its equivalent). For example, the switching device is configured and operated to quickly turn on and off a voltage source connected thereto, such that the switching device itself effectively becomes a source of a pulsed voltage, as referred to herein as a "pulsed voltage source". The switching device can be embodied in the form of an electronic switch, such as a switching circuit, a transistor (e.g., a bipolar, a metal oxide semiconductor field effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), etc.), a silicon controlled rectifier (SCR), a triode alternating current (TRIAC) switch, a diode alternating current (DIAC) switch, an electromagnetic switch (such as a relay), a mechanical switch (such as a toggle switch), a push button switch, and a physical property-related switch (i.e., having different physical properties (such as pressure, temperature, magnetic field, light intensity), such as in a pressure-related switch, a temperature-related switch, a magnetic field-related switch, a light intensity-related switch, etc.), etc. Alternatively, the pulse voltage of the pulse voltage source is generated by a current flowing through a resistive element (e.g., a resistor, a resistive load, a sensing resistor, etc.), thereby generating a voltage drop across the resistive element. The resistive element constitutes an internal resistance (R s (on)) (e.g., the internal resistance of a switching device such as a silicon (Si) MOSFET, a gallium nitride (GaN) high electron mobility transistor (HEMT), etc.). According to a specific implementation, the current passes through the sensing resistor due to the sensing resistor being connected in series with the switching device (i.e., the pulse voltage source). To further explain this exemplary implementation in detail, reference is now made to Figure 5 , which is a schematic diagram of an exemplary implementation of an apparatus 100 for derivative voltage and current sensing of and through a transistor-implemented switch for switching voltage and current in accordance with the disclosed technology, generally designated 150 .

[0056] Figure 5 The device 100 ( Figure 4A ) in which the input of the device 100 is coupled to a transistor-implemented switch, which in this example (and without loss of generality) is embodied as a MOSFET 152. The MOSFET 152 includes a gate terminal (G), a body terminal (B), a drain terminal (D), and a source terminal (S). Specifically, the input terminal 114 of the device 100 is coupled to a transistor-implemented switch, which in this example (and without loss of generality) is embodied as a MOSFET 152. 1 and 114 2 ( Figure 4A) are connected to the drain terminal and source terminal of MOSFET 152, respectively. The drain terminal of MOSFET 152 is connected to voltage source 154, and the source terminal is grounded. MOSFET 152 is configured and operated to supply a gate-source voltage (V GS ) signal (not shown) between an 'OFF' state (i.e., no conduction between the drain and source terminals) and an 'ON' state (i.e., conduction between the drain and source terminals). During the OFF state of MOSFET 152, there is no conduction between the drain and source terminals, and in effect, the switch is open, and no current flows between voltage source 154 and ground and no current flows to device 100. When MOSFET 152 transitions from the OFF state to the ON state, current begins to flow between the drain and source terminals, and device 100 is switched to an 'ON' state via input terminal 114. 1 and 114 2 ( Figure 4A ) receives a voltage pulse 156 as an input (e.g., having a rising edge from a 'high' voltage (absolute) value to a 'low' voltage (absolute) value). When MOSFET 152 is in the ON state, it is characterized as represented herein as R DS The internal resistance of the transistor (on) depends on the type of transistor and other factors such as temperature. Typical exemplary values ​​are R DS (on)≈1mΩ. The internal resistance R during the ON state DS (on) causes a voltage drop V on MOSFET 152 DS (on), and then the device 100 senses the voltage drop via the rising edge of the voltage pulse 156. Similarly, when the MOSFET 152 transitions from the ON state to the OFF state, the current stops flowing between the drain terminal and the source terminal, and the voltage pulse 156 changes from 'high' to 'low'. The device 100 senses the sudden voltage drop across the MOSFET 152 via the falling edge of the voltage pulse 156. As described above in conjunction with Figure 4A , Figure 4B and Figure 4C As illustrated, the voltage pulse 156 ( Figure 5 ) generates a corresponding voltage V at the terminals of the inductor 108 d =V L1 , and then a corresponding induced pulse voltage is induced at the secondary inductor 110, and the induced pulse voltage indicates the voltage pulse 156. According to the disclosed technology, by using equation (3), where V S ≡V DS , it is known that the resistor 112 has a R DS (on) and R values, and by measuring the output 158V from the device 100O , the current I flowing through MOSFET 152 is determined according to:

[0057] V Omax =I·R 方程 (on)·n≈I·R DS (on)·n=V DS (on)·n (4),

[0058] Where R 方程 (on) means including R DS (on), R (of resistor 112) and R L1 (the ohmic resistance of inductor 108, which is usually very low). Typically, R DS (on) is approximately equal to R 方程 (on)(i.e., R DS (on)≈R 方程 (on)), which is about one thousandth of a percent. For example, given R DS (on) = 1mΩ, R = 100Ω and R L1 =1Ω, resulting in: R 方程 (on)≈0.999 mΩ. In practice, since MOSFET 152 has a relatively low (on) resistance value, most of the current I flows through it. In this way, the device 100 is configured and operated to perform derivative voltage and current sensing of the voltage pulse 156 of the pulse voltage source (in this case, the switching device, which is MOSFET 152). Therefore, the output voltage 158 also indicates the current flowing through the pulse voltage source (i.e., in this example, the transistor-implemented switch, which is MOSFET 152).

[0059] In an alternative configuration (not shown), MOSFET 152 is replaced by a small value sense resistor (not shown) such that voltage source 154 is connected to the sense resistor and is configured and operated as a pulsed voltage source. The sense resistor may be selected to have an R value corresponding to the value of MOSFET 152. DS The same principles of the disclosed technology detailed above apply to this alternative configuration.

[0060] According to another embodiment of the disclosed technology, a direct current (DC) filter and a high voltage (HV) protector are provided, which are configured to be coupled between the resistor of the differentiator and the pulse voltage source. The DC filter and the HV protector can be implemented, for example, by capacitors. To further explain this embodiment in detail, reference is now made to Fig. 6A , Figure 6B and Figure 6C . Fig. 6Ais a schematic diagram of an apparatus for derivative voltage and current sensing of a pulsed voltage source, constructed and operative in accordance with another embodiment of the disclosed technology. Figure 6B is constructed and operated according to embodiments of the disclosed technology Fig. 6A Schematic diagram of the simulation circuit of the device shown. Figure 6C Is Figure 6B 200 ( Fig. 6A ) and device 100( Figure 4A ) is substantially the same as the device 200, except that the capacitor 204 is included in the device 200, and all components and their corresponding reference numerals remain unchanged. The device 200 is implemented for performing derivative voltage and current sensing of a pulsed voltage source (or its equivalent, such as a switching device). The device 200, including peripheral components such as the voltage source 120 and the external load 122, is Fig. 6A 112 and 120. The capacitor 204 is a DC decoupler configured and operated to filter out (i.e., block) the DC component (i.e., at least one non-time-varying DC component) of the pulse voltage source signal (thus avoiding potential transformer saturation), and to provide (to a certain extent) HV protection for the device 200. Since the differentiator 106 effectively generates an output signal according to the time-varying signal component of the pulse voltage source signal, the time-varying DC component of the pulse voltage source signal can be filtered out by the capacitor 204. If the pulse voltage source signal includes a high voltage DC component, the capacitor is configured and operated to provide HV protection to the rest of the device 200 by filtering out such high voltage. The capacitance value of the capacitor 204 is selected according to the characteristics of the pulse voltage source signal. In addition, in the case where the pulse voltage source 120 generates repeated periodic pulses, the capacitive reactance of the capacitor 204 (labeled as X c ) is inversely proportional to the frequency f of the pulse voltage source 120 according to the following terms: c = -1 / (2πfC), so that its reactance (absolute term) decreases with increasing frequency. Therefore, for a given capacitance C of capacitor 204, the higher the value of f, the greater the X c The value of will be larger.

[0061] Figure 6B The device 200 ( Fig. 6A ) of the simulation circuit (generally labeled 220), wherein the pulse voltage source 120 is simulated by I1 and resistor R5, and the capacitor 204 ( Fig. 6A ) by capacitor C1( Figure 6B ) simulation, transformer 104 ( Fig. 6A ) is composed of the transformer equivalent simulation subcircuit TRF1( Figure 6B) simulation, and the load resistor 122 ( Fig. 6A ) is simulated by resistor R3. The transformer equivalent simulation subcircuit includes inductor L1 (simulating primary winding 108), inductor L2 (simulating secondary inductor 110), resistor R1 (simulating resistor 112), resistor R2, capacitor C2, and resistor R4 (simulating the resistance and parasitic capacitance of transformer 104). It should be further noted that a real (i.e., non-ideal) transformer typically exhibits a coupling coefficient of k<1. Figure 6B Typical values ​​used in simulation circuit 220 are shown. Figure 6C The graph 230 in FIG. Figure 6B The simulation circuit of "V OUT The peak-to-peak output voltage is approximately 1.6 volts, so there is no need for an amplifier at the output.

[0062] Output voltage V OUT ( Figure 6B ) usually depends on the input voltage V of the pulse voltage source l1 IN To further illustrate this correlation, now refer to Fig.6D , which is Figure 6B The simulation circuit is changed by the input voltage V of the pulse voltage source IN The rise time of the output voltage V OUT Graph 250 shows the output voltage (in millivolts) on the vertical axis as a function of time (in nanoseconds) on the horizontal axis. The data points in graph 250 indicate the output voltage V of the pulse voltage source l1 as a function of the input voltage V IN The rise time of the output voltage V OUT A fairly flat (i.e., slight) sensitivity or dependence of . In general, the output voltage decreases as the rise time of the (input) pulse voltage source increases. Fig.6D In the exemplary graph shown, the value of capacitor C1 is 30 pF. Although the rise time shown ranges from 0 (excluding 0) to 300 ns (i.e., t r =(0,300]), but the device 100 of the disclosed technology ( Figure 4A ) and device 200( Fig. 6A ) can usually be used with a very narrow range of rise times, for example, t r =(0,100ns.], where the output voltage depends only slightly on the rise time.

[0063] Output voltage V OUT ( Figure 6B ) is also usually dependent on capacitor 204 ( Fig. 6A ) or C1( Figure 6B) value. To further illustrate this correlation, now further refer to Fig. 6E , which is Figure 6B The simulation circuit is Fig. 6A The value of the capacitor of the device changes the output voltage V OUT , generally designated 260. Graph 260 shows the capacitance of capacitor C1 ( Figure 6C ) in picofarads versus the output voltage (in volts) on the vertical axis. The data points in graph 260 generally indicate that the output voltage decreases as the capacitance value of capacitor C1 increases. Fig. 6E In the exemplary curve diagram shown, the rise time of the pulse voltage source l1 is t r =100ns.

[0064] According to a further embodiment of the disclosed technology, there is thus provided a differentiator coupled to a primary winding of a transformer, wherein the differentiator is separate from the transformer (i.e., the differentiator and the transformer have no shared components, i.e., they are mutually exclusive in terms of components). To further elaborate on this embodiment, reference is now made to Figure 7 , which is a schematic diagram of an apparatus for derivative voltage and current sensing of a pulsed voltage source, generally designated 270, constructed and operative in accordance with further embodiments of the disclosed technology.

[0065] Device 270 is similar to device 100 ( Figure 4A ), except that the differentiator 282 is separated from the transformer 104 (i.e., the differentiator 282 and the transformer 104 do not have shared components), the same components and their corresponding reference numerals remain the same. Specifically, the device 280 includes a differentiator 282 and a transformer 104, wherein the differentiator 282 is of RC type, which includes at least one capacitive element represented by a capacitor 284 (C3) (interchangeably represented) and at least one resistive element represented by a resistor 286 (R6) (interchangeably represented). One terminal of the capacitor 284 is connected to the resistor 286, and the other terminal is connected to the pulse voltage source 120. The transformer 104 includes a primary winding 108 and a secondary winding 110. The primary winding 108 is electromagnetically coupled with the secondary winding 110 of the transformer 104, so that the primary winding 108 is configured to electromagnetically induce an induced pulse voltage on the secondary winding 110. The primary winding 108 is connected in parallel to the resistor 286. Differentiator 282 forms a closed circuit with pulse voltage source 120. Device 280 is implemented for derivative voltage and current sensing of pulse voltage source 120 (or its equivalent, such as a switching device). Device 280 including peripheral components such as voltage source 120 and external load 122 is Figure 7 They are uniformly marked as 270.

[0066] The voltage source 120 is configured to apply a pulsed time-varying (input) signal (not shown) to the differentiator 282, which in turn is configured and operative to receive the pulsed time-varying input signal and differentiate it, thereby generating a derivative signal (not shown). The transformer 104 is configured and operative to up-convert (i.e., step up voltage conversion) the derivative signal received by the primary winding 108, thereby generating a derivative signal at the output terminal 116. 1 and 116 2 The up-converted differential signal (not shown) output at V O (t) indicates. The device 280 measures the output voltage of the indicating pulse voltage source to realize the control of the output terminal 116 1 and 116 2 Alternatively, the device 280 implements the detection and measurement of the up-converted differential signal at the sensing device 122 (eg, by the resistor R L The sensing device is connected to the output terminal 116. 1 and 116 2 coupling.

[0067] Therefore, according to another embodiment of the disclosed technology, a resonant circuit is provided, the resonant circuit having a capacitor, the capacitor forming an LC circuit with the primary winding of the transformer. The resonant circuit forms a closed circuit with the pulse voltage source. The primary winding of the transformer induces an induced pulse voltage on the secondary winding of the transformer, so that the induced pulse voltage indicates the pulse voltage source. To further explain this embodiment in detail, reference is now made to Fig. 8A , Figure 8B and Figure 8C . Fig. 8A is a schematic diagram of an apparatus for derivative voltage and current sensing of a pulsed voltage source utilizing a resonant circuit, generally designated 300, constructed and operative in accordance with another embodiment of the disclosed technology. Figure 8B yes Fig. 8A A schematic diagram of a basic exemplary equivalent circuit of a resonant circuit of a device in FIG. 1 is generally designated 330 . Figure 8C 3 is a schematic diagram of a characteristic curve diagram of an input time-dependent pulse signal of a pulse voltage source, generally designated 350. Except that (device 300) does not include resistor 112 and includes capacitor C4 in device 300 ( Fig. 8A ), the device 300 ( Fig. 8A ) all parts and devices 100 ( Figure 4A ) are substantially the same. Apparatus 300 is implemented for derivative voltage and current sensing of a pulsed voltage source (or its equivalent, such as a switching device).

[0068] Specifically, the device 300 includes a transformer 304 and a resonant circuit 306. The transformer 304 includes a primary winding 308 and a secondary winding 310, wherein the primary winding and the secondary winding are electromagnetically coupled to each other. Figure 4A ), the primary winding 308 includes N P windings, and having a first inductance L 3 (non-zero). The secondary winding 310 of the device 300 includes N S winding, and has a second inductance L 4 (may be zero). The resonant circuit 306 comprises at least one (electrical) capacitive (capacitive) element represented by capacitor 305 coupled to a primary winding 308, thereby forming a parallel LC circuit. Therefore, the main inductive component in the resonant circuit 306 is the same as the primary winding 308 of the transformer 304. The at least one capacitive element is typically embodied in the form of a capacitor. Alternatively, the at least one capacitive element is embodied as a plurality of capacitors coupled together (not shown) to form an effective capacitor. The device 300 also includes an input terminal 314 1 and 314 2 And output terminal 316 1 and 316 2 ,like Fig. 8A The device 300 is configured and operated to receive a signal via the input terminal 314. 1 and 314 2 is coupled to an external time-varying voltage source 320, which is pulsed ("pulsed voltage source"), represented by Vs(t) and characterized as having an internal resistance (ie, R) during the 'ON' state. s (on)). Fig. 8A As shown, the resonant circuit 306, and in particular the parallel coupled pair of capacitor 305 and primary winding 308, is configured to form a closed circuit with a pulse voltage source 320. The primary winding 308 is configured and operated to electromagnetically induce an induced pulse voltage on the secondary winding 310, so that the induced pulse voltage indicates the pulse voltage source 320, as will be described in detail herein below. Moreover, the device 300 and in particular the secondary inductor 310 are configured and operated to generate an induced pulse voltage via the output terminal 316. 1 and 316 2 With external load 322(R L ) (i.e., a "sense load"), current can flow through the external load and can be detected and measured. The device 300 including peripheral components such as a voltage source 320 and an external load 322 is Fig. 8AIt is collectively labeled as 302. It should be noted that at least one capacitive element is represented by capacitor 305, which can be embodied as a single capacitor, multiple capacitors (in series and / or in parallel), or an equivalent device exhibiting capacitance, which is compatible with the principles of the disclosed technology.

[0069] The pulse voltage source 320 is configured to apply a pulse signal 324 to the resonant circuit 306 , thereby forming a closed circuit with the pulse voltage source 320 . Figure 8C A characteristic curve graph 350 of a time-dependent input pulse signal of a pulse voltage source 320 is shown. The pulse signal 324 is a time-varying signal, and the waveform of the time-varying signal includes a rising edge and a falling edge. Without loss of generality, in order to illustrate the disclosed technology, the pulse signal 324 is selected as a square pulse with a duty cycle of 50%. The principle of the disclosed technology is also applicable to other types of pulse signals with different duty cycles (for example, a rectangular pulse with a duty cycle of 20%).

[0070] Figure 8B The resonant circuit 306 ( Fig. 8A ) is a basic model of an equivalent circuit 330. The equivalent circuit 330 includes: an inductor 332, which simulates the primary winding 308 ( Fig. 8A ) of non-zero inductance; capacitor 334, the capacitor simulating capacitor 305 ( Fig. 8A ) capacitance; resistor 336, the resistor simulating the resistance of the primary winding 308; and resistor 338, the resistor simulating the resonant circuit 306 ( Fig. 8A ). Resistors 336 and 338 represent the actual dissipative (resistive) elements of resonant circuit 306.

[0071] Given the time-dependent pulse signal 324 applied to the pulse voltage source 320, the accompanying time-dependent voltage drop (generally represented as V d (t)) usually depends on Figure 8B The values ​​of the components in the equivalent circuit 330 are shown. To further illustrate this correlation, further reference is now made to Fig.8D , Fig. 8E and Fig.8F . Fig.8D is based on Fig. 8A A schematic diagram of a characteristic graph of a time-dependent voltage drop across a primary winding of a transformer for a first model configuration of a resonant circuit of an apparatus is generally designated 360 . Fig. 8E is based on Fig. 8A A graphical representation of a characteristic graph of a time-dependent voltage drop across a primary winding of a transformer for a second model configuration of a resonant circuit of an apparatus is generally designated 370 . Fig.8FAccording to the embodiments of the disclosed technology Fig. 8A A characteristic graph of the output voltage of the device over time is generally designated 380 .

[0072] Without loss of generality, and in order to illustrate the principles of the disclosed technology, two exemplary configurations of resonant circuits will now be described. Fig.8D , the curve 360 ​​shows that according to the device 300 ( Fig. 8A ) of the resonant circuit 306 (given by the equivalent circuit 330 ( Figure 8B ) simulation) in response to the applied time-dependent pulse signal 324 ( Figure 8C )'s time-varying voltage drop (denoted as V d1 (t)) of the characteristic waveform 362. According to the first model configuration of the resonant circuit 306, the value of the resistor 336 has a relatively small resistance value (0.1Ω) so as to produce a slight dissipative effect on the resonant circuit 306. Other exemplary values ​​of the components are: capacitor 334 is 10μF; inductor 332 is 0.2μH; and resistor 338 is 0Ω. The frequency of the applied (periodic) pulse signal 324 is selected to be consistent with the resonant frequency f of the resonant circuit 306. 0 matching, the frequency is given by:

[0073]

[0074] According to another general alternative preference, select L 3 and C 3 Make the period T 0 =1 / f 0 is greater than the pulse width of the applied (periodic) pulse signal 324. The maximum voltage drop across the primary winding 308 (given by V d1max ) is usually determined by the maximum voltage of the pulse voltage signal 324 (denoted by V Smax ) and other factors, such as the rise and fall times of the pulse voltage signal 324.

[0075] According to the second exemplary model configuration, the resonant circuit 306 exhibits a greater dissipative effect, wherein the resistor 336 exhibits a value that is an order of magnitude greater (ie, 1Ω) than the first exemplary model of the resonant circuit 306 . Fig. 8E Shown according to Fig. 8A360. In this second exemplary model configuration, except for the change in the value of resistor 336, all values ​​of other components (including the resonant frequency) remain unchanged. The larger dissipative effect of resistor 336 affects the maximum voltage drop across primary winding 308, which is determined by V d2max indicates that the voltage drop is usually less than V d1max .

[0076] In response to the current flowing through the primary winding 308 (ie, having a non-zero inductance, L 3 ≠0), is configured and operated to generate a changing magnetic field, which propagates through the electromagnetically coupled secondary inductor 310, which in turn is output at the output terminal 316. 1 With 316 2 A time-varying output voltage V' is generated between O (t). Induction pulse output voltage V' O (t) indicates the input pulse voltage source Vs(t). Maximum output voltage V' Omax (Right now, ) is (ideally) given by:

[0077]

[0078] Where n is the turns ratio of transformer 304, and is the maximum voltage on the primary winding 308. Therefore, the maximum output voltage V' Omax is n times the voltage across inductor 308 , which in turn dictates the voltage of pulse voltage source 320 . Fig.8F An embodiment according to the disclosed technology is shown. Fig. 8A 380. In order to measure the current flowing through the secondary winding 310, the device 300 is configured and operated to output the current through the output terminal 316. 1 and 316 2 With a sensing load R, for example in the form of a sensing resistor 322 L The current flowing through the closed circuit formed by the secondary winding 310 and the sense resistor 322 can be detected and measured via a current measuring device (not shown) (e.g., a current detection circuit, an ammeter, etc.). (Note that Fig.8F Not relative to Fig. 8E and 8E shown to scale).

[0079] The device of the disclosed technology, specifically the device 100 ( Figure 4A )、device 200( Fig. 6A )、device 280( Figure 7 ) and device 300( Fig. 8A ) are each configured and operative to be used in conjunction with various auxiliary output detection and measurement circuits configured and operative to at least sense, condition or modify an output signal from each of the devices 100, 200, 280 and 300 in order to generate data at least related to or indicative of a pulsed voltage source. Reference is now made to Fig.9A , Fig. 9B and Fig. 9C . Fig.9A is a schematic diagram of an exemplary implementation of an auxiliary output detection and measurement comparator circuit configured for use in conjunction with an apparatus of the disclosed technology, generally designated 400 . Fig. 9B is a schematic diagram of another exemplary implementation of an auxiliary output peak / envelope detector circuit, generally designated 420, configured for use with an apparatus of the disclosed technology. Fig. 9C is a schematic diagram of another exemplary implementation of an auxiliary output comparator peak detector configured for use in conjunction with an apparatus of the disclosed technology, generally designated 440 .

[0080] refer to Fig.9A , the exemplary implementation 400 includes a pulse voltage source 402 (similar to 120 ( Figure 4A ); Alternatively, similar to MOSFET 152 ( Figure 5 ); alternatively, a shunt resistor), the pulse voltage source is input to the device 100 ( Figure 4A )、device 200( Fig. 6A ) and device 300( Fig. 8A ) (denoted by "100 / 200 / 300" in this article), sense (load) resistor R L 404 and any of the comparators 406. The pulse voltage source 402 provides a pulse source (or a pulse) to the input of any of the devices 100, 200, 280 and 300, and its output V1 out 4. A pulse voltage source 402 according to the above-described embodiment of the corresponding device is indicated. A sense resistor 404 is coupled to an output terminal of one of the devices 100, 200, 280, and 300, and one terminal of a comparator 406 is coupled to the output terminal (and the sense resistor 404), as shown in FIG. Fig.9A The comparator 406 is configured and operates to convert the output voltage V1 out With reference voltage V ref. Compare and generate output V1 aux-out , the output depends on V1 out With V ref.For example, if V1 out >V ref , then V1 aux-out =V1 out , and if V1 out <V ref , then V1 aux-out =V ref. (i.e., alternatively, V1 aux-out =0). Alternatively, comparator 406 is configured and operated to compare between currents (not shown). Thus, implementation 400 may indicate whether the output of one of devices 100, 200, 280, and 300 is greater than or less than a reference value (e.g., a threshold value, a user-selected value, etc.). Comparator 406 may be implemented by an operational amplifier, a dedicated integrated comparator, a comparator circuit including a transistor, an integrated circuit (IC), etc.

[0081] Fig. 9B An exemplary implementation 420 is shown, which includes a pulse voltage source 422 (similar to pulse voltage source 402 ( )) input to one of the devices 100, 200, 280, and 300, a buffer 424, a diode 426, a capacitor 428, a resistor 430, and a switch 432. Fig.9A )). The pulse voltage source 422 provides a pulse source to the input of any one of the devices 100, 200, 280 and 300, and its output V2 out Indicates a pulse voltage source 402 according to the above embodiment of the corresponding device. Output V2 out 426 is used as an input to buffer 424. Buffer 424 is connected to the anode terminal of diode 426. The cathode terminal of diode 426 is connected to one terminal of capacitor 428, while the other terminal of capacitor 428 is grounded. Switch 432 is configured and operated to switch between utilization and non-utilization of resistor 430. When switch 432 is closed, an RC circuit is formed by capacitor 428 and resistor 430. Buffer 424 is generally configured and operated to have its input V2 out Buffer 424 outputs an output signal (not shown) that serves as an input to diode 426, which in turn rectifies the output signal, the output of which charges capacitor 428. When switch 432 is open, diode 426 and capacitor 428 are configured and operate to act as a peak (voltage) detector, such that when capacitor 428 is charged to a peak voltage, output V2 aux-out The output is maintained at the voltage peak. Diode 426 conducts only when forward biased, causing capacitor 428 to charge to a new peak value (taking into account the forward voltage drop of the diode, which depends on the diode, e.g., 0.6V). When V2aux-out When the output voltage peak (at the cathode) is greater than the input voltage (at the anode), diode 426 is reverse biased and does not conduct current from capacitor 428 to the input, thus maintaining or retaining the output at the same output voltage peak (i.e., minus the forward voltage drop of the diode). Alternatively, the polarity of diode 426 is reversed, thereby acting as a negative voltage peak detector (not shown) in conjunction with non-polarized capacitor 428.

[0082] When switch 432 is closed, diode 426, capacitor 428 and resistor 430 are configured and operate as an envelope detector whose output V2 is increased when the input voltage (at the anode) drops below the output voltage. aux-out The voltage V2 decreases according to the characteristic RC time constant, which determines the decay time. Conversely, when the input voltage rises above the output voltage (taking into account the forward voltage drop of the diode), the diode 426 is forward biased and V2 aux-out increases, thus outputting the envelope of the input signal.

[0083] Fig. 9C An exemplary implementation 440 is shown, which includes a pulse voltage source 442 (similar to pulse voltage source 422 ( )) input to one of the devices 100, 200, 280, and 300, a buffer 444, a capacitor 446, a switch 448, and a capacitor 450. Fig. 9B )). The pulse voltage source 442 provides a pulse source to the input of any one of the devices 100, 200, 280 and 300, and its output V3 out Indicates a pulse voltage source 422 according to the above embodiment of the corresponding device. Output V3 out Serves as an input to the buffer 444 and one terminal of the comparator 446. The other terminal of the comparator 446 is connected via a reference voltage V ref. When switch 448 is closed, the output of buffer 444 is connected to one terminal of capacitor 450, which constitutes output V3 aux-out , while the other terminal of capacitor 450 is grounded. Implementation 440 includes a sample and hold (i.e., interchangeably referred to as a "follow and hold") circuit (i.e., having a "sample" and "hold" mode). Implementation 440 is more efficient than implementation 420 ( Fig. 9B ) exhibits a relatively faster following response voltage envelope. In the sampling mode, switch 448 is closed, and buffer 444 charges capacitor 450, discharges capacitor 450, or keeps the capacitor voltage constant so as to be equal to V3 outIn the hold mode, switch 448 is open and capacitor 450 retains its charge (i.e., eventually and typically it may discharge due to leakage current). Comparator 446 is configured to out With V ref. and is configured to operate according to the comparison result (and V ref. ) to control the switching operation of switch 448. Alternatively, other auxiliary devices may be coupled to the output terminals (not shown) of at least one of devices 100, 200, 280, and 300, such as a voltage analyzer, a current analyzer, a voltmeter, an ammeter, more generally a voltage / current sensing device, etc.

[0084] The devices 100, 200, 280, 300 of the disclosed technology can be used to quickly detect short circuit conditions or events in monitored devices. Generally, the ability of modern wide bandgap semiconductor power transistors to withstand avalanche operation modes is limited compared to silicon (Si) counterparts. The devices 100, 200, 280 and 300 of the disclosed technology can be incorporated into protection circuits with very short reaction times, which enable power transistor-based power electronic systems to be saved and protected in the event of a short circuit event. Specifically, due to the input V IN The rise time of the pulse signal and the corresponding output induction pulse voltage signal V OUT The fast response between the output inductive pulse voltage and the short circuit condition or event (i.e., characterized by a rise in current (fast rate of rise) and a corresponding fall in voltage (fast rate of fall)) can be indicated.

[0085] Now further reference Fig.10 , which is a schematic block diagram of a method for performing at least one of detecting and measuring a pulse voltage source according to the principles of the disclosed technology, generally designated as 500. The method 500 starts with process 502. In process 502, a pulse signal of a pulse voltage source is differentiated via a differentiator, the inductive component of the differentiator being a winding of a transformer, thereby generating a differential signal. Figure 4A , Figure 4B and Figure 4C , through the differentiator 106 to the pulse voltage source 120 ( Figure 4A ) of the pulse signal V s (t) is differentiated to generate a differential signal (V d (t)), the inductive component of the differentiator is the winding (primary winding) 108 of the transformer 104 ( Figure 4A and Figure 4B ). According to the substitution process ( Fig.10(not shown), the pulse signal of the pulse voltage source is differentiated via a differentiator, and the RC circuit of the differentiator is separated from the transformer (mutually exclusive components). Figure 7 The differentiator 282 differentiates the pulse signal of the pulse voltage source 120 to generate a differential signal V d (t).

[0086] In process 504, the differential signal is up-converted via a transformer, thereby generating an up-converted differential signal. Figure 4A , Figure 4B and Figure 4C , through transformer 104 ( Figure 4A and Figure 4B ) to the differential signal (V d (t)) is up-converted ("boosted") to generate an up-converted differential signal V O (t), Figures (4A, 4B and 4C). Specifically, Figure 4C shows the output voltage V varying with time O A characteristic curve graph of (t) (denoted by 144) is plotted together with a curve graph of the time-dependent input pulse signal 142 of the pulse voltage source 120.

[0087] In process 506, at least one of detecting and measuring the up-converted differential signal is performed, wherein the up-converted differential signal is indicative of the pulse voltage source. Figure 4A , Figure 4B and Figure 4C , the up-converted differential signal V O (t) can be connected via an external resistive load 122 (R L) (i.e., "sense load," "sense resistor," all of which are interchangeable terms) detects and measures the voltage drop of the indicating pulse voltage source across a resistive load 122 and the current of the indicating pulse voltage source flowing through the resistive load 122, the external resistive load being coupled to the output terminal 116 of the device 200. 1 and 116 2 . Further references Fig.9A , Fig. 9B and Fig. 9C , auxiliary output detection and measurement comparator circuit 400 ( Fig.9A ) is used to detect and measure the pulse voltage source 402. Alternatively, the auxiliary output peak / envelope detector 420 ( Fig. 9B ) is used to detect the pulse voltage source 422. Further alternatively, the auxiliary output comparator peak detector 440 ( Fig. 9C ) is used to measure and detect the pulse voltage source 442.

[0088] Those skilled in the art will appreciate that the disclosed technology is not limited to the technology that has been specifically shown and described above. Instead, the scope of the disclosed technology is limited only by the appended claims.

Claims

1. A device (100) for detecting and measuring at least one of a pulse current flowing through a transistor (152), the device include: The transistor exhibits an on-resistance value; a transformer (104) having a primary winding (108) electromagnetically coupled to a secondary winding (110), the primary winding having a first inductance; and a differentiator (106), the differentiator (106) comprising at least one resistive element (112) coupled to the primary winding, thereby forming a closed circuit with the transistor, the differentiator being configured to electromagnetically induce an induced pulse voltage on the secondary winding; The induced pulse voltage indicates the pulse current and is related to the on-resistance value.

2. The apparatus of claim 1, wherein the at least one resistive element is a resistor coupled in series with the primary winding. The apparatus of claim 1 , wherein the first inductance is non-zero.

4. The apparatus of claim 1, wherein the secondary winding has an output terminal configured to couple to an external resistive load (122).

5. The device according to claim 1, wherein a maximum value of the induced pulse voltage is proportional to a turns ratio between the secondary winding and the primary winding.

6. The apparatus of claim 5, wherein the maximum value is of the order of a single volt.

7. The apparatus of claim 4, wherein the external resistive load enables measurement of current flowing therethrough.

8. The apparatus of claim 1 , wherein the transistor is selected from the list consisting of: Universal switch; Mechanical switch; Electronic switches; Solenoid switches; and Electromechanical switch.

9. The apparatus of claim 1, wherein the apparatus further comprises a direct current (DC) decoupler (204) between the at least one resistive element and the transistor, the DC decoupler being configured to block at least one time-invariant DC component of the pulse current.

10. The apparatus of claim 9, wherein the DC decoupler is a capacitor coupled in series with the at least one resistive element.

11. The apparatus of claim 9, wherein the DC decoupler is configured to provide high voltage (HV) protection to the apparatus.

12. The apparatus of claim 9, wherein the peak-to-peak voltage of the induced pulse voltage exceeds one volt.

13. The device of claim 1, wherein the secondary winding has an output terminal configured to couple with at least one auxiliary device selected from the list consisting of: Comparator; Peak / envelope detectors; Comparator peak detector; Voltage analyzer; Current analyzer; voltmeter; and Ammeter.

14. The device of claim 1, wherein the transistor is a metal oxide semiconductor field effect transistor (MOSFET).

15. A device (300) for at least one of detecting and measuring a pulse current flowing through a transistor (152), the device include: The transistor exhibits an on-resistance value; a transformer (304) having a primary winding (308) electromagnetically coupled to a secondary winding (310), the primary winding having a first inductance; and a resonant circuit (306), the resonant circuit (306) comprising at least one capacitive element (305) forming an LC circuit with the primary winding, the resonant circuit forming a closed circuit with the transistor, the primary winding being configured to electromagnetically induce an induced pulse voltage on the secondary winding; The induced pulse voltage indicates the pulse current and is related to the on-resistance value.

16. The apparatus of claim 15, wherein the at least one capacitive element is a capacitor coupled in parallel with the primary winding. The apparatus of claim 16 , wherein the first inductance is non-zero.

18. The apparatus of claim 16, wherein the secondary winding has an output terminal configured to couple to an external resistive load.

19. The device of claim 15, wherein a maximum value of the induced pulse voltage is proportional to a turns ratio between the secondary winding and the primary winding.

20. The apparatus of claim 19, wherein the maximum value is on the order of a single volt.

21. The apparatus of claim 18, wherein the external resistive load enables measurement of current flowing therethrough.

22. The apparatus of claim 15, wherein the transistor is selected from the list consisting of: Universal switch; Mechanical switch; Electronic switches; Solenoid switches; and Electromechanical switch.

23. The apparatus of claim 15, wherein a resonant frequency of the resonant circuit is selected to match a frequency of the pulsed current.

24. The device of claim 15, wherein the secondary winding has an output terminal configured to couple with at least one auxiliary device selected from the list consisting of: Comparator; Peak / envelope detectors; Comparator peak detector; Voltage analyzer; Current analyzer; voltmeter; and Ammeter.

25. A device (280) for at least one of detecting and measuring a pulse current flowing through a transistor (152), the device include: The transistor exhibits an on-resistance value; a differentiator (282) comprising at least one resistive element (286) coupled to at least one capacitive element (284) thereby forming a closed circuit with the transistor, and A transformer (104), the transformer (104) having a primary winding (108) electromagnetically coupled to a secondary winding (110), the primary winding being connected in parallel with the at least one resistive element, the primary winding being configured to electromagnetically induce an induced pulse voltage on the secondary winding; The induced pulse voltage indicates the pulse current and is related to the on-resistance value.

26. The device of claim 25, wherein the at least one resistive element is a resistor.

27. The apparatus of claim 25, wherein the at least one capacitive element is a capacitor coupled in series with the at least one resistive element.

28. The apparatus of claim 25, wherein the first inductance is non-zero.

29. The device of claim 25, wherein a maximum value of the induced pulse voltage is proportional to a turns ratio between the secondary winding and the primary winding.

30. The apparatus of claim 29, wherein the maximum value is on the order of a single volt.

31. A method (500) for at least one of detecting and measuring a pulsed current flowing through a transistor, the transistor exhibiting an on-resistance value, the method include: regulating the switching of the pulse current by the transistor; Differentiating the pulse signal of the pulse current via a differentiator (502) to generate a differential signal, wherein the inductive component of the differentiator is a winding of a transformer; Up-converting the differential signal via the transformer (504) to generate an up-converted differential signal; as well as Performing (506) at least one of detecting and measuring the up-converted differential signal, wherein the up-converted differential signal is indicative of the pulse current and is related to the on-resistance value.

32. The method of claim 31 , wherein the winding is a primary winding of the transformer.

33. The method of claim 31, wherein the up-conversion has an up-conversion factor that is proportional to a turns ratio of the transformer.

34. The method of claim 33, wherein a maximum value of the induced pulse voltage is proportional to the turns ratio.

35. The method of claim 34, wherein the maximum value is on the order of a single volt.

36. The method of claim 31 , wherein at least one of the detecting and measuring is of a current flowing through a sensing device coupled to a secondary winding of the transformer.

37. The method according to claim 31, further comprising: include: At least one time-varying direct current (DC) component of the pulse current is blocked before differentiating the pulse signal of the pulse current.

38. The method of claim 37, wherein the blocking enables high voltage (HV) protection for the device.

39. The method of claim 37, wherein the peak-to-peak voltage of the induced pulse voltage exceeds one volt.

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