A power mixer capable of suppressing the third harmonic of local oscillator

Abstract

The present application discloses a power mixer capable of suppressing the third harmonic of a local oscillator, comprising a local oscillator signal link circuit, a transconductance stage circuit, a switch stage circuit, a common source and common gate stage circuit, and a load circuit. The local oscillator signal link circuit provides two local oscillator signals for the switch stage circuit, one of which has a phase difference of 60 degrees relative to the other. There are two transconductance stage circuits for converting a baseband voltage signal into a current signal. There are two switch stage circuits, which respectively use the two local oscillator signals to up-convert the current signals from the two transconductance stage circuits into the required radio frequency signals. There are two common source and common gate stage circuits, which respectively transmit the radio frequency signals output by the two switch stage circuits to the load circuit. The load circuit realizes the addition of the output currents of the two common source and common gate stage circuits. The function of suppressing the third harmonic is realized through a special phase difference design.

Description

Technical Field

[0001] The present application relates to the technical field of wireless communication, and in particular to a power mixer capable of suppressing the third harmonic of a local oscillator for a radio frequency transmitter. Background Art

[0002] In RF transceivers, the third harmonic of the local oscillator signal will deteriorate the performance of the wireless communication system. For example, after the third harmonic of the mixer output passes through the power amplifier, it will be mixed back to the required frequency band due to the nonlinearity of the power amplifier itself, deteriorating the efficiency and linearity of the RF transmitter. The third harmonic of the mixer local oscillator signal in the RF transmitter may also fall into the frequency band where the RF receiver is located, deteriorating the performance of the RF receiver.

[0003] See also Figure 1 The existing RF transmitter with direct up-conversion architecture includes a driver B1, a divider DT, a driver B2, an active mixer M1, a pre-power amplifier PPA, a power amplifier PA and an antenna from left to right. Drivers B1 and B2 are both composed of inverters to improve the signal driving capability. The local oscillator input signal f LO (t) After the signal is enhanced by driver B1, it is passed through the two-way frequency divider DT to generate four IQ orthogonal signals, and then the four signals are further enhanced by driver B2, which are four orthogonal local oscillator signals LO_0, LO_90, LO_180, and LO_270 with phase shifts of 0, 90, 180, and 270 degrees, respectively, and provided to the active mixer M1. The active mixer M1 up-converts the baseband signal from the baseband chip BB into a radio frequency signal, which is amplified by the pre-power amplifier PPA and the power amplifier PA respectively, and then transmitted through the antenna.

[0004] Figure 1 The RF transmitter of the direct up-conversion architecture shown has the following variations, each of which has certain defects.

[0005] Solution 1: Gilbert mixer and power amplifier. The disadvantage is that it does not have the function of suppressing the third harmonic of the local oscillator; the mixer load is an inductor, and the pre-power amplifier load also requires an inductor, the layout area is large, and the design cost is high.

[0006] Solution 2: Using 8-phase local oscillator drive A total of 3 passive harmonic rejection mixers and power amplifiers, such as Figure 2 shown. Figure 2 The 4-way divider DF is used to replace Figure 1 The two-way divider DT in the Figure 2 Two drives B2 and B3 are used to replace Figure 1 One of the drives in B2, Figure 2 The three passive mixers M1 to M3 replace Figure 1 An active mixer M1 in Figure 2 A 25% duty cycle circuit LOG1 is added between the driver B2 and the three passive mixers M1 to M3. Figure 2 A 25% duty cycle circuit LOG2 is added between the driver B3 and the three passive mixers M1 to M3. The disadvantages are: the traditional 8-phase local oscillator signal design requires at least one four-way divider to generate, and the divider design is more complex; three mixers are used to achieve the harmonic suppression function, and the layout area is large; The design is difficult and the local oscillator signal mismatch is sensitive to its load, that is, the mixer design is difficult; a passive mixer is used, so a 25% duty cycle circuit is required; the subsequent circuit needs to add a power amplifier to transmit power, which further increases the design difficulty.

[0007] Solution 3: Use a passive mixer and power amplifier that cancels the third harmonic of the local oscillator by inductor and capacitor resonance. The disadvantage is that it requires a large area of ​​passive components, a large layout area, and a power amplifier is also needed to transmit power.

[0008] In summary, designing a harmonic suppression power mixer that integrates frequency conversion, amplification and harmonic suppression functions, that is, one circuit simultaneously realizes the functions of mixer, power amplifier and harmonic suppression, will simplify the design of RF transmitter circuits and reduce costs. Summary of the invention

[0009] The technical problem to be solved by the present application is to propose a power mixer that can suppress the third harmonic of the local oscillator, that is, a circuit that combines the functions of a Gilbert mixer, a power amplifier and harmonic suppression.

[0010] In order to solve the above technical problems, the present application proposes a power mixer capable of suppressing the third harmonic of the local oscillator, comprising a local oscillator signal link circuit, a transconductance stage circuit, a switch stage circuit, a common source and common gate stage circuit and a load circuit. The local oscillator signal link circuit provides two local oscillator signals for the switch stage circuit, one of which has a phase difference of 60 degrees relative to the other. There are two transconductance stage circuits for converting the baseband voltage signal into a current signal. There are two switch stage circuits, which respectively use the two local oscillator signals to up-convert the current signals from the two transconductance stage circuits into the required radio frequency signals. There are two common source and common gate stage circuits, which respectively transmit the radio frequency signals output by the two switch stage circuits to the load circuit. The load circuit realizes the addition of the output currents of the two common source and common gate stage circuits. The function of suppressing the third harmonic is realized through a special phase difference design.

[0011] Furthermore, an active mixer is used in the switch stage circuit, and its gain is equal to the product of the transconductance of the transconductance stage and the load, and the product value is greater than 1.

[0012] Furthermore, the local oscillator signal link circuit includes a delay module, a plurality of drivers, two frequency dividers, a plurality of capacitors and a plurality of resistors. LO (t) is divided into two paths; one path remains unchanged and is called the local oscillator signal f LO (t), the other path passes through the delay module to obtain the delayed local oscillator signal f LO (tT / 6), where T represents the local oscillator signal f LO (t) period. The local oscillator signal f LO (t) After being amplified by a driver signal, a two-way frequency divider generates four IQ orthogonal signals. These four IQ orthogonal signals are further amplified by four drivers and then connected to four DC blocking capacitors. The output ends of these four DC blocking capacitors are four orthogonal local oscillator signals LOIp, LOIn, LOQp and LOQn, and finally connected to the mixer module; four resistors provide DC bias voltage to the four orthogonal local oscillator signals LOIp, LOIn, LOQp and LOQn. Delayed local oscillator signal f LO (tT / 6) After being enhanced by a driver signal, four IQ orthogonal signals are generated through a divider. The four IQ orthogonal signals are further enhanced by four drivers and then connected to four DC blocking capacitors. The output ends of the four DC blocking capacitors are four orthogonal local oscillator signals LOQp_60, LOQn_60, LOIn_60 and LOIp_60 with a delay of 60 degrees, and finally connected to the mixer module; four resistors provide DC bias voltage to the four orthogonal local oscillator signals LOQp_60, LOQn_60, LOIn_60 and LOIp_60 with a delay of 60 degrees.

[0013] Furthermore, the local oscillator signal f LO The Fourier expansion of (t) is Delayed local oscillator signal f LO The Fourier expansion of (tT / 6) is where ω LO Represents the angular frequency of the local oscillator signal; the sum of the two cancels out the third harmonic of the local oscillator signal.

[0014] Furthermore, the entirety of the transconductance stage circuit, the switch stage circuit and the common-source and common-gate stage circuit is composed of a mixer module consisting of two IQ orthogonal power mixers.

[0015] Further, the local oscillator signal LOIp is connected to the Ip terminal of mixer one, and the local oscillator signal LOIp_60 is connected to the Ip terminal of mixer two, at which time the phase difference between the input of the Ip terminal of mixer one and the input of the Ip terminal of mixer two is 60 degrees. The local oscillator signal LOIn is connected to the In terminal of mixer one, and the local oscillator signal LOIn_60 is connected to the In terminal of mixer two, at which time the phase difference between the input of the In terminal of mixer one and the input of the In terminal of mixer two is 60 degrees. The local oscillator signal LOQp is connected to the Qp terminal of mixer one, and the local oscillator signal LOQp_60 is connected to the Qp terminal of mixer two, at which time the phase difference between the input of the Qp terminal of mixer one and the input of the Qp terminal of mixer two is 60 degrees. The local oscillator signal LOQn is connected to the Qn terminal of mixer one, and the local oscillator signal LOQn_60 is connected to the Qn terminal of mixer two, at which time the phase difference between the input of the Qn terminal of mixer one and the input of the Qn terminal of mixer two is 60 degrees.

[0016] Furthermore, each IQ quadrature power mixer is composed of an I mixer and a Q mixer; in the I mixer, the transconductance stage circuit includes a pair of NMOS differential pairs of tubes, which convert the input differential baseband signal from the voltage domain to the current domain respectively; the transconductance stage circuit outputs the baseband signal current and enters the switch stage circuit, and the switch stage circuit includes two groups of NMOS differential pairs of tubes; under the drive of four orthogonal local oscillator signals, the pairs of tubes in the switch stage circuit sample the input signal from the transconductance stage circuit to achieve up-conversion; the RF signal output by the switch stage circuit enters the common source and common gate stage circuit, and the common source and common gate stage circuit is composed of two NMOS tubes, and its gate terminal is connected to a DC bias voltage; the Q mixer has the same structure as the I mixer; the I and Q mixers connect the up-converted RF current signals through the output ends of their respective common source and common gate circuits to achieve the addition of the I and Q signals in the current domain.

[0017] Furthermore, transistors in the switch-stage circuit of an IQ quadrature power mixer are driven by four orthogonal local oscillator signals LOIp, LOIn, LOQp and LOQn; transistors in the switch-stage circuit of another IQ quadrature power mixer are driven by four orthogonal local oscillator signals LOQp_60, LOQn_60, LOIn_60 and LOIp_60 delayed by 60 degrees; and the output currents of the two IQ quadrature power mixers have a 60-degree delay in the time domain.

[0018] Furthermore, the load circuit includes an adjustable capacitor, a transformer and a load resistor; the current signal passes through a resonant frequency selection network composed of the transformer and the adjustable capacitor, and the power is transferred to the load resistor through the impedance transformation ratio of the transformer to achieve signal output.

[0019] Furthermore, the adjustable capacitor is a programmable capacitor array, and different frequency bands are covered by adjusting the programmable capacitor array.

[0020] The technical effect achieved by the present application is good harmonic suppression effect, low design complexity, high efficiency and high linearity. BRIEF DESCRIPTION OF THE DRAWINGS

[0021] Figure 1 It is a schematic diagram of the structure of a traditional direct up-conversion transmitter.

[0022] Figure 2 It is an existing 8-phase local oscillator drive Schematic diagram of the structure of a direct up-conversion transmitter with a total of 3 passive harmonic rejection mixers.

[0023] Figure 3 The overall structure of the power mixer proposed in this application that can suppress the third harmonic of the local oscillator is shown in FIG. Figure 1 .

[0024] Figure 4 The overall structure of the power mixer proposed in this application that can suppress the third harmonic of the local oscillator is shown in FIG. Figure 2 .

[0025] Figure 5 It is a waveform diagram of the local oscillator signal and the delayed local oscillator signal.

[0026] Figure 6 It is a schematic diagram of the spectrum of the local oscillator signal and the delayed local oscillator signal.

[0027] Figure 7 It is a schematic diagram of the circuit structure of a single IQ quadrature power mixer.

[0028] Figure 8 It is a schematic diagram comparing the third harmonic suppression effect of the local oscillator signal of the present application and that of a common power mixer.

[0029] Explanation of the reference numerals in the figure: local oscillator signal chain circuit 1, transconductance stage circuit 2, switch stage circuit 3, common source and common gate stage circuit 4, load circuit 5. DETAILED DESCRIPTION

[0030] See also Figure 3The power mixer that can suppress the third harmonic of the local oscillator proposed in this application includes a local oscillator signal link circuit 1, a transconductance stage circuit 2, a switch stage circuit 3, a common source and common gate stage circuit 4 and a load circuit 5. The local oscillator signal link circuit 1 provides two local oscillator signals for the switch stage circuit 3, one of which has a phase difference of 60 degrees relative to the other. The input of the local oscillator signal link circuit 1 is the local oscillator input signal received from the frequency synthesizer. There are two transconductance stage circuits 2, which are used to convert the baseband voltage signal into a current signal. There are two switch stage circuits 3, which use the two local oscillator signals to up-convert the current signals from the two transconductance stage circuits 2 into the required radio frequency signals. There are two common source and common gate stage circuits 4, which respectively transmit the radio frequency signals output by the two switch stage circuits to the load circuit, so as to improve the reliability of the circuit and prevent the circuit from being broken down; and improve the local oscillator suppression. The load circuit 5 realizes the addition of the output currents of the two common source and common gate stage circuits 4, thereby realizing the third harmonic suppression of the local oscillator signal, and transmits the power to the output load to realize high power output. The switch stage circuit 3 adopts an active mixer, whose gain is equal to the product of the transconductance of the transconductance stage and the load. In the design, the value of this product is greater than 1, that is, the signal is amplified.

[0031] See also Figure 4 The delay module B0, drivers B1 and B6, dividers DT1 and DT2, drivers B2 to B5 and B7 to B10, capacitors C1 to C8, and resistors R1 to R8 constitute the local oscillator signal chain circuit 1. The mixer module composed of two mixer units M1 and M2 constitutes the transconductance stage circuit 2, the switch stage circuit 3 and the common source and common gate stage circuit 4. The adjustable capacitor C, transformer T1 and load resistor R behind the mixer module L A load circuit 5 is constructed.

[0032] See also Figure 4 In the local oscillator signal chain circuit 1, the local oscillator input signal f with a duty cycle of 50% LO (t) is divided into two paths. One path remains unchanged and is called the local oscillator signal f LO (t), the other path passes through the delay module B0 to obtain the delayed local oscillator signal f LO (tT / 6), where T represents the local oscillator signal f LO (t) period. The local oscillator signal f LO The Fourier expansion of (t) is Delayed local oscillator signal f LO The Fourier expansion of (tT / 6) is The expansion here omits higher harmonics above the third order, where ω LO Represents the angular frequency of the local oscillator signal. From the Fourier expansion of these two signals, it can be seen that the sum of the two can cancel the third harmonic of the local oscillator signal. Figure 5 and Figure 6 The following are the waveform and spectrum diagrams of the two signals respectively. Figure 6 In, f LO Represents the frequency of these two signals, 3f LO Indicates the frequency of the third harmonic of these two signals.

[0033] Local oscillator signal f LO (t) After the signal is enhanced by the driver B1, four IQ orthogonal signals are generated by the divider DT1. The four IQ orthogonal signals are further enhanced in driving capability by the drivers B2 to B5, and then connected to the DC blocking capacitors C1 to C4, respectively. The output ends of the DC blocking capacitors C1 to C4 are four orthogonal local oscillator signals LOIp, LOIn, LOQp and LOQn, respectively, and finally connected to the mixer module. The coupling mode of the local oscillator signal link circuit 1 and the mixer module is AC coupling, and the DC bias voltage LOvbias is provided to the four orthogonal local oscillator signals LOIp, LOIn, LOQp and LOQn respectively through the large resistors R1 to R4.

[0034] Delayed local oscillator signal f LO (tT / 6) path and local oscillator signal f LO The path of (t) is similar. The delayed local oscillator signal f LO (tT / 6) After the signal is enhanced by driver B6, four IQ orthogonal signals are generated by divider DT2. The four IQ orthogonal signals are further enhanced in driving capability by drivers B7 to B10, and then connected to DC blocking capacitors C5 to C8 respectively. The output ends of DC blocking capacitors C5 to C8 are four orthogonal local oscillator signals LOQp_60, LOQn_60, LOIn_60 and LOIp_60 with a delay of 60 degrees, and finally connected to the mixer module. Large resistors R5 to R8 provide the DC bias voltage LOvbias to the four orthogonal local oscillator signals LOQp_60, LOQn_60, LOIn_60 and LOIp_60 with a delay of 60 degrees. However, in this delayed signal path, after passing through divider DT2, the delay value of the four IQ orthogonal signals is f LO (tT / 12), that is, the delay value becomes T / 12. Therefore, it is necessary to swap the phases of the four orthogonal local oscillator signals LOQp_60, LOQn_60, LOIn_60 and LOIp_60 before entering the mixer module. LO (tT / 12+T / 4)=f LO(t+T / 6), so that the required T / 6 delay signal can be achieved. Phase exchange is achieved by changing the connection mode of the local oscillator signal link circuit 1 and the mixer module. Specifically, the structures of mixers M1 and M2 are exactly the same, and both have four local oscillator inputs. It can be seen from the previous deduction that when the local oscillator signal LOIp output by capacitor C1 is connected to the Ip end of mixer M1, and the local oscillator signal LOIp_60 output by capacitor C8 is connected to the Ip end of mixer M2, the phase difference between the Ip end input of mixer M1 and the Ip end input of mixer M2 is 60 degrees. The output end of capacitor C8 is selected as LOIp_60 because if the phase exchange is not performed, the output end of capacitor C8 should be consistent with the output end of capacitor C4, that is, it is also Qn. However, according to the previous deduction, capacitor C5 and capacitor C8 have a delay of T / 4. At this time, the output of capacitor C8 is used as LOIp_60, which realizes a delay of -T / 12+T / 4=T / 6. Similarly, the phase exchange of the other three terminals is similar, such as Figure 4 shown.

[0035] Figure 4 The mixer module in the embodiment is composed of two IQ quadrature power mixers as power mixer units M1 and M2. One advantage of the IQ quadrature power mixer is that it has a good image suppression effect.

[0036] See also Figure 7, the IQ quadrature power mixer M1 is composed of an I mixer and a Q mixer. For the I mixer of the IQ quadrature power mixer M1, the NMOS differential pair MI1 and MI2 of the transconductance stage circuit convert the input baseband signals BBIp and BBIn from the voltage domain to the current domain respectively. In order to take into account both efficiency and linearity, the transistors MI1 and MI2 in the transconductance stage circuit are generally biased in the AB class state. The baseband signal current output by the transconductance stage circuit enters the switch stage circuit, which is composed of two groups of NMOS differential pairs MI3 and MI4, as well as MI5 and MI6. In general, the swing amplitude of the four orthogonal local oscillator signals LOIp, LOIn, LOQp and LOQn should be large enough so that the pairs in the switch stage circuit have good switching characteristics, thereby sampling the input signal from the transconductance stage circuit, realizing up-conversion, and reducing the influence of the switch stage circuit on the circuit linearity and noise. The fully differential circuit structure used in this application can improve the suppression of the mixer, that is, reduce the leakage from the local oscillator to the RF end and the local oscillator to the baseband input end. The RF signal output by the switch-level circuit enters the common source and common gate circuit. The common source and common gate circuit is composed of NMOS tube MC1 and NMOS tube MC2, and its gate terminal is connected to the DC bias voltage Vcas to improve the circuit reliability. The conventional power supply voltage is 1.8V. In order to improve the speed response of the circuit, the transconductance stage circuit and the switch-level circuit use transistors with a maximum withstand voltage of 1.2V, so it is necessary to add an NMOS tube that can withstand 1.8V in the common source and common gate circuit to share the voltage drop. Because the power mixer has the characteristics of a linear power amplifier, that is, the drain voltage of the NMOS tube MC1 and the NMOS tube MC2 can reach 2 times the power supply voltage of 3.6V under ideal conditions, so adding a common source and common gate circuit reduces the possibility of transistor breakdown in the transconductance stage circuit and the switch-level circuit, and at the same time reduces the leakage of the local oscillator signal to the RF output end, that is, improves the isolation of the circuit. The Q mixer in the IQ quadrature mixer M1 is the same as the I mixer, and the functions of the NMOS tubes MQ1 to MQ6, MC3 and MC4 in the Q mixer are the same as those of the NMOS tubes MI1 to MI6, MC1 and MC2 in the I mixer. The I and Q mixers connect the up-converted RF current signals through the output ends (drains) of MC1, MC2, MC3 and MC4, thus realizing the addition of the I and Q signals in the current domain, thereby suppressing the image signal. The IQ synthesized current signal forms a resonant frequency selection network through the load-level transformer T1 and the programmable capacitor array C, and transfers the power of the primary coil of the transformer T1 to the secondary coil load resistor R through the appropriate impedance transformation ratio of the transformer T1. LBy adjusting the programmable capacitor array C, the sub-1GHz (below 1GHz) frequency band can be covered, and it can be used in wireless communication systems such as 2G, LTE (Long Term Evolution), NB-IoT (Narrowband Internet of Things), LoRa (Long Distance Wide Area Network), etc. By changing the load, such as adjusting the value of the inductor and capacitor, or using a resistive load, it can also be applied to the sub-6GHz (below 6GHz) frequency band.

[0037] The structure of the IQ quadrature power mixer M2 is the same as that of the IQ quadrature power mixer M1, and the only difference is the signal. The transistors in the switch-stage circuit of the IQ quadrature power mixer M2 are driven by four orthogonal local oscillator signals LOQp_60, LOQn_60, LOIn_60 and LOIp_60 with a delay of 60 degrees. Finally, the output currents of the IQ quadrature power mixers M1 and M2 have a 60-degree delay in the time domain, that is, their current spectrum is consistent with the spectrum of the local oscillator signal, and the amplitudes at the third harmonic of the local oscillator signal are equal but the phase difference is 180 degrees. The addition of the two currents achieves the suppression of the third harmonic of the local oscillator signal, that is, the use of a 1:1 harmonic suppression power mixer is achieved.

[0038] See also Figure 8 , where the solid line corresponds to the vertical axis on the left, and the dotted line corresponds to the vertical axis on the right. Under the conditions of sub-1GHz frequency band and the same output power, the harmonic suppression power mixer of the present application (solid line above) can improve the third harmonic suppression effect of the local oscillator signal by 25.35dB to 34.59dB, compared with the mixer without harmonic suppression function (solid line below), which effectively improves the linearity of the power mixer. At the same time, within the above-mentioned frequency band, the efficiency value of the harmonic suppression power mixer of the present application can reach 7.71% to 10.78%, which is relatively high for the circuit in front of the power amplifier, indicating that the present application has good efficiency.

[0039] Compared with the prior art, the power mixer capable of suppressing the third harmonic of the local oscillator proposed in the present application has the following beneficial effects.

[0040] First, the Gilbert mixer and the pre-power amplifier are combined together, and the pre-power amplifier is omitted. The load of the pre-power amplifier is generally composed of a transformer and a capacitor array, so the present application only needs one transformer T1, which reduces the layout area and reduces the cost.

[0041] Second, it has good harmonic suppression function, which can suppress the third harmonic of the local oscillator signal. It can be used as a driver in LTE, 5G and other communication systems to provide good linearity for subsequent circuits and reduce the impact on other frequency bands. The third harmonic suppression of this application can reach 49.2dB to 59.3dB in the sub-1GHz bandwidth, which is 25dB to 34.6dB higher than that of the mixer without harmonic suppression. At the same time, the linearity of the output signal is effectively improved by harmonic suppression.

[0042] Third, only two identical power mixer units are needed and no By approximating, harmonic suppression can be achieved, the multiplexing rate is large, and the circuit design is simplified.

[0043] Fourth, only a two-frequency divider is needed in the local oscillator signal chain circuit, and no four-frequency divider is needed, which reduces the design difficulty. At the same time, the local oscillator signal chain circuit does not need a 25% duty cycle generation circuit, which further simplifies the design.

[0044] The invention discloses a low-cost harmonic suppression power mixer which has a simple structure and can suppress the third harmonic of a local oscillator signal. Harmonic suppression can be achieved and the signal can be amplified by using only a simple divider and two power mixer units.

[0045] The above are only preferred embodiments of the present application and are not intended to limit the present application. For those skilled in the art, the present application may have various changes and variations. Any modifications, equivalent substitutions, improvements, etc. made within the spirit and principles of the present application should all be included in the protection scope of the present application.

Claims

1. A power mixer capable of suppressing the third harmonic of a local oscillator, characterized in that: It includes a local oscillator signal link circuit, a transconductance stage circuit, a switch stage circuit, a common source and common gate stage circuit and a load circuit; The local oscillator signal link circuit divides the local oscillator input signal into two paths, one path is called the local oscillator signal, and the other path is called the delayed local oscillator signal after passing through the delay module, wherein one path has a phase difference of 60 degrees relative to the other path; the two local oscillator signals respectively pass through a two-way frequency divider to generate four IQ orthogonal signals respectively; There are two transconductance stage circuits, which are used to convert the baseband voltage signal into a current signal; There are two switch-stage circuits, which respectively use the two local oscillator signals to up-convert the current signals from the two transconductance-stage circuits into the required radio frequency signals; There are two common-source and common-gate circuits, which respectively transmit the radio frequency signals output by the two switch-stage circuits to the load circuit; The load circuit realizes the addition of output currents of two common-source and common-gate circuits; The entirety of the transconductance stage circuit, the switch stage circuit and the common source and common gate stage circuit is composed of a mixer module consisting of two IQ orthogonal power mixers; each IQ orthogonal power mixer is composed of an I mixer and a Q mixer; In the I-channel mixer, the transconductance stage circuit includes a pair of NMOS differential pair transistors, which convert the input differential baseband signal from the voltage domain to the current domain; The baseband signal current output by the transconductance stage circuit enters the switch stage circuit, which includes two groups of NMOS differential pairs of tubes. Driven by four orthogonal local oscillator signals, the pairs of tubes in the switch stage circuit sample the input signal from the transconductance stage circuit to achieve up-conversion. The RF signal output by the switch stage circuit enters the common-source and common-gate stage circuit, which consists of two NMOS tubes, and the gate terminals of which are connected to a DC bias voltage. The Q mixer has the same structure as the I mixer; The I and Q mixers connect the up-converted RF current signals through the output ends of their respective common-source and common-gate circuits to achieve the addition of the I and Q signals in the current domain.

2. The power mixer capable of suppressing the third harmonic of the local oscillator according to claim 1, characterized in that: The switch stage circuit adopts an active mixer, whose gain is equal to the product of the transconductance of the transconductance stage and the load, and the product value is greater than 1.

3. The power mixer capable of suppressing the third harmonic of the local oscillator according to claim 1, characterized in that: The local oscillator signal link circuit includes a delay module, a plurality of drivers, two frequency dividers, a plurality of capacitors and a plurality of resistors; Local oscillator input signal f LO (t) is divided into two paths; one path remains unchanged and is called the local oscillator signal f LO (t), the other path passes through the delay module to obtain the delayed local oscillator signal f LO (tT / 6), where T represents the local oscillator signal f LO (t) period; Local oscillator signal f LO (t) After being amplified by a driver signal, four IQ orthogonal signals are generated through a two-way frequency divider. The four IQ orthogonal signals are further amplified by four drivers respectively, and then connected to four DC blocking capacitors respectively. The output ends of the four DC blocking capacitors are four orthogonal local oscillator signals LOIp, LOIn, LOQp and LOQn respectively, and finally connected to the mixer module; four resistors provide DC bias voltage to the four orthogonal local oscillator signals LOIp, LOIn, LOQp and LOQn respectively; Delayed local oscillator signal f LO (tT / 6) After being enhanced by a driver signal, four IQ orthogonal signals are generated through a divider. The four IQ orthogonal signals are further enhanced by four drivers and then connected to four DC blocking capacitors. The output ends of the four DC blocking capacitors are four orthogonal local oscillator signals LOQp_60, LOQn_60, LOIn_60 and LOIp_60 with a delay of 60 degrees, and finally connected to the mixer module; four resistors provide DC bias voltage to the four orthogonal local oscillator signals LOQp_60, LOQn_60, LOIn_60 and LOIp_60 with a delay of 60 degrees.

4. The power mixer capable of suppressing the third harmonic of the local oscillator according to claim 3, characterized in that: Local oscillator signal f LO The Fourier expansion of (t) is Delayed local oscillator signal f LO The Fourier expansion of (tT / 6) is where ω LO Represents the angular frequency of the local oscillator signal; the sum of the two cancels out the third harmonic of the local oscillator signal.

5. The power mixer capable of suppressing the third harmonic of the local oscillator according to claim 3, characterized in that: The local oscillator signal LOIp is connected to the Ip terminal of mixer one, and the local oscillator signal LOIp_60 is connected to the Ip terminal of mixer two. At this time, the phase difference between the input of the Ip terminal of mixer one and the input of the Ip terminal of mixer two is 60 degrees; The local oscillator signal LOIn is connected to the In terminal of mixer one, and the local oscillator signal LOIn_60 is connected to the In terminal of mixer two. At this time, the phase difference between the In terminal input of mixer one and the In terminal input of mixer two is 60 degrees; The local oscillator signal LOQp is connected to the Qp terminal of mixer one, and the local oscillator signal LOQn_60 is connected to the Qp terminal of mixer two. At this time, the phase difference between the input of the Qp terminal of mixer one and the input of the Qp terminal of mixer two is 60 degrees; The local oscillator signal LOQn is connected to the Qn terminal of mixer one, and the local oscillator signal LOQn_60 is connected to the Qn terminal of mixer two. At this time, the phase difference between the input of the Qn terminal of mixer one and the input of the Qn terminal of mixer two is 60 degrees.

6. The power mixer capable of suppressing the third harmonic of the local oscillator according to claim 3, characterized in that: The transistors in the switch-stage circuit of one IQ quadrature power mixer are driven by four orthogonal local oscillator signals LOIp, LOIn, LOQp and LOQn; the transistors in the switch-stage circuit of another IQ quadrature power mixer are driven by four orthogonal local oscillator signals LOQp_60, LOQn_60, LOIn_60 and LOIp_60 with a delay of 60 degrees; the output currents of the two IQ quadrature power mixers have a 60-degree delay in the time domain.

7. The power mixer capable of suppressing the third harmonic of the local oscillator according to claim 1, characterized in that: The load circuit includes an adjustable capacitor, a transformer and a load resistor; the current signal passes through a resonant frequency selection network composed of the transformer and the adjustable capacitor, and the power is transferred to the load resistor through the impedance transformation ratio of the transformer to achieve signal output.

8. The power mixer capable of suppressing the third harmonic of the local oscillator according to claim 7, characterized in that: The adjustable capacitor is a programmable capacitor array, and different frequency bands can be covered by adjusting the programmable capacitor array.

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