Isolated Resonant Converter and Its Control Method

Abstract

The present disclosure provides an isolated resonant converter and a control method therefor. The control method enables the converter to have a wide voltage conversion ratio range by substantially narrowing the switching frequency range, thereby improving its performance. By combining variable duty cycle, variable frequency, and delay time control to control the output voltage or current, the switching frequency range can be narrowed.

Description

Technical Field

[0001] The present disclosure relates to an isolated resonant DC / DC converter and a method for controlling the converter, particularly to a single-phase or multi-phase isolated resonant converter and a method for controlling the isolated resonant converter, and thereby increasing the range of input voltage and / or output voltage. Background Art

[0002] The power supply industry continuously demands converters with high efficiency, high power density, and low cost to achieve less energy consumption, smaller installation space, and better cost-effectiveness. In addition, in many newly developed applications (such as electric vehicles and data centers), higher power handling is required. By using converters with high rated power, the charging time of electric vehicles and the size of power cabinets in data centers can be significantly reduced. Generally, resonant converters use a resonant cavity circuit to shape the waveforms of voltage and / or current, thereby minimizing switching losses and enabling high-frequency operation without affecting conversion efficiency. Therefore, resonant converters are widely used in existing power supplies that can provide the highest efficiency and power density.

[0003] Figure 1A A typical full-bridge topology of an isolated resonant power converter 100 is shown, and its resonant cavity circuit includes a resonant inductor L P and a resonant capacitor C P and C S . Since the components L P , C P and C S of the resonant cavity are connected in series with each other through a transformer TR, the converter 100 in Figure 1A is a series resonant converter (SRC). The inductor L P can be located on the primary side and / or the secondary side, and the inductor L P can be arranged together with the capacitor C P and / or C S such that at least one inductor and one capacitor are connected in series to form a resonant cavity circuit. To minimize conduction losses, a metal-oxide-semiconductor field effect transistor (MOSFET) with a low on-resistance can be used to replace the diode as the secondary side rectifier. In addition, since the MOSFET as the secondary side rectifier allows the power current to flow from the secondary side to the primary side, the converter 100 can operate bidirectionally.

[0004] When the converter 100 is in the case of an SRC, the magnetizing inductance L m of the transformer TR is much larger than the inductor L​P (e.g., more than 10 times). When the excitation inductance L of the transformer TR m is only several times that of the inductance L P (e.g., 2 to 10 times), such a converter is called an LLC resonant converter. Since the transformer TR of the LLC resonant converter has a relatively small excitation inductance, the circulating current flowing through the excitation inductance is greater than the circulating current in the SRC. Due to the relatively large circulating current, the LLC converter achieves wide-range zero voltage switching (ZVS) at the cost of increased conduction losses.

[0005] Figure 1B are the switching control signals for the primary-side switch Q P1-P4 and the secondary-side switch Q S1-S4 and the waveform diagram of the primary-side current i P . In the primary side, the switching control signals of Q P1 and Q P4 are complementary to the switching control signals of Q P2 and Q P3 . The duty cycle of each switching control signal is usually 50%, thereby obtaining a symmetric i P waveform. Due to the resonance among the components L P , C P and C S of the resonant cavity, the waveform of the primary-side current is a sine wave. To achieve ZVS, the switching frequency is slightly greater than the resonant frequency, where the resonant frequency depends on the components L P , C P and C S of the resonant cavity. Thereby, at the moment when the primary-side switch Q P1-P4 switches (i.e., the moments t2 and t3 in Figure 1B ), the primary-side current i P becomes the ZVS current of the primary-side switch Q P1-P4 . The primary-side current i P is transmitted to the secondary side and is divided by the turns ratio of the transformer TR (i.e., i S = i P / n, where n = N P / N S , and N P and N S represent the number of turns of the primary-side winding and the secondary-side winding respectively).

[0006] The switching control signal of the secondary-side switch Q S1-S4 depends on the polarity of i P . When i P is positive, the switches Q S1 and Q S4 conduct, and the switch QS2 and Q S3 turn off; conversely, when i P is negative, switch Q S1 and Q S4 turn off, and switch Q S2 and Q S3 turn on. To achieve ZVS in practical applications, the rising edges of all switch control signals (such as at times t0, t1, t2, and t3) can be delayed so that the two complementary switches in the same arm are turned off simultaneously for a short time, and this simultaneously turned-off time is called the dead time. During this dead time, the primary-side current turns from the switch in the off state to the anti-parallel diode in the complementary switch, thus creating conditions for the subsequent ZVS turn-on of the complementary switch.

[0007] The output current value supplied to the load is the average value of the secondary-side current i S . Since the secondary-side current i S has a sinusoidal waveform, the peak value of the secondary-side current i S is always greater than the output current, resulting in a relatively large root-mean-square (RMS) current in converter 100, and further increasing the temperature of the components due to conduction losses. Therefore, the relatively large RMS current also becomes a drawback of the resonant converter, especially in applications with large current or high power. Since the heat capacity of the components is physically limited, the relatively high conduction losses and the increase in the temperature of the components will limit the maximum power transfer capability of the converter. In applications with large current or high power, temperature control is a key issue closely related to the reliability of the converter.

[0008] Figure 2A Fig. shows a typical three-phase isolated SRC 200, which includes three phases, namely Phase 1, Phase 2, and Phase 3. Each phase of the three-phase isolated SRC 200 includes two primary-side switches, a plurality of resonant cavity components, a transformer, and two secondary-side switches. Taking Phase 1 as an example, switches Q P1 and Q P2 are the primary-side switches, inductor L P1 and capacitors C P1 and C S1 are the resonant cavity components, transformer TR1 is the transformer of Phase 1, and switches Q S1 and Q S2 are the secondary-side switches. Similarly, for Phase 2, switches Q P3 and Q P4 are the primary-side switches, inductor L P2 and capacitors C P2 and C S2 are the resonant cavity components, transformer TR2 is the transformer of Phase 2, and switches Q S3 and Q S4is the secondary side switch. In addition, for phase 3, switches Q P5 and Q P6 are the primary side switches, inductor L P3 and capacitor C P3 and C S3 are the resonant cavity components, transformer TR3 is the transformer for phase 3, and switches Q S5 and Q S6 are the secondary side switches.

[0009] The greatest advantage of the multiphase SRC is that each phase transmits a portion of the total output power. In the three-phase SRC 200, the current stress in any one phase is Figure 1A one-third of the current stress in the single-phase SRC 100 shown, which means that the RMS current stress on each resonant cavity component also becomes one-third. Considering that the resistive conduction loss is proportional to the square of the RMS current, the conduction loss of each resonant cavity component becomes one-ninth of the conduction loss of each resonant cavity component in the single-phase SRC. Therefore, the three-phase SRC has a much higher power transmission capacity than the single-phase SRC.

[0010] Figure 2B are the waveforms of the switching control signals for the primary side switches Q P1 , Q P3 and Q P5 and the secondary side switch Q S1 with respect to the primary side currents i P1 , i P2 and i P3 . For the sake of simplicity of the diagram, the switching control signals of switches Q P2 , Q P4 , Q P6 , Q S2 , Q S3 , Q S4 , Q S5 and Q S6 are omitted. The switching control signals of switches Q P2 , Q P4 and Q P6 are complementary to the switching control signals of switches Q P1 , Q P3 and Q P5 respectively. Similar to the single-phase SRC, in each phase, the secondary side switch is turned on according to the polarity of the primary side current. Specifically, when i P1 , i P2 or i P3 is positive, the corresponding switches Q S1 , Q S3 or Q S5 are turned on; while when i P1 , i P2 or iP3 When it is negative, the corresponding switch Q S2 , Q S4 or Q S6 conducts. The way to achieve ZVS is similar to that of single-phase SRC. As Figure 2B shown, the switch Q P1 , Q P3 and Q P5 have their switching control signals phase-shifted from each other by a phase shift angle of 120 degrees (a T S / 3 offset in the time domain). Thereby, the RMS current on the filter capacitor C O can be significantly reduced. Due to the above reasons, SRC can improve its maximum power transfer capacity by forming a multi-phase structure.

[0011] By changing the switching frequency, the voltage conversion ratio (V OUT / V IN ) of the resonant converter can be controlled. For example, SRC can provide the maximum voltage conversion ratio when operating at the resonant frequency (i.e., when the switching frequency is equal to the resonant frequency), and the voltage conversion ratio will decrease as the switching frequency increases. Variable switching frequency control is generally regarded as a shortcoming of resonant converters, especially in applications with a wide input voltage range and / or a wide output voltage range. In order to cover a wider voltage conversion ratio range, the maximum switching frequency of the resonant converter increases, thus increasing the losses in driving the magnetic components and the turn-off losses during switching. Therefore, it should be noted that for high-power applications with a wide input voltage range and / or a wide output voltage range (such as high-power electric vehicle charging applications), the research on methods for expanding the voltage conversion ratio range of multi-phase resonant converters is crucial. Summary of the Invention

[0012] To overcome the deficiencies of the aforementioned prior art, the present disclosure provides a multi-phase isolated resonant converter and its control method. This control method enables the multi-phase isolated resonant converter to have a wide input voltage range and / or a wide output voltage range by substantially narrowing the switching frequency range. By combining variable duty cycle, variable frequency, and delay time control to control the output voltage or current, the switching frequency range can be narrowed. The switching frequency and duty cycle can be used to control the primary-side switches of the multi-phase isolated resonant converter, while the delay time control can be used to control the secondary-side switches. By delaying the turn-off of the corresponding secondary-side switches relative to the zero-crossing moment of the primary-side or secondary-side current or relative to the turn-off moment of the primary-side switches, the delay time control of the secondary-side switches can be achieved. The delay time control of the present disclosure can be extended to dual delay time control when there is a full-bridge rectifier, thereby obtaining a wider output voltage range. Furthermore, the delay time control of the present disclosure can also be used for active current sharing in each phase.

[0013] According to the concept of one aspect of the present disclosure, the present disclosure provides an isolated resonant converter including one or more phases and a control circuit. Each phase includes a transformer, a plurality of resonant elements, a plurality of primary-side switches electrically coupling an input terminal to the transformer via the plurality of resonant elements, and a plurality of secondary-side switches electrically coupling the transformer to an output terminal. The control circuit is electrically coupled to the one or more phases, wherein the control circuit is configured to: sense an input voltage on the input terminal and an output voltage on the output terminal; determine a first control signal for the primary-side switches and a second control signal for the secondary-side switches based on a plurality of parameters, wherein the plurality of parameters includes physical characteristics of the plurality of resonant elements, the input voltage, the output voltage, and a reference voltage; output the first control signal including a switching frequency and a first duty cycle to the primary-side switches; and output the second control signal including the switching frequency and a second duty cycle to the secondary-side switches. The second duty cycle of the first secondary-side switch among the plurality of secondary-side switches is greater than the first duty cycle of the first primary-side switch among the corresponding plurality of primary-side switches.

[0014] In some embodiments, the second duty cycle of the second secondary-side switch among the plurality of secondary-side switches is defined according to the turn-off time of the second primary-side switch among the plurality of primary-side switches.

[0015] In some embodiments, the control circuit is further configured to sense the current flowing through each phase.

[0016] In some embodiments, the second duty cycle of the second secondary-side switch among the plurality of secondary-side switches is defined according to the zero-crossing time when the current changes from positive to negative or from negative to positive.

[0017] In some embodiments, the control circuit includes: a sensing and scaling circuit configured to receive the input and output voltages and convert the input and output voltages into scaled input and output voltages; a subtraction circuit configured to receive the scaled output voltage and generate an error signal by subtracting the reference voltage from the scaled output voltage; an error amplifier configured to receive the error signal and generate an amplified and compensated error signal; and a processing circuit configured to receive the scaled input voltage and the amplified and compensated error signal and generate the first and second control signals based on the scaled input voltage and the amplified and compensated error signal.

[0018] In some embodiments, the control circuit further includes a zero-current sensor configured to sense the current signal flowing through each phase.

[0019] In some embodiments, the processing circuit is further configured to receive the current signal and generate the first and second control signals based on the scaled input voltage, the amplified and compensated error signal, and the current signal.

[0020] In some embodiments, the one or more phases include at least two phases, and the control circuit further includes a current balancing circuit configured to adjust the second control signal before it is transmitted to the secondary side switch so that the currents flowing through different phases are balanced with each other.

[0021] In some embodiments, the current balancing circuit includes: a current sensing, scaling, and averaging circuit configured to obtain an average value of the currents flowing through each phase; and a delay time adder configured to determine the delay time for each phase based on the difference between the currents of any two phases and adjust the second control signal by adding the delay time to the duty cycle of the second control signal.

[0022] According to another aspect of the present disclosure, the present disclosure provides a method for controlling an isolated resonant converter, where the isolated resonant converter has one or more phases, and each phase includes a transformer, a plurality of resonant elements, a plurality of primary side switches electrically coupling an input end to the transformer via the resonant elements, and a plurality of secondary side switches electrically coupling the transformer to an output end. The method includes: sensing an input voltage on the input end and an output voltage on the output end; determining a first control signal for the primary side switches and a second control signal for the secondary side switches based on a plurality of parameters, where the plurality of parameters includes physical characteristics of the resonant elements, the input voltage, the output voltage, and a reference voltage; outputting a first control signal including a switching frequency and a first duty cycle to the primary side switches; and outputting a second control signal including a switching frequency and a second duty cycle to the secondary side switches. The second duty cycle of the first secondary side switch among the plurality of secondary side switches is greater than the first duty cycle of the first primary side switch among the corresponding plurality of primary side switches.

[0023] In some embodiments, the second duty cycle of the second secondary side switch among the plurality of secondary side switches is defined according to the turn-off time of the second primary side switch among the plurality of primary side switches.

[0024] In some embodiments, the method further includes sensing the current flowing through each phase.

[0025] In some embodiments, the second duty cycle of the second secondary side switch among the plurality of secondary side switches is defined according to the zero-crossing time when the current changes from positive to negative or from negative to positive.

[0026] In some embodiments, the method further includes adjusting the second control signal before it is transmitted to the secondary side switch so that the currents flowing through different phases are balanced with each other.

[0027] According to another concept of the present disclosure, the present disclosure provides an isolated resonant converter, comprising: a transformer, a plurality of resonant elements, a primary full-bridge circuit, a secondary full-bridge circuit, and a control circuit. The primary full-bridge circuit includes a first arm and a second arm, and electrically couples an input terminal to the transformer via the resonant elements. The secondary full-bridge circuit includes a third arm and a fourth arm, and electrically couples the transformer to an output terminal. The control circuit is electrically coupled to the primary full-bridge circuit and the secondary full-bridge circuit, wherein the control circuit is configured to: sense an input voltage on the input terminal and an output voltage on the output terminal; determine control signals for the first, second, third, and fourth arms according to an amplified and compensated error signal, wherein all the control signals include the duty cycle of at least one of the first, second, third, and fourth arms; and output all the control signals to the primary full-bridge circuit and the secondary full-bridge circuit. When the amplified and compensated error signal is less than a threshold, the isolated resonant converter is a buck converter, and when the amplified and compensated error signal is greater than the threshold, the isolated resonant converter is a boost converter.

[0028] In some embodiments, when the amplified and compensated error signal is less than the threshold, the duty cycle of the first arm is between 0.0 and 0.5, and the duty cycle of the second arm is 0.0.

[0029] In some embodiments, when the amplified and compensated error signal is less than the threshold, the duty cycle of the first arm is 0.5, and the duty cycle of the second arm is between 0 and 0.5.

[0030] In some embodiments, when the amplified and compensated error signal is greater than the threshold, the duty cycles of the first and second arms are 0.5, the duty cycle of the third arm is between 0.5 and 1.0, and the duty cycle of the fourth arm is 0.5.

[0031] In some embodiments, when the amplified and compensated error signal is greater than the threshold, the duty cycles of the first and second arms are 0.5, the duty cycle of the third arm is 1.0, and the duty cycle of the fourth arm is between 0.5 and 1.0.

[0032] In some embodiments, as the amplified and compensated error signal increases, the duty cycle increases monotonically.

[0033] In some embodiments, as the amplified and compensated error signal increases, the duty cycle increases linearly. BRIEF DESCRIPTION OF THE DRAWINGS

[0034] Figure 1A AND Figure 1B respectively show a typical full-bridge topology of an isolated resonant power converter and a timing diagram of switch control signals for its ZVS operation.

[0035] Figure 2A ANDFigure 2B A typical three-phase isolated series resonant converter and the timing diagram of its switching control signals operating in ZVS are respectively shown.

[0036] Figures 3A to 3D Isolated multiphase resonant converters of various preferred embodiments of the present disclosure are respectively shown.

[0037] Figure 4A and Figure 4B An isolated multiphase series resonant converter connected to a controller in a preferred embodiment of the present disclosure is shown.

[0038] Figure 5A 、 Figure 5B 、 Figure 5C and Figure 5D Waveform examples of the switching control signals and the primary side current for controlling a three-phase converter in a preferred embodiment of the present disclosure are shown.

[0039] Figure 6A and Figure 6B An exemplary embodiment of an isolated multiphase resonant converter having a diode rectifier on the secondary side is shown.

[0040] Figure 7A and Figure 7B An exemplary embodiment of an isolated resonant converter having a full-bridge rectifier on the secondary side is shown.

[0041] Figure 8A and Figure 8B A single-phase resonant converter in a preferred embodiment of the present disclosure is shown.

[0042] Figure 9A An isolated multiphase resonant converter in a preferred embodiment of the present disclosure, which is coupled to a controller to perform active current sharing in the converter, is shown.

[0043] Figure 9B is Figure 9A an enlarged schematic diagram of the controller in

[0044] Figure 10 A single-phase isolated series resonant converter in a preferred embodiment of the present disclosure, which adopts a full-bridge configuration on both the primary side and the secondary side, is shown.

[0045] Figure 11 A single-phase isolated series resonant converter in a preferred embodiment of the present disclosure, which is coupled to a control circuit, is shown in Figure 10

[0046] Figure 12 is a schematic diagram of the relationship between V EA in a preferred embodiment of the present disclosure and the duty cycle signals D1, D2, D3, and D4.

[0047] Figure 13A and Figure 13B respectively show the equivalent circuit and key operating waveforms when the converter of Figure 10 adopts the control method of the D1 region of Figure 11 . The labels Q P1 , Q P2 , Q P3 and Q P4 respectively represent the switching control signals of the corresponding primary-side switches. The labels Q S1 , Q S2 , Q S3 , Q S4 , Q S5 and Q S6 respectively represent the switching control signals of the corresponding secondary-side switches. The labels i P1 , i P2 and i P3 represent the primary-side currents of each phase.

[0048] Figure 14A and Figure 14B respectively show the equivalent circuit and key operating waveforms when the converter of Figure 10 adopts the control method of the D2 region of Figure 11 .

[0049] Figure 15A and Figure 15B respectively show the equivalent circuit and key operating waveforms when the converter of Figure 10 adopts the control method of the D3 region of Figure 11 .

[0050] Figure 16A and Figure 16B respectively show the equivalent circuit and key operating waveforms when the converter of Figure 10 adopts the control method of the D4 region of Figure 11 .

[0051] Figure 17A and Figure 17B respectively show the schematic diagram of the relationship between V EA and the duty cycle signals D1, D2, D3 and D4 in other preferred embodiments of the present disclosure.

[0052] Figures 18A to 18E respectively show the single-phase isolated series resonant converters of various preferred embodiments of the present disclosure.

[0053] Among them, the reference numerals are explained as follows:

[0054] 100: Converter

[0055] L P : Resonant inductor

[0056] CP , C S : Resonant capacitor

[0057] TR: Transformer

[0058] L m : Excitation inductance

[0059] Q P1 , Q P2 , Q P3 , Q P4 : Primary side switch

[0060] Q S1 , Q S2 , Q S3 , Q S4 : Secondary side switch

[0061] i P : Primary side current

[0062] i S : Secondary side current

[0063] N P , N S : Winding

[0064] t0, t1, t2, t3, t 0’ , t 1’ : Moment

[0065] 200: Converter

[0066] Q P5 , Q P6 : Primary side switch

[0067] Q S5 , Q S6 : Secondary side switch

[0068] L P1 , L P2 , L P3 : Inductance

[0069] C P1 , C S1 , C P2 , C S2 , C P3 , C S3 : Capacitor

[0070] i P1 , i P2 , i P3 , i PN : Primary side current

[0071] i S1 , i S2, i S3 , i SN : Secondary-side current

[0072] C O : Filter capacitor

[0073] V OUT : Output voltage

[0074] V IN : Input voltage

[0075] 300: Converter

[0076] Q P(2N-1) , Q P(2N) : Primary-side switch

[0077] Q S(2N-1) , Q S(2N) : Secondary-side switch

[0078] C PN , C SN : Resonant capacitor

[0079] L PN , L SN : Resonant inductor

[0080] TR1, TR2, TR N : Transformer

[0081] N P1 , N P2 , N PN : Primary-side winding

[0082] N S1 , N S2 , N SN : Secondary-side winding

[0083] 400: Controller

[0084] f S : Switching frequency

[0085] T S : Switching period

[0086] D: Duty cycle

[0087] T D : Delay time

[0088] 410: Zero-current sensor

[0089] 420: Sensing and scaling circuit

[0090] 430: Subtraction circuit

[0091] 440: Error amplifier

[0092] 450: Processing circuit

[0093] V REF : Reference voltage

[0094] V E : Error signal

[0095] V EA : Amplified and compensated error signal

[0096] I IN : Input current

[0097] I OUT : Output current

[0098] V OUT(SCLD) 、V IN(SCLD) : Voltage

[0099] I OUT(SCLD) 、I IN(SCLD) : Current

[0100] D SA1 、D SA2 、D SB1 、D SB2 、D SC1 、D SC2 : Diode

[0101] Q PA1 、Q PA2 、Q PB1 、Q PB2 、Q PC1 、Q PC2 : Primary side switch

[0102] Q SA1 、Q SA2 、Q SB1 、Q SB2 、Q SC1 、Q SC2 : Secondary side switch

[0103] C RA 、C RB 、C RC : Resonant capacitor

[0104] C BA 、C BB 、C BC : Resonant inductor

[0105] i LRA 、i LRB 、i LRC : Current

[0106] 700: Converter

[0107] 800: Controller

[0108] 810: Current balance circuit

[0109] 811: Current sensing, scaling, and averaging circuit

[0110] 812: Delay time adder

[0111] |i P1 | (AVG) 、|i P2 | (AVG) 、|i P3 | (AVG) : Current magnitude

[0112] T D_1 : Delay time

[0113] 1000: Converter

[0114] 1100: Controller

[0115] 1110: Sensing and scaling circuit

[0116] 1120: Error amplifier

[0117] 1130: Leg controller

[0118] B1, B2, B3, B4: Half-bridge legs

[0119] D1, D2, D3, D4: Duty cycle signals

[0120] 1202: Transition point Detailed implementation manners

[0121] Some exemplary embodiments embodying the features and advantages of the present disclosure will be described in detail in the following description. It should be understood that the present disclosure can have various variations in different aspects, all of which do not depart from the scope of the present disclosure, and the descriptions and illustrations therein are for illustrative purposes in nature and not intended to limit the present disclosure.

[0122] Figures 3A to 3D The isolated multiphase resonant converter 300 of each preferred embodiment of the present disclosure is shown respectively. As Figures 3A to 3D shown, the converter 300 has N phases, where N is a natural number (i.e., having phase 1, phase 2,..., phase N). The converter 300 includes primary-side switches Q P1 、Q P2 、…、Q P(2N) 、secondary-side switches Q S1 、Q S2 、…、Q S(2N) 、resonant capacitors C located on the primary sideP1 , C P2 , …, C PN , the resonant inductor L located on the primary side P1 , L P2 , …, L PN , transformers TR1, TR2, …, TR N , the resonant capacitor C located on the secondary side S1 , C S2 , …, C SN and an output filter capacitor C O . In some embodiments, the converter 300 has three phases (i.e., N = 3).

[0123] Please refer to Figure 3A . Each phase of the converter 300 includes two switches forming a half-bridge on the primary side, a plurality of resonant cavity elements, a transformer, and two switches forming a half-bridge on the secondary side. The transformers TR1, TR2, …, TR N have primary side windings N P1 , N P2 , …, N PN located on the primary side and secondary side windings N S1 , N S2 , …, N SN . In this embodiment, one ends of the primary side windings N P1 , N P2 , …, N PN are connected to the same point (or common node), while the other ends of the primary side windings N P1 , N P2 , …, N PN are connected to their corresponding half-bridges. Similarly, one ends of the secondary side windings N S1 , N S2 , …, N SN are connected to the same point (or common node), while the other ends of the secondary side windings N S1 , N S2 [[ID=6)2]], …, N SN are connected to their corresponding half-bridges. Figure 3A The connection configuration of the transformers TR1, TR2, …, TR N in N is generally referred to as a Y-Y connection configuration. It can be understood that the transformers TR1, TR2, …, TR Figure 3B can be coupled to each other in any suitable connection configuration, such as a Y-Δ connection configuration (as shown in Figure 3C ), a Δ-Y connection configuration (as shown in Figure 3D ) and a Δ-Δ connection configuration (as shown in ). The control method of the present disclosure is substantially not affected by the connection configuration of the transformers.

[0124] Figure 4A and Figure 4B illustrates an isolated multiphase series resonant converter 300 connected to a controller 400 in a preferred embodiment of the present disclosure. Except that Figure 4B the controller 400 in Figure 4B further includes a zero-current detector (ZCD), Figure 4A the controller 400 in Figure 4A and Figure 4B is substantially the same as the controller 400 in S f S = 1 / T D ) control, duty cycle (D) control, and delay time (T P1-P(2N) and secondary side switch Q S1-S(2N) , duty cycle control is applicable to the primary side switch Q P1-P(2N) , and delay time control is applicable to the secondary side switch Q S1-S(2N) .

[0125] Figure 5A , Figure 5B , Figure 5C and Figure 5D illustrate waveform examples of switch control signals and primary side current for controlling the three-phase converter 300 in a preferred embodiment of the present disclosure, including the switch control signals of primary side switches Q Figure 4B , Q P1 , Q P2 , …, Q P6 and secondary side switches Q S1 , Q S2 , …, Q S6 and the waveforms of primary side currents i P1 , i P2 and i P3 , and the converter 300 has three phases (i.e., N = 3). As Figure 5A and Figure 5B shown, all switches Q P1-P6 and Q S1-S6 operate at the same switching frequency f S (or period T S = 1 / f S ). The primary side switch Q P1-P6 operates at a duty cycle D (e.g., 50%). The delay time T D (e.g., from t1 to t2) is used to control the secondary side switch Q based on the primary side switches Q P1 , Q P3 and Q P5The turn-off moments respectively control the secondary-side switches Q S1 , Q S3 and Q S5 's turn-off moments. The switching control signals of each phase are mutually offset by approximately T S / 3 (or 120 degrees). In the case of N phases, the phases can be mutually offset by approximately T S / N. As Figure 5C and Figure 5D shown, the turn-off moments of the secondary-side switches Q S1 , Q S3 and Q S5 can be determined according to the zero-crossing moment of the primary-side current.

[0126] As Figure 4B shown, if the zero-current sensor 410 is included in the controller 400, the delay time T D can be defined according to the zero-crossing moment when the primary-side or secondary-side current changes from positive to negative. It should be noted that the delay time T D can also be defined by the time difference between the off states of the upper switches Q P1 , Q P3 and Q P5 on the primary side and the upper switches Q S1 , Q S3 and Q S5 on the secondary side.

[0127] For example, as Figure 5A and[[ID=·47]] ​ shown, the delay time T D is defined as the time period from t1 (when the upper switch Q P1 [[ID=·52]]on the primary side turns off) to t2 (when the upper switch Q S1 on the secondary side turns off). Relative to the turn-off moments of the upper switches Q P3 [[ID=·56]]and Q P5 on the primary side, the turn-off moments of the upper switches Q S3 and Q S5 on the secondary side are delayed by the same time length as the delay time T D . Since the switching frequencies of all switches are the same, and the control signals of the upper and lower switches are complementary, the delay time applied to the upper switches Q S1 , Q S3 and Q S5 can suppress or reduce the conduction time of the lower switches Q S2 , Q S4 and Q S6 .

[0128] ​ The waveforms in P1The zero-crossing point from negative to positive, lower switch Q S2 turns off, so that from t3 to t 0’ the charge is drawn from the output filter capacitor C O Conversely, the waveform in ​ can be applied under heavy load conditions, such as ​ as shown, the lower switches Q P2 and Q S2 on the primary and secondary sides turn off simultaneously (i.e., synchronous turn-off). Similarly, the lower switches Q P4 and Q S4 turn off simultaneously, and the lower switches Q P6 and Q S6 turn off simultaneously (i.e., synchronous turn-off). If the delay time T D is defined by the time difference between the off states of the lower switches Q P2 , Q P4 and Q P6 on the primary side and the lower switches Q S2 , Q S4 and Q S6 on the secondary side, then the delay time T D can be defined relative to or synchronized with the zero-crossing moment of the primary or secondary side current from negative to positive (as shown in ​ ) (as shown in ​ ).

[0129] Please refer to ​ and ​ . The sensing and scaling circuit 420 of the controller 400 receives the output voltage V OUT from the converter 300 and scales the output voltage V OUT up or down proportionally to the output voltage V OUT(SCLD) . The subtraction circuit 430 of the controller 400 receives the scaled output voltage V OUT(SCLD) and generates an error signal V REF by subtracting the scaled output voltage V OUT(SCLD) from the reference voltage V E (i.e., the reference output voltage). The error amplifier 440 of the controller 400 receives the error signal V E and provides the amplified and compensated error signal V EA to the processing circuit 450. The switching frequency f S (= 1 / T S ), duty cycle D and delay time T D are determined in the processing circuit 450.

[0130] Depending on the specific application and / or implementation aspect, the input voltage V IN , output voltage V OUTThe input current I IN and the output current I OUT Any one or a combination of more than one of them, and provide them to the processing circuit 450. To achieve ZVS, a short dead time is introduced between the on and off times of the complementary switches. In some embodiments, the zero current sensor 410 shown in ​ can be used to measure the primary side currents i P1 , i P2 and i P3 , to determine the time points when the primary side currents i P1 , i P2 and i P3 are zero, and thereby turn off the secondary side switches Q S2 , Q S4 and Q S6 (as shown in ​ ).

[0131] The delay time control enables the resonant converter to provide a boost conversion ratio. On the contrary, the traditional variable switching frequency control can only provide a buck conversion ratio in a series resonant converter. Specifically, as shown in ​ and ​ , when the traditional series resonant converter operates without delay time control, the resonant cavity elements and the transformer are always located between the input voltage source and the output voltage capacitor. The voltages on the resonant cavity elements and the transformer actually become V IN -V OUT . Assuming that the output voltage is greater than the input voltage, the resonant inductor current cannot be formed. For these reasons, the traditional series resonant converter can only provide a buck conversion ratio.

[0132] In contrast, as shown in ​ and ​ , when delay time control is adopted, during the delay time T D , the resonant cavity elements and the transformer are separated from the output voltage capacitor, where the delay time T D can be defined, for example, as the time period from t1 to t2. During the delay time T D , the voltages on the resonant cavity elements and the transformer substantially become –V IN –V OUT . Therefore, the formation speed of the resonant inductor current is much faster than that of the traditional control method. Since both V IN and V OUT help to form the resonant inductor current in the same direction, the resonant inductor current can be formed regardless of the output voltage. Thus, when the output voltage is higher than the input voltage, the delay time control can provide an output current, and thus has the ability to provide a boost conversion ratio.

[0133] Without delay time control, traditional variable switching frequency control has to cover all ranges of the voltage conversion ratio. Since the voltage conversion ratio varies with the switching frequency, in applications with a wide voltage conversion ratio range, the switching frequency range is also wide. A wide switching frequency range will result in larger drive losses and switching losses and increase the difficulty of optimizing magnetic components.

[0134] Compared with the ability of delay time control to provide a boost conversion ratio, the voltage conversion ratio range covered by traditional variable switching frequency control is narrower. Therefore, with delay time control, drive losses and switching losses can be reduced, and a better-designed magnetic element can be adopted.

[0135] The control method of the present disclosure can also be applied to a multiphase resonant converter, where the multiphase resonant converter includes a secondary side rectifier combining a diode and a controllable switch, as ​ and ​ shown. For example, ​ the upper switches Q S1 , Q S3 ,..., Q S(2N-1) of the secondary side rectifier in ​ can operate as a diode rectifier. Therefore, these upper switches can be replaced by the diodes in ​ Similarly, S2 the lower switches Q S4 , Q S(2N) of the secondary side rectifier in ​ can be replaced by the diodes in ​ and ​ It can be understood that the converters in

[0136] operate unidirectionally.

[0136] In some embodiments, the control method of the present disclosure can be extended to a multiphase or single-phase resonant converter with a full-bridge rectifier. By having a full-bridge rectifier on the secondary side, dual delay time control can be applied to each phase of the multiphase resonant converter. Given this high degree of freedom, the voltage conversion ratio range of the resonant converter can be further extended.

[0137] ​ and ​ show embodiments of an isolated three-phase resonant converter. As ​ shown, multiple transformers on the primary side are connected to each other in a Y-type configuration. On the secondary side, the transformer windings are not connected to form multiple independent full-bridge rectifiers. As ​ shown, multiple transformers on the primary side are connected to each other in a Δ-type configuration. ​ and ​ The converters shown can be extended to a multiphase converter, and can also be replaced by switches instead of diodes D SA1 , D SA2 , DSB1 , D SB2 , D SC1 and D SC2 to achieve bidirectional operation. In addition, the double-delay time control can also be applied to ​ and ​ the single-phase resonant converter shown, which will be described in detail later.

[0138] The delay time control of the present disclosure can also be used to control active current sharing in a multiphase resonant converter, thereby balancing the current magnitudes in each phase. ​ An isolated multiphase resonant converter 700 of a preferred embodiment of the present disclosure is shown, which is coupled to a controller 800 to perform active current sharing in the converter 700. ​ is ​ an enlarged schematic diagram of the controller 800 in

[0139] As ​ and ​ shown, to balance the current magnitudes in each phase, the current balance circuit 810 of the controller 800 can be used to apply an additional delay time control to each phase. Through the current sensing, scaling, and averaging circuit 811 of the current balance circuit 810, the currents i P1 , i P2 and i P3 in each phase can be sensed, scaled, and averaged to obtain the current magnitudes |i P1 | (AVG) , |i P2 | (AVG) and |i P3 | (AVG) in each phase. Based on the current difference between any two of the three phases, an additional delay time can be determined, and the additional delay time is added to the general delay time T D using one or more delay time adders 812. The total delay time is applied to the corresponding leg in the secondary side. For the first phase corresponding to the switches Q S1 and Q S2 , the additional delay time T D_1 is added to the delay time T D to balance |i P1 | (AVG) and |i P2 | (AVG) . The additional delay time T D_1 is provided by an error amplifier according to the compensation amount of the error signal, where the compensation amount is generated by subtracting |i P2 | (AVG) from |i P1 | (AVG) . Thus, T D_1 can be used to balance |i P1 |(AVG) and |i P2 | (AVG) For the second phase, it can also be balanced in the same way |i P2 | (AVG) and |i P3 | (AVG) Finally, for the third phase, it can also be balanced in the same way |i P3 | (AVG) and |i P1 | (AVG) Since |i P1 | (AVG) , |i P2 | (AVG) and |i P3 | (AVG) are all balanced with each other, the three-phase resonant converter 700 can achieve active current sharing.

[0140] ​ Shows a single-phase isolated series resonant converter 1000 with full-bridge configurations on both the primary side and the secondary side according to a preferred embodiment of the present disclosure. ​ Shows the ​ single-phase isolated series resonant converter 1000 coupled to the controller 1100 in a preferred embodiment of the present disclosure. As ​ and ​ shown, the converter 1000 includes primary-side switches Q P1 , Q P2 , Q P3 and Q P4 , secondary-side switches Q S1 , Q S2 , Q S3 and Q S4 , a resonant capacitor C P on the primary side, a transformer TR, a resonant inductor L P on the primary side, a resonant capacitor C S on the secondary side, a resonant inductor L S on the secondary side, and an output filter capacitor C O . The primary-side switches Q P1 and Q P2 form a half-bridge arm B1. The primary-side switches Q P3 and Q P4 form a half-bridge arm B2. The secondary-side switches Q S1 and Q S3 form a half-bridge arm B3. The secondary-side switches Q S2 and Q S4 form a half-bridge arm B4.

[0141] Referring simultaneously to ​ and ​In the sensing and scaling circuit 1110, the input voltage V IN And the output voltage V OUT can be sensed and scaled to V IN(SCLD) and V OUT(SCLD) . Set the reference voltage V REF Subtract V OUT(SCLD) Can generate error signal V E (i.e. V E =V REF -V OUT(SCLD) ). Then, V E The error signal V is fed into the error amplifier 1120 with compensation function to generate an amplified and compensated error signal V EA In addition, V IN(SCLD) and V OUT(SCLD) is fed into the error amplifier 1120 with compensation function and used for the signal V output by the error amplifier 1120 EA Furthermore, if necessary, the input current or output current of the converter 1000 can be further sensed and used to determine V EA . Then, V EA is transmitted to the bridge arm controller 1130. The bridge arm controller 1130 converts V EA Converted into four duty cycle signals D1, D2, D3 and D4. Duty cycle signal D1 determines the switch Q P1 The duty cycle of the switch Q P2 The control signal and switch Q P1 The duty cycle signal D2 determines the switch Q P3 The duty cycle of the switch Q P4 The control signal and switch Q P3 The duty cycle signal D3 determines the switch Q S1 The duty cycle of the switch Q S3 The control signal makes the switch Q S3 As a synchronous rectifier. The duty cycle signal D4 determines the switch Q S2 The duty cycle of the switch Q S4 The control signal makes the switch Q S4 as a synchronous rectifier.

[0142] ​ V is a preferred embodiment of the present disclosure EA Schematic diagram of the relationship between the duty cycle signals D1, D2, D3 and D4. In order to continuously provide voltage conversion ratio in a wide range, when V EA When V EAStarting from zero and increasing, the duty cycle signal D1 linearly increases from zero to 0.5. It can be understood that the duty cycle signal D1 can also increase non-linearly. Since the main control variable is the duty cycle signal D1, this region is called the D1 region. In the D1 region, V EA changes will cause the duty cycle signal D1 to change accordingly. After the duty cycle signal D1 reaches 0.5, the duty cycle signal D1 remains at 0.5, and the duty cycle signal D2 linearly increases from zero to 0.5. It can be understood that the duty cycle signal D2 can also increase non-linearly as long as it is monotonically increasing. This region is called the D2 region. In the D2 region, V EA changes will cause the duty cycle signal D2 to change in the region where it changes. In the D1 and D2 regions, since the output voltage V OUT is less than the input voltage V IN , the converter 1000 is a buck converter. When both the duty cycle signals D1 and D2 reach 0.5, the input voltage V IN is equal to the output voltage V OUT , so the D1 and D2 regions can also be called the "buck region".

[0143] When the duty cycle signal D2 reaches 0.5 at the transition point 1202, as V EA further increases, the converter 1000 becomes a boost converter. The duty cycle signals D1 and D2 remain at 0.5, while the duty cycle signal D3 increases from 0.5 to 1.0. This region is called the D3 region. In the D3 region, V EA changes will cause the duty cycle signal D3 to change accordingly. After the duty cycle signal D3 reaches 1.0, the duty cycle signals D1 and D2 remain at 0.5, the duty cycle signal D3 remains at 1.0, and D4 increases from 0.5. This region is called the D4 region. In the D4 region, V EA changes will cause the duty cycle signal D4 to change accordingly. In the D3 and D4 regions, since the output voltage V OUT is greater than the input voltage V IN , the converter 1000 is a boost converter, so the D3 and D4 regions can also be called the "boost region".

[0144] ​ and ​ respectively show the equivalent circuit and key operating waveforms when the control method of the D1 region of the converter 1000 for ​ is adopted ​ . As ​ shown, the main control variable is the duty cycle signal D1, so the half-bridge arm B1 is actuated. The duty cycle signal D2 is zero, which means that the switch Q P3 remains in the off state while the switch Q P4Maintained in the conducting state. The half-bridge arms B3 and B4 form a synchronous rectifier, so its operation is exactly the same as that of a diode rectifier. In some embodiments, the synchronous rectifier means that the gate signal of the MOSFET switch is only turned on when the current direction is from the anode to the cathode of the antiparallel diode. In other embodiments, the switches in the half-bridge arms D3 and D4 can be replaced by diodes. As ​ shown, the switching period T S is a constant and very close to the resonance period, where the resonance period is determined by the resonant cavity components. From t0 to t1 (with a duration of D1T S ), the switch Q P1 is turned on, and the resonant current i P (i.e., the primary side current) is transferred to the output side via the transformer TR and the secondary side rectifier. After t1, the switch Q P1 is turned off, and the resonant current i P decreases but remains positive. After t2, the resonant current i P becomes zero, and negative resonance occurs. The switch Q P2 is turned on for a long time so that the negative resonant current is transmitted for half the duration of the resonant period, and at the start of the next switching period, the resonant current naturally approaches zero.

[0145] ​ and ​ respectively show the equivalent circuit and key operating waveforms when the control method of the D2 region of the converter 1000 is adopted for ​ . As ​ shown, the duty cycle of the half-bridge arm B1 is fixed at 0.5. The main control variable is the duty cycle signal D2, so the half-bridge arm B2 is actuated. The half-bridge arms B3 and B4 form a synchronous rectifier, so its operation is exactly the same as that of a diode rectifier. As ​ shown, the switching period T ​ is still a constant and very close to the resonance period determined by the resonant cavity components. From t0 to t1 (with a duration of 0.5T S ), the switches Q S and Q P1 are turned on, and the resonant current i P4 is transferred to the output side via the transformer TR and the secondary side rectifier. The switches Q P and Q P1 are turned on for a long time so that the resonant current i P4 is transmitted for half the duration of the resonant period. When the switches Q P and Q P1 are turned off at t1, the resonant current i P4 is close to zero. At t1, the switches Q P and Q P1 are turned off, and the switch Q P4 is turned off, and the switch QP2 and Q P3 conducts. The resonant current i P becomes negative and is transferred to the secondary side. When the switches Q P3 and Q P4 turn off at t2, the magnitude of the resonant current i P decreases and is transferred to the secondary side. After the resonant current i P becomes zero at t3, a small ripple current flows through the switches Q P2 and Q P4 and the transformer TR until the switches Q P2 and Q P4 turn off to start a new switching cycle.

[0146] ​ and ​ respectively show the equivalent circuit and key operating waveforms when the control method of the D3 region of the ​ converter 1000 is adopted. As ​ shown, the duty cycles of the half-bridge arms B1 and B2 are fixed at 0.5. The main control variable is the duty cycle signal D3, so the switch Q S1 is represented as an active switch rather than a diode. Since the switches Q S2 , Q S3 and Q S4 form a synchronous rectifier and its operation is exactly the same as that of a diode rectifier, so it is represented by a diode. As ​ shown, the switching period T ​ is still a constant and is very close to the resonant period determined by the resonant cavity components. From t0 to t1 (with a duration of 0.5T S ), the switches Q S and Q P1 conduct, and the resonant current i P4 is transferred to the output side via the transformer TR, the switch Q P and other secondary-side rectifier components. The switches Q S1 and Q P1 conduct for a long time so that the resonant current is transferred for half of the resonant period, and when the switches Q P4 and Q P1 turn off at t1, the resonant current approaches zero. At t1, the switches Q P4 and Q P1 turn off, and the switches Q P4 and Q P2 conduct. Since the duty cycle signal D3 is greater than 0.5, the switch Q P3 remains conducting, so the current i S1 is not transferred to the output filter capacitor C P , and will rise rapidly. When the switch Q O ​S1 After turning off at t2, the rising current i P is transferred to the output filter capacitor C O , and the magnitude of the current i P decreases. After the current i P becomes zero at t3, a small fluctuating current flows through the switches Q P2 and Q P3 and the transformer TR until the switches Q P2 and Q P3 turn off to start a new switching cycle.

[0147] ​ And ​ respectively show the equivalent circuit and key operating waveforms when the control method of the D4 region of the ​ converter 1000 is adopted. As ​ shown, the duty cycles of the half-bridge arms B1 and B2 are fixed at 0.5, and the duty cycle of the half-bridge arm B3 is fixed at 1.0. Therefore, the switch Q ​ is maintained in the on state, and the switch Q S1 is maintained in the off state. The main control variable is the duty cycle signal D4, so the switch Q S3 is represented by an active switch rather than a diode. Since the switch Q S2 forms a synchronous rectifier and its operation is exactly the same as that of a diode, it is represented by a diode. As ​ shown, the switching period T is still a constant and very close to the resonant period determined by the resonant cavity components. From t0 to t1 (with a duration of 0.5T S ), the switches Q S , Q P1 , and Q P4 are on, and the resonant current i S2 rises through the transformer TR and the switches Q P and Q S1 and Q S2 . After the switch Q S2 turns off at t1, the rising resonant current is transferred to the output filter capacitor C S4 through the switch Q O . After the resonant current i P becomes zero at t2, a small fluctuating current flows through the switches Q P1 and Q P4 . After the switches Q P1 and Q P4 turn off at t3, the switches Q P2 , Q P3 , and Q S2 are on, and the resonant current i P rises through the transformer TR and the switches Q S1 and QS2 while for C S charging.

[0148] ​ and ​ respectively show the relationship diagrams between V of other preferred embodiments of the present disclosure EA and the duty cycle signals D1, D2, D3, and D4. An object of the present disclosure is to continuously provide a voltage conversion ratio corresponding to V EA Therefore, as long as at least one of the control variables D1, D2, D3, and D4 continuously changes with V EA changing, the order of these control variables can be changed or even mixed. As ​ shown, in their respective buck or boost regions, the order of D1, D2, D3, and D4 can be mixed or even changed. ​ shows another variation of the control method. As long as at least one control variable changes with V EA changing, the maximum or minimum value of D1, D2, D3, and D4 can be any value. The maximum and minimum values of D1 and D2 can be set between 0.0 and 0.5, and the maximum and minimum values of D3 and D4 can be set between 0.5 and 1.0.

[0149] ​ respectively show single-phase isolated series resonant converters of multiple preferred embodiments of the present disclosure, where the number of switch bridge arms in each converter is different. According to different gain ranges, the number of bridge arms in the converter can also be changed. ​ and ​ show an embodiment including half-bridge arms B1, B3, and B4. As ​ shown, in this embodiment, the switch Q of bridge arm B2 P3 is always in the off state, and the switch Q of bridge arm B2 P4 is always in the on state. As ​ shown, in this embodiment, the switches Q of bridge arm B2 P3 and Q P4 are respectively replaced by capacitors C P1 and C P2 respectively. ​ and ​ show an embodiment including half-bridge arms B1, B2, and B4. As ​ shown, in this embodiment, the switch Q of bridge arm B3 S3 is always in the off state, and the switch Q of bridge arm B3 S1 is always in the on state. As ​ shown, in this embodiment, the switches Q of bridge arm B3 S3 and Q S1 are respectively replaced by capacitors C S1 and CS2 Replacement. ​ An embodiment including only half - bridge arms B1 and B4 is shown. It can be understood that, among the half - bridge arms B1, B2, B3, and B4, any combination of two bridge arms (any primary - side bridge arm and any secondary - side bridge arm) can apply the control method shown and described in the present disclosure.

[0150] The present disclosure provides various control methods for a multiphase converter, enabling the converter to provide a wider range of voltage conversion ratios, thereby enhancing its performance. Specifically, the control method of the present disclosure enables a single - phase or multiphase converter to have a wide input voltage range and / or a wide output voltage range by substantially narrowing the switching - frequency range, thereby enhancing its performance. By combining variable duty - cycle, variable frequency, and delay - time control to control the output voltage or current, the switching - frequency range can be narrowed.

[0151] According to a preferred embodiment of the present disclosure, variable duty - cycle and variable - frequency control can be used to control the primary - side switches and secondary - side switches of a multiphase isolated resonant converter, while delay - time control can be used to control the secondary - side switches replacing the diode rectifier. In any phase of the multiphase resonant converter, by sensing the secondary - side current and / or primary - side current in that phase and delaying the turn - off moment of the corresponding secondary - side switch relative to the zero - crossing point of the secondary - side current or primary - side current in that phase, a switching control signal for the secondary - side switch in that phase can be obtained.

[0152] In delay - time control, since the switching control signal of the secondary - side switch is asymmetrically delayed, the relevant current zero - crossing point can be from negative to positive or from positive to negative, but cannot exist simultaneously. On the other hand, the turn - off moment of the corresponding secondary - side switch can be simply delayed relative to the turn - off moment of the primary - side switch to achieve delay - time control. The primary - side switches and secondary - side switches operate at substantially the same switching frequency, but the duty - cycle of each primary - side or secondary - side switch may vary depending on the designer's choice and the delay time.

[0153] It should be noted that delay - time control is only applied to one switch in the bridge arm of the secondary - side rectifier. If delay - time control is applied to one switch in the bridge arm of the secondary - side rectifier, delay - time control will no longer be applied to the other switch in that bridge arm to minimize the circulating current. That is, the turn - off moment is the earlier of the zero - crossing point of the current and the turn - off moment of the corresponding primary - side switch. To achieve ZVS operation, a short dead - time is introduced between the turn - off moment of any switch and the turn - on moment of its complementary switch in the primary - side and secondary - side.

[0154] For ease of explanation and definition of the technical content of the present disclosure, terms such as "substantially", "about", "slightly", "relatively", etc. are used to represent an inherent degree of uncertainty, which may be caused by factors such as quantitative comparison, numerical values, sensing, etc. These terms generally mean that the deviation from a given value or range is within 10%, 5%, 1%, or 0.5%, and such deviation does not affect the basic function of the corresponding technical feature. Unless otherwise specifically stated, the numerical parameters stated in the present disclosure can be regarded as specific numerical values or numerical values within their error ranges.

[0155] It should be noted that the above are only preferred embodiments proposed for the purpose of explaining the present disclosure. The present disclosure is not limited to the described embodiments, and the scope of the present disclosure is determined by the appended claims. And the present disclosure can be variously modified by those skilled in the art, but all are not beyond what is intended to be protected by the appended claims.

Claims

1. An isolated resonant converter, comprising: One or more phases, each of which includes a transformer, a plurality of resonant elements, a plurality of primary side switches electrically coupling an input terminal to the transformer via the resonant elements, and a plurality of secondary side switches electrically coupling the transformer to an output terminal; and a control circuit electrically coupled to the one or more phases, wherein the control circuit is configured to: sense an input voltage on the input terminal and an output voltage on the output terminal; determine a first control signal for the primary side switches and a second control signal for the secondary side switches based on a plurality of parameters, wherein the plurality of parameters includes physical characteristics of the resonant element, the input voltage, the output voltage, and a reference voltage; output the first control signal including a switching frequency and a first duty cycle to the primary side switches; and output the second control signal including the switching frequency and a second duty cycle to the secondary side switches, wherein, the second duty cycle of the first of the plurality of secondary side switches is greater than the first duty cycle of the first of the corresponding plurality of primary side switches, wherein the control circuit includes: a sensing and scaling circuit configured to receive the input and output voltages and convert the input and output voltages into scaled input and output voltages; a subtraction circuit configured to receive the scaled output voltage and generate an error signal by subtracting the reference voltage from the scaled output voltage; an error amplifier configured to receive the error signal and generate an amplified and compensated error signal; and a processing circuit configured to receive the scaled input voltage and the amplified and compensated error signal and generate the first control signal and the second control signal based on the scaled input voltage and the amplified and compensated error signal.

2. The isolated resonant converter as claimed in claim 1, wherein, The second duty cycle of the second of the plurality of secondary side switches is defined according to a turn-off time of the second of the plurality of primary side switches.

3. The isolated resonant converter according to claim 1, wherein, The control circuit is further configured to sense a current flowing through each of the phases.

4. The isolated resonant converter according to claim 3, wherein, The second duty cycle of the second of the plurality of secondary side switches is defined according to a zero-crossing time when the current changes from positive to negative or from negative to positive.

5. The isolated resonant converter according to claim 1, wherein, The control circuit further includes a zero current sensor configured to sense a current signal flowing through each of the phases.

6. The isolated resonant converter according to claim 5, wherein, The processing circuit is further configured to receive the current signal and generate the first control signal and the second control signal based on the scaled input voltage, the amplified and compensated error signal, and the current signal.

7. The isolated resonant converter according to claim 1, wherein, The one or more phases include at least two phases, and the control circuit further includes a current balance circuit configured to adjust the second control signal before the second control signal is transmitted to the secondary side switches to balance the currents flowing through different phases of the at least two phases.

8. The isolated resonant converter according to claim 7, wherein, The current balance circuit includes: a current sensing, scaling, and averaging circuit configured to obtain an average value of the current flowing through each of the phases; and a delay time adder configured to determine a delay time for each of the phases based on a difference in the currents of any two of the phases and adjust the second control signal by adding the delay time to a duty cycle of the second control signal.

9. A method for controlling an isolated resonant converter, wherein, The isolated resonant converter has one or more phases, each phase including a transformer, a plurality of resonant elements, a plurality of primary side switches electrically coupling an input terminal to the transformer via the resonant elements, and a plurality of secondary side switches coupling the transformer to an output terminal. The method includes: Sensing an input voltage on the input terminal and an output voltage on the output terminal; Determining a first control signal for the primary side switches and a second control signal for the secondary side switches based on a plurality of parameters, where the plurality of parameters includes physical characteristics of the resonant elements, the input voltage, the output voltage, and a reference voltage; Outputting the first control signal including a switching frequency and a first duty cycle to the primary side switches; and Outputting the second control signal including the switching frequency and a second duty cycle to the secondary side switches, wherein the second duty cycle of the first of the plurality of secondary side switches is greater than the first duty cycle of the first of the plurality of corresponding primary side switches, wherein the method includes: Receiving the input and output voltages and converting the input and output voltages to scaled input and output voltages; Receiving the scaled output voltage and generating an error signal by subtracting the scaled output voltage from the reference voltage; Receiving the error signal and generating an amplified and compensated error signal; and Receiving the scaled input voltage and the amplified and compensated error signal and generating the first control signal based on the scaled input voltage and the amplified and compensated error signal, and generating the second control signal based on the scaled input voltage and the amplified and compensated error signal.

10. The method according to claim 9, wherein, The second duty cycle of the second of the plurality of secondary side switches is defined according to a turn-off time of the second of the plurality of primary side switches.

11. The method according to claim 9, further comprising sensing a current flowing through each phase.

12. The method according to claim 11, wherein, The second duty cycle of the second of the plurality of secondary side switches is defined according to a zero-crossing time when the current changes from positive to negative or from negative to positive.

13. The method according to claim 9, further comprising adjusting the second control signal before the second control signal is transmitted to the secondary side switches to balance currents flowing through different phases of at least two phases.

14. An isolated resonant converter, comprising: A transformer; A plurality of resonant elements; A primary side full-bridge circuit including a first arm and a second arm and electrically coupling an input terminal to the transformer via the resonant elements; A secondary side full-bridge circuit including a third arm and a fourth arm and coupling the transformer to an output terminal; and A control circuit electrically coupled to the primary side full-bridge circuit and the secondary side full-bridge circuit, wherein the control circuit is configured to: Sense an input voltage on the input terminal and an output voltage on the output terminal; Determine the control signals for the first arm, the second arm, the third arm, and the fourth arm according to an amplified and compensated error signal, where the control signals include the duty cycles of at least one of the first arm, the second arm, the third arm, and the fourth arm; and Output the control signals to the primary side full-bridge circuit and the secondary side full-bridge circuit, where when the amplified and compensated error signal is less than a threshold, the isolated resonant converter is a buck converter, where when the amplified and compensated error signal is greater than the threshold, the isolated resonant converter is a boost converter.

15. The isolated resonant converter according to claim 14, wherein, When the amplified and compensated error signal is less than the threshold, the duty cycle of the first arm is between 0.0 and 0.5, and the duty cycle of the second arm is 0.

0.

16. The isolated resonant converter according to claim 14, wherein, When the amplified and compensated error signal is less than the threshold, the duty cycle of the first arm is 0.5, and the duty cycle of the second arm is between 0 and 0.

5.

17. The isolated resonant converter according to claim 14, wherein, When the amplified and compensated error signal is greater than the threshold, the duty cycles of the first arm and the second arm are 0.5, the duty cycle of the third arm is between 0.5 and 1.0, and the duty cycle of the fourth arm is 0.

5.

18. The isolated resonant converter according to claim 14, wherein, When the amplified and compensated error signal is greater than the threshold, the duty cycles of the first arm and the second arm are 0.5, the duty cycle of the third arm is 1.0, and the duty cycle of the fourth arm is between 0.5 and 1.

0.

19. The isolated resonant converter according to claim 14, wherein, As the amplified and compensated error signal increases, the duty cycle increases monotonically.

20. The isolated resonant converter as claimed in claim 19, wherein, As the amplified and compensated error signal increases, the duty cycle increases linearly.

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