Hybrid control method for dead-time control of LLC series resonant converter

Abstract

The application provides a mixed control LLC series resonant converter dead-time control method, which respectively calculates the current of phase leading and lagging bridge arm switch time and the shortest dead-time required for realizing ZVS (zero voltage switching), and uses a parameter fitting method to simplify the calculation amount, and controls the dead-time of two bridge arm switches in real time on an embedded controller. The application considers the different dead-time required for realizing zero voltage switching of the two bridge arm switches of the primary side of the mixed control LLC series resonant converter, and controls in real time, so that the primary side switch loses zero voltage switching due to working condition change is prevented, and the reverse conduction time of the switch is reduced as much as possible, and the loss is reduced. The application has good application effect in the mixed control LLC series resonant converter with high frequency and wide gain range.

Description

TECHNICAL FIELD

[0001] The present application relates to the field of power electronics, in particular to a variable dead-time control method for a hybrid control LLC series resonant converter. BACKGROUND

[0002] LLC series resonant DC / DC converter is widely concerned due to its simple circuit structure, superior switching performance and other advantages. It is easy to realize the zero-voltage switching (ZVS) of the primary side switch tube and the zero-current switching (ZCS) of the secondary side rectifier tube, which is convenient to improve the switching frequency and the power density.

[0003] The LLC series resonant converter generally adopts a variable frequency control strategy to obtain the corresponding gain by changing the switching frequency to stabilize the output voltage. However, when the input and output voltage range is wide, the frequency variation range is large, which leads to difficulties in magnetic element design and EMI design. Therefore, a hybrid control strategy of variable frequency + phase shift is generally used to reduce the frequency variation range. However, when the phase shift angle gradually increases, the turn-on current of the phase-lead bridge arm gradually increases, and the turn-off current of the phase-lag bridge arm gradually decreases. In order to ensure that the primary side switch tube realizes ZVS (zero voltage switching), a suitable dead-time is given to the primary side bridge arm, so that the bridge arm turn-off current can charge and discharge the parasitic output capacitor of the switch tube within the dead-time.

[0004] The traditional method fixedly uses the maximum dead-time required by the primary side switch tube to realize ZVS, and does not distinguish between the two bridge arms. However, in the high-frequency hybrid control LLC series resonant converter, when the phase shift angle is large, the dead-time required by the phase-lead bridge arm switch tube is small, and the reverse conduction time of its body diode will be prolonged, which will cause large reverse conduction loss. The existing technology analyzes the variable dead-time control of the LLC resonant converter under variable frequency control. This method can dynamically adjust the dead-time of the primary side switch tube according to the switching frequency, load condition and other parameters of the circuit during operation, but this method is not suitable for LLC resonant converters under hybrid control of variable frequency and phase shift. SUMMARY

[0005] Therefore, it is necessary to propose a dead-time control method for a hybrid control LLC series resonant converter, which can dynamically adjust the dead-time of the two bridge arm switch tubes according to the operation of the circuit, prevent the switch tube from losing the zero voltage turn-on state due to the short dead-time, or prevent the switch tube from having large reverse conduction loss due to the long dead-time, so as to improve the overall operation efficiency of the circuit.

[0006] The purpose of the present application is to disclose a variable dead-time control method for a hybrid control LLC series resonant converter, which improves the problem of unreasonable setting of the dead-time of the phase-lead and phase-lag bridge arm switch tubes, and reduces the operation efficiency of the circuit. The present application has the advantages of accurate calculation results, small calculation amount, high operation efficiency, easy implementation and the like.

[0007] This invention calculates the current at the switching moments of the leading and lagging bridge arms, as well as the shortest dead time required to achieve zero-voltage turn-on. It simplifies the calculations using parameter fitting and calculates and controls the dead time of the two bridge arm switches in real time on an embedded controller. This invention considers the different dead times required for the two primary-side bridge arm switches to achieve zero-voltage turn-on in a hybrid-controlled LLC series resonant converter and implements real-time control to prevent the primary-side switches from losing zero-voltage turn-on due to changes in operating conditions. It also minimizes the reverse conduction time of the switches, thus reducing losses. This invention shows good application results in high-frequency, wide-gain-range hybrid-controlled LLC series resonant converters.

[0008] The present invention specifically adopts the following technical solution:

[0009] A method for variable dead-time control of a hybrid control LLC series resonant converter, characterized by comprising the following steps:

[0010] Step S1: Obtain the circuit hardware parameters and input / output parameter ranges, including: resonant inductor. L r resonant capacitor C r ,transformer T x primary excitation inductance L m Transformer turns ratio n Primary-side switching transistor output capacitor C oss Input voltage V i Variation range, output voltage V o Variation range and output current I o Range of variation;

[0011] Step S2: Using an improved time-domain analysis method, calculate the current at the turn-off time of the bridge arm switches with leading and lagging phases of the drive signal for the hybrid control LLC resonant converter under different input and output parameters. i off ;

[0012] The improved time-domain analysis method includes: listing the resonant current at different time periods within half a switching cycle. i Lr Excitation current i Lm With resonant capacitor voltage v Cr By considering the equations satisfied and the symmetry of the resonant waveform of the full-bridge LLC resonant converter in steady state, the resonant current at any given time can be solved. iLr Excitation current i Lm With resonant capacitor voltage v Cr ;

[0013] Step S3: Based on the output capacitance of the primary-side switching transistor C oss and the shut-off current i off The shortest dead time required for the primary-side switch to achieve zero-voltage turn-on under different input and output parameters was calculated. t dead_min ;

[0014] The method for calculating the shortest dead time is as follows: , C oss The output capacitor of the primary-side switching transistor;

[0015] Step S4: Use a simple fitting function model f ( f s , θ Fitting the shortest dead time t dead_min With switching frequency f s and phase shift angle θ The functional relationship is used to solve for the undetermined parameters in the fitted function model;

[0016] Step S5: Apply the solved fitting function model f ( f s , θ The data is written into the embedded controller and calculated in real time during circuit operation to obtain the shortest dead time required for each primary-side switch to achieve zero-voltage turn-on under this operating condition.

[0017] Step S6: After adding an appropriate margin to the shortest dead time calculated in step S5, the final dead time signal is transmitted to the PWM generator, and the dead time of the primary-side switching transistor is controlled by the drive circuit.

[0018] Furthermore, in the improved time-domain analysis method, in the magnetizing inductor... L m resonant inductor L r With resonant capacitor C r The time period t of common resonance sp Within this context, the resonant current is considered to change linearly, and the rate of this linear change is t. sp The average rate of change of the resonant current at the endpoint.

[0019] Further, the improved time domain analysis method specifically includes:

[0020] Step S21: Obtain specific software and hardware parameters of the controlled LLC series resonant converter in operation, including:

[0021] Resonant inductance L r Excitation inductance L m Resonant capacitance C r Transformer ratio n, switching frequency f s Phase shift angle θ Input voltage V i Output voltage V o Output current I o ; wherein, V i 、V o 、I o At least two of them should be obtained;

[0022] Step S22: Calculate the initial value of the resonant current and the resonant capacitor voltage under the condition:

[0023] (1)

[0024] wherein, i Lr denotes the resonant current; v Cr denotes the resonant capacitor voltage; t 0~2 and t 2~3 respectively denote the time from t0 to t2 and the time from t2 to t3, t0 is the initial time of a switching period, t1 is the time when the leading bridge arm is turned off, t2 is the time when the secondary side current is interrupted in the first half of the switching period, t3 is the middle time of the switching period; k 23 denotes the rate of change of the resonant inductance current in the time period from t2 to t3;

[0025] Solve equation set (1) to obtain the initial value of the resonant current i Lr (t0) and the initial value of the resonant capacitor voltage v Cr (t0);

[0026] Step S23: According to the circuit operation, list i Lr and v CrThe differential equation satisfied in the period from t0 to t3, in combination with the initial value condition in formula (1), is solved to obtain the time-domain expression of i Lr and v Cr The corresponding differential equation is

[0027] (2)

[0028] Step S24: According to the boundary condition satisfied by i Lr and v Cr , substitute into the time-domain expression of i Lr and v Cr , eliminate the unknowns therein to obtain the final expression, and the corresponding boundary condition is

[0029] (3)

[0030] Step S25: Use a numerical solution method to obtain the turn-off currents of the two bridge arms under different operating conditions, including: the lead bridge arm turn-off current i Lr (t1) and the lag bridge arm turn-off current i Lr (t3).

[0031] Further, the simple fitting function model used in step S4 is

[0032]

[0033] wherein, f s denotes the switching frequency; θ denotes the phase shift angle, which is greater than or equal to 0, C 0 -C 5 is a parameter to be fitted; the phase shift angle θ ; the dead time under different load resistances is fitted to simplify the dead time as a simple function of the switching frequency and the phase shift angle.

[0034] Further, the shortest dead time required for the phase lag bridge arm switch to achieve zero-voltage turn-on is greater than or equal to the shortest dead time required for the phase lead bridge arm switch to achieve zero-voltage turn-on.

[0035] Further, a voltage and current sampling circuit is connected to the load of the LLC series resonant converter to collect the output voltage V o and the output currentI o The information is embedded into the control signal; four switch tubes of the LLC series resonant converter are connected by a driving circuit to provide a driving signal; the embedded controller is connected with a voltage and current sampling circuit and the driving circuit.

[0036] Compared with the prior art, the scheme provided by the application and the preferred schemes thereof can dynamically adjust the dead time of the different bridge arm switch tubes of the primary side according to the change of the input and output without increasing the hardware circuit resources, so that the switch tubes work in a state that can realize ZVS, the ZVS condition of part of the switch tubes is prevented from being lost due to the change of the working condition, and the reverse conduction loss of the switch tubes of the primary side is greatly reduced when working at a high frequency and a large phase shift angle. The scheme has the advantages of less operation amount, easy implementation, high circuit operation efficiency and the like. BRIEF DESCRIPTION OF DRAWINGS

[0037] The application will be further described in detail below in combination with the drawings and specific embodiments:

[0038] Figure 1 It is a full-bridge LLC series resonant converter topology diagram of the embodiment of the application.

[0039] Figure 2 It is a time domain waveform diagram of the phase shift frequency conversion hybrid control LLC series resonant converter of the embodiment of the application.

[0040] Figure 3 It is an equivalent circuit diagram of the resonant cavity when the full-bridge LLC series resonant converter of the embodiment of the application works in different periods.

[0041] Figure 4 It is an equivalent circuit diagram of the turn-off moment of the upper switch tube of the phase lead bridge arm of the full-bridge LLC series resonant converter of the embodiment of the application.

[0042] Figure 5 It is an equivalent circuit diagram of the turn-off moment of the lower switch tube of the phase lag bridge arm of the full-bridge LLC series resonant converter of the embodiment of the application.

[0043] Figure 6 It is a principle diagram of the variable dead time control method of the hybrid control LLC series resonant converter of the embodiment of the application.

[0044] Figure 7 It is a comparison diagram of the PSIM simulation waveforms of the existing dead time control method and the variable dead time control method of the application. DETAILED DESCRIPTION

[0045] In order to make the features and advantages of the patent more obvious and easy to understand, the following specific embodiments are taken as examples and described in detail as follows.

[0046] It should be noted that the following detailed description is exemplary in nature and is intended to provide further description of the application. Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application belongs.

[0047] It is to be understood that the terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of example embodiments in accordance with the present application. As used herein, the singular forms "a", "an" and "the" are intended to include the plural forms as well, unless the context clearly indicates otherwise. It will be further understood that the terms "comprises" and / or "comprising," when used in this specification, specify the presence of stated features, steps, operations, elements, components, and / or groups thereof, but do not preclude the presence or addition of one or more other features, steps, operations, elements, components, and / or groups thereof.

[0048] In the scheme provided by the present application:

[0049] In the first aspect, an improved time domain analysis method is provided, which is applied to a phase-shift and frequency-modulation hybrid control full-bridge LLC series resonant converter to obtain the off current of the phase-advanced and phase-lagged bridge arms. The full-bridge LLC series resonant converter sequentially comprises an input capacitor, an inverter bridge, a resonant cavity, and a rectification filter circuit. The inverter bridge comprises two half-bridge arms each composed of four switching tubes. If the phase difference between the driving signals of the two bridge arms is greater than zero, the bridge arm with the phase-advanced driving signal is the phase-advanced bridge arm, and vice versa. θ The resonant cavity comprises a resonant inductor, a transformer, and a resonant capacitor and is formed by cascading them. The rectification filter circuit comprises a full-wave, full-bridge, or voltage-doubling rectification circuit and a capacitor filter circuit, which is used to convert the alternating current output by the transformer into direct current.

[0050] The method is specifically implemented as follows:

[0051] Step S21: Obtain the specific hardware and software parameters of the LLC series resonant converter during operation, including:

[0052] resonant inductor L r excitation inductance L m resonant capacitor C r transformer turns ratio n, switching frequency f s phase shift angle θ input voltage V i output voltage V o output current I o Among them, V i 、V o、I o At least two of them should be obtained.

[0053] Step S22: Calculate the resonance current and the resonance capacitor voltage initial value conditions:

[0054] (1)

[0055] Wherein, i Lr I (t) refers to the resonance current; v Cr V (t) refers to the resonance capacitor voltage; t 0~2 And t 2~3 t0 is the initial time of a switching cycle, t1 is the time when the leading bridge arm is turned off, t2 is the time when the secondary side current is intermittent in the first half of the switching cycle, t3 is the middle time of the switching cycle; k 23 I' (t) refers to the rate of change of the resonance inductor current in the period from t2 to t3.

[0056] Solving equation set (1) can obtain the resonance current initial value i Lr (t0) and the resonance capacitor voltage initial value v Cr (t0).

[0057] Step S23: According to the circuit operation, list i Lr And v Cr The differential equation satisfied in the period from t0 to t3, combined with the initial value condition in formula (1), is solved to obtain i Lr And v Cr The time domain expression of. The corresponding differential equation is:

[0058] (2)

[0059] It is noted that the time domain expression obtained at this time contains unknown t2, and V i 、V o 、I o One of them.

[0060] Step S24: According to the boundary conditions satisfied by i Lr And v Cr , bring into i Lr And vCr The unknowns in the time-domain expression are eliminated to obtain the final expression. The corresponding boundary conditions are:

[0061] (3)

[0062] Step S25: using a numerical solution to obtain the turn-off current of the two bridge arms under different operating conditions, including: the leading bridge arm turn-off current i Lr (t1), the lagging bridge arm turn-off current i Lr (t3).

[0063] By the improved time-domain analysis method of the first aspect, the DC gain of the LLC series resonant converter can be accurately calculated, and parameters such as resonant current at any time can be obtained. Compared with the traditional method, the calculation complexity is slightly increased, but the calculation accuracy is greatly increased, and the accuracy of the dead time control is ensured.

[0064] In the second aspect, a dead time calculation and numerical fitting method is provided for accurately calculating the dead time required by the LLC series resonant converter to achieve ZVS of the two bridge arms, and using a simple fitting function to fit the functional relationship between the required dead time and the circuit hardware and software parameters.

[0065] The method is implemented as follows:

[0066] Step A1: obtaining the distributed parameters of the primary side switch tube of the LLC series resonant converter to be controlled, including the output capacitance of the primary side switch tube C oss .

[0067] Step A2: combining the improved time-domain analysis method of the first aspect to obtain the minimum dead time required by the switch tube.

[0068] (4)

[0069] wherein, i off the bridge arm turn-off current, the leading bridge arm turn-off current is i Lr (t1), and the lagging bridge arm turn-off current is i Lr (t3).

[0070] Step A3: using mathematical software (such as mathcad, etc.) to fit the dead time under different load resistances, and simplifying the dead time as a simple function of the switching frequency and the phase shift angle. The fitting function is:

[0071] (5)

[0072] wherein, f s denotes the switching frequency, θ denotes the phase shift angle, C 0 -C 5 is the parameter to be fitted.

[0073] By the dead time calculation and fitting method of the second aspect, the dead time required for the switching tube to achieve ZVS in the hybrid control LLC series resonant converter can be accurately calculated. After parameter fitting using the function, the complexity of the dead time calculation is greatly reduced, and the method is convenient to use in embedded controllers.

[0074] In a third aspect, a hybrid control LLC series resonant converter bridge arm variable dead time control method is provided for accurately controlling the dead time of two bridge arms to adapt to changes in circuit operating conditions. The method comprises:

[0075] Step B1: Sample the output voltage and output current, and adjust the output voltage to be stable through closed-loop control.

[0076] Step B2: According to the control parameters (switching frequency, phase shift angle) stored in the controller, the optimal dead time of the leading and lagging bridge arms is calculated respectively by using the fitting function.

[0077] Step B3: On the basis of the optimal dead time of the two bridge arms, a small amount of margin is added respectively, and the corresponding driving signals are output to the driving circuit to control the switching of the switching tube.

[0078] The present application and the corresponding principles are further described below in conjunction with the accompanying drawings:

[0079] Figure 1 is a full-bridge LLC series resonant converter topology, wherein V i is the input voltage; Q1 - Q4 is a switching tube, which constitutes a full-bridge inverter circuit; C q1 - C q4 is the parasitic output capacitance of the switching tube; L r is a resonant inductor, C r is a resonant capacitor, T x is a transformer with a center tap, and the transformer excitation inductance is L m , D1 , D2 is a rectifier diode, which constitutes a full-wave rectifier circuit,C o For output filter capacitor, R o The load resistor is used. In this embodiment, preferably, a GaN HEMT is used as the switching transistor. It has no parasitic body diode, but due to the symmetry of its structure, it can still conduct in reverse and does not have the reverse recovery process of a normal diode. The disadvantage is that its reverse conduction voltage drop is large, and long-term reverse conduction will cause large losses.

[0080] Figure 2 The above-mentioned full-bridge LLC resonant converter operates in phase-shift frequency modulation hybrid control mode. In this embodiment, the secondary rectifier current is discontinuous. Figure 3 The equivalent circuit diagrams of the resonant cavity at different time periods are shown below. The analysis method is described below by time period, combining the time-domain waveform diagrams and the equivalent circuit diagrams. For ease of analysis, time t0 indicates t=0.

[0081] [t0-t1]: Switching transistor Q1 , Q4 When the circuit is turned on, the input voltage to the resonant cavity is the power supply voltage, the secondary diode is turned on, the magnetizing inductor is clamped by the output voltage, and only the resonant inductor and resonant capacitor participate in resonance. The equivalent circuit diagram for this period is as follows: Figure 3 As shown in (a), the resonant current, excitation current, and resonant capacitor voltage satisfy the following equation:

[0082] (6)

[0083] [t1-t2]: Switching transistor Q1 Turn off, Q3 , Q4 The circuit is switched on. With the resonant cavity input short-circuited, the secondary rectifier diode continues to carry current, and the resonant current drops rapidly. The equivalent circuit diagram for this period is as follows: Figure 3 As shown in (b), the resonant current, excitation current, and resonant capacitor voltage satisfy the following equations:

[0084] (7)

[0085] [t2-t3]: At time t2, the current of the secondary rectifier diode drops to zero, and the diode naturally turns off. The magnetizing inductor is no longer clamped by the output voltage and participates in resonance. The equivalent circuit diagram for this period is as follows: Figure 3 As shown in (c). Traditional analysis methods assume that the magnetizing inductance is much larger than the resonant inductance, therefore treating the magnetizing current during this period as a constant value. However, this is not always true when the phase shift angle is large. L m / L rThe smaller the time, the greater the analysis error. There are also methods of directly using differential equations for calculation, but the overall calculation is large and not intuitive. The present application improves the traditional analysis method, linearizes the excitation current in this period, and takes the average of the resonance current change rate at t2 and t3 as the rate of resonance current change in the whole period. In the t2-t3 period, according to the equivalent circuit Figure 3 (c) shown, the resonance current and the resonance capacitor voltage satisfy the relationship:

[0086] (8)

[0087] Therefore, the rate k of change of the resonance current at any time is:

[0088] (9)

[0089] Then the average k of the resonance current change rate at t2 and t3 is: 23

[0090] (10)

[0091] It is assumed that in the t2-t3 period, the resonance current changes linearly at a rate of k 23 , then the equation in this period is simplified to:

[0092] (11)

[0093] In addition, since the system is stable, the resonance waveform of the adjacent half switching period is the same and symmetric about the time axis, so:

[0094] (12)

[0095] In the above equation, L m ;L r ;C r are circuit hardware parameters, which are determined during design and are known parameters; t0 and t2 can be obtained by known parameters switching frequency f s and phase shift angle θ .

[0096] Therefore, by combining equations (6), (7), (10), (11), (12), the numerical solution of the DC gain of the hybrid control LLC series resonant converter M = nV o / V i can be obtained by mathematical software. If it is set that V o and​V i One of them, then, can be further explained above the equation to get the value of the resonant current, the excitation current, the resonant capacitor voltage at any time.

[0097] Further, according to Figure 2 The waveform diagram shown, t1 and t3 time of the resonant current is the leading leg and the lagging leg of the off current. In t1 and t3 time, the current flow in the main circuit is respectively as Figure 4 And Figure 5 As shown in the figure, after the switch tube is turned off, the resonant current will be the output capacitor of the two switch tubes of this bridge arm. In order to ensure that the LLC resonant converter can reliably realize zero voltage turn-on, it is necessary to ensure that the output capacitor of the switch tube is completely charged and discharged in the dead time. Therefore, the shortest dead time of the leading and lagging bridge arms is:

[0098] (13)

[0099] Where, C oss The output capacitor of the primary side switch tube; i off The off current of the switch tube, the leading and lagging bridge arms i off The resonant current at t1 and t3 time respectively i Lr ( t 1 )、 i Lr ( t 3 )。

[0100] Because the phase leading bridge arm switch tube Q1 、 Q3 The off current i Lr ( t 1 ) is greater than the phase lagging bridge arm switch tube Q2 、 Q4 The off current i Lr ( t 3 ), therefore, the leading bridge arm switch required dead time is shorter, the lagging bridge arm switch required dead time is larger. The driving signal of the two bridge arms switch tube is given different dead time, which can reduce the reverse conduction loss of the switch tube, and further reduce the loss.

[0101] The above method can accurately calculate the dead time required by the switch tube, but the overall calculation amount is large. In practical application, although it can be calculated again after several switching cycles, the embedded controller is still difficult to bear. In order to facilitate the use of the above variable dead time control method in the embedded controller, the parameter fitting method is further introduced below to reduce the calculation amount.

[0102] When the switching frequency f s , the phase shift angle θ and the load resistance R o remain unchanged, the circuit DC gain M remains unchanged, the shape of the resonant cavity current waveform is independent of the input and output voltage, and the current size is proportional to the input and output voltage. According to equation (13), the minimum dead time required by the switch tube is also proportional to the input voltage. Therefore, the minimum dead time required by the switch tube is independent of the input and output voltage, and is determined by the switching frequency f s , the phase shift angle θ and the load resistance R o .

[0103] When the output resistance does not change, the minimum dead time required by the switch tube is a function of the switching frequency f s and the phase shift angle θ . By using the improved time domain analysis method, the minimum dead time required by the switch tube under different switching frequency f s and phase shift angle θ is calculated. Further, the fitting function is used to fit the function as follows:

[0104] (14)

[0105] wherein, C 0 -C 5 is the parameter to be fitted.

[0106] Further, for the case of load resistance change, multiple possible load resistance values can be calculated to obtain different fitting functions. When the circuit is running, the actual load resistance value is detected first, and the linear interpolation method is used to estimate the minimum dead time required by the switch tube.

[0107] Based on the improved time domain analysis method and the bridge arm optimal dead time calculation and parameter fitting method, the bridge arm variable dead time control method of the hybrid control LLC series resonant converter can be obtained. The principle is shown in Figure 6 , which includes:

[0108] Step 1: According to the hardware parameters and indicators, the optimal dead time under different working conditions is calculated offline.

[0109] Step 2: The fitting function is used to fit the dead time function, and the calculated fitting function is written into the embedded controller.

[0110] Step 3: When the circuit is initially running, the output voltage and output current are collected in real time, and the voltage and current closed-loop control is normally performed. At this time, the circuit dead time is large, which prevents the primary side switch tube from losing zero voltage turn-on.

[0111] Step 4: After the circuit is basically stable, the switching frequency, phase shift angle and other parameters are obtained from the closed-loop control module, and combined with the output voltage, output current and other parameters, the required dead time of the leading bridge arm and the lagging bridge arm is calculated by the fitting function.

[0112] Step 5: The dead time data calculated in step 4 is increased by a suitable margin and transmitted to the PWM generator to output a drive signal containing the target dead time, thereby reducing the reverse conduction loss of the switch tube.

[0113] Step 6: When the dead time calculator detects that the circuit output voltage, output current, switching frequency, phase shift angle and other parameters change greatly, steps 4 and 5 are repeated again to ensure that the primary side switch tube can achieve zero voltage turn-on.

[0114] Therefore, the dead time calculator only needs to recalculate when the circuit working condition changes greatly, which further reduces the calculation amount of the embedded controller.

[0115] The PSIM simulation circuit waveform diagrams of the existing dead zone control method and the variable dead zone control method of the present application are shown in (a) and (b) of Figure 7 respectively. The transformer works at a switching frequency of 700 kHz and a phase shift angle of 100°. In order to make both bridge arm switch tubes achieve zero voltage turn-on, the phase leading and lagging bridge arm switch tube dead time in the existing method is consistent, both being 20 ns. Under the same conditions, after using the variable dead zone control method of the present application, the leading bridge arm switch tube dead time is reduced to about 3 ns, and the lagging bridge arm switch tube dead time is reduced to about 17 ns, and both bridge arm switch tubes achieve zero voltage turn-on. This method ensures that the switch tube can achieve zero voltage turn-on under different circuit operating conditions, and greatly reduces the reverse conduction time of the switch tube and the loss in the switching process.

[0116] The above merely describes preferred embodiments of the present application, but is not intended to limit the present application to other forms. Any person skilled in the art can make changes or modifications to the above disclosed technical contents to obtain equivalent embodiments. However, any simple modification, equivalent change and modification made to the above embodiments without departing from the technical solution of the present application and according to the technical essence of the present application shall still fall within the protection scope of the present application.

[0117] The present application is not limited to the above preferred embodiments, and any person can derive other various forms of hybrid control LLC series resonant converter dead-time control methods based on the present application. Any equivalent change and modification made within the scope of the present application shall fall within the scope of the present application.

Claims

1. A hybrid control LLC series resonant converter dead-time control method, characterized in that, The method comprises the following steps: Step S1: Obtain the circuit hardware parameters and input / output parameter ranges, including: resonant inductor. L r resonant capacitor C r ,transformer T x primary excitation inductance L m Transformer turns ratio n Primary-side switching transistor output capacitor C oss Input voltage V i Variation range, output voltage V o Variation range and output current I o Range of variation; Step S2: using the improved time domain analysis method, the current at the turn-off time of the bridge arm switch tube with phase lead and lag of the driving signal of the hybrid control LLC resonant converter is calculated under different input and output parameters respectively i off ; The improved time domain analysis method includes: column write out half a switching period, different time period resonant current i Lr , excitation current i Lm and resonant capacitor voltage v Cr The equation is satisfied, and the symmetry of the resonant waveform of the full-bridge LLC resonant converter in the steady state is combined to solve the resonant current at any time i Lr , excitation current i Lm and resonant capacitor voltage v Cr ; in the improved time domain analysis method, the excitation inductance L m , resonant inductance L r and resonant capacitor C r Common resonance time period t sp , the resonant current is regarded as linear change, and the linear change rate is t sp The average value of the change rate of the resonant current at two end points. Step S3: calculating the shortest dead time of the primary switch according to the output capacitance of the primary switch C oss and the turn-off current i off , and the shortest dead time of the primary switch required for realizing zero-voltage turn-on under different input and output parameters is calculated t dead_min ; Among them, the shortest dead zone time calculation method is , C oss Cin is the output capacitance of the primary-side switch tube Step S4: Use a simple fitting function model f ( fs , θ Fit the shortest dead time for the lead and lag bridge arms respectively. t dead_min With switching frequency f s and phase shift angle θ The functional relationship is used to solve for the undetermined parameters in the fitted function model; Step S5: write the solved fitting function model into the embedded controller and calculate in real time when the circuit is running to obtain the shortest dead time required for each switch tube of the primary side to realize zero-voltage turn-on under this working condition f f s θ ​​​ Step S6: After adding a suitable margin to the shortest dead-time obtained in step S5, a final dead-time signal is transmitted to the PWM generator, and the primary-side switch is controlled by the driving circuit according to the dead-time.

2. The hybrid control LLC series resonant converter dead-time control method of claim 1, wherein: The improved time domain analysis method specifically comprises: Step S21: Obtain specific software and hardware parameters of the LLC series resonant converter in operation, including: resonant inductance L r , excitation inductance L m , resonant capacitance C r , transformer ratio n, switching frequency f s , phase shift angle θ , input voltage V i , output voltage V o , output current I o ; wherein V i 、V o 、I o at least two of which should be obtained; Step S22: Calculate the initial conditions of the resonant current and the resonant capacitor voltage: （1） wherein, i Lr denotes the resonant current; v Cr denotes the resonant capacitor voltage; 0~2 and 2~3 denotes the time from t0 to t2 and the time from t2 to t3, respectively, t0 being the initial time of a switching period, t1 being the time of the turn-off of the leading bridge arm, t2 being the time of the discontinuity of the secondary-side current in the first half of the switching period, t3 being the middle time of the switching period, k being a constant; 23 denotes the rate of change of the resonant inductance current in the time interval from t2 to t3. Solving the system of equations (1) to obtain the resonant current initial value i Lr and the resonant capacitor voltage initial value v Cr (t0); Step S23: According to the circuit operation, list i Lr And v Cr The differential equation satisfied in the period t0 to t3, combined with the initial value condition in formula (1), is solved to obtain i Lr And v Cr The time domain expression of, and the corresponding differential equation is: （2） Step S24: According to i Lr and v Cr the boundary conditions satisfied by i Lr and v Cr are substituted into the time-domain expressions, the unknowns are eliminated, and the final expression is obtained, with the corresponding boundary conditions being: （3） Step S25: using numerical solution to obtain the turn-off current of two bridge arms under different working conditions, including: the turn-off current of the leading bridge arm i Lr the turn-off current of the lagging bridge arm i Lr (t3).

3. The hybrid control LLC series resonant converter dead-time control method of claim 1, wherein: The simple fitting function model used in step S4 is: ； wherein, f s denotes the switching frequency; θ denotes the phase shift angle, being greater than or equal to 0, C 0 -C 5 is a parameter to be fitted; the phase shift angle θ ; the dead-time is fitted for different load resistances and is simplified to a simple function of the switching frequency and the phase shift angle.

4. The hybrid control LLC series resonant converter dead-time control method of claim 1, wherein: In step S5, the dead-time of the drive signal phase lag bridge arm and the phase lead bridge arm is calculated separately, and the shortest dead-time required for the drive signal phase lag bridge arm switch to realize zero-voltage turn-on is greater than or equal to the shortest dead-time required for the phase lead bridge arm switch to realize zero-voltage turn-on.

5. The mixed control LLC series resonant converter dead-time control method of claim 1, wherein: Adopt voltage current sampling circuit to connect LLC series resonant converter's load to collect output voltage V o And output current I o Information; The driving circuit is used to connect the four switches of the LLC series resonant converter to provide a driving signal. The embedded controller is connected to the voltage and current sampling circuit and the driving circuit.

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