Low-carrier-ratio current prediction control system and method for high-power permanent magnet synchronous motor

Abstract

The present invention discloses a high-power permanent magnet synchronous motor low-carrier ratio current PWM prediction control system and control method. In the process of current prediction model modeling, it not only compensates for the inherent delay of a control cycle in the digital control system, but also takes into account the zero-order holder characteristics of the inverter and the position change of the rotor within a control cycle, compensates for the model error caused by the high-power low-carrier ratio working condition, and improves the model accuracy of the current prediction model. And the decoupling current error integral compensation method can eliminate the parameter mismatch error to a certain extent, greatly improving the dynamic performance of the current loop. Both the simulation results and the experimental verification results show that the speed response and current response of this system under the high-power low-carrier ratio working condition can quickly follow the command and be stable, the three-phase stator current distortion is small, the harmonic content is low, and the system has a good control effect.

Description

Technical Field

[0001] The invention relates to a low-carrier ratio current PWM prediction control system and method for a high-power permanent magnet synchronous motor, belonging to the field of permanent magnet synchronous motor control technology. Background Art

[0002] For high-power permanent magnet synchronous traction permanent magnet synchronous motor speed control system, in order to reduce switching losses and avoid damage to switching devices, rail transit traction permanent magnet synchronous motors need to operate at an extremely low carrier ratio, which is the ratio of the switching frequency to the operating frequency of the permanent magnet synchronous motor.

[0003] In the control of permanent magnet synchronous motors, the basic theory of vector control is usually used to design the control strategy. For low power and high carrier ratio conditions, the dual closed-loop PI control strategy can generally meet the requirements of high-performance speed regulation. Under high power and low carrier conditions, the switching frequency will be limited to a lower level, which makes the digital control delay and inverter delay larger. When the carrier is low, it will cause serious cross-coupling in the system, resulting in serious stator current distortion, large torque pulsation, and deterioration of system control performance. The current PWM predictive control method (PPC for short) inputs the reference stator current and the current sampled current at this moment into the current prediction model, so as to directly calculate the command voltage that makes the permanent magnet synchronous motor reach the reference current at the next moment. The control process is simple and the dynamic response is fast. Applying the current PPC method in the current inner loop design of the permanent magnet synchronous speed regulation system will greatly improve the control performance of the system under low carrier ratio.

[0004] However, the traditional current PPC method does not meet the high-performance speed regulation of high-power permanent magnet synchronous motors at low carrier ratios. Since the forward Euler method used by the traditional current PPC directly discretizes the mathematical model of the continuous permanent magnet synchronous motor, there is a large discrete error at low switching frequency, and the influence of digital control delay and inverter delay is not considered. The system control effect is relatively poor, and there will be a large current static error in the current loop, and even serious oscillation will occur, causing system instability.

[0005] Therefore, it is necessary to design a high-power permanent magnet synchronous motor low-carrier ratio current PWM predictive control system to meet the high-performance speed regulation under high-power and low-carrier conditions. Summary of the invention

[0006] In order to solve the many problems existing in the low-carrier control of high-power permanent magnet synchronous motors, the purpose of the present invention is to design a low-carrier current PWM predictive control system and control method for high-power permanent magnet synchronous motors, and eliminate the influence of several main factors on the system current loop control effect under such working conditions, such as discretization error, delay in the system, change in rotor position, etc. These factors have a particularly significant impact on the system control performance under high-power low-carrier ratio control, and the system designed by the present invention fully considers several main influencing factors during the modeling process, and can achieve good control effects of the current PWM predictive control strategy under high-power low-carrier ratio working conditions.

[0007] In order to achieve the above object, the present invention adopts the following technical solution:

[0008] A high-power permanent magnet synchronous motor low-carrier current PWM prediction control system, comprising a speed outer loop PI controller, a current prediction model, an inverse Park transformation module, an SVPWM modulation module, an inverter, a permanent magnet synchronous motor, an encoder, and an abc-dq coordinate transformation module;

[0009] The input of the speed outer loop PI controller is the given speed n* and real-time speed n of the permanent magnet synchronous motor at time k, and the output is the given current i of the q axis. q * (k);

[0010] The input of the current prediction model is the given stator current dq-axis component i at time k d * (k) and i q * (k), and the sampled current i at time k d (k) and i q (k), the output is such that the sampling current i at time k+2 dq (k+2) accurately follows the current given i dq * (k) Command voltage u dq * (k+1);

[0011] The input of the inverse Park transformation module is the dq axis voltage command and the permanent magnet synchronous motor rotor position angle θ, and the output is the α-β axis command voltage u αβ * (k).

[0012] The input of the SVPWM modulation module is the α-β axis command voltage, and the output is the inverter switch control signal;

[0013] The inverter is used to generate a stator voltage for controlling the permanent magnet synchronous motor according to the switch control signal.

[0014] The encoder is used to obtain the rotor position angle θ of the permanent magnet synchronous motor and the real-time speed n of the permanent magnet synchronous motor;

[0015] The input quantity of the abc-dq coordinate transformation module is the real-time current i of the AB phase of the permanent magnet synchronous motor stator collected at time k. a (k), i b (k) and the permanent magnet synchronous motor rotor position angle θ, the output is the stator current dq axis component i d (k), i q (k).

[0016] Preferably, in a high-power permanent magnet synchronous motor low-carrier ratio current PWM predictive control system, the conversion equation of the abc-dq coordinate transformation module is: In the formula, θ e is the angle of the stator current vector to the α axis, θ is the rotor position angle, n p is the number of pole pairs of permanent magnet synchronous motor.

[0017] Preferably, the current prediction model includes a current PPC algorithm operation module and a decoupling current error integral compensation module;

[0018] The current PPC algorithm operation module includes two modules: current prediction algorithm I and current prediction algorithm II; the input quantity of current prediction algorithm I is the current instruction i dq * (k) and the predicted current value i at time k+1 dq (k+1), where i dq (k+1) is calculated by the current prediction algorithm II at time k, and the output is the command voltage u at time k+1 dqpre (k+1); the input quantity in the current prediction algorithm II is the sampled current i at time k dq (k) and command voltage u dq * (k), the output is the current prediction value i dq (k+1); Finally, this method can be used to make the current value i at time k+2 dq (k+2) accurately follows the current instruction i dq * (k), eliminating the effect of one-beat delay in the digital control system.

[0019] The decoupling current error integral compensation module is to deduct the real-time stator current i collected at the time k from the given stator current dq axis component at the time k. d (k), i q (k), and obtain the dq axis stator current error Δi d (k), Δiq (k), according to the command voltage equation, the decoupled command voltage error Δu can be obtained d (k+1),Δu q (k+1), the command voltage error obtained after decoupling is passed through the integral regulator to output the dq axis compensation voltage u dcom (k+1),u qcom (k+1).

[0020] Preferably, the conversion equation of the inverse Park transform module is as follows: In the formula, θ e is the angle of the stator current vector to the α axis, θ is the rotor position angle, n p is the number of pole pairs of permanent magnet synchronous motor.

[0021] The present invention also provides a control method for a low-carrier ratio current PWM predictive control system of a high-power permanent magnet synchronous motor, comprising the following steps:

[0022] Step 1) At time k, the encoder obtains the real-time speed n and rotor position angle θ of the permanent magnet synchronous motor;

[0023] Step 2) At time k, input the given speed n * and the real-time speed n of the permanent magnet synchronous motor to the speed outer loop PI controller, and the reference stator current q-axis component i is calculated and output. q * (k);

[0024] Step 3) At time k, the rotor position angle θ and the collected AB phase stator current i a (k), i b (k) is input to the abc-dq coordinate transformation module, and the dq axis component i of the sampled current at time k is output d (k), i q (k);

[0025] Step 4) At time k, the d-axis component of the reference stator current is converted into d * =0 control strategy is set to 0 input current prediction model, at the same time, the reference stator current q-axis component i output by the speed outer loop is q * (k) and the real-time stator current i d (k), i q (k) is input into the current prediction model, and after the calculation of the current PPC algorithm, current prediction algorithm I and current prediction algorithm II, the command voltage u at time k+1 is output. dpre (k+1),u qpre (k+1);

[0026] Step 5) At time k, the reference stator current dq axis components i d * (k), i q * (k) minus the sampling current i d (k), i q (k), and the corresponding dq-axis stator current error Δi is obtained d (k), Δi q (k) Use the decoupling current error integral compensation method to obtain the decoupled command voltage error Δu d (k+1),Δu q (k+1);

[0027] Step 6) At time k, after passing through the integral regulator, the dq axis compensation voltage u is output dcom (k+1),u qcom (k+1).

[0028] Step 7) At time k, the output u of the current PPC algorithm operation module is dpre (k+1),u qpre (k+1) and the output of the decoupling current error integral compensation module Δu d (k+1),Δu q (k+1) is added to obtain the dq axis command voltage u after current error integral compensation d * (k+1),u q * (k+1).

[0029] Step 8) Input the dq axis command voltage and the rotor position angle θ in step 7) into the inverse Park transformation module, and output the α-β axis command voltage u α * (k),u β * (k);

[0030] Step 9) inputting the α-β axis command voltage in step 8) into the SVPWM modulation module to obtain a switch control signal for controlling the inverter;

[0031] In step 10), the inverter receives the switch control signal in step 9) to obtain a three-phase symmetrical permanent magnet synchronous motor stator voltage.

[0032] Preferably, the current prediction algorithm I formula in step 4) is:

[0033] Current Prediction Algorithm The current prediction algorithm II formula is:

[0034] in,

[0035] In the formula, R s is the stator resistance of the permanent magnet synchronous motor, L s is the stator inductance of the permanent magnet synchronous motor, τ s is the electromagnetic time constant of the permanent magnet synchronous motor, and τ s =L s / R s , T s is the switching period, ψ f is the permanent magnet rotor flux. dpre (k+1),u qpre (k+1) is the dq axis command voltage at time k+1, i d * (k), i q * (k) is the reference stator current dq axis component, i d (k), i q (k) are the dq axis components of the sampled current at time k, i d (k+1), i q (k+1) are the dq axis components of the current prediction value at time k+1 respectively.

[0036] Preferably, the formula of the decoupling current error integral compensation module in step 5) is:

[0037] in, In the formula, R0 and L0 are the stator resistance and stator inductance used in the current prediction model, respectively, and are not the actual operating parameters of the permanent magnet synchronous motor. s0 =L0 / R0,T s is the switching period, Δu d (k+1),Δu q (k+1) is the dq axis command voltage error after decoupling, Δi d (k), Δi q (k) is the dq axis current error, u dpre (k+1),u qpre (k+1) is the dq-axis command voltage at time k+1.

[0038] The beneficial effects achieved by the present invention are as follows: in the process of modeling the current prediction model, not only the inherent delay of a control cycle in the digital control system is compensated, but also the zero-order holder characteristics of the inverter and the position change of the rotor within a control cycle are taken into account, the model error caused by the high-power and low-carrier ratio working conditions is compensated, and the model accuracy of the current prediction model is improved. And the decoupling current error integral compensation method can eliminate the parameter mismatch error to a certain extent, greatly improving the dynamic performance of the current loop. Both the simulation results and the experimental verification results show that the speed response and current response of this system under the high-power and low-carrier ratio working conditions can quickly follow the instructions and be stable, the three-phase stator current distortion is small, the harmonic content is low, and the system has a good control effect. BRIEF DESCRIPTION OF THE DRAWINGS

[0039] Figure 1 It is a principle block diagram of a low-carrier ratio current PWM predictive control system for a high-power permanent magnet synchronous motor in the present invention;

[0040] Figure 2 It is the control block diagram of the decoupling current error integral error compensation method;

[0041] Figure 3 This is the principle block diagram of the scalar model of permanent magnet synchronous motor;

[0042] Figure 4 It is a flowchart for changing the discrete signal of dq axis command voltage into a continuous signal;

[0043] Figure 5 It is the speed simulation waveform of the current PWM prediction control system of the permanent magnet synchronous motor under different carrier ratios;

[0044] Figure 6 The simulation waveform diagram of the stator current dq axis component of the current PWM prediction control system of the permanent magnet synchronous motor under different carrier ratios;

[0045] Figure 7 This is the stator current FFT harmonic analysis diagram of the current PWM predictive control output system of the permanent magnet synchronous motor under low carrier;

[0046] Figure 8 This is the experimental waveform of the speed mutation of the current PWM prediction control system of the permanent magnet synchronous motor under different carrier ratios;

[0047] Fig. 9 The waveform diagram of the load mutation experiment of the current PWM prediction control system of the permanent magnet synchronous motor under different carrier ratios;

[0048] Fig.10 This is the parameter sensitivity analysis experiment diagram of the current PWM predictive control system of the permanent magnet synchronous motor when the carrier ratio is 5. DETAILED DESCRIPTION

[0049] The present invention is further explained below in conjunction with the accompanying drawings. The following embodiments are only used to more clearly illustrate the technical solution of the present invention, and cannot be used to limit the protection scope of the present invention.

[0050] like Figure 1 As shown, a high-power permanent magnet synchronous motor low carrier ratio current PWM prediction control system includes a speed outer loop PI controller, a current prediction model, an inverse Park transformation module, an SVPWM modulation module, an inverter, a permanent magnet synchronous motor, an encoder, and an abc-dq coordinate transformation module. Specifically:

[0051] The encoder is used to obtain the rotor position angle θ of the permanent magnet synchronous motor and the real-time speed n of the permanent magnet synchronous motor;

[0052] The input of the speed outer loop PI controller is the given speed n* and real-time speed n of the permanent magnet synchronous motor at time k, and the output is the given current i of the q axis. q * (k);

[0053] The input of the abc-dq coordinate transformation module is the real-time current i of the AB phase of the permanent magnet synchronous motor stator collected at time k. a (k), i b (k) and the permanent magnet synchronous motor rotor position angle θ, the output is the stator current dq axis component i d (k), i q (k);

[0054] The input of the current prediction model is the given stator current dq-axis component i at time k d * (k) and i q * (k), and the sampled current i at time k d (k) and i q (k), the output is such that the sampling current i at time k+2 dq (k+2) accurately follows the current given i dq * (k) Command voltage u dq * (k+1);

[0055] The input of the inverse Park transformation module is the dq axis voltage command and the permanent magnet synchronous motor rotor position angle θ, and the output is the α-β axis command voltage u αβ * (k).

[0056] The input of the SVPWM modulation module is the α-β axis command voltage, and the output is the inverter switch control signal;

[0057] The inverter is used to generate a stator voltage for controlling the permanent magnet synchronous motor according to the switch control signal.

[0058] will i d (k), i q (k), i d * (k), i q * (k) The four variables are substituted into the current prediction model, and the dq axis command voltage u at time k+1 is obtained through the current PPC prediction algorithm. dpre (k+1),u qpre (k+1).

[0059] In order to compensate for the system model error and improve the system control performance. Figure 2 As shown, in the present invention, a decoupling current error integral compensation module is added to the system, and at time k, the given stator current dq axis component i d * (k), i q * (k) minus the sampling current i at that moment d (k), i q (k), and obtain the dq axis stator current error Δi d (k), Δi q (k) Use the decoupling current error integral compensation method to obtain the decoupled command voltage error Δu d (k+1),Δu q (k+1), and then through the integral regulator, the voltage command compensation u is obtained dcom (k+1),u qcom (k+1), and finally u dpre (k+1),u qpre (k+1) and u dcom (k+1),u qcom (k+1) is added to output the dq axis command voltage u d * (k+1),u q * (k+1).

[0060] Combine the following Figure 3 The scalar model principle diagram of the permanent magnet synchronous motor is used to introduce the derivation process of the current prediction model command voltage equation used in the present invention:

[0061] Depend on Figure 3 The voltage equation of permanent magnet synchronous motor can be obtained:

[0062]

[0063] Write equation (1) in the form of current state equation:

[0064]

[0065] For ease of analysis, equation (2) can be written in matrix vector form:

[0066]

[0067] Where:

[0068] Formula (3) is actually a differential equation in matrix-vector form. Solving it yields t∈[kT s ,(k+1)T s The analytical solution of ] is:

[0069]

[0070] Where: the solution of the differential equation can be divided into two parts, the free component i1 and the forced component i2; i1 is the solution of the equation when the control quantity is zero, that is, the current value in the dq coordinate system corresponding to the action of the zero control quantity; i2 represents the current value in the dq coordinate system corresponding to the action of the non-zero control quantity, and I is the unit matrix; u(τ) is the dq-axis stator voltage vector acting on the permanent magnet synchronous motor side;

[0071] When kT s ≤t<(k+1)T s If the position change of the rotor within a control cycle is ignored and the zero-order holding characteristic of the inverter is taken into account, the control quantity u can be expressed as u(t)=u(kT s ); If the rotor position change is considered, the signal flow diagram of the dq axis command voltage discrete signal changing to a continuous signal is as follows Figure 4 As shown;

[0072] Can Figure 4 It is expressed as formula (5):

[0073]

[0074] In the formula: u(t)=[u d (t)u q (t)] T ,u(kT s )=[u d * (k)u q * (k)] T u d (t),u q (t) is the continuous quantity of dq-axis stator voltage actually applied to the permanent magnet synchronous motor side within one control cycle;

[0075] According to the current PWM predictive control principle, it is necessary to predict the value at the next moment through the state quantity at the current moment. Let the current moment be kT s moment, the next moment is (k+1)T s At this moment, equation (4) can be rewritten as follows:

[0076]

[0077] Where:

[0078] The scalar equation of the prediction model in the dq coordinate system can be derived from formula (6). Now a simple derivation process is given;

[0079] First, we analyze equation (1) of equation (6):

[0080]

[0081] Where:

[0082] Further analysis of equation (2) of equation (6) yields:

[0083]

[0084] Where:

[0085] By adding the free component of equation (7) and the forced component of equation (8), we can get the scalar equation of the current prediction model, which can be called the time-domain discrete current PPC model considering the rotor position, and can be expressed as equation (9):

[0086]

[0087] By adding control delay compensation to equation (9), we can get the time domain discrete current PPC system model with delay compensation and considering the rotor position. This is the low carrier ratio current PWM predictive control system model of the high-power permanent magnet synchronous motor designed in this paper, as shown in equations (10) and (11):

[0088]

[0089]

[0090] Where: R s is the stator resistance of the permanent magnet synchronous motor, L s is the stator inductance of the permanent magnet synchronous motor, τ s is the electromagnetic time constant of the permanent magnet synchronous motor, and τ s =Ls / R s , T s is the switching period, ψ f is the permanent magnet rotor flux. dpre (k+1),u qpre (k+1) is the dq axis command voltage at time k+1, i d * (k), i q * (k) is the reference stator current dq axis component, i d (k), i q (k) are the dq axis components of the sampled current at time k, i d (k+1), i q (k+1) are the dq axis components of the current prediction value at time k+1 respectively.

[0091] In order to eliminate the current static error caused by the system model error, a decoupled current error integral compensation method is proposed.

[0092] The formula of the decoupling current error integral compensation module is: in, In the formula, R0 and L0 are the stator resistance and stator inductance used in the current prediction model, respectively, and are not the actual operating parameters of the permanent magnet synchronous motor. s0 =L0 / R0,T s is the switching period, Δu d (k+1),Δu q (k+1) is the dq axis command voltage error after decoupling, Δi d (k), Δi q (k) is the dq axis current error, u dpre (k+1),u qpre (k+1) is the dq axis command voltage at time k+1, u d * (k+1),u q * (k+1) is the dq-axis command voltage at time k+1 after current integral compensation.

[0093] The above is the derivation process of the command voltage equation of the time domain discrete current PPC model with delay compensation and considering the rotor position.

[0094] In order to verify that the low carrier ratio current PWM predictive control method in the present invention can achieve the purpose of the present invention, a permanent magnet synchronous motor MATLAB / simulink simulation is built, the inverter switching frequency is set to 500Hz, it is started with rated load, and changes to a light load state after reaching the rated speed (carrier ratio is 5). The simulation parameters are shown in Table 1:

[0095] Table 1

[0096] name Numeric Stator resistance 0.26Ω Stator inductance 7.5mH Rated Power 600kW Rated torque 2865N·m Rated speed 2000r / min Moment of Inertia <![CDATA[0.85kg·m 2 ]]> Pole pairs 3

[0097] Combine the following Figure 5 , Figure 6 , Figure 7 , Figure 8 To introduce the simulation results of the above simulation.

[0098] Figure 5 The figure is a simulation waveform of the speed of the current PWM prediction control system of the permanent magnet synchronous motor under different carrier ratios. The simulation results show that the speed of the permanent magnet synchronous motor can quickly follow the instruction and stabilize under high or low carrier ratios by using the control method of the present invention, the speed ripple of the permanent magnet synchronous motor is small, and it still has a fast dynamic response speed after the system suddenly reduces the load, and the overshoot is also small.

[0099] Figure 6 The simulation waveform of the stator current dq axis component of the current PWM prediction control system of the permanent magnet synchronous motor under different carrier ratios is shown in the figure. Figure 6 (a) is the q-axis component of the stator current, Figure 6 (b) is the d-axis component of the stator current. It can be observed that the q-axis current has basically no static error under low carrier ratio, the adjustment time is short, the overshoot is small, it has good dynamic performance, and it still has high stability when the system suddenly reduces the load.

[0100] Figure 7 This is the stator current FFT harmonic analysis diagram of the current PWM predictive control output system of the permanent magnet synchronous motor under low carrier. It can be seen from the figure that the stator current harmonic content is relatively small, indicating that the degree of stator current distortion is relatively light, and it will not generate too many harmonics to cause large harmonic torque.

[0101] From the simulation results, it can be seen that the high-power permanent magnet synchronous motor low-carrier ratio current PWM predictive control system designed in the present invention is suitable for high-power low-carrier ratio control conditions, and can achieve high-performance speed regulation of the permanent magnet synchronous motor under such conditions.

[0102] In order to further verify the superiority of the system designed by the present invention under high-power and low-carrier ratio control, a permanent magnet synchronous motor experimental platform is built based on the NI cRIO-9033 controller, and a permanent magnet synchronous motor speed regulation experiment under low-carrier ratio is completed using a small-power surface-mounted permanent magnet synchronous motor. The inverter switching frequency is set to 500Hz, and the permanent magnet synchronous motor parameters are shown in Table 2:

[0103] Table 2:

[0104] parameter Numeric Stator rated voltage 380V Stator rated current 5.8A Rated frequency 100Hz Rated Power 3kW Rated load 19.1N Pole pairs 4 <![CDATA[Stator inductance L s > 8mH <![CDATA[Stator resistance R s > 1.3Ω

[0105] The experiment contents include speed mutation experiment, load mutation experiment, parameter sensitivity analysis experiment, such as Figure 8 , Fig. 9 , Fig.10 As shown in Figure 3, the experimental results are consistent with the simulation results, which verifies the feasibility of the algorithm.

[0106] Figure 8 This is a speed mutation experiment diagram of the permanent magnet synchronous motor speed control system. From the figure, it can be seen that the stator current waveform is basically undistorted, the stator current harmonic content is small, the speed response is very fast, the overshoot is small, and the current error of the stator current dq axis component can quickly stabilize near 0.

[0107] Fig. 9 This is a diagram of the load mutation experiment of the permanent magnet synchronous motor speed control system. It can be seen from the figure that the dq component of the stator current can still maintain a high stability after the system is suddenly loaded and unloaded, and the overshoot of the dq axis current waveform is small and the adjustment time is short.

[0108] Fig.10 This is the experimental diagram of parameter sensitivity analysis of permanent magnet synchronous motor speed control system. Fig.10 (a) is the resistance mismatch test waveform. Fig.10 (b) is the waveform of the inductance mismatch experiment. Fig.10 (c) is the rotor flux mismatch test waveform. It can be seen from the figure that the resistance mismatch has little effect on the system control performance, while the inductance mismatch and the rotor flux mismatch have a relatively large impact on the system control performance. However, in general, the high-power permanent magnet synchronous motor low-carrier ratio current PWM predictive control system designed by the present invention has good robustness when the motor parameters are mismatched.

[0109] The experimental results are consistent with the simulation results, which verify that the high-power permanent magnet synchronous motor low-carrier ratio current PWM predictive control system designed in the present invention is suitable for high-power low-carrier ratio control conditions, and can achieve high-performance speed regulation of the permanent magnet synchronous motor under such conditions.

[0110] The above is only a preferred embodiment of the present invention. It should be pointed out that for ordinary technicians in this technical field, several improvements and modifications can be made without departing from the principle of the present invention. These improvements and modifications should also be regarded as the scope of protection of the present invention.

Claims

1. A low-carrier ratio current PWM predictive control system for a high-power permanent magnet synchronous motor, characterized in that: Including speed outer loop PI controller, current prediction model, inverse Park transformation module, SVPWM modulation module, inverter, permanent magnet synchronous motor, encoder, abc-dq coordinate transformation module; The input of the speed outer loop PI controller is the given speed n* and real-time speed n of the permanent magnet synchronous motor at time k, and the output is the given current i of the q axis. q * (k); The input of the current prediction model is the given stator current dq-axis component i at time k d * (k) and i q * (k), and the sampled current i at time k d (k) and i q (k), the output is the command voltage u d * (k+1) and u q * (k+1),u d * (k+1) makes the sampling current i at time k+2 d (k+2) accurately follows the current given i d * (k),u q * (k+1) makes the sampling current i at time k+2 q (k+2) accurately follows the current given i q * (k); The input of the inverse Park transformation module is the dq axis voltage command and the permanent magnet synchronous motor rotor position angle θ, and the output is the α-β axis command voltage u αβ * (k); The SVPWM modulation module input is the α-β axis command voltage, and the output is the inverter switch control signal; The inverter is used to generate a stator voltage for controlling the permanent magnet synchronous motor according to the switch control signal; The encoder is used to obtain the rotor position angle θ of the permanent magnet synchronous motor and the real-time speed n of the permanent magnet synchronous motor; The input quantity of the abc-dq coordinate transformation unit is the real-time current i of the AB phase of the permanent magnet synchronous motor stator collected at time k. a (k), i b (k) and the permanent magnet synchronous motor rotor position angle θ, the output is the stator current dq axis component i d (k), i q (k); The current prediction model includes a current PPC algorithm operation module and a decoupling current error integral compensation module; The current PPC algorithm operation module includes two modules: current prediction algorithm I and current prediction algorithm II; the input quantity of current prediction algorithm I is the current instruction i d * (k), i q * (k) and the predicted current value i at time k+1 d (k+1), i q (k+1), where i d (k+1) and i q (k+1) is calculated by the current prediction algorithm II at time k, and the output is the command voltage u at time k+1 dpre (k+1) and u qpre (k+1); the input quantity in the current prediction algorithm II is the sampled current i at time k d (k), i q (k) and command voltage u d * (k) and u q * (k), the output is the current prediction value i d (k+1), i q (k+1); Finally, this method can be used to make the current value i at time k+2 d (k+2), i q (k+2) respectively accurately follows the current instruction i d * (k), i q * (k), eliminating the effect of one-beat delay in digital control systems; The decoupling current error integral compensation module is to deduct the real-time stator current i collected at the time k from the given stator current dq axis component at the time k. d (k), i q (k), and obtain the dq axis stator current error Δi d (k), Δi q (k), according to the command voltage equation, the decoupled command voltage error Δu can be obtained d (k+1),Δu q (k+1), the command voltage error obtained after decoupling is passed through the integral regulator to output the dq axis compensation voltage u dcom (k+1),u qcom (k+1).

2. A high-power permanent magnet synchronous motor low-carrier ratio current PWM predictive control system according to claim 1, characterized in that: The conversion equation of the abc-dq coordinate transformation module is: In the formula, θ e is the angle of the stator current vector to the α axis, θ is the rotor position angle, n p is the number of pole pairs of permanent magnet synchronous motor.

3. A high-power permanent magnet synchronous motor low-carrier ratio current PWM predictive control system according to claim 2, characterized in that: The conversion equation of the inverse Park transform module is as follows: In the formula, θ e is the angle of the stator current vector to the α axis, θ is the rotor position angle, n p is the number of pole pairs of permanent magnet synchronous motor.

4. A high-power permanent magnet synchronous motor low-carrier ratio current PWM predictive control system according to claim 1, characterized in that: The current prediction algorithm I is: The current prediction algorithm II is: in, In the formula, R s is the stator resistance of the permanent magnet synchronous motor, L s is the stator inductance of the permanent magnet synchronous motor, τ s is the electromagnetic time constant of the permanent magnet synchronous motor, and τ s =L s / R s , T s is the switching period, ψ f is the permanent magnet rotor flux; u dpre (k+1),u qpre (k+1) is the dq axis command voltage at time k+1, u d * (k) and u q * (k) is the reference stator voltage dq axis component, i d * (k), i q * (k) is the reference stator current dq axis component, i d (k), i q (k) are the dq axis components of the sampled current at time k, i d (k+1), i q (k+1) are the dq axis components of the current prediction value at time k+1, ω e (k) is the angular velocity of the permanent magnet synchronous motor at the current moment.

5. A high-power permanent magnet synchronous motor low-carrier ratio current PWM predictive control system according to claim 1, characterized in that: The decoupling current error integral compensation module is: in, In the formula, R 0、 L0 is the stator resistance and stator inductance used in the current prediction model, which are not the actual operating parameters of the permanent magnet synchronous motor. s0 =L0 / R0,T s is the switching cycle, U dpre (k+1) and U qpre (k+1) are the dq components of the command voltage output by the current prediction model respectively.

6. A control method for a low-carrier ratio current PWM predictive control system of a high-power permanent magnet synchronous motor based on any one of claims 1-5, characterized in that: The steps include: Step 1: At time k, the encoder obtains the real-time speed n and rotor position angle θ of the permanent magnet synchronous motor; Step 2: At time k, input the given speed n * and the real-time speed n of the permanent magnet synchronous motor to the speed outer loop PI controller, and the reference stator current q-axis component i is calculated and output. q * (k); Step 3: At time k, the rotor position angle θ and the collected AB phase stator current i a (k), i b (k) is input to the abc-dq coordinate transformation module, and the dq axis component i of the sampled current at time k is output d (k), i q (k); Step 4: At time k, set the stator current to i dq * (k) and the sampling current dq axis component i at time k d (k), i q (k) is input into the current prediction model, and the dq axis command voltage u after current error integral compensation at time k+1 is obtained. d * (k+1),u q * (k+1); Step 5: Input the dq axis command voltage and the rotor position angle θ in step 4) into the inverse Park transformation module, and output the α-β axis command voltage u α * (k) and u β * (k); Step 6: input the α-β axis command voltage in step 5) into the SVPWM modulation module to obtain a switch control signal for controlling the inverter; Step 7: The inverter receives the switch control signal in step 6) to obtain a three-phase symmetrical permanent magnet synchronous motor stator voltage.

7. The control method of a high-power permanent magnet synchronous motor low-carrier ratio current PWM predictive control system according to claim 6, characterized in that: Step 4 includes: Step 4.1: Set the d-axis component of the reference stator current at the next moment to i d * =0 control strategy is set to 0 input current prediction model, at the same time, the reference stator current q-axis component i output by the speed outer loop is q * (k) and the real-time stator current i d (k), i q (k) is input into the current prediction model, and after the calculation of the current prediction algorithm I and the current prediction algorithm II, the command voltage u at time k+1 is output. dpre (k+1),u qpre (k+1); Step 4.2: At time k, the reference stator current dq axis components i d * (k), i q * (k) minus the sampling current i d (k), i q (k), and the corresponding dq-axis stator current error Δi is obtained d (k), Δi q (k) Use the decoupling current error integral compensation method to obtain the decoupled command voltage error Δu d (k+1),Δu q (k+1), after passing through the integral regulator, the output dq axis compensation voltage u dcom (k+1),u qcom (k+1); Step 4.3, u dpre (k+1),u qpre (k+1) and u dcom (k+1),u qcom (k+1) is added, and the dq axis command voltage u after current error integral compensation at time k+1 is output. d * (k+1),u q * (k+1).

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