A control method for a high-frequency isolated DC-DC converter with output current source
By employing a specific topology and control method in the output current source type high-frequency isolated DC-DC converter, the voltage spike problem during the commutation process is solved, and zero-current or zero-voltage turn-on of the switching transistors is achieved, thereby improving the reliability and safety of the converter.
Patent Information
- Application Number
- CN202211071913.3
- Authority / Receiving Office
- CN · China
- Patent Type
- Patents(China)
- Current Assignee / Owner
- Filing Date
- 2022-09-02
- Publication Date
- 2025-10-28
- Estimated Expiration
- 2042-09-02
AI Technical Summary
The output current source type high-frequency isolated DC-DC converter has voltage spike problems during the commutation process, which affects the safety and reliability of the converter.
The topology consists of a voltage source inverter bridge, a high-frequency isolation transformer, a clamping capacitor, and a current source rectifier bridge. By using a control method that combines phase-shift control and clamping capacitor, a transition process is added between the energy transmission and current freewheeling stages of the converter. The clamping capacitor clamps the transformer leakage inductance voltage to the negative value of the capacitor voltage, ensuring that the switching transistor turns on and off with zero current or zero voltage.
It effectively eliminates voltage spikes in the converter during the commutation process, achieves zero-current or zero-voltage turn-on of the switching transistors across the entire power range, reduces power loss, and improves the reliability and safety of the converter.
Smart Images
Figure CN115395790B_ABST
Abstract
Description
Technical Field
[0001] This invention relates to the field of power supply technology, specifically a control method for an output current source type high-frequency isolated DC-DC converter. Background Technology
[0002] High-frequency isolated full-bridge DC-DC converters can be divided into two categories: voltage-source full-bridge DC-DC converters and current-source full-bridge DC-DC converters. Voltage-source full-bridge converters are more widely used, but this topology suffers from problems such as high current stress and large DC-side current ripple, making it difficult to apply in applications with high current requirements. Current-source converters, based on the placement of the energy storage inductor, can be divided into input current-source and output current-source types. Output current-source converters have a large inductor connected in series on the output side, resulting in an approximately constant output current, small current ripple, and the peak current of the switching transistor equal to the output current, leading to low current stress. However, the voltage spike problem limits the application of output current-source converters.
[0003] A current-source full-bridge converter mainly consists of two operating phases within one duty cycle: an energy transfer phase and a current freewheeling phase. During the energy transfer phase, energy is transferred from the input side to the output side, with the energy storage inductor connected in series with the transformer. In steady state, the transformer leakage current equals the output current. During the current freewheeling phase, the output current freewheels through the secondary full-bridge switches. At this time, the primary and secondary full-bridge circuits operate independently, and the transformer leakage current has a steady-state value of zero.
[0004] When the output current source converter switches from the energy transfer stage to the current freewheeling stage, the transformer leakage inductance current cannot change abruptly. Before it decreases to the steady-state value of the leakage inductance current in the current freewheeling stage, the leakage inductance will charge the parasitic capacitance of the switching transistor, generating a voltage spike. When the converter switches from the current freewheeling stage to the energy transfer stage, the output inductor will charge the parasitic capacitance of the switching transistor before it increases to the steady-state value of the leakage inductance current in the energy transfer stage, generating a large voltage spike. This voltage spike is caused by the constant current source charging the parasitic capacitance and is an active excitation type voltage spike. Its amplitude is much larger than the resonant voltage spike of the voltage source converter. If it is not suppressed, it will damage the switching devices and reduce the safety and reliability of the converter. Summary of the Invention
[0005] The purpose of this invention is to provide an output current source type high-frequency isolated DC-DC converter and control method to solve the problems mentioned in the background art.
[0006] To achieve the above objectives, the present invention provides the following technical solution:
[0007] A current-source type high-frequency isolated DC-DC converter comprises a voltage-source inverter bridge, a high-frequency isolation transformer, a clamping capacitor, and a current-source rectifier bridge. The voltage-source inverter bridge consists of two half-bridges connected in parallel. Each half-bridge has an upper and lower arm composed of a switching transistor. The midpoint of the two half-bridges is the output terminal of the inverter bridge, which is connected to the primary side of the high-frequency isolation transformer. The current-source rectifier bridge consists of two half-bridges connected in parallel. Each half-bridge has an upper and lower arm composed of a switching transistor and a diode connected in series. A large inductor is connected in series on the output side of the current-source rectifier bridge, and its input side is the midpoint of the two half-bridges. The clamping capacitor is connected in series between the secondary side of the high-frequency isolation transformer and the input side of the current-source rectifier bridge.
[0008] As a further technical solution of the present invention: the switching transistors of the voltage source inverter bridge and the current source rectifier bridge are insulated gate bipolar transistors or metal-oxide-semiconductor field-effect transistors.
[0009] As a further technical solution of the present invention: the voltage source inverter bridge is composed of switching transistors S1-S4, the current source rectifier bridge is composed of switching transistors S5-S8, and the clamping capacitor is capacitor C. c .
[0010] A control method for a current-source type high-frequency isolated DC-DC converter, using the aforementioned converter, includes the following steps: The voltage-source inverter bridge employs phase-shift control; the duty cycle of the drive signals for switches S1-S4 is 50%; switches S1 and S3 are complementary to S2 and S4 respectively; the ratio of the phase difference between the drive signals of S3 and S1 to half a working cycle is the phase shift ratio D; the turn-on and turn-off times of the current-source rectifier bridge switches S5, S6, S7, and S8 are respectively referenced to S2... The turn-on and turn-off times of switches S4, S1, and S3 are the same, with the duty cycle of the drive signal being greater than 50%. Switch S5 turns off after S2 turns on, delayed by T1, and turns on ahead of T2 before S2 turns off. Switch S6 turns on ahead of T1 before S4 turns off, and turns off after S4 turns on, delayed by T2. Switch S7 turns off after S1 turns on, delayed by T1, and turns on ahead of T2 before S1 turns off. Switch S8 turns on ahead of T1 before S3 turns off, and turns off after S3 turns on, delayed by T2.
[0011] As a further technical solution of the present invention: the control method adds a transition process between the energy transmission stage and the current freewheeling stage of the current source converter. Considering the dead time of the voltage source full-bridge switch, this transition process can be regarded as the sum of the dead time and T2. Taking the positive half-cycle of the converter as an example, the period from S4 to S8 is the transition process of the converter switching from the energy transmission stage to the current freewheeling stage; the period from S7 to S2 is the transition process of the converter switching from the current freewheeling stage to the energy transmission stage.
[0012] As a further technical solution of the present invention: T1 is set after the start and before the end of the energy transmission stage of the converter, and its function is to ensure that the switching transistors S6 and S8 can achieve zero-current turn-on. The value range of T1 is...
[0013] Within the range of values, T1 should take a smaller value. Taking a larger value will increase the lower limit of the shift ratio D. In the formula, T s The switching cycle.
[0014] As a further technical solution of the present invention: T2 is set in the transition process of the converter, and its function is to reserve a certain time for the transformer leakage inductance current to transition from the steady-state value of the leakage inductance current in one stage to the steady-state value of the leakage inductance current in another stage. The value of T2 should meet the following constraints.
[0015] T2≥(2~3)△t
[0016] Its value can be obtained by the following formula.
[0017]
[0018] In the formula, Δt is the actual time taken for the transformer leakage inductance current to increase or decrease from its steady-state value in one stage to its steady-state value in another stage, T s For the switching cycle, C c L is the capacitance value of the clamping capacitor. k D is the shift ratio to account for the leakage inductance referred to the secondary side of the transformer.
[0019] As a further technical solution of the present invention: the clamping capacitor, in conjunction with the control method, clamps the transformer leakage inductance voltage to a negative value during the transition process, so that the transformer leakage inductance current reaches the steady-state value of the leakage inductance current in the next working stage in advance. Its rated voltage should be greater than the maximum voltage within one working cycle. The capacitance value of the clamping capacitor and its maximum voltage can be obtained by the following formula.
[0020]
[0021] In the formula, Δt is the actual time taken for the transformer leakage inductance current to increase or decrease from its steady-state value in one stage to its steady-state value in another stage, T s For the switching cycle, C c L is the capacitance value of the clamping capacitor. k To account for the leakage inductance referred to the secondary side of the transformer, D is the shift ratio, V c_max I is the maximum value of the clamping capacitor voltage. o This represents the average output current.
[0022] Compared with the prior art, the beneficial effects of the present invention are:
[0023] The control method for a current-source type high-frequency isolated DC-DC converter described in this invention adds a transition process between the energy transfer stage and the current freewheeling stage of the converter. During the transition process, the clamping capacitor cooperates with the control method to ensure that the transformer leakage inductance current reaches the steady-state value of the leakage inductance current in the next operating stage in advance. This eliminates the voltage spikes caused by the difference in the steady-state values of the leakage inductance current between the two operating stages during commutation of the current-source type converter. This invention effectively solves the voltage spike problem in the commutation process of the current-source type full-bridge DC-DC converter, eliminates the backflow power during converter operation, and achieves zero-current turn-on and zero-current turn-off of the current-source type rectifier bridge-side switches and zero-current turn-on or zero-voltage zero-current turn-off of the voltage-source type inverter bridge-side switches across the entire power range. This reduces converter power loss, decreases voltage and current stress on the switches, and improves the reliability and safety of the converter during operation. Attached Figure Description
[0024] Figure 1 Topology diagram of a high-frequency isolated DC-DC converter with output current source;
[0025] Figure 2 A schematic diagram of the working waveform considering the dead time of the switching transistors on the voltage source inverter bridge side;
[0026] Figure 3 Schematic diagram of the operating modes of the output current source type high-frequency isolated DC-DC converter (1-9);
[0027] Figure 4 The graph shows the relationship between T2 and Δt under the condition that the clamping capacitor value is fixed and the shift ratio is changed;
[0028] Figure 5 The graph shows the relationship between T2 and Δt under the condition that the displacement ratio is fixed and the clamping capacitor value is changed.
[0029] Explanation of symbols and labels in the attached drawings: L k —Transformer leakage inductance referred to the secondary side; C c —Clamping capacitor; L o —Output-side energy storage inductor; R o —Equivalent load on the output side; C in —Input-side filter capacitor; v ab —Voltage at the midpoint of the two half-bridges in a voltage source type full-bridge; i pr —Transformer primary current; v cd —Voltage at the midpoint of the two half-bridges in a current-source full-bridge configuration; v se —Transformer secondary voltage; i se This refers to the secondary current of the transformer; v c —Clamping capacitor voltage; i o —Output current; V in—Input voltage; D—Ratio of the phase difference between S1 and S3 drive signals to half a switching cycle; T s —Switching cycle; T d —Dead time of the switching transistor on the voltage source full-bridge side; T1—Reserved time to achieve zero-current turn-on or turn-off of the switching transistor; T2—Reserved time for the transformer leakage inductance current to transition from the steady-state value of one stage to the steady-state value of another stage; Δt—Actual time taken for the transformer leakage inductance current to increase or decrease from the steady-state value of one stage to the steady-state value of another stage; V c_max —Maximum voltage of clamping capacitor, I o —Average output current. Detailed Implementation
[0030] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.
[0031] Example 1, as Figure 1 As shown, a high-frequency isolated DC-DC converter with an output current source has the following topology. Figure 1 As shown, this topology consists of a voltage source inverter bridge, a high-frequency isolation transformer, and a series clamping capacitor C. c A current-source rectifier bridge consists of two half-bridges connected in parallel. Each half-bridge has an upper and lower arm composed of a switching transistor. The midpoint of the two half-bridges is the inverter bridge output, connected to the primary side of the high-frequency transformer. A current-source rectifier bridge also consists of two half-bridges connected in parallel. Each half-bridge has an upper and lower arm composed of a switching transistor and a diode connected in series. A large inductor is connected in series on the output side of the rectifier bridge, and its input side is the midpoint of the two half-bridges. Clamping capacitor C c It is connected in series between the secondary side of the high-frequency transformer and the input side of the rectifier bridge.
[0032] The output current source type high-frequency isolated DC-DC converter control method of the present invention is as follows: the voltage source type inverter bridge adopts phase shift control, the duty cycle of the drive signals of the switching transistors S1-S4 is 50%, the switching transistors S1 and S3 are complementary to S2 and S4 respectively, and the ratio of the phase difference of the drive signals of S3 and S1 to half of the working cycle is the phase shift ratio D. The turn-on and turn-off times of the current source rectifier bridge switches S5, S6, S7, and S8 are referenced from the turn-on and turn-off times of S2, S4, S1, and S3, respectively. The duty cycle of the drive signal for each switch is the same, all greater than 50%. Switch S5 turns off after S2 turns on with a delay of T1, and turns on ahead of T2 before S2 turns off. Switch S6 turns on ahead of T1 before S4 turns off, and turns off after S4 turns on with a delay of T2. Switch S7 turns off after S1 turns on with a delay of T1, and turns on ahead of T2 before S1 turns off. Switch S8 turns on ahead of T1 before S3 turns off, and turns off after S3 turns on with a delay of T2.
[0033] The schematic diagram of the drive signals and related operating waveforms of the switching transistors S1-S8 are as follows: Figure 2 As shown, in order to prevent shoot-through of the upper and lower switching transistors on the voltage source inverter bridge side during actual operation, Figure 2 The drive signal diagram takes dead time into account. Under this control method, the operating mode of the output current source type high-frequency isolated DC-DC converter during the positive half-cycle is as follows: Figure 3 As shown.
[0034] Example 2: Based on the example, the switching transistors of the voltage source inverter bridge and the current source rectifier bridge can be insulated gate bipolar transistors or metal-oxide-semiconductor field-effect transistors.
[0035] The working principle is as follows:
[0036] Figure 3 The working principle of each mode is as follows:
[0037] Mode 1 [t0-t1]: At time t0, S1 is turned on, while S4, S5, S7, and S8 remain on. The converter begins operating in the energy transfer phase, at which time v ab =V in Clamping capacitor voltage v c The voltage is negative, and the transformer secondary current i se The current is equal to the output current, and the current in the branch where S7 is located is zero. Within t0-t1, the transformer secondary current i... se To clamp capacitor C c Charging, v cd With the clamping capacitor voltage v c It decreases as v decreases. cd The voltage remains positive, causing D7 to be cut off due to reverse voltage, and the secondary current only flows through C. cS5, D5, S8, and D8 supply power to the load. The circuit operates as follows: Figure 3 As shown in (a).
[0038] Mode 2 [t1-t2]: At time t1, S7 is turned off. Since the current in the branch containing S7 is zero, S7 achieves zero-current turn-off. During t1-t2, the transformer secondary current i se To clamp capacitor C c Charging causes the voltage across its terminals to become a positive voltage, v cd With v c The secondary current decreases as the C increases. c S5, D5, S8, and D8 supply power to the load. The circuit operates as follows: Figure 3 As shown in (b).
[0039] Mode 3 [t2-t3]: At time t2, S6 is turned on, but due to v cd With a positive voltage, D6 is cut off due to reverse voltage, so the current in the branch containing D6 is zero, and S6 achieves zero-current turn-on. During this stage, the converter still operates in the energy transfer stage, and its operation is the same as in mode 2. The circuit operating state is as follows: Figure 3 As shown in (c).
[0040] Mode 4 [t3-t4]: At time t3, S4 is turned off. Since the leakage inductance current cannot change abruptly, v cd =-v c D6 is turned on under a positive voltage. At this time, the potentials at points d and c are equal. Let v cd =0; The transformer primary current freewheels through the anti-parallel diodes S1 and S3, causing v ab =0, the transformer leakage inductance voltage is clamped to -V c The current flowing through the transformer begins to decrease. During the period t3-t4, the secondary current i... se During the reduction process, the clamping capacitor is charged, the capacitor voltage increases slowly, and the output current I... o By C c S5, D5, L o R o S8, D8 circuit to S5, D5, L o R o During the commutation process in circuits S6 and D6, the current in the branch containing S6 increases, while the current in the branch containing S8 decreases. The circuit operation is as follows: Figure 3 As shown in (d).
[0041] Mode 5 [t4-t5]: At time t4, the current in the branch containing S8 and the current flowing through the transformer decrease to zero, and the output current I... o Through S5, D5, L o R oThe S6 and D6 circuits continue to flow, and the commutation process ends. During t4-t5, S1, S5, S6, and S8 remain in the on-state, and the clamping capacitor voltage v... c The circuit remains unchanged. The circuit's operating state is as follows: Figure 3 As shown in (e).
[0042] Mode 6 [t5-t6]: At time t5, S3 is turned on. Since the current in its branch is zero, S3 is turned on with zero current. During t5-t6, S1, S5, S6, and S8 remain on, and the output current flows through S5, D5, and L. o R o S6 and D6 freewheeling. The circuit's operating state is as follows: Figure 3 As shown in (f).
[0043] Mode 7 [t6-t7]: At time t6, S8 is turned off because the current in its branch is zero, resulting in zero-current turn-off. During t6-t7, S1, S3, S5, and S6 remain on, and the output current flows through S5, D5, and L. o R o S6 and D6 freewheeling. The circuit's operating state is as follows: Figure 3 As shown in (g).
[0044] Mode 8 [t7-t8]: At time t7, S7 is turned on. Since the energy storage inductor current cannot change abruptly, the current in the branch containing D5 is still I. o Therefore, D7 is turned on under a positive voltage. Let the potentials at points d and c be equal. At this time, v cd =0. Because v ab =0, the transformer leakage inductance voltage is clamped to -V c The current flowing through the transformer begins to increase in reverse. During the period t7-t8, the primary current flows through the anti-parallel diode S1 and freewheeling diode S3, while the secondary current charges the clamping capacitor as it increases in reverse, causing the capacitor voltage to decrease slowly. The output current flows from S5, D5, and L... o R o S6, D6 loop to S7, D7, L o R o S6, D6, C c Circuit commutation. During the commutation process, the current in the branch containing S5 decreases, while the current in the branch containing S7 increases. At time t8, the transformer secondary current increases to -I. o The output current passes through S7, D7, and L. o R o S6, D6, C c The freewheeling and commutation processes have ended. The circuit's operating state is as follows: Figure 3 As shown in (h).
[0045] Mode 9 [t8-t9]: At time t8, S1 is turned off. Since the voltage across it is zero, S1 can achieve zero-voltage turn-off. During t8-t9, the primary current flows through the anti-parallel diode S1 and freewheeling diode S3, while the secondary current charges the clamping capacitor, reducing its voltage. The output current flows through S7, D7, and L. o R o S6, D6, C c Freewheeling current. The circuit's operating state is as follows: Figure 3 As shown in (i).
[0046] The control method described in this invention adds a transition process between the energy transfer stage and the current freewheeling stage of the output current source converter. Considering the dead time of the voltage source full-bridge switch, this transition process can be regarded as the sum of the dead time and T2. Taking the positive half-cycle of the converter as an example, the period from S4 to S8 is the transition process of the converter switching from the energy transfer stage to the current freewheeling stage (modes 4 to 6); the period from S7 to S2 is the transition process of the converter switching from the current freewheeling stage to the energy transfer stage (modes 8 to 9).
[0047] Based on the above modal analysis, when the converter is in the transition process from the energy transmission stage to the current freewheeling stage, the transformer leakage inductance voltage in mode 4 is clamped at -V. c This reduces the current flowing through the transformer to zero, allowing the steady-state value of the leakage inductance current to be reached earlier when the converter is operating in the current freewheeling stage, thus eliminating voltage spikes caused by the difference in leakage inductance current between the two operating stages during converter commutation.
[0048] When the converter is in the transition process from the current freewheeling stage to the energy transfer stage, the transformer leakage inductance voltage in mode 8 is clamped at -V. c This causes the transformer leakage inductance current to increase to -I. o This allows the leakage inductance current to reach its steady-state value earlier when the converter is operating in the energy transmission stage, eliminating voltage spikes caused by the difference in leakage inductance current between the two stages during converter commutation.
[0049] Based on the above modal analysis and Figure 2It can be seen that during one operating cycle of the converter, the current flowing through the transformer has the same polarity as the primary side voltage, and there is no stage where the polarity of the transmitted power is opposite to the average transmitted power. When the current source rectifier bridge switches S5 and S7 are in the off state, their maximum voltage is equal to the maximum voltage of the clamping capacitor. When the switches S6 and S8 are in the off state, their maximum voltage can reach zero. When the diodes D5-D8 are in the cutoff state, their maximum voltage is the sum of the transformer secondary voltage and the absolute value of the clamping capacitor voltage at time t0. Therefore, the backflow power during converter operation is eliminated, effectively reducing the voltage and current stress on the switches.
[0050] In the output current source type high-frequency isolated DC-DC converter control method of the present invention, T1 is set after the start and before the end of the energy transmission conversion stage, which ensures zero-current turn-on of switching transistors S6 and S8, and its value range is [missing value].
[0051]
[0052] Within the range of values, T1 should take a smaller value; taking a larger value will increase the lower limit of the shift ratio D.
[0053] In the output current source type high-frequency isolated DC-DC converter control method described in this invention, T2 is i se The reserved time for the transition from the steady-state value in one stage to the steady-state value in another stage should be greater than i. se The actual time Δt taken for the steady-state value of one stage to increase or decrease to the steady-state value of another stage is set during the transient process of the converter, and its constraints should satisfy...
[0054] T2≥(2~3)△t
[0055] The value of T2 can be obtained by the following formula.
[0056]
[0057] Simulation of the above equation using MATLAB yields the following result. Figure 4 and Figure 5 .
[0058] Figure 4 The graph shows the relationship between T2 and Δt under the condition that the clamping capacitor value is fixed at 66nF and the shift ratio is changed. It can be seen from the graph that Δt gradually decreases as T2 and the shift ratio D increase.
[0059] Figure 5 The graph shows the relationship between T2 and Δt when the shift ratio is fixed at 0.5 and the clamping capacitor value is changed. It can be seen from the graph that when T2 remains unchanged, Δt decreases as the clamping capacitor value decreases.
[0060] Therefore, when determining the value of T2, in order to satisfy the above constraints, it is necessary to comprehensively consider the relationship between the three parameters: T2, Δt, and the clamping capacitor value.
[0061] In the output current source type high-frequency isolated DC-DC converter and its control method described in this invention, the series clamping capacitor, in conjunction with the control method, clamps the transformer leakage inductance voltage to a negative value during the transition process, allowing the transformer leakage inductance current to reach the steady-state value of the leakage inductance current in the next operating stage earlier. Its rated voltage should be greater than the maximum voltage within one operating cycle. The capacitance value of the clamping capacitor and its maximum voltage can be obtained by the following formula.
[0062]
[0063] It will be apparent to those skilled in the art that the present invention is not limited to the details of the exemplary embodiments described above, and that the invention can be implemented in other specific forms without departing from its spirit or essential characteristics. Therefore, the embodiments should be considered in all respects as exemplary and non-limiting, and the scope of the invention is defined by the appended claims rather than the foregoing description. Thus, all variations falling within the meaning and scope of equivalents of the claims are intended to be included within the present invention. No reference numerals in the claims should be construed as limiting the scope of the claims.
[0064] Furthermore, it should be understood that although this specification describes embodiments, not every embodiment contains only one independent technical solution. This narrative style is merely for clarity. Those skilled in the art should consider the specification as a whole, and the technical solutions in each embodiment can also be appropriately combined to form other embodiments that can be understood by those skilled in the art.
Claims
1. A control method for a high-frequency isolated DC-DC converter with an output current source, characterized in that, This current-source type high-frequency isolated DC-DC converter consists of a voltage-source type inverter bridge, a high-frequency isolation transformer, a clamping capacitor, and a current-source type rectifier bridge. The voltage-source type inverter bridge is composed of two half-bridges connected in parallel. Each half-bridge has an upper and lower arm consisting of a switching transistor. The midpoint of the two half-bridges is the inverter bridge output terminal, connected to the primary side of the high-frequency isolation transformer. The current-source type rectifier bridge is also composed of two half-bridges connected in parallel. Each half-bridge has an upper and lower arm consisting of a switching transistor and a diode connected in series. A large inductor is connected in series on the output side of the current-source type rectifier bridge, and its input side is the midpoint of the two half-bridges. The clamping capacitor is connected in series between the secondary side of the high-frequency isolation transformer and the input side of the current-source type rectifier bridge. The voltage source inverter bridge consists of switches S1-S4, the current source rectifier bridge consists of switches S5-S8, and the clamping capacitor is capacitor C. c ; The converter described above includes the following steps: The voltage source inverter bridge employs phase-shift control; the duty cycle of the drive signals for switches S1-S4 is 50%; switches S1 and S3 are complementary to S2 and S4, respectively; the ratio of the phase difference between the drive signals of S3 and S1 to half a working cycle is the phase shift ratio D; the turn-on and turn-off times of the current source rectifier bridge switches S5, S6, S7, and S8 are referenced to the turn-on and turn-off times of S2, S4, S1, and S3, respectively. During the interruption period, the duty cycle of the drive signals of each switch is the same, all greater than 50%; switch S5 turns off after S2 turns on and is delayed by T1, and turns on ahead of T2 before S2 turns off; switch S6 turns on ahead of T1 before S4 turns off, and turns off after S4 turns on and is delayed by T2; switch S7 turns off after S1 turns on and is delayed by T1, and turns on ahead of T2 before S1 turns off; switch S8 turns on ahead of T1 before S3 turns off, and turns off after S3 turns on and is delayed by T2. Switch S1 and switch S2 are on the same half-bridge of the voltage source inverter bridge, and the upper half-bridge of this half-bridge is the switch S... 1， The lower half of the bridge is the switching transistor S2; Switches S3 and S4 are on the same half-bridge of the voltage source inverter bridge, with the upper half-bridge consisting of switch S4. 3， The lower half of the bridge is the switching transistor S4; Switch S5 and switch S6 are on the same half of the current source rectifier bridge, and the upper half of this half is the switch S... 5， The lower half of the bridge is the switching transistor S6; Switch S7 and switch S8 are on the same half of the current source rectifier bridge, and the upper half of this half is the switch S7. 7， The lower half of this half-bridge is the switching transistor S8.
2. The control method for an output current source type high-frequency isolated DC-DC converter according to claim 1, characterized in that, The switching transistors of the voltage source inverter bridge and the current source rectifier bridge are insulated gate bipolar transistors or metal-oxide-semiconductor field-effect transistors.
3. The control method for an output current source type high-frequency isolated DC-DC converter according to claim 1, characterized in that, This control method adds a transition process between the energy transfer stage and the current freewheeling stage of the current source converter. Considering the dead time of the voltage source full-bridge switch, this transition process can be regarded as the sum of the dead time and T2. Taking the positive half-cycle of the converter as an example, the period from S4 to S8 is the transition process of the converter switching from the energy transfer stage to the current freewheeling stage; the period from S7 to S2 is the transition process of the converter switching from the current freewheeling stage to the energy transfer stage.
4. The control method for a high-frequency isolated DC-DC converter with output current source type according to claim 1, characterized in that, T1 is set at the beginning and end of the energy transfer phase of the converter. Its function is to ensure that switches S6 and S8 can achieve zero-current turn-on. The value range of T1 is... Within the range of values, T1 should take a smaller value. Taking a larger value will increase the lower limit of the shift ratio D. In the formula, T s The switching cycle.
5. The control method for a high-frequency isolated DC-DC converter with output current source type according to claim 1, characterized in that, T2 is set during the converter's transient process. Its function is to allow sufficient time for the transformer leakage inductance current to transition from its steady-state value in one stage to its steady-state value in another stage. The value of T2 should meet the following constraints. T2≥(2~3)Δt Its value is obtained by the following formula. In the formula, Δt is the actual time taken for the transformer leakage inductance current to increase or decrease from its steady-state value in one stage to its steady-state value in another stage, T s For the switching cycle, C c L is the capacitance value of the clamping capacitor. k D is the shift ratio to account for the leakage inductance referred to the secondary side of the transformer.
6. The control method for a high-frequency isolated DC-DC converter with output current source type according to claim 1, characterized in that, During the transition process, the clamping capacitor, in conjunction with the control method, clamps the transformer leakage inductance voltage to a negative value, allowing the transformer leakage inductance current to reach the steady-state value of the leakage inductance current in the next operating stage earlier. Its rated voltage should be greater than the maximum voltage within one operating cycle. The capacitance value of the clamping capacitor and its maximum voltage can be obtained using the following formula. In the formula, Δt is the actual time taken for the transformer leakage inductance current to increase or decrease from its steady-state value in one stage to its steady-state value in another stage, T s For the switching cycle, C c L is the capacitance value of the clamping capacitor. k To account for the leakage inductance referred to the secondary side of the transformer, D is the shift ratio, V c_max I is the maximum value of the clamping capacitor voltage. o This represents the average output current.