A Hybrid Three-Level Dual Active Bridge Extended Phase-Shift Optimization Modulation Method

Abstract

The present invention discloses a hybrid three-level dual active bridge extended phase-shifted optimization modulation method, specifically: according to the given output voltage and output power requirements, calculate the real-time transmission power P<subgt;0< / subgt;<supgt;*< / supgt; and the real-time voltage conversion ratio M, determine the working mode of the hybrid three-level dual active bridge converter according to the given conditions, and determine the corresponding optimal solution expression according to the specific working conditions; obtain the corresponding optimal D<subgt;P0< / subgt>, D<subgt;P1< / subgt> and D<subgt;SS< / subgt> to drive the 10 switching tubes of the hybrid three-level dual active bridge converter, so as to realize the control of the hybrid three-level dual active bridge extended phase-shifted optimization modulation method. The present invention minimizes the inductor current stress within the full power range, takes into account the optimization of the switching loss and conduction loss of the hybrid three-level dual active bridge converter, and improves the efficiency of the hybrid three-level dual active bridge converter within the full power range.

Description

Technical Field

[0001] The present invention belongs to the technical field of converter control, and particularly relates to a hybrid three-level dual active bridge extended phase-shift optimization modulation method. Background Art

[0002] The diode-clamped hybrid three-level dual active bridge converter (Neutral Point Clamped Hybrid Three-level dual active bridge, NPCH3L-DAB) as shown in Figure 1 is composed of a primary full bridge formed by 8 switching tubes (S1 - S8) and 4 diodes (D1 - D4), a secondary full bridge formed by 4 switching tubes (Q1 - Q4), three voltage-regulating and filtering capacitors (C1, C2, and C3), a high-frequency transformer T, and an equivalent leakage inductance L. The input-side DC voltage source is V1, the output-side DC voltage source is V2, the transformer turns ratio is n:1, and the capacitance values of C1 and C2 are the same. The diode-clamped hybrid three-level dual active bridge converter (Neutral Point Clamped Hybrid Three-level dual active bridge, NPCH3L-DAB) as shown in Figure 1 is composed of a primary full bridge formed by 8 switching tubes (S1 - S8) and 4 diodes (D1 - D4), a secondary full bridge formed by 4 switching tubes (Q1 - Q4), three voltage-regulating and filtering capacitors (C1, C2, and C3), a high-frequency transformer T, and an equivalent leakage inductance L. The input-side DC voltage source is V1, the output-side DC voltage source is V2, the transformer turns ratio is n:1, and the capacitance values of C1 and C2 are the same.

[0003] The simplest and most traditional control strategy for the hybrid three-level dual active bridge converter is single-phase shift control (SPS). It controls the direction and magnitude of the transmitted power by controlling the phase-shift angle between the primary and secondary sides of the converter, but it has only one control degree of freedom. When applying SPS control to the diode-clamped hybrid three-level dual active bridge converter, zero-voltage switching can be achieved under heavy load, and full-range zero-voltage switching can be achieved when M = nV2 / V1 = 1 and M = 0.5. However, when the voltages on both sides of the transformer do not match, that is, when M is not equal to 1 and M is not equal to 0.5, the ZVS characteristic will be lost under light load, and there are also disadvantages such as large current stress, large reflux power, and low converter efficiency.

[0004] For example, in Reference [1], the hybrid three-level dual active bridge converter uses a single-phase-shift modulation strategy, and its operating modes are similar to those of the two-level DAB, without reflecting the more flexible characteristics of the three-level topology in modulation strategies. In References [2, 3], a three-level NPC-type bridge arm is introduced into the primary side of the DAB to form a three-level full-bridge converter. As Figure 1 shown, both of its two bridge arms are of NPC-type topologies, which are suitable for high-power and high-voltage applications. However, it has more switches and complex control.

[0005] To solve the above problems, a hybrid three-level dual active bridge converter composed of an NPC bridge arm and a half-bridge arm is introduced, and the extended phase-shift optimization modulation method is applied to this converter. Based on the traditional single-phase-shift control, the converter is regulated by increasing the phase shift of the diagonal switches in the H-bridge on one side of the transformer. This not only increases the control freedom of the system but also reduces the current stress and the reflux power of the converter, improving the system efficiency.

[0006] Since the primary side of the diode-clamped hybrid three-level dual active bridge converter is a three-level full bridge, both of its two bridge arms are of NPC-type topologies, consisting of eight switches and four diodes. Compared with the traditional two-level DAB converter, the number of switches in the high-voltage side bridge arm is doubled, which reduces the voltage stress of the switches by half under the same input voltage. There are more types of switches available, relatively reducing the cost of the converter. However, it has the largest number of devices and is very complex to control. Therefore, based on the above diode-clamped hybrid three-level dual active bridge, two switches and two diodes are reduced to form a hybrid three-level dual active bridge converter, whose topology includes an NPC bridge arm and a half-bridge arm. As Figure 2 shown, it has the same number of output levels as the full-bridge converter, without changing the input and output characteristics of the converter. However, the number of switching devices is reduced by 4, reducing the switching loss and making the control simpler, which is suitable for wide voltage range scenarios.

[0007] At the same time, when using SPS to control the hybrid three-level dual active bridge converter, the control strategy is proposed for 1 control variable, which cannot make full use of the topology advantages. The obtained control strategy is not the optimal solution, and when M is not equal to 1 and M is not equal to 0.5, the control effect is poor under light load, and the efficiency optimization cannot be achieved.

[0008] References:

[0009] [1] Jing Penghui, Wang Cong, Jiang Wei, et al. Performance Analysis of Isolated Three-level Half bridge Bidirectional DC / DC Converter. in: 7th International Power Electronics Motion Control Conference, 2012, 1527-1531.

[0010] [2] Moonem M.A., Krishnaswami H. Analysis and Control of Multi-level Dual Active Bridge DC-DC Converter. in: IEEE Energy Conversion Congress Exposition (ECCE), 2012, 1556-1561.

[0011] [3] Ortiz Gabriel, Uemura Hirofumi, Bortis Dominik, et al. Modeling of Soft-Switching Losses of IGBTs in High-Power High-Efficiency Dual-Active-Bridge DC / DC Converters. IEEE Transactions on Electron Devices, 2013, 60(2): 587-597. Summary of the Invention

[0012] The present invention aims to optimize the inductor current stress of a hybrid three-level dual active bridge, and provides an extended phase-shifted optimized modulation method for the hybrid three-level dual active bridge.

[0013] In an extended phase-shifted optimized modulation method for a hybrid three-level dual active bridge of the present invention, it is assumed that the conduction period of switching tubes S1, S2, and S5 is D p0 times the switching period T s , and the conduction period of switching tubes S1, S2, and S6 is D p1 times the switching period T s ; the difference between the starting times of the turn-on of switching tube S1 and Q1 is D ss times the switching period T s ; that is, D ss represents the relative phase shift between the primary and secondary sides, D p0 and D p1respectively represent the duty cycles of the original-side full-bridge 0 level and ±V1 levels; the secondary-side v s The duration of the high level nV2 or the low level -nV2 in one switching period is 1 / 2T s ; By controlling the phase shift of the original-side and secondary-side bridge arms, i.e., D ss to control the power flow direction. When D ss ranges from 0 to 1, the power is transmitted from the input side to the output side; when D ss is in the range of -1 to 0, the power is transmitted from the output side to the input side.

[0014] Specify the reference current I b and the reference power P b as:

[0015]

[0016] where, V1 is the input voltage, f s is the switching frequency, and L is the auxiliary inductor.

[0017] Calculate the voltage conversion ratio M as:

[0018]

[0019] [[ID=3ed]]

[0019] where, n is the transformer turns ratio, and V2 is the output voltage.

[0020] Conduct time-domain analysis on the light-load mode and heavy-load mode to obtain the inductor current stress value and the expression of the transmitted power.

[0021] When 0 ≤ M < 0.5:

[0022] Obtain the normalized inductor current stress value in the light-load mode and the expression of the normalized transmitted power P1 * as:

[0023]

[0024]

[0025] Obtain the normalized inductor current stress value in the heavy-load mode and the normalized transmitted power as:

[0026]

[0027]

[0028] Based on the Lagrange multiplier method, with the inductor current stress as the objective function and the transmitted power as the equality constraint condition, the optimal solution for the light-load mode is:

[0029]

[0030] Among them, is the real-time benchmark power.

[0031] The optimal solution for the heavy load mode is:

[0032]

[0033] The benchmark critical power point between the light load mode and the heavy load mode is defined as:

[0034]

[0035] Among them, P c1 is the critical power between the light load mode and the heavy load mode when 0 ≤ M < 0.5.

[0036] When 0.5 < M < 1:

[0037] The benchmark inductor current stress value of the light load mode is obtained and the benchmark transmission power of the expression:

[0038]

[0039]

[0040] The benchmark inductor current stress value of the heavy load mode is obtained and the benchmark transmission power of the expression:

[0041]

[0042]

[0043] Based on the Lagrange multiplier method, with the inductor current stress as the objective function and the transmission power as the equality constraint condition, the optimal solution of the light load mode is obtained:

[0044]

[0045] The optimal solution for the heavy load mode is:

[0046]

[0047] The benchmark critical power point between the light load mode and the heavy load mode is defined as:

[0048]

[0049] Among them, Pc2 It is the critical power between the light load mode and the heavy load mode when 0.5 < M < 1.

[0050] When M = 0.5, the optimal solution is:

[0051]

[0052] When M = 1, the optimal solution is:

[0053]

[0054] According to different voltage conversion ratios M, the corresponding optimal D P0 , D P1 and D SS are obtained to drive the 10 switching tubes of the hybrid three-level dual active bridge converter, realizing the control of the hybrid three-level dual active bridge extended phase-shifted optimization modulation method.

[0055] The beneficial technical effects of the present invention are as follows:

[0056] Through a novel hybrid three-level dual active bridge extended phase-shifted optimization modulation method, the present invention minimizes the inductor current stress within the full power range. The modulation method of the present invention takes into account the optimization of the switching loss and conduction loss of the hybrid three-level dual active bridge converter, improving the efficiency of the hybrid three-level dual active bridge converter within the full power range. Description of the Drawings

[0057] Figure 1 It is the circuit structure of the diode-clamped hybrid three-level dual active bridge converter.

[0058] Figure 2 It is the circuit structure of the hybrid three-level dual active bridge converter.

[0059] Figure 3 It is the typical waveform of the light load mode when 0 ≤ M < 0.5.

[0060] Figure 4 It is the typical waveform of the heavy load mode when 0 ≤ M < 0.5.

[0061] Figure 5 It is the typical waveform of the light load mode when 0.5 < M < 1.

[0062] Figure 6 It is the typical waveform of the heavy load mode when 0.5 < M < 1.

[0063] Figure 7 It is the flow chart of the hybrid three-level dual active bridge extended phase-shifted optimization modulation method of the present invention.

[0064] Figure 8This is the schematic diagram of the hybrid three-level dual active bridge extended phase-shifted optimization modulation method of the present invention. Specific embodiments

[0065] The present invention will be further described in detail below with reference to the accompanying drawings and specific implementation methods.

[0066] For a hybrid three-level dual active bridge extended phase-shifted optimization modulation method of the present invention, let the conduction period of switching transistors S1, S2, and S5 be D p0 times the switching period T s , and the conduction period of switching transistors S1, S2, and S6 be D p1 times the switching period T s ; the difference between the starting times of turning on switching transistor S1 and Q1 is D ss times the switching period T s ; that is, D ss represents the relative phase shift between the primary and secondary sides, D p0 and D p1 respectively represent the duty cycles of the 0 level and the ±V1 levels of the primary full bridge, and the duty cycle of ±V1 / 2 is 1 - D p0 - D p1 , so it is necessary to satisfy 1 - D p0 - D p1 > 0; the duration of the high level nV2 or the low level -nV2 of the secondary side v s in one switching period is 1 / 2T s ; the power flow is controlled by controlling the phase shift of the primary and secondary bridge arms, that is, D ss . When the range of D ss is 0 to 1, the power is transmitted from the input side to the output side; when D ss is -1 to 0, the power is transmitted from the output side to the input side.

[0067] Specify the reference current I b and the reference power P b as:

[0068]

[0069] where V1 is the input voltage, f s is the switching frequency, and L is the auxiliary inductor.

[0070] Calculate the voltage conversion ratio M as:

[0071]

[0072] where n is the transformer turns ratio and V2 is the output voltage.

[0073] Perform time-domain analysis on the light load mode and the heavy load mode to obtain the inductance current stress value and the transmission power expression.

[0074] When \(0\leq M\lt0.5\), the typical waveforms of its light - load mode and heavy - load mode are as shown in Figure 3 and Figure 4 shown

[0075] According to Figure 3 the normalized inductor current stress value of the light - load mode is obtained and the normalized transmission power \(P_1\) * The expressions are

[0076]

[0077]

[0078] According to Figure 4 the normalized inductor current stress value of the heavy - load mode is obtained and the normalized transmission power The expressions are

[0079]

[0080]

[0081] Based on the Lagrange multiplier method, taking the inductor current stress as the objective function and the transmission power as the equality constraint condition, the optimal solution of the light - load mode is

[0082]

[0083] where is the real - time normalized power (i.e., the per - unit value of the actual power of the circuit).

[0084] The optimal solution of the heavy - load mode is

[0085]

[0086] The normalized critical power point between the light - load mode and the heavy - load mode is defined as

[0087]

[0088] where \(P\) c1 is the critical power between the light - load mode and the heavy - load mode when \(0\leq M\lt0.5\).

[0089]

[0089] When \(0.5\lt M\lt1\), the typical waveforms of its light - load mode and heavy - load mode are as shown in Figure 5 and Figure 6 shown

[0090] According to Figure 5 the normalized inductor current stress value of the light - load mode is obtained and the normalized transmission power The expression of:

[0091]

[0092]

[0093] According to Figure 6 The normalized inductive current stress value of the heavy load mode is obtained and the normalized transmission power The expression of:

[0094]

[0095]

[0096] Based on the Lagrange multiplier method, with the inductive current stress as the objective function and the transmission power as the equality constraint condition, the optimal solution of the light load mode is obtained:

[0097]

[0098] The optimal solution of the heavy load mode is:

[0099]

[0100] The normalized critical power point between the light load mode and the heavy load mode Is defined as:

[0101]

[0102] Wherein, P c2 Is the critical power between the light load mode and the heavy load mode when 0.5 < M < 1.

[0103] When M = 0.5, the optimal solution is:

[0104]

[0105] When M = 1, the optimal solution is:

[0106]

[0107] The flow chart of the hybrid three-level dual-active-bridge extended phase-shift optimization modulation method of the present invention is as shown in Figure 7 According to the given output voltage and output power requirements, the real-time transmission power P0 * And the real-time voltage conversion ratio M are calculated, the working mode of the hybrid three-level dual-active-bridge converter is judged according to the given conditions, and the corresponding optimal solution expression is specifically adopted according to the specific working conditions.

[0108] The schematic diagram of the hybrid three-level dual-active-bridge extended phase-shift optimization modulation method of the present invention is as follows Figure 8 shown. A simple PI controller is used to control the required voltage. The real-time output voltage V2 obtained by sampling is used to obtain the real-time voltage conversion ratio M according to Expression (2). At the same time, the voltage error passes through the PI controller to obtain the real-time normalized power P0* and input it into the Figure 8 hybrid three-level dual-active-bridge extended phase-shift optimization modulation method shown as follows. According to the modulation method, the optimal D p1 , D p0 , D ss are obtained to drive the 10 switching tubes of the hybrid three-level dual-active-bridge converter to achieve the control of the hybrid three-level dual-active-bridge extended phase-shift optimization modulation method.

[0109] Based on Matlab / Simulink, a simulation platform for the hybrid three-level dual-active-bridge converter based on the extended phase-shift optimization modulation method and the traditional single-phase-shift modulation method is built. The simulation parameters are shown in Table 1:

[0110] Table 1 Simulation platform parameters of the hybrid three-level dual-active-bridge converter

[0111] <![CDATA[Input voltage V1]]> 380V <![CDATA[Output voltage V2]]> 114V (M = 0.6) 76V (M = 0.4) Transformer turns ratio n 2:1 Auxiliary inductor L 60 μH <![CDATA[Switching frequency f s > 100K <![CDATA[Maximum transmission power P b > 1800W

[0112] By giving the same set of normalized power P0* and voltage conversion ratio M, the test conditions of the method proposed in the present invention and the single-phase-shift modulation method are ensured to be the same, and the peak value and effective value of the inductor current of the two modulation methods are measured.

[0113] Table 2 Simulation data of the present invention and the traditional modulation method (M = 0.4)

[0114]

[0115] Table 3 Simulation data of the present invention and the traditional modulation method (M = 0.6)

[0116]

[0117] According to the simulation results in Table 2 and Table 3, it can be seen that the modulation method of the present invention can effectively reduce the inductor current stress and effective value in the full power range compared with the single-phase-shift control, effectively reducing the conduction loss. Therefore, the modulation method of the present invention effectively reduces the switching loss while reducing the conduction loss.

Claims

1. A hybrid three-level dual-active bridge extended phase-shift optimization modulation method, characterized in that, Let the conduction time period of switching transistors S1, S2, and S5 be D p0 times the switching period T s , and the conduction time period of switching transistors S1, S2, and S6 be D p1 times the switching period T s ; the difference between the starting times of turn-on of switching transistor S1 and Q1 is D ss times the switching period T s ; that is, D ss represents the relative phase shift between the primary and secondary sides, D p0 and D p1 respectively represent the duty cycles of the 0 level and the ±V1 levels of the primary full-bridge; the duration of the high level nV2 or the low level -nV2 of the secondary side v s in one switching period is 1 / 2T s ; by controlling the phase shift of the primary and secondary side bridge arms, that is, D ss , the power flow is controlled. When D ss is in the range of 0 to 1, the power is transmitted from the input side to the output side; when D ss is in -1 to 0, the power is transmitted from the output side to the input side; Specify the reference current I b and the reference power P b as follows: Among them, V1 is the input voltage, f s is the switching frequency, and L is the auxiliary inductor; The calculation of the voltage conversion ratio M is as follows: where n is the transformer turns ratio and V2 is the output voltage; Perform time-domain analysis on the light-load mode and heavy-load mode to obtain the inductor current stress value and the expression of the transmitted power; When 0 ≤ M < 0.5: Obtain the benchmark inductor current stress value in the light load mode and the benchmark transfer power P1 * The expression of: Obtain the benchmark inductor current stress value of the overload mode and the benchmark transmission power expression: Based on the Lagrange multiplier method, with the inductor current stress as the objective function and the transmitted power as the equality constraint condition, the optimal solution for the light-load mode is obtained as: Among them, is the real-time benchmark power; The optimal solution for the heavy-load mode is: The benchmark critical power point between the light load mode and the heavy load mode Is defined as: Among them, P c1 is the critical power between the light load mode and the heavy load mode when 0 ≤ M < 0.5; When 0.5 < M < 1: Obtain the normalized inductor current stress value in the light load mode and the expression of the normalized transmission power P3 * as follows: Obtain the benchmark inductor current stress value of the overload mode and the benchmark transmission power The expression of: Based on the Lagrange multiplier method, with the inductor current stress as the objective function and the transmitted power as the equality constraint condition, the optimal solution for the light-load mode is obtained: The optimal solution for the heavy-load mode is: The benchmark critical power point between the light load mode and the heavy load mode is defined as: Among them, P c2 is the critical power between the light load mode and the heavy load mode when 0.5 < M < 1; When M = 0.5, the optimal solution is: When M = 1, the optimal solution is: According to different voltage conversion ratios M, the corresponding optimal D is obtained P0 , D P1 and D SS , to drive the 10 switching tubes of the hybrid three-level dual-active bridge converter and realize the control of the hybrid three-level dual-active bridge extended phase-shifted optimization modulation method.

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