A High Dynamic and Fast Synchronization Method for Non-Spread Spectrum Continuous Wave Signals

Abstract

The present invention relates to the field of wireless communication technology, and more particularly to a high-dynamic fast synchronization method for non-spread spectrum continuous wave signals. The present invention comprises the following steps: S1, down-converting a received signal R(t) to a baseband to obtain a digital baseband signal r(n); S2, filtering the digital baseband signal r(n); extracting the filtered digital baseband signal and obtaining a first extracted signal r(n); N (n); S3, the first extraction signal r N The residual frequency deviation of the coarse synchronization signal (n) is synchronized to the symbol rate Rs level, and a coarse synchronization signal S4 is obtained. The residual frequency deviation of the coarse synchronization signal is synchronized to the Hz level, and a fine synchronization signal S5 is obtained. The fine synchronization signal is time-synchronized using a bit synchronization algorithm to obtain an initial synchronization signal. S6, the frequency deviation of the initial synchronization signal is tracked and the frequency deviation of the initial synchronization signal is compensated to obtain a target synchronization signal. The present invention achieves high-dynamic and fast synchronization of non-spread spectrum continuous wave signals with low algorithmic complexity.

Description

Technical Field

[0001] This invention relates to the field of wireless communication technology, and specifically to a high dynamic and fast synchronization method for non-spread spectrum continuous wave signals. Background Art

[0002] In high-dynamic scenarios, the ability to achieve rapid and accurate synchronization of transmitted signals will affect the performance of the entire communication system. Furthermore, issues such as Doppler frequency shift and Doppler rate of change caused by high dynamics can severely impact receiver demodulation.

[0003] Currently, most common fast synchronization methods for transmitted signals are based on spread spectrum systems. The core idea is to complete the coarse acquisition of Doppler frequency offset during the code acquisition stage of the spread spectrum signal. Examples include sliding correlation method, matched filtering method, independent channel method, FFT-based time-frequency two-dimensional acquisition method, and phase-locked loop method.

[0004] However, non-spread spectrum receivers cannot remove modulation from the received signal using code acquisition, thus making the aforementioned high-dynamic-range fast acquisition algorithms unusable. Existing carrier extraction and recovery algorithms for non-spread spectrum receivers assume a small frequency offset between the transmitted and received signals, making it difficult to overcome the Doppler frequency offset. Furthermore, when the non-spread spectrum receiver is a continuous wave, it requires not only high-precision frequency offset synchronization but also the ability to continuously track the signal. Summary of the Invention

[0005] The purpose of this invention is to provide a high-dynamic-range fast synchronization method for non-spread spectrum continuous wave signals, which can solve the problem of fast synchronization of non-spread spectrum continuous wave signals in high-dynamic scenarios, achieve high-precision acquisition and tracking of Doppler, and adapt to scenarios where the Doppler frequency offset is much greater than the symbol rate.

[0006] A high-dynamic-range fast synchronization method for non-spread spectrum continuous wave signals includes the following steps:

[0007] S1, downconvert the received signal R(t) to baseband to obtain the digital baseband signal r(n);

[0008] S2, the digital baseband signal r(n) is filtered; the filtered digital baseband signal is extracted to obtain the first extracted signal r. N (n);

[0009] S3, the first extraction signal r N The residual frequency offset of (n) is synchronized to the symbol rate Rs level, and a coarse synchronization signal is obtained.

[0010]

[0011] S4, the coarse synchronization signal The residual frequency offset is synchronized to the Hz level, and a fine synchronization signal is obtained.

[0012] S5, employing a bit synchronization algorithm to refine the synchronization signal. Perform timed synchronization to obtain the initial synchronization signal;

[0013] S6, perform frequency offset tracking on the initial synchronization signal and frequency offset compensation on the initial synchronization signal to obtain the target synchronization signal.

[0014] Furthermore, in step S3, the first extracted signal r is first processed. N (n) Perform frequency offset compensation to obtain a frequency offset compensation signal, and then synchronize the frequency offset compensation signal to the symbol rate Rs level.

[0015] Furthermore, step S3 specifically includes the following steps:

[0016] S31, perform multiple FFT transformations on the frequency offset compensation signal to obtain multiple sets of search operation results for Doppler frequency offset in the frequency domain; sum all search operation results and determine the peak-to-average power ratio;

[0017] S32, when the peak-to-average power ratio is greater than the preset threshold value Cor, the magnitude of the Doppler frequency offset F1 is determined according to the position P1 of the maximum value of the accumulated result;

[0018] S33, the Doppler frequency offset F1 is compensated onto the frequency offset compensation signal to obtain a coarse synchronization signal.

[0019] Furthermore, in step S31, all search results are accumulated using the differential coherent accumulation method, specifically including the following steps:

[0020] S311, Perform an M1-point FFT transform on the frequency offset compensation signal and search for the Doppler frequency offset in the frequency domain;

[0021] S312, the FFT result a of the current point M1. l The FFT result a of the previous set of M1 points l-1 Perform conjugate multiplication and sum them to obtain the l-th differential coherent summation result.

[0022]

[0023] Among them, A 1～l This represents the result of the first to the l-th difference coherent summation. Represents a l Take conjugate;

[0024] S313, after the l-th differential coherent accumulation is completed, determine whether the accumulation count l has reached the preset maximum value MaxCnt;

[0025] S314, when the cumulative count is greater than or equal to MaxCntt, the cumulative count is reset to zero, and the peak-to-average ratio R of the cumulative result is calculated. The formula for calculating the peak-to-average ratio is:

[0026] R = max(abs(A) 1～MaxCnt )) / mean(abs(A 1～MaxCnt ))

[0027] Where max(u) represents taking the maximum value of the array u in parentheses; abs(u) represents taking the modulus of the vector u in parentheses; mean(u) represents taking the average value of the array u in parentheses;

[0028] If the cumulative count is less than MaxCnt, then the cumulative count is incremented by 1, and steps S311 to S313 are repeated for the last M1 points of the frequency offset compensation signal.

[0029] Furthermore, in step S32, the formula for calculating the Doppler frequency offset F1 is as follows:

[0030] F1 = P / M*CLK / N1,

[0031] Where CLK is the sampling clock and N1 is the decimation factor of the filtered digital baseband signal.

[0032] Furthermore, in step S33, the coarse synchronization signal rN1(n) is determined by the following formula:

[0033]

[0034] Where π is the mathematical constant pi, j is the imaginary unit, and n is a positive integer.

[0035] Furthermore, step S4 specifically includes the following steps:

[0036] S41, regarding the coarse synchronization signal Divide the signal into segments of length M2 to obtain multiple segmented signals;

[0037] S42, each segment of the signal is filtered; the filtered segment of the signal is extracted to obtain the second extracted signal.

[0038] S43, extract each segment of the second decimation signal All signals undergo demodulation processing to obtain multiple demodulated signal segments; each demodulated signal segment is then subjected to an M3-point FFT transform to obtain the corresponding segment's abs(fft(S)). 2)) and determine abs(fft(S) in all segments 2 Find the maximum value of )) and determine the location P2 where the maximum value is located;

[0039] S44, Based on position P2, determine the residual frequency offset value of the corresponding segment's demodulation signal, and compensate the residual frequency offset value into the corresponding segment's demodulation signal to obtain a fine synchronization signal.

[0040] Furthermore, in step S43, when the signal modulation mode is BPSK, a square demodulation method is used, which is as follows:

[0041]

[0042] Among them, S 2 This is the demodulated signal.

[0043] Furthermore, in step S44,

[0044] when At that time, the formula for calculating the residual frequency offset value F2 is,

[0045] F2=(M3-P2) / M3×CLK / N1 / N2 / 2

[0046] when At that time, the formula for calculating the residual frequency offset F2 is,

[0047] F2 = -P2 / M3 × CLK / N1 / N2 / 2

[0048] The frequency control word derived from the calculated residual frequency offset value F2 is fed back to the carrier NCO module to compensate for the current M3 points, thereby obtaining the precise synchronization signal.

[0049]

[0050] Where CLK is the sampling clock, N1 is the decimation factor of the filtered digital baseband signal, π is pi, j is the imaginary unit, and n is a positive integer.

[0051] Furthermore, in step S5, the fine synchronization signal The sampling rate value is consistent with the symbol rate value Rs.

[0052] The beneficial effects of this invention are as follows:

[0053] This invention provides a high-dynamic-range, fast synchronization method for non-spread spectrum continuous wave signals. In this invention, the received signal R(t) is first transformed into a digital baseband signal r(n), and then the digital baseband signal r(n) is filtered to remove high-frequency components and out-of-band noise, while preventing spectral aliasing caused by subsequent signal decimation. Simultaneously, decimation of the filtered digital baseband signal reduces resource consumption and the computational load of FFT during frequency search. Next, the residual frequency offset of the first decimated signal rN(n) is synchronized to the symbol rate Rs level, and then the residual frequency offset of the coarse synchronization signal rN1(n) is synchronized to the Hz level. Finally, a bit synchronization algorithm is used to synchronize the fine synchronization signal. The invention performs timing synchronization to obtain an initial synchronization signal. Specifically, it acquires the initial synchronization signal through two rounds of carrier acquisition: "coarse synchronization" and "fine synchronization." Finally, it compensates for the frequency offset of the initial synchronization signal using frequency offset tracking to obtain the target synchronization signal.

[0054] In summary, this invention employs a two-round carrier acquisition combined with frequency offset tracking to achieve high dynamic and fast synchronization of non-spread spectrum continuous wave signals, while also possessing low algorithm complexity. Attached Figure Description

[0055] Figure 1 This is a flowchart illustrating this embodiment.

[0056] Figure 2 This is a schematic diagram of the carrier coarse synchronization and carrier fine synchronization process in this embodiment.

[0057] Figure 3 This is a schematic diagram of the carrier coarse synchronization process based on differential coherent accumulation in the embodiment.

[0058] Figure 4 This is a schematic diagram of the carrier fine synchronization process for BPSK signals in the embodiment.

[0059] Figure 5 This is a schematic diagram illustrating the cumulative result of a single coarse synchronization frequency search for the carrier in the embodiment.

[0060] Figure 6 This is a schematic diagram showing the results of four cumulative carrier coarse synchronization frequency searches in the embodiment.

[0061] Figure 7 This is a schematic diagram showing the results of 16 cumulative coarse synchronization frequency searches for the carrier in the embodiment.

[0062] Figure 8 This is the search result for the carrier fine synchronization frequency in the embodiment.

[0063] Figure 9This is the residual frequency offset result after passing through the coarse capture module in 500 random independent cases as shown in the example.

[0064] Figure 10 This is the residual frequency offset result after passing through the fine capture module in 500 random independent cases as shown in the example. Detailed Implementation

[0065] The present invention will be further described in detail below with reference to experimental examples and specific embodiments. However, this should not be construed as limiting the scope of the above-mentioned subject matter of the present invention to the following embodiments; all technologies implemented based on the content of the present invention fall within the scope of the present invention.

[0066] like Figures 1 to 10 As shown, this embodiment provides a high-dynamic-range fast synchronization method for non-spread spectrum continuous wave signals, which can solve the problem of fast synchronization of non-spread spectrum continuous wave signals in high-dynamic scenarios, achieve high-precision acquisition and tracking of Doppler, and adapt to scenarios where the Doppler frequency offset is much greater than the symbol rate.

[0067] In this embodiment, the symbol rate of the communication system is denoted as Rs, and the Doppler effect caused by high dynamic range is denoted as f. D In [-f D ,+f D Random within the range. Among them, the maximum Doppler frequency offset f D It is much greater than Rs.

[0068] This embodiment specifically includes the following steps:

[0069] S1, the receiver down-converts the received signal R(t) to a digital baseband signal r(n) after passing it through an analog-to-digital converter chip with a sampling clock of CLK. The interpolation coefficients of the digital baseband signal r(n) are OSR:

[0070]

[0071] S2, Filter the digital baseband signal r(n); Extract the filtered digital baseband signal to obtain the first extracted signal r. N (n). In this embodiment, the received signal R(t) is first transformed into a digital baseband signal r(n), and then the digital baseband signal r(n) is filtered to remove high-frequency components and out-of-band noise, and to prevent spectral aliasing caused by subsequent signal decimation. At the same time, decimation of the filtered digital baseband signal can also reduce resource consumption and the amount of FFT computation during frequency search.

[0072] In step S2, this embodiment performs N1-fold decimation on the filtered digital baseband signal to obtain the first decimated signal r. N(n), the oversampling factor is OSR / N1. At this time, the processing clock is reduced from the original CLK to CLK / N1, where...

[0073] CLK / N1≥2·Rs.

[0074] S3, the first extraction signal r N The residual frequency offset of (n) is synchronized to the symbol rate Rs level, and a coarse synchronization signal is obtained.

[0075]

[0076] In step S3, when the Doppler frequency offset range is tens or even hundreds of times the symbol rate Rs, direct low-pass filtering will reduce the first decimated signal r to... N (n) Filtering. Therefore, in this embodiment, the first extracted signal r is first filtered. N (n) Perform frequency offset compensation to obtain a frequency offset compensation signal, and then synchronize the frequency offset compensation signal to the symbol rate Rs level.

[0077] like Figure 3 As shown, step S3 in this embodiment specifically includes the following steps:

[0078] S31, perform multiple FFT transformations on the frequency offset compensation signal to obtain multiple sets of search operation results for Doppler frequency offset in the frequency domain; sum all search operation results and determine the peak-to-average power ratio.

[0079] Since the signal-to-noise ratio is often low in practical communication systems, a single FFT cannot accurately identify the position of the Doppler frequency offset. Therefore, multiple accumulations are performed on the frequency offset search operation. This embodiment uses differential coherent accumulation. Step S31 in this embodiment also includes the following steps:

[0080] S311 performs an M1-point FFT transform on the frequency offset compensation signal and searches for the Doppler frequency offset in the frequency domain.

[0081] S312, the FFT result a of the current point M1. l The FFT result a of the previous set of M1 points l-1 Perform conjugate multiplication and sum them to obtain the l-th differential coherent summation result.

[0082]

[0083] Among them, A 1～l This represents the result of the first to the l-th difference coherent summation. Represents a l Take conjugate;

[0084] S313, after the l-th differential coherent accumulation is completed, determine whether the accumulation count l has reached the preset maximum value MaxCnt;

[0085] S314, when the cumulative count is greater than or equal to MaxCntt, the cumulative count is reset to zero, and the peak-to-average ratio R of the cumulative result is calculated. The formula for calculating the peak-to-average ratio is:

[0086] R = max(abs(A) 1～MaxCnt )) / mean(abs(A 1～MaxCnt ))

[0087] Where max(u) represents taking the maximum value of the array u in parentheses; abs(u) represents taking the modulus of the vector u in parentheses; mean(u) represents taking the average value of the array u in parentheses;

[0088] If the cumulative count is less than MaxCnt, then the cumulative count is incremented by 1, and steps S311 to S313 are repeated for the last M1 points of the frequency offset compensation signal.

[0089] S32, when the peak-to-average power ratio is greater than the preset threshold Cor, the magnitude of the Doppler frequency offset F1 is determined according to the position P1 of the maximum value of the accumulated result;

[0090] S33, the Doppler frequency offset F1 is compensated onto the frequency offset compensation signal to obtain the coarse synchronization signal.

[0091] Specifically, in this embodiment, step S3 is referred to as carrier coarse synchronization. During carrier coarse synchronization, when no frequency offset compensation value is calculated, the carrier frequency entering the complex multiplier is 0; when coarse acquisition is successful, the carrier NCO will calculate the frequency control word based on the acquisition result, thereby outputting the frequency offset compensation carrier.

[0092] After passing through the complex multiplier, the first decimation signal r is obtained. N The 2048 points in (n) are subjected to FFT transformation to obtain frequency domain information a of length 2048. l , where l is the number of iterations. This embodiment uses differential coherent accumulation to process the current 2048-point FFT result a. l Compared with the previous FFT result a of 2048 points l-1 Perform conjugate multiplication and sum them to obtain the l-th differential coherent summation result:

[0093]

[0094] Among them, A 1～l This represents the result of the first to the l-th differential coherent summation. Represents a l Take conjugate.

[0095] After the l-th differential coherent accumulation is completed, it is determined whether the accumulation count l has reached the preset maximum value MaxCnt.

[0096] If the number of accumulations is less than MaxCnt, then the number of accumulations is incremented by 1, and the last 2048 points of the frequency offset compensation signal are taken, and steps S311 to S313 are repeated.

[0097] If the cumulative count is greater than or equal to MaxCnt, the cumulative count is reset to zero, and the peak-to-average ratio (PAR) R of the cumulative result is calculated. The PAR calculation formula is as follows:

[0098] R = max(abs(A) 1～MaxCnt )) / mean(abs(A 1～MaxCnt ))

[0099] Where max(u) represents taking the maximum value of the array u in parentheses; abs(u) represents taking the modulus of the vector u in parentheses; and mean(u) represents taking the average value of the array u in parentheses.

[0100] If the peak-to-average power ratio R is less than the threshold Cor for coarse carrier synchronization, then the coarse carrier acquisition is deemed to have failed, and the system will attempt coarse acquisition again.

[0101] If the peak-to-average power ratio (PAPR) R is greater than the carrier coarse synchronization threshold Cor, then the carrier coarse acquisition is considered successful, and the system determines the success of the differential coherent accumulation result A. 1～MaxCnt At position P1 with the largest modulus, the coarse acquisition estimate F1 of the Doppler frequency offset is calculated, and then the frequency control word is calculated. The formula for calculating the coarse acquisition estimate F1 of the Doppler frequency offset is:

[0102] F1 = P1 / M * CLK / N1

[0103] The carrier NCO generates a frequency offset compensation based on the frequency control word and applies it to the frequency offset compensation signal to obtain the coarse synchronization signal.

[0104]

[0105] Where π is the mathematical constant pi, j is the imaginary unit, and n is a positive integer.

[0106] At this point, coarse carrier synchronization is complete, and the first decimation signal r is now obtained. N The residual frequency offset of (n) is synchronized to the symbol rate Rs level.

[0107] S4 will use the coarse synchronization signal The residual frequency offset is synchronized to the Hz level, and a fine synchronization signal is obtained.

[0108] In this embodiment, step S4 is referred to as carrier fine synchronization. Figure 4 As shown, step S4 in this embodiment specifically includes the following steps:

[0109] S41, for coarse synchronization signal The signal is divided into multiple segments by length M2.

[0110] S42, each segment of the signal is filtered; the filtered segment of the signal is extracted to obtain the second extracted signal.

[0111] If the decimation factor in a single pass is too high, the corresponding decimation filter order will also be high, consuming significant resources. Therefore, multiple low-multiple decimations are used instead of a single high-multiple decimation. In carrier precision synchronization, the total decimation factor is denoted as N2, and the second decimated signal is obtained through decimation filtering. At this point, the oversampling factor drops to CLK / N1 / N2, and the processing clock also decreases again from CLK / N1 to N2 times. The following relationship still holds:

[0112] CLK / N1 / N2≥2·Rs

[0113] S43, extract each segment of the second decimation signal All signals undergo demodulation processing to obtain multiple demodulated signal segments; each demodulated signal segment is then subjected to an M3-point FFT transform to obtain the corresponding segment's abs(fft(S)). 2 )) and determine abs(fft(S) in all segments 2 The maximum value of )) and the location P2 where the maximum value is located.

[0114] Due to the second extraction signal As the signal is modulated and has a certain bandwidth, the Doppler magnitude cannot be accurately extracted directly using FFT. Therefore, the second decimated signal is first processed... Demodulation. When the signal modulation method is BPSK, the square demodulation method can be used, as follows:

[0115] S 2 =r N2 (n) 2

[0116] Among them, S 2 This is the demodulated signal.

[0117] For the demodulated signal S 2 Perform a 2048-point FFT to obtain 2048 points of one-dimensional frequency domain data. Calculate the modulus of the FFT result to obtain abs(fft(S)). 2 Compare abs(fft(S) one by one. 2 Find the value of )) and the maximum value, as well as the position P2 where the maximum value is located.

[0118] S44, Based on position P2, determine the residual frequency offset value of the corresponding segment's demodulation signal, and compensate the residual frequency offset value into the corresponding segment's demodulation signal to obtain a fine synchronization signal.

[0119] The carrier fine synchronization frequency offset is obtained based on the position of the maximum value. Since the residual frequency offset of the corresponding segment of the demodulated signal is doubled during squared demodulation, the residual frequency offset is calculated based on the position of the maximum value P2, as follows:

[0120] Since the number of FFT points in fine synchronization is 2048, the position of the maximum value ranges from 1 to 2048. When P2 is greater than 1024, the formula for calculating the residual frequency offset F2 is:

[0121] F2=(2048-P2) / 2048×CLK / N1 / N2 / 2

[0122] When P2 is less than or equal to 1024, the formula for calculating the residual frequency offset F2 is:

[0123] F2 = -P2 / 2048 × CLK / N1 / N2 / 2

[0124] The frequency control word is derived based on the calculated residual frequency offset value F2 and fed back to the carrier NCO module to compensate for the current 2048 points, thus obtaining the precise synchronization signal.

[0125]

[0126] Where CLK is the sampling clock, N1 is the decimation factor of the filtered digital baseband signal, π is pi, j is the imaginary unit, and n is a positive integer.

[0127] At this point, fine carrier synchronization is complete, and the coarse synchronization signal is ready. The residual frequency offset is reduced to the Hz level, such as Figure 10 As shown.

[0128] S5 uses a bit synchronization algorithm to finely synchronize the signal. Perform timing synchronization to obtain an initial synchronization signal. In step S5, the fine synchronization signal... The sampling rate value is consistent with the symbol rate value Rs. For BPSK signals, the Garden open-loop algorithm can be used.

[0129] S6 performs frequency offset tracking on the initial synchronization signal and compensates for the frequency offset to obtain the target synchronization signal. In continuous wave systems, residual frequency offset causes the received signal phase to change over time. Even after coarse and fine carrier synchronization, and although the residual frequency offset has been compensated to the Hz level, the received signal will still become out of phase over time, affecting the reception of the phase-modulated signal. Therefore, frequency offset tracking of the initial synchronization signal is necessary. Differential dismodulation can be used directly to track and compensate for the frequency offset of the initial synchronization signal.

[0130] This embodiment provides a high-dynamic-range, fast synchronization method for non-spread spectrum continuous wave signals. In this embodiment, the received signal R(t) is first transformed into a digital baseband signal r(n), and then the digital baseband signal r(n) is filtered to remove high-frequency components and out-of-band noise, and to prevent spectral aliasing caused by subsequent signal decimation. Simultaneously, decimating the filtered digital baseband signal reduces resource consumption and the amount of FFT computation during frequency search. Next, this embodiment decimates the first signal r... N The residual frequency offset of (n) is synchronized to the symbol rate Rs level, and then the coarse synchronization signal is... The residual frequency offset is synchronized to the Hz level, and the fine synchronization signal is achieved through a bit synchronization algorithm. Timing synchronization is performed to obtain an initial synchronization signal. That is, this embodiment obtains the initial synchronization signal through two rounds of carrier acquisition: "coarse synchronization" and "fine synchronization." Finally, this embodiment compensates for the frequency offset of the initial synchronization signal using frequency offset tracking to obtain the target synchronization signal.

[0131] In summary, this embodiment uses a two-round carrier acquisition combined with frequency offset tracking to achieve high dynamic and fast synchronization of non-spread spectrum continuous wave signals, while also having low algorithm complexity.

[0132] The above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included within the protection scope of the present invention.

Claims

1. A high-dynamic-range fast synchronization method for non-spread spectrum continuous wave signals, characterized in that: Includes the following steps, S1, downconvert the received signal R(t) to baseband to obtain the digital baseband signal r(n); S2, the digital baseband signal r(n) is filtered; Extract the filtered digital baseband signal to obtain the first decimation signal r. N (n); S3, the first extraction signal r N The residual frequency offset of (n) is synchronized to the symbol rate Rs level, and a coarse synchronization signal is obtained. S4, the coarse synchronization signal The residual frequency offset is synchronized to the Hz level, and a fine synchronization signal is obtained. S5, employing a bit synchronization algorithm to refine the synchronization signal. Perform timed synchronization to obtain the initial synchronization signal; S6, perform frequency offset tracking on the initial synchronization signal and frequency offset compensation on the initial synchronization signal to obtain the target synchronization signal; In step S3, the first extracted signal r is first processed. N (n) Perform frequency offset compensation to obtain a frequency offset compensation signal, and then synchronize the frequency offset compensation signal to the symbol rate Rs level; Step S3 details Includes the following steps, S31, perform multiple FFT transformations on the frequency offset compensation signal to obtain multiple sets of search operation results for Doppler frequency offset in the frequency domain; sum all search operation results and determine the peak-to-average power ratio; S32, when the peak-to-average power ratio is greater than the preset threshold value Cor, the magnitude of the Doppler frequency offset F1 is determined according to the position P1 of the maximum value of the accumulated result; S33, the Doppler frequency offset F1 is compensated onto the frequency offset compensation signal to obtain a coarse synchronization signal.

2. The high dynamic fast synchronization method for non-spread spectrum continuous wave signals according to claim 1, characterized in that: In step S31, all search results are accumulated using the differential coherent accumulation method. Specifically... Includes the following steps, S311, Perform an M1-point FFT transform on the frequency offset compensation signal and search for the Doppler frequency offset in the frequency domain; S312, the FFT result a of the current M1 point l The FFT result a of the previous set of M1 points l-1 Perform conjugate multiplication and sum them to obtain the l-th differential coherent summation result. Among them, A 1～l This represents the result of the first to the l-th difference coherent summation. Represents a l Take conjugate; S313, after the l-th differential coherent accumulation is completed, determine whether the accumulation count l has reached the preset maximum value MaxCnt; S314, when the cumulative count is greater than or equal to MaxCntt, the cumulative count is reset to zero, and the peak-to-average ratio R of the cumulative result is calculated. The formula for calculating the peak-to-average ratio is: R=max(abs(A 1～MaxCnt )) / mean(abs(A 1～MaxCnt )) Where max(u) represents taking the maximum value of the array u in parentheses; abs(u) represents taking the modulus of the vector u in parentheses; mean(u) represents taking the average value of the array u in parentheses; If the cumulative count is less than MaxCnt, then the cumulative count is incremented by 1, and steps S311 to S313 are repeated for the last M1 points of the frequency offset compensation signal.

3. The high dynamic fast synchronization method for non-spread spectrum continuous wave signals according to claim 2, characterized in that: In step S32, the formula for calculating the Doppler frequency offset F1 is as follows: F1 = P1 / M1 * CLK / N1, Where CLK is the sampling clock and N1 is the decimation factor of the filtered digital baseband signal.

4. The high dynamic fast synchronization method for non-spread spectrum continuous wave signals according to claim 3, characterized in that: In step S33, the coarse synchronization signal Determined by the following formula, Where π is the mathematical constant pi, j is the imaginary unit, and n is a positive integer.

5. The high dynamic fast synchronization method for non-spread spectrum continuous wave signals according to claim 1, characterized in that: Step S4 specific Includes the following steps, S41, regarding the coarse synchronization signal Divide the signal into segments of length M2 to obtain multiple segmented signals; S42, each segment of the signal is filtered; the filtered segment of the signal is extracted to obtain the second extracted signal. S43, extract each segment of the second decimation signal All signals undergo demodulation processing to obtain multiple demodulated signal segments; each demodulated signal segment is then subjected to an M3-point FFT transform to obtain the corresponding segment's abs(fft(S)). 2 )) and determine abs(fft(S) in all segments 2 Find the maximum value of )) and determine the location P2 where the maximum value is located; S44, Based on position P2, determine the residual frequency offset value of the corresponding segment's demodulation signal, and compensate the residual frequency offset value into the corresponding segment's demodulation signal to obtain a fine synchronization signal.

6. The high dynamic fast synchronization method for non-spread spectrum continuous wave signals according to claim 5, characterized in that: In step S43, when the signal modulation mode is BPSK, a square demodulation method is used, which is as follows: Among them, S 2 This is the demodulated signal.

7. The high dynamic fast synchronization method for non-spread spectrum continuous wave signals according to claim 5, characterized in that: In step S44, when At that time, the formula for calculating the residual frequency offset value F2 is, F2=(M3-P2) / M3×CLK / N1 / N2 / 2 when At that time, the formula for calculating the residual frequency offset F2 is, F2 = -P2 / M3 × CLK / N1 / N2 / 2 The frequency control word derived from the calculated residual frequency offset value F2 is fed back to the carrier NCO module to compensate for the current M3 points, thereby obtaining the precise synchronization signal. Where CLK is the sampling clock, N1 is the decimation factor of the filtered digital baseband signal, π is pi, j is the imaginary unit, and n is a positive integer.

8. The high dynamic fast synchronization method for non-spread spectrum continuous wave signals according to any one of claims 1-7, characterized in that: In step S5, the fine synchronization signal The sampling rate value is consistent with the symbol rate value Rs.

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