A quasi-constant power-constant voltage wireless charging device and method based on a power-adjustable PPS-S topology

Abstract

The present invention discloses a quasi-constant power-constant voltage wireless charging system and method based on a power-adjustable PPS-S topology. The system includes a power supply circuit, a resonant coupling circuit, and a signal monitoring and driving circuit. The power supply circuit supplies power to the resonant coupling circuit. The resonant coupling circuit includes a transmitting end and a receiving end. The signal monitoring and driving circuit is used to monitor the voltage and current at the transmitting end, and control the resonant coupling circuit to switch from the PPS-S topology to the LCC-S topology according to the voltage and current at the transmitting end. Under the PPS-S topology, the receiving end operates in the constant current output mode CC, and under the LCC-S topology, the receiving end operates in the constant voltage output mode CV, realizing quasi-constant power-constant voltage charging. The present invention can more accurately control the charging current and charging voltage, can largely maintain the stability of the charging state, shorten the charging time and improve the energy utilization rate, reduce the complexity of the control algorithm, reduce the switching loss, and has higher flexibility.

Description

Technical Field

[0001] The present invention relates to a wireless power transmission device and method, and particularly to a quasi-constant power-constant voltage wireless charging device and method based on a power-adjustable PPS-S topology. Background Art

[0002] With the unique characteristics of convenience, safety, and automatic operation, wireless charging technology has been widely applied in fields such as biomedical implants and electrified transportation. Subsequently, as the heart of power equipment, how to quickly and safely replenish the battery, that is, the electrical energy, has become a research hotspot for scholars. In the context of wireless charging technology, the safe and fast charging strategy of lithium batteries has become the focus of people's attention.

[0003] Compared with traditional charging methods, CP charging provides a constant output power for the battery while controlling the output power not to exceed the rated value, ensuring the maximum power transmission during the charging process, greatly shortening the charging duration required for full charge, and at the same time ensuring the stable change of power on the power supply side and the load side during the charging process. It has a wide range of application scenarios and is particularly suitable for devices such as unmanned aerial vehicles and intelligent inspection vehicles for unmanned operations. In practical applications, CC-CV charging requires designing different charging strategies for different types and specifications of batteries, increasing the complexity of design and application. However, CP charging can flexibly adapt to different types and specifications of batteries by adjusting the power settings, with higher compatibility and flexibility, simplifying the design and application of the charging system.

[0004] In addition, the charging temperature of the battery is also an important indicator for battery charging. On the premise of high-current charging, as the charging voltage of the battery increases, the problem of battery heating is inevitable. The temperature rise of the battery is often accompanied by the risks of battery swelling and explosion. Therefore, most current electrical devices are equipped with temperature warnings, and when the battery temperature is too high, the charging power will be forced to decrease to promote battery cooling. This measure ensures the charging safety of the battery but limits the charging rate. Due to the charging characteristics of the battery in CP charging, the battery voltage continuously increases while the battery current continuously decreases, which can effectively improve the heating situation of the battery, resulting in a continuous reduction in the energy dissipated in the form of heat, improving the energy utilization rate, and at the same time ensuring the charging safety.

[0005] As an emerging technology, constant power charging has the potential to improve charging speed, reduce heat loss, and extend battery life. This is not only crucial for the development of electric vehicles and mobile devices but also poses new challenges and opportunities for infrastructure and energy management systems. In the existing wireless charging field, the research on CP charging is relatively scarce. The existing research mainly focuses on aspects such as PT symmetry and semi-active rectifiers, which require relatively complex control algorithms and will generate certain switching losses. At the same time, there are certain deficiencies in the judgment of the battery charging state. Traditional CC-CV charging is relatively simple in terms of control algorithms and has a relatively mature judgment of the battery state. In addition, most of the current research on CP charging focuses on the process of power stability, and there are certain deficiencies in the regulation of the output power. Summary of the Invention

[0006] Object of the Invention: One object of the present invention is to provide a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology that can achieve adaptive switching between QCP-CV modes. Another object of the present invention is to provide a charging method based on the said device.

[0007] Technical Solution: A quasi-constant power-constant voltage wireless charging system based on a power-adjustable PPS-S topology according to the present invention includes a power supply circuit, a resonant coupling circuit, and a signal monitoring and driving circuit. The power supply circuit supplies power to the resonant coupling circuit. The resonant coupling circuit includes a transmitting end and a receiving end. The transmitting end includes n inductive switching branches, n capacitive switching branches, a constant voltage inductive switching branch, a constant voltage capacitive switching branch, a first compensation inductor L1, a first compensation capacitor C1, a second compensation capacitor C2, a third compensation capacitor C3, and a transmitting coil L T , when the n inductive switching branches, the n capacitive switching branches, and the constant voltage capacitive switching branch are not conducting, the resonant coupling circuit is in the PPS-S topology; when the n inductive switching branches, the n capacitive switching branches, and the constant voltage capacitive switching branch are all conducting, the resonant coupling circuit is in the LCC-S topology; the signal monitoring and driving circuit is used to monitor the voltage and current at the transmitting end and control the resonant coupling circuit to switch from the PPS-S topology to the LCC-S topology according to the voltage and current at the transmitting end. Under the PPS-S topology, the receiving end operates in the constant current output mode CC to charge the load; under the LCC-S topology, the receiving end operates in the constant voltage output mode CV to charge the load; achieving quasi-constant power-constant voltage charging.

[0008] Optionally, the n inductive switching branches have the same structure, each including an inductor and a switching switch connected in series; the n capacitive switching branches have the same structure, each including a capacitor and a switching switch connected in series; the constant voltage inductive switching branch includes an inductor and a switching switch connected in series; the constant voltage capacitive switching branch includes a capacitor and a switching switch connected in series.

[0009] Optionally, the first compensation inductor L1 is connected in parallel with n inductor switching branches, one end is connected to one end of the high-frequency inverter circuit, and the other end is connected to one end of the first compensation capacitor C1; the first compensation capacitor C1 is connected in parallel with n capacitor switching branches and the constant-voltage inductor switching branch, one end is connected to the other end of the first compensation inductor L1, and the other end is connected to the other end of the high-frequency inverter circuit; the second compensation capacitor C2 and the third compensation capacitor C3 are connected in series and then connected in parallel with the constant-voltage capacitor switching branch. After being connected in parallel, one end is connected to the other end of the first compensation inductor L1, and the other end is connected to one end of the transmitting coil L T One end; the other end of the transmitting coil is connected to the other end of the high-frequency inverter circuit; both ends of the transmitting coil are connected to the signal monitoring and driving circuit.

[0010] Optionally, for the PPS-S topology, the input current I1, the output current I4, and the output power P are expressed by the following formulas:

[0011]

[0012] where ω represents the system operating angular frequency, C1 is the first compensation capacitor, L2 is the second compensation inductor, R is the equivalent resistance of the battery and the rectifier circuit, U is the AC equivalent input voltage, and M is the mutual inductance between the transmitting coil and the receiving coil;

[0013] For the LCC-S topology, the expressions for the output voltage and the output power are as follows:

[0014]

[0015] where U R is the output voltage, and C5 is the overall capacitance of the parallel branch of the first compensation inductor L1 and n inductor switching branches.

[0016] Optionally, the parameter configuration scheme of each component in the resonant coupling circuit is as follows:

[0017]

[0018] where r is the value of the equivalent resistance, a to e are constants and not equal to 0, and A to E respectively represent the proportionality coefficients of the output power P and the equivalent resistance R of the PPS-S topology when the first compensation capacitor C1 and the second compensation inductor L2 take different values; P base is the CP reference value, △ is the accuracy, and (1 + △)P base is the maximum value of the variable power;

[0019] There is a certain proportional relationship among the coefficients A to E, and the solution is:

[0020]

[0021] To meet the QCP-CV charging requirement, the compensation components at all levels need to satisfy the following relationships:

[0022]

[0023] Among them, C1 is the first compensation capacitor, and C Pi represents the parallel value of the i-level compensation capacitor, and L Pi represents the parallel value of the i-level compensation inductor, where i = 0, 1,..., n - 1; C CV represents the value of C5 after switching from the CC mode to the CV mode, L2 is the second compensation inductor, R is the equivalent resistance of the battery and the rectification circuit, ω represents the system operating angular frequency, and M is the mutual inductance between the transmitting coil and the receiving coil.

[0024] Optionally, the input impedance of the system in the CC mode and the CV mode is calculated by the following formula:

[0025]

[0026] Among them, Z CC and Z CV represent the input impedances of the PPS-S topology and the LCC-S topology respectively, I in_CC is the input current of the PPS-S topology, and I in_CV is the input current of the LCC-S topology. U is the AC equivalent input voltage, ω represents the system operating angular frequency, M is the mutual inductance between the transmitting coil and the receiving coil, C1 is the first compensation capacitor, L2 is the second compensation inductor, R is the equivalent resistance of the battery and the rectification circuit, and C5 is the overall capacitance of the parallel branch of the first compensation inductor L1 and n inductance switching branches.

[0027] Optionally, a rectification circuit and a filter capacitor are also provided between the transmitting end of the resonant coupling circuit and the load.

[0028] Optionally, zero-voltage switching is adopted for the switching devices in the system.

[0029] The charging method using the above system includes the following steps:

[0030] S1. Initialize the system: Clear the internal counter CNT of the first MCU in the signal monitoring and driving circuit. The first MCU controls the system to be in the PPS-S topology. When charging starts, a constant charging current is provided to the load;

[0031] S2. CP stage: After the load is connected, the system enters the charging stage and starts charging in the constant current charging mode. The voltage at the transmitting end is monitored by the first MCU. As the charging process progresses, the voltage at the transmitting end will continuously rise. When it exceeds the preset value V Ti , the internal counter CNT of the first MCU increments by 1. When the count is i, the corresponding inductor switching switch Sia and the capacitor switching switch S ib act, corresponding to disconnecting the circuit connections of the compensation inductor and the compensation capacitor in the resonant coupling circuit;

[0032] S3, CV stage: When the count reaches n, S na and S nb are closed, and the circuit switches to the terminal CV charging topology. As the charging progresses, the voltage at the transmitting end will slowly decrease. When it is monitored that the voltage drops to the preset value V Tmin , the counter resets, the charging state ends, and the system returns to the initial state.

[0033] Furthermore, the system satisfies soft switching, and the specific steps are as follows:

[0034] (1) Sample and monitor VT, and pre-judge the VT value. When the VTRMS value is greater than the threshold, enter the timing monitoring;

[0035] (2) Call the internal clock of the MCU to monitor the PWM timing. After completing the cycle synchronization of sampling and monitoring, call the ADC pin of the MCU to monitor the voltage condition of the PWM waveform in real time for feedback;

[0036] (3) According to the timing logic, judge whether the real-time output of PWM is 0. According to the ADC monitoring, judge whether the PWM voltage value received in the current feedback link is 0. If it is accurate, record and synchronize the switch signal clock, prepare for conduction, and return to the internal clock part to repeat the verification. After the switch signal clock is synchronized 3 times, send a conduction signal to the Gate pole of the MOS tube to trigger the MOS tube to conduct.

[0037] Beneficial effects: Compared with the prior art, the significant advantages of the present invention are: (1) Simplify the control logic and algorithm, reduce the risk of system disorder, and improve system stability; (2) Rely on the designed PPS-S circuit to realize the adjustment of the output power by adjusting the value of a certain circuit element, simplifying the user operation; (3) Meet the quasi-ZPA operation, with less reactive power and higher charging efficiency; (4) Simple structure, lower component cost, perfect charging strategy, clear parameter configuration method, and the power supply device and the power-consuming device applying the wireless charging system transfer energy through the electromagnetic field, without physical direct contact, which can effectively cope with the influence of bad weather, humid environment, etc. on the charging of the drone, and has high power supply reliability. Description of the Drawings

[0038] Figure 1 is the equivalent circuit diagram of a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology provided by this embodiment;

[0039] Figure 2Simplified diagram of the PPS-S / LCC-S topology of a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology provided in this embodiment, where (a) is a schematic diagram of power supply equivalent transformation, (b) is the equivalent circuit of the SS topology, (c) is the equivalent circuit of the PPS-S topology, and (d) is the equivalent circuit of the LCC-S topology;

[0040] Figure 3 Schematic diagram of the ideal output power of the charging strategy of a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology provided in this embodiment;

[0041] Figure 4 Flowchart of the charging method of a quasi-constant power-constant voltage wireless charging method based on a power-adjustable PPS-S topology provided in this embodiment;

[0042] Figure 5 Flowchart of the realization of soft switching of a quasi-constant power-constant voltage wireless charging method based on a power-adjustable PPS-S topology provided in this embodiment;

[0043] Figure 6 Simulation curve of the output power of a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology provided in this embodiment, where (a) is the simulation curve of the output power of the system in State 1, (b) is the simulation curve of the output power of the system in State 2, (c) is the simulation curve of the output power of the system in State 3, and (d) is the simulation curve of the output power of the system in State 4;

[0044] Figure 7 Variation of output current, output voltage and output power of a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology provided in this embodiment under four states;

[0045] Figure 8 Simulation curves of G(ωCC), E(ωCV) and the corresponding input impedance angle under different load conditions provided in this embodiment;

[0046] Figure 9 Experimental prototype of a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology provided in this embodiment;

[0047] Figure 10 Variation of charging parameters of a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology provided in this embodiment without applying the QCP-CV charging controller, where (a) is the variation of charging voltage and current during charging, and (b) is the variation of charging power during charging;

[0048] Figure 11 The charging parameter variation of a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology provided in this embodiment under the application of a QCP-CV charging controller;

[0049] Figure 12 The state change of a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology provided in this embodiment before and after the switch action. Among them, (a) is the voltage change of the MOS tube before and after the driving signal is released, (b) is the voltage change of the compensation element before and after the driving signal is released, (c) is the circuit transient waveform before the driving signal is released, and (d) is the circuit transient waveform after the driving signal is released;

[0050] Figure 13 The continuous waveform diagram of the switching between two CC charging topologies of a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology provided in this embodiment under the QCP-CV charging strategy;

[0051] Figure 14 The continuous waveform diagram of the switching from the CC charging topology to the CV charging topology in the QCP-CV charging strategy of a quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology provided in this embodiment. Detailed implementation manners

[0052] The present invention will be described in detail below with reference to the accompanying drawings and specific embodiments. The accompanying drawings are only for illustrative purposes and cannot be construed as limiting the present invention; for better illustrating this embodiment, some components in the drawings will be omitted, enlarged or reduced, which does not represent the size of the actual product; for those skilled in the art, it is understandable that some well-known structures and their descriptions in the drawings may be omitted; the same or similar reference numerals correspond to the same or similar components; other different forms of changes or variations can be made based on the following description, and it is not necessary and impossible to enumerate all the implementation manners here. Any modifications, equivalent replacements, and improvements made within the spirit and principle of the present invention shall be included in the protection scope of the claims of the present invention.

[0053] Combining the advantages of CC-CV charging and CP charging, the present invention proposes a quasi-constant power-constant voltage (QCP-CV) charging method. The battery is quickly charged through the front-end CP charging, and automatically switches to the CV mode to protect the battery after the battery power reaches the ideal value, avoiding the safety hazards caused by overcharging the battery. In addition, considering the differences in the charging power required by different loads, the present invention innovatively proposes a parallel-parallel-series-series (PPS-S) topology with adjustable output power, which can achieve precise specified power output by adjusting the size of the compensation inductor. This circuit structure satisfies the quasi-zero phase angle (Quasi zero phase angle, QZPA), and the reactive power loss is almost zero, which can achieve a high transmission efficiency.

[0054] As Figure 1 shown, the quasi-constant power-constant voltage wireless charging device based on the power-adjustable PPS-S topology of the present invention includes several main parts: a power supply circuit, a resonant coupling circuit, a signal monitoring and driving circuit, and a battery charging circuit;

[0055] The power supply circuit includes a DC power supply V in , a high-frequency full-bridge inverter circuit Q1-Q4, and a high-frequency drive signal circuit; the resonant coupling circuit includes a transmitting end and a receiving end. The transmitting end includes a transmitting coil L T , a first compensation inductor L1, a first branch compensation inductor L 11 , a second branch compensation inductor L 12 , ……, an nth branch compensation inductor L 1n , a first compensation capacitor C1, a first branch compensation capacitor C 11 , a second branch compensation capacitor C 12 , ……, an nth branch compensation capacitor C 1n , a second compensation inductor L2, a second compensation capacitor C2, a third compensation capacitor C3, a constant voltage compensation capacitor C T , a first inductor switching switch S 1a , a second inductor switching switch S 2a , ……, an nth inductor switching switch S na , a first capacitor switching switch S 1b , a second inductor switching switch S 2b , ……, an nth inductor switching switch S nb , a first constant voltage switching switch S nc , a second constant voltage switching switch S nd ; the receiving end includes a receiving coil L R and a receiving compensation capacitor C4. Among them, the transmitting coil L T and the receiving coil L RCoupling, with a mutual inductance of M; the signal monitoring and driving circuit includes a first MCU controller and a current and voltage monitoring circuit, which is used to monitor the current and voltage of the transmitting coil. When the switching value is reached, an action signal is sent to the first MCU controller. After receiving the signal, the first MCU controller calls the internal counter and sends a driving signal to the GPIO port corresponding to the switch to be switched; the battery charging circuit includes a rectifier circuit D1 - D4, a filter capacitor C, and a battery equivalent resistance R BAT 。

[0056] The resonant coupling circuit includes a segmented constant current (CC) and constant voltage (CV) output mode, and overall constitutes a quasi-constant power - constant voltage (QCP - CV) output. The circuit structure is switched from a PPS - S topology to an LCC - S topology;

[0057] In the PPS - S / LCC - S topology, the first branch compensation inductor L 11 is connected in series with the first inductor switching switch S 1a to form a first inductor switching branch. The second branch compensation inductor L 12 is connected in series with the second inductor switching switch S 2a to form a second inductor switching branch,..., the nth branch compensation inductor L 1n is connected in series with the nth inductor switching switch S na to form the nth inductor switching branch. The first branch compensation capacitor C 11 is connected in series with the first capacitor switching switch S 1b to form a first capacitor switching branch. The second branch compensation capacitor C 12 is connected in series with the second capacitor switching switch S 2b to form a second capacitor switching branch,..., the nth branch compensation capacitor C 1n is connected in series with the nth capacitor switching switch S nb to form the nth capacitor switching branch. The second compensation inductor L2 is connected in series with the first constant voltage switching switch S nc to form a constant voltage inductor switching branch. The constant voltage compensation capacitor C T is connected in series with the second constant voltage switching switch S ndForm a constant-voltage capacitance switching branch in series. The first compensation inductor L1 is connected in parallel with the first inductor switching branch, the second inductor switching branch, ……, the nth inductor switching branch, one end is connected to one end of the high-frequency inverter circuit, and the other end is connected to one end of the first compensation capacitor. The first compensation capacitor is connected in parallel with the first capacitance switching branch, the second capacitance switching branch, ……, the nth capacitance switching branch, the constant-voltage inductor switching branch, one end is connected to the other end of the first compensation inductor, and the other end is connected to the other end of the high-frequency inverter circuit. The second compensation capacitor C2 is connected in series with the third compensation capacitor C3, and the whole is connected in parallel with the constant-voltage capacitance switching branch. After parallel connection, one end is connected to the other end of the first compensation inductor, and the other end is connected to one end of the transmitting coil. The other end of the transmitting coil is connected to the other end of the high-frequency inverter circuit. Both ends of the transmitting coil are connected to a current and voltage monitoring circuit for determining the circuit switching point.

[0058] When selecting the topology, first analyze the CC output mode. In the SS topology, the system output current is:

[0059]

[0060] where U is the system input voltage, I is the system output current, M is the coupling mutual inductance between the transmitting coil and the receiving coil, and ω is the operating angular frequency. When the system M, U, and ω are fixed, the output current of the SS topology is a constant value. To adjust the output current, it is necessary to change the number of turns and other characteristics of the transmitting coil L T and the receiving coil L R , the system input voltage U, or the operating angular frequency ω, which causes great damage to the stability of the system. Therefore, although the SS topology is for CC output, due to its poor controllability, the circuit design needs to be optimized. The redesigned topology equivalent circuit is as shown in Figure 2 . Figure 2 In (a), it is a schematic diagram of the equivalent transformation of a current source and a voltage source. Figure 2 In (b), the resonant coupling circuit is the SS topology equivalent circuit. In the figure, C R is the receiving coil compensation capacitor in the SS topology. Figure 2 In (c), the resonant coupling circuit is the PPS-S topology equivalent circuit. A voltage source in series with an inductor and then in parallel with a capacitor is equivalent to a current source in parallel with an inductor and then in parallel with a capacitor. A current source in parallel with an inductor and then in series with a capacitor is equivalent to a voltage source in series with an inductor and then in series with a capacitor. On this basis, by setting the resonance condition, the PPS-S topology can be equivalently obtained on the basis of the SS topology in Figure 2 In (b), as shown in Figure 2 In (c). According to Kirchhoff's law, the following equations can be listed:

[0061]

[0062] where X A= jωL1 - j / ωC1, X B = jωL2 - j / ωC1, X C = jωL2 + jωL TX - j / ωC2 - j / ωC3, X D = jωL RX - j / ωC4 + R. L1, L2 are compensation inductors, C1, C2, C3, C4 are compensation capacitors, L TX is the self - inductance of the transmitting coil, L RX is the self - inductance of the receiving coil, M is the coupling mutual inductance between the receiving coil and the transmitting coil, R BAT is the equivalent resistance of the battery, R is the equivalent resistance of the battery charging circuit. I1 - I4 are the four mesh currents, V in is the DC input voltage, U is the system input voltage. From the above analysis of the equivalent power supply, it can be seen that the system needs to satisfy the following resonance relationship:

[0063]

[0064] The mesh current I1 is the input current of the system, the mesh current I4 is the output current of the resonant coupling circuit. In the CC mode, the input current of the system, the output current of the resonant coupling circuit, and the output power P of the resonant coupling circuit CC can be expressed by the following formula:

[0065]

[0066] For the topology in the CV mode, an LCC - S topology similar to the PPS - S structure is selected, and its equivalent circuit is as shown in Figure 2 (d) in the figure. In the figure, L3 is the CV compensation inductor, C5, C6 are the CV compensation capacitors. The expressions for the output voltage and output power under this topology are:

[0067]

[0068] Among them, U R is the battery charging voltage in the CV mode, P CV is the output power of the resonant coupling circuit in the CV mode, and C5 is the overall capacitance of the parallel branch of the first compensation inductor L1 and the first inductor switching branch, the second inductor switching branch,..., the nth inductor switching branch.

[0069] Obviously, the output power P under the PPS - S topology is proportional to the equivalent resistance R, and the regulation of the output current can be achieved by segmentally adjusting the magnitudes of C1 and L2. The output power P under the LCC - S topology is inversely proportional to the equivalent resistance R, and the regulation of the output voltage can be achieved by adjusting the magnitude of C5, so as to limit the output power in a relatively stable range.

[0070] Based on the above topological design, a QCP-CV charging strategy is designed, and the ideal output power variation is as Figure 3 shown. Figure 3 where r is the resistance value of the equivalent resistance R, which is a non-zero constant, a - e are constants and not zero, and A - E respectively represent the proportionality coefficients of the output power P of the PPS-S topology with different values of C1 and L2 to the equivalent resistance R, P a ~P e are the output power variations for different values of A - E, CV_line is the output power variation in the constant voltage charging mode, and CP_line is the output power variation in the quasi-constant power mode. P base is the CP reference value, △ is the accuracy, and (1 + △)P base is the maximum value of the variable power. To achieve the ideal goal, a numerical relationship needs to be established among the topologies, as shown in the following formula:

[0071]

[0072] There is a certain proportional relationship among the coefficients A - E, and the solution is:

[0073]

[0074] In the setting of the present invention, L2 is a fixed quantity. Therefore, on the premise of system stability, when and only when C1 in the formula changes, the output power will be adjusted accordingly. Therefore, to meet the QCP-CV charging, the following relationships need to be satisfied for each level of compensation components:

[0075]

[0076] where i = 0, 1, …, n - 1; C Pi represents the parallel value of the i-level compensation capacitor, L Pi represents the parallel value of the i-level compensation inductor, and C CV represents the value of C5 in (5) after switching from the CC mode to the CV mode.

[0077] In circuit design, in order to reduce reactive power loss and improve the overall transmission efficiency, the PPS-S / LCC-S topology circuit should satisfy zero-phase-angle operation (Zero-Phase-Angle, ZPA). The input impedance of the system in the CC mode and the CV mode can be calculated by the following formula:

[0078]

[0079] where Z CC and Z CV represent the input impedances in the CC mode and the CV mode respectively, I in_CC is the input current of the PPS-S topology, I in_CVThe current input to the LCC-S topology.

[0080] To achieve ZPA operation, the input resistance of the circuit should be made as purely resistive as possible. It can be seen from (9) that under the resonance condition of the designed circuit of the present invention, Z CV already meets the requirement of pure resistance and can achieve ZPA operation. For the CC mode, it is necessary to satisfy 1 - ω 2 C1L2 = 0, that is, C1 and L2 resonate, in order to ensure the realization of ZPA operation. However, if this condition is met, C1 and L2 resonate open circuit, and L2 resonates short circuit with C2 at the same time, and the system will return to the SS topology again, which cannot meet the requirements for the adjustable characteristics of the circuit mentioned above. Therefore, this circuit scenario cannot fully meet the ZPA operation. In order to minimize the reactive power loss, it is necessary to satisfy 1 - ω 2 C1L2 approaches 0. In the parameter analysis, C1 has been used to adjust the output power and cannot be changed arbitrarily. Therefore, when determining the value of L2, L2 should be made to resonate with the overall parallel value of C1 and its subsequent compensation capacitors at all levels as much as possible to perform QZPA operation.

[0081] The signal monitoring and driving circuit includes a first MCU controller and a current and voltage monitoring circuit; the current and voltage monitoring circuit is used to monitor the current and voltage of the transmitting coil. When the switching value is reached, it sends an action signal to the first MCU controller. After receiving the signal, the first MCU controller calls the internal counter and sends a driving signal to the GPIO port corresponding to the switch to be switched (S1, S2,..., Sn).

[0082] Based on the same inventive concept, as Figure 4 shown, the charging method using the above charging system includes the following steps:

[0083] S1: Initialize the system, clear the internal counter CNT of the first MCU controller, initialize the switching switch, set S na 、S nb and S nd to be off, and the rest of the switches to be on. At this time, the compensation inductor L 1n 、compensation capacitor C 1n 、C T are not connected to the circuit, and the system is in the PPS-S topology. When charging starts, a constant charging current is provided for the load.

[0084] S2: CP stage (consisting of multiple CC stages). After the load is connected, the system enters the charging stage and starts charging in the constant current charging mode. The voltage across the transmitting coil is monitored by the first MCU controller. As the charging process progresses, the voltage across L T will continuously rise. When the current and voltage monitoring circuit monitors that the voltage across L T exceeds the preset value V TiWhen a signal is sent to the TIM_ETR counter port of the first MCU controller, the internal counter CNT of the first MCU controller increments by 1. When the count reaches i (0 < i < n), the i-th GPIO port sends a high-level signal, and the corresponding switch S i operates to disconnect L 1i and C 1i from the circuit, switching the circuit to the next state.

[0085] S3: CV stage. When the count reaches n, S n closes, and at this time the circuit switches to the CV charging topology. Due to the CV charging characteristic, as the charging progresses, the current flowing through L T will slowly decrease. When the current-voltage monitoring circuit monitors that the current flowing through L T drops to the preset value I Tmin , a signal is sent to the first MCU controller, the counter is reset, the charging state ends, and the system returns to the initial state.

[0086] For a quasi-constant power-constant voltage wireless charging method based on a power-adjustable PPS-S topology according to the present invention, to reduce switching losses and protect switching devices, zero-voltage switching should be satisfied to achieve soft switching. The specific steps are as follows, as Figure 5 shown:

[0087] S1: Sample and monitor the voltage V Ti across the transmitting coil, and pre-judge the V Ti value. When the root mean square V TiRMS value of the voltage across the transmitting coil is greater than the threshold, enter the timing monitoring.

[0088] S2: Call the internal clock of the first MCU controller to monitor the PWM timing. After completing the cycle synchronization of sampling and monitoring, call the ADC pin of the MCU to monitor the voltage condition of the PWM waveform in real time for feedback.

[0089] S3: According to the timing logic, judge whether the real-time PWM output is 0. According to the ADC monitoring, judge whether the PWM voltage value received in the current feedback link is 0. If it is accurate, record and synchronize the switch signal clock, prepare for conduction, and return to the internal clock part to repeat the verification. After the switch signal clock is synchronized 3 times, send a conduction signal to the Gate pole of the corresponding switching switch MOS transistor (S1, S2,..., S n ) to trigger the MOS transistor to conduct.

[0090] To verify the rationality of the theory of the present invention, a corresponding model is established in the finite element simulation software for simulation verification. Taking n = 3 as an example, the simulation parameters are shown in Table 1.

[0091] Table 1 System simulation parameters

[0092]

[0093] In the table, f is the system operating frequency, and R BAT_min is the minimum internal resistance of the battery, and R BAT_max is the maximum internal resistance of the battery. The rest of the capacitors and inductors are compensation capacitors and inductors. As the switch toggles, the circuit is divided into four stages, divided according to the response switch: after the start of charging and before the response of S1 is State 1; after the response of S1 and before the response of S2 is State 2; after the response of S2 and before the response of S3 is State 3; after the response of S3 and before the end of charging is State 4. Among them, State 1, State 2, and State 3 are in CC mode, and State 4 is in CV mode.

[0094] The output impedance angles of the system in the four states of State 1 - State 4 are as Figure 6 shown in (a) to (d). It can be seen that in State 1 - State 3, the system has a relatively small input impedance angle, achieving an approximate ZPA operation with relatively small reactive power loss; in State 4, the input impedance angle is 0, achieving a complete ZPA operation without reactive power loss.

[0095] As the charging process progresses, the output power generated by the system in the four states and the change in output power after adding the switching logic are as Figure 7 shown. The change in the battery charging voltage, charging current, and system output power is as Figure 8 shown. It can be seen from the figure that during the transition of the system from State 1 to State 3, the charging current decreases step by step, the charging voltage increases linearly, and the fluctuation amplitude of the output power is significantly clamped. Compared with State 1, it avoids damage to the battery caused by excessive voltage in the middle and late stages of charging; compared with State 2 and State 3, it not only improves the charging speed in the early stage but also ensures the charging safety of the battery; compared with State 4, it makes up for the safety hazard of excessive charging current in the early stage of CV charging. At the same time, the power fluctuation is controlled within an acceptable range, ensuring the stability of charging, which can not only improve the charging rate but also extend the battery life.

[0096] To verify the authenticity of the above theory and simulation, an experimental prototype as Figure 9 shown was built, and the prototype parameters are shown in Table 2.

[0097] Table 2 Experimental Prototype Parameters

[0098]

[0099] The components of the experimental prototype mainly include a high-power DC power supply (DH1798-10, 0-160V / 0-60A, 3000W), a high-frequency inverter (f = 85kHz), a coupler, a QCP-CV charging control circuit, an electronic load (DH27605B 500V / 200A, 5000W), and an oscilloscope (Tektronix 5Series MSO).

[0100] When the QCP-CV charging method is not applied, the variations of the load charging current, charging voltage, and charging power are as shown in Figure 10 (a) and (b) below. The maximum charging power of the load is set to 6.72W, and the maximum charging voltage is 16V. In State1, State2, and State3, when the load changes to 13Ω, 19Ω, and 26Ω respectively, safety hazards will occur due to exceeding the safety threshold. In State 4, safety hazards exist before the load changes to 23Ω. The variations of the load charging current, charging voltage, and charging power after applying the QCP-CV charging controller are as shown in Figure 11 shown below.

[0101] The present invention selects an N-channel power MOSFET (CSD18536KCS) as the switching device. The threshold voltage VGS(th) of this power MOS is 1.8V, which can be directly driven by the MCU, and the on-resistance is in the mΩ level, resulting in relatively small on-loss and off-loss, and having a relatively small impact on the transmission efficiency. Connect the Source electrodes of two CSD18536KCS as the switch, and simultaneously receive the driving signal from the MCU. During the switching process of the MOS transistor, in order to reduce the switching loss, zero voltage switching (ZVS) usually needs to be considered, that is, before the MOS transistor is turned on, it is necessary to ensure that the voltage VDS between the Drain and Source electrodes must be zero. In order to achieve ZVS, the switching drive principle needs to be further refined. From the analysis of the ZPA operation above, it can be seen that the topology circuit designed by the present invention satisfies the QZPA operation, that is, the input voltage, input current, the voltage of the inductor-side MOS transistor, and the PWM driving signal are in the same phase, and the phase of the voltage of the capacitor-side MOS transistor differs from the PWM driving signal by 90°. Then the internal clock of the MCU can be called to achieve ZVS. The state changes before and after the switching action are as shown in Figure 12 shown below. Figure 12 (a) below shows the voltage change of the MOS transistor before and after the driving signal is released, Figure 12 (b) below shows the voltage change of the compensation element before and after the driving signal is released, Figure 12 (c) below shows the circuit transient waveform before the driving signal is released, Figure 12 (d) below shows the circuit transient waveform after the driving signal is released.

[0102] Taking the example that when the load value reaches 13 Ω, the system switches from State1 to State2, and the system switches from Topology 3 to the CV topology, the continuous waveform change diagram is as Figure 13 and Figure 14 shown, which reflects the waveform transformation of each electrical parameter at the moment when the switch occurs.

[0103] The experimental prototype verifies the feasibility of the QCP-CV charging strategy of the present invention. Taking a 20V voltage environment as an example, when the maximum output power is limited to 6.72W, the QCP-CV charging strategy designed by the present invention can stabilize the output power between 6.3 - 6.7W. Compared with the traditional CC, CV, and CC-CV charging methods, the charging stability has been significantly improved, and the charging power is stabilized at the peak power with fewer control logics. On the premise of ensuring charging safety, the charging time required is greatly reduced.

[0104] The present invention can take into account the CC and CV output characteristics of the system and the overall transmission efficiency, and has higher application value; compared with the traditional constant current and constant voltage charging method, it can more accurately control the charging current and charging voltage, can largely maintain the stability of the charging state, shorten the charging time and improve the energy utilization rate; compared with the existing constant power wireless charging strategy, it reduces the complexity of the control algorithm and the switching loss; at the same time, the output power of this charging system can be adjusted by adjusting the component parameters, which has higher flexibility compared with the traditional method. The present invention can achieve zero phase angle and soft-switching operation, with less reactive power loss in the system and improved charging efficiency. Using this method can solve the inherent problems of the traditional wireless charging system such as low efficiency, high control cost, and high maintenance cost, can significantly improve the charging stability, and stabilize the charging power at the peak power with fewer control logics. While ensuring charging safety, the charging time required is greatly reduced. In the wireless charging field involving lithium battery charging such as electric vehicles, drones, and portable power electronic devices, the present invention has clear judgment logic, stable switching control method, efficient charging strategy, and relatively high charging efficiency, and has good economy and practicality.

Claims

1. A quasi-constant power-constant voltage wireless charging device based on a power-adjustable PPS-S topology, characterized in that, It includes a power supply circuit, a resonant coupling circuit, and a signal monitoring and driving circuit. The power supply circuit powers the resonant coupling circuit. The resonant coupling circuit includes a transmitting end and a receiving end. The transmitting end includes n inductive switching branches, n capacitive switching branches, a constant-voltage inductive switching branch, a constant-voltage capacitive switching branch, a first compensation inductor L1, a first compensation capacitor C1, a second compensation capacitor C2, a third compensation capacitor C3, and a transmitting coil L T , the first compensation inductor L1 is connected in parallel with the n inductive switching branches, one end is connected to one end of the high-frequency inverter circuit, and the other end is connected to one end of the first compensation capacitor C1; the first compensation capacitor C1 is connected in parallel with the n capacitive switching branches and the constant-voltage inductive switching branch, one end is connected to the other end of the first compensation inductor L1, and the other end is connected to the other end of the high-frequency inverter circuit; the second compensation capacitor C2 and the third compensation capacitor C3 are connected in series and then connected in parallel with the constant-voltage capacitive switching branch. After parallel connection, one end is connected to the other end of the first compensation inductor L1, and the other end is connected to one end of the transmitting coil L T 's one end; the other end of the transmitting coil is connected to the other end of the high-frequency inverter circuit; both ends of the transmitting coil are connected to the signal monitoring and driving circuit; when the n inductive switching branches, the n capacitive switching branches, and the constant-voltage capacitive switching branch are not conducting, the resonant coupling circuit is in the PPS-S topology; when the n inductive switching branches, the n capacitive switching branches, and the constant-voltage capacitive switching branch are all conducting, the resonant coupling circuit is in the LCC-S topology; the signal monitoring and driving circuit is used to monitor the voltage and current at the transmitting end, and control the resonant coupling circuit to switch from the PPS-S topology to the LCC-S topology according to the voltage and current at the transmitting end. In the PPS-S topology, the receiving end operates in the constant-current output mode CC to charge the load; in the LCC-S topology, the receiving end operates in the constant-voltage output mode CV to charge the load; achieving quasi-constant power-constant voltage charging; The n inductive switching branches have the same structure, each including a compensation inductor L connected in series 1i and an inductive switching switch S ia ; The n capacitive switching branches have the same structure, each including a compensation capacitor C connected in series 1i and a capacitive switching switch S ib ; The constant-voltage inductive switching branch includes a second compensation inductor L2 and a first constant-voltage switching switch S connected in series nc ; The constant-voltage capacitive switching branch includes a constant-voltage compensation capacitor C connected in series T and a second constant-voltage switching switch S nd , where i = 1, 2, …, n.

2. The quasi-constant power-constant voltage wireless charging device based on the power-adjustable PPS-S topology according to claim 1, wherein, For the PPS-S topology, the input current I1, the output current I4, and the output power P are expressed as follows: Where ω represents the system operating angular frequency, C1 is the first compensation capacitor, L2 is the second compensation inductor, R is the equivalent resistance of the battery and the rectification circuit, U is the AC equivalent input voltage, and M is the mutual inductance between the transmitting coil and the receiving coil; For the LCC-S topology, the expressions for the output voltage and the output power are as follows: Among them, U R is the output voltage, and C5 is the overall capacitance of the parallel branch of the first compensation inductor L1 and the n inductor switching branches.

3. The quasi-constant power-constant voltage wireless charging device based on the power-adjustable PPS-S topology according to claim 1, wherein, The parameter configuration scheme of each component in the resonant coupling circuit is as follows: Among them, r is the value of the equivalent resistance, a to e are constants and not zero, and A to E respectively represent the proportionality coefficients of the output power P of the PPS-S topology with different values of the first compensation capacitor C1 and the second compensation inductor L2 to the equivalent resistance R; P base is the CP reference value, △ is the accuracy, and (1 + △)P base is the maximum value of the variable power; There is a certain proportional relationship among coefficients A to E, and the solution is: To meet QCP-CV charging, each stage of compensation components need to satisfy the following relationship: Among them, C1 is the first compensation capacitor, C Pi represents the parallel value of the i-level compensation capacitor, L Pi represents the parallel value of the i-level compensation inductor, i = 0, 1, ……, n - 1; C CV represents the value of C5 after switching from the CC mode to the CV mode, L2 is the second compensation inductor, R is the equivalent resistance of the battery and the rectifier circuit, ω represents the system operating angular frequency, and M is the mutual inductance between the transmitting coil and the receiving coil.

4. The quasi-constant power-constant voltage wireless charging device based on the power-adjustable PPS-S topology according to claim 1, wherein, The input impedance of the system in CC mode and CV mode is calculated by the following formula: Among them, Z CC and Z CV represent the input impedances of the PPS-S topology and the LCC-S topology respectively, I in_CC is the input current of the PPS-S topology, I in_CV is the input current of the LCC-S topology, U is the AC equivalent input voltage, ω represents the system operating angular frequency, M is the mutual inductance between the transmitting coil and the receiving coil, C1 is the first compensation capacitor, L2 is the second compensation inductor, R is the equivalent resistance of the battery and the rectifier circuit, and C5 is the overall capacitance of the parallel branch of the first compensation inductor L1 and n inductive switching branches.

5. The quasi-constant power-constant voltage wireless charging device based on the power-adjustable PPS-S topology according to claim 1, characterized in that, A rectification circuit and a filter capacitor are also provided between the transmitting end of the resonant coupling circuit and the load.

6. The quasi-constant power-constant voltage wireless charging device based on the power-adjustable PPS-S topology according to claim 1, wherein, Zero-voltage switching is adopted for the switching devices in the system.

7. A quasi-constant power-constant voltage wireless charging method based on a power-adjustable PPS-S topology, characterized in that, Using the device according to any one of claims 1 to 6, the method includes the following steps: S1. Initialize the system: Clear the internal counter CNT of the first MCU in the signal monitoring and driving circuit. The first MCU controls the system to be in the PPS-S topology. When charging starts, provide a constant charging current for the load; S2. CP Phase: After the load is connected, the system enters the charging phase and starts charging in the constant current charging mode. The voltage of the transmitting end is monitored by the first MCU. As the charging process progresses, the voltage of the transmitting end will continuously rise. When it exceeds the preset value V Ti , the internal counter CNT of the first MCU increments by 1. When the count is i, the corresponding inductance switching switch S ia and the capacitance switching switch S ib act, and the compensation inductor and compensation capacitor in the resonant coupling circuit disconnect the circuit connection; S3. CV stage: When the count reaches n, S na and S nb are pulled in, the circuit switches to the end CV charging topology. As the charging progresses, the voltage at the transmitting end will slowly decrease. When it is monitored that the voltage drops to the preset value V Tmin , the counter is reset, the charging state ends, and the system returns to the initial state; The system satisfies soft switching, and the specific steps are as follows: (1) Sample and monitor VT, and make a preliminary judgment on the VT value. When the VTRMS value is greater than the threshold, enter the timing monitoring; (2) Call the internal clock of the MCU to monitor the PWM timing. After completing the cycle synchronization of sampling and monitoring, call the ADC pin of the MCU to monitor the voltage condition of the PWM waveform in real time for feedback; (3) According to the timing logic, judge whether the PWM real-time output is 0, and according to the ADC monitoring, judge whether the PWM voltage value received in the current feedback link is 0. If it is accurate, record and synchronize the switch signal clock, prepare for conduction, and return to the internal clock part to repeat the verification; When the synchronization of the switch signal clock is repeated 3 times, send a conduction signal to the Gate electrode of the MOS transistor to trigger the MOS transistor to conduct.

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