Double-fed wind turbine wide frequency oscillation suppression method based on adaptive impedance reshaping

Abstract

The application provides a double-fed fan wide-frequency oscillation suppression method based on adaptive impedance remodeling, for the sub / super synchronous frequency band resonance suppression, an adaptive virtual impedance controller is added to the RSC current control link, a five-order Butterworth low-pass filter is selected and the low-pass filter is adaptively improved, so that the response characteristic in the passband can be adaptively increased with the frequency, and the characteristic of adapting to the oscillation frequency change is achieved; for the resonance suppression of the middle frequency band, an active damping control is added in the GSC current control link, and the parameter deviation and uncertainty of the controlled object are considered, so that the impedance remodeling effect of the GSC is not affected by the parameters of the converter itself, and the phase compensation can be adaptively performed when the parameters of the converter change. Through the frequency band control, the negative damping characteristics of the DFIG in the sub / super synchronous frequency band and the middle frequency band are improved, and the grid-connected system can adapt to different power grid conditions, and effectively suppresses the oscillation in a wide frequency range.

Description

TECHNICAL FIELD

[0001] The present application relates to the technical field of power system wide frequency oscillation, in particular to a double-fed wind turbine wide frequency oscillation suppression method based on adaptive impedance remodeling. BACKGROUND

[0002] With the increasing penetration of new energy in new power systems, the power system presents the characteristics of high power electronics, the dynamic process inside the system is increasingly complex, and the power system oscillation stability problem is gradually highlighted. Unlike traditional low-frequency oscillation, sub- / super-synchronous oscillation, large-scale double-fed wind turbines (DFIG) will make the power system stability problem present the characteristics of wide frequency domain, strong time-varying and wide-area propagation. The complex control loop of wind turbines leads to the dynamic interaction process between them and the AC power grid, and the coupling effect is more obvious. The wideband oscillation of this coupling characteristic has become a new form of oscillation.

[0003] At present, the wide frequency oscillation is defined as the periodic oscillation of electrical quantities caused by the interaction between power electronic devices and power grid, and the dynamic process of the oscillation frequency changing in the range of several Hz to several thousand Hz. The main reason for the wide frequency oscillation is the dynamic interaction between a large number of heterogeneous power electronic devices and the power grid. The specific performance is the resonance type wide frequency oscillation caused by the interaction between power electronic devices and series compensation and shunt compensation capacitors, and the medium and high frequency resonance caused by the interaction between power electronic phase-locked loop and weak power grid. According to the impedance stability theory, the stability of the grid-connected system depends on the phase margin of the intersection of the amplitude of its output impedance and the grid impedance. If the phase margin is insufficient at this time, the harmonic resonance of the grid-connected voltage will occur, which will worsen the stability of the power grid and the operation performance of the wind power system, and even lead to disconnection. Therefore, how to suppress the wide frequency oscillation of double-fed wind turbines connected to the grid has become a key problem.

[0004] In the prior art, the means for suppressing wide-frequency oscillation of wind turbines mainly include parameter optimization adjustment and additional damping control. For the parameter optimization adjustment mode, the control parameters are optimized to prevent and control the wide-frequency oscillation. For example, the system is subjected to sensitivity analysis, and according to the analysis result, it is pointed out that the dominant parameters affecting the system stability are the current loop, the phase-locked loop proportional coefficient and the grid capacitance, and a parameter adjustment strategy is formulated. Such means are mainly applied to provide parameter design basis for converter manufacturers before the wind turbine is connected to the grid. In addition, in the process of parameter optimization, new oscillation modes may be excited or the damping of other oscillation modes may be weakened, which is also not conducive to system stability, so this method has certain limitations. For the additional damping control strategy, the control is added to the converter to improve the damping of the wind power generator. For example, a current feedback control of an improved band filter is designed, and a voltage feedforward is added to the current loop control link to realize impedance reshaping. Although the additional damping control method has the advantages of economy and flexibility, the existing control means cannot cope with the time-varying characteristics of the wide-frequency oscillation, and it is difficult to effectively improve the damping of the converter in a wide frequency range.

[0005] At present, there are problems such as complex and variable grid operation conditions, and new energy stations usually transmit power through series compensation or parallel compensation lines over long distances, so the impedance characteristics of the grid side have strong time-varying characteristics, and most of the existing control methods usually have a narrow frequency range, which is difficult to effectively suppress; most of the current suppression strategies need to be designed for specific unit parameters and control structures, but the capacity, parameters and control structures of different wind turbines usually have great differences, and when the unit parameters deviate, the suppression effect will also weaken. SUMMARY

[0006] In view of the technical problems that the existing wide-frequency oscillation suppression method will weaken the suppression effect when facing complex and variable grid operation conditions and unit parameter deviation, the purpose of the present application is to propose a double-fed wind turbine wide-frequency oscillation suppression method based on adaptive impedance reshaping, so that the DFIG grid-connected system can adapt to different grid operation conditions and effectively suppress the oscillation in a wide frequency range.

[0007] To achieve the above purpose, the present application proposes a double-fed wind turbine wide-frequency oscillation suppression method based on adaptive impedance reshaping, comprising the following steps:

[0008] Step 1, the three-phase currents flowing into the stator port and the rotor winding are sampled by current sensors and output, respectively, and are denoted as i sa,sb,sc and i ra,rb,rc ; the grid-side filter inductance current is sampled by a current sensor and output, denoted as i La,Lb,Lc ; the three-phase stator voltage is sampled by a voltage sensor and output, denoted as three-phase stator voltage v sa,sb,sc ;

[0009] Step 2, according to the phase-locked angle θ PLL , the three-phase current i flowing into the stator port and the three-phase current i flowing into the rotor winding are calculated by Park transformation sa,sb,sc ra,rb,rc the current i in the dq rotating coordinate system sdq and i rdq the current i of the grid-side filter inductor in the dq rotating coordinate system Ldq , and the three-phase stator voltage v sa,sb,sc the output voltage v in the dq rotating coordinate system sdq ;

[0010] Step 3, according to the dq-axis rotor output current i rd , i rq and the d-axis stator output voltage v sd , the machine-side dq-axis modulation signals m Rd , m Rq are generated by current decoupling control and stator voltage feedforward control

[0011] Step 4, according to the dq-axis rotor output current i rd , i rq , the modulation correction signals △m d1 , △m q1 are generated after frequency selection and phase correction by an adaptive virtual impedance controller, and are superimposed into the machine-side dq-axis modulation signals m Rd , m Rq respectively, to obtain the modified machine-side dq-axis modulation signals

[0012] Step 5, according to the dq-axis current i flowing into the stator port sd , i sq , the grid-side dq-axis modulation signals m Gd , m Gq are generated by current decoupling control

[0013] Step 6, according to the dq-axis current i flowing into the stator port Ld , i Lq , the modulation correction signals △m d2 , △m q2 are generated after phase correction and noise isolation by an adaptive active damping controller, and are superimposed into the grid-side dq-axis modulation signals m Gd , m Gq respectively, to obtain the modified grid-side dq-axis modulation signals

[0014] Step 7, according to the phase-locked angle θ PLL , the machine-side dq-axis modulation signals m Rd , m​Rq and grid-side dq-axis modulation signal m Gd , m Gq After inverse Park transformation, the machine-side converter switching signal and the grid-side converter switching signal in three-phase stationary coordinate system are obtained, and the impedance reshaping is performed on the DFIG sub / super synchronous frequency band and the medium frequency band by changing the converter output of the machine side and the grid side.

[0015] In further embodiments, in the step 2, the phase-locked angle θ PLL is obtained by the phase-locked loop transfer function of the grid-side voltage q-axis component, and is expressed as:

[0016]

[0017] In the formula, k pp , k pi respectively represent the proportional coefficient and the integral coefficient of the phase-locked loop, v q represents the grid-side voltage q-axis component.

[0018] In further embodiments, in the step 3, according to the dq-axis rotor output current i rd , i rq and the d-axis stator output voltage v sd , the machine-side dq-axis modulation signal m Rd , m Rq is generated by current decoupling control and stator voltage feedforward control, including:

[0019] The machine-side dq-axis modulation signal m Rd , m Rq is calculated according to the following method:

[0020]

[0021] In the formula, K respectively represent the proportional and integral coefficients of the d-axis and q-axis current controllers of the machine side, K rd represents the decoupling gain coefficient, and K f represents the voltage feedforward gain coefficient.

[0022] In further embodiments, in the step 4, the transfer function H fa (s) of the adaptive virtual impedance controller is expressed as:

[0023]

[0024] In the formula, H f (s) is a five-order Butterworth low-pass filter, represents the normalized Laplacian operator, ω c represents the cut-off angular frequency, and takes 2π×100 rad / s; kf represents a virtual impedance coefficient;

[0025] In the further embodiment, the amplitude-frequency characteristic of the adaptive virtual impedance controller linearly increases with the increase of frequency in the sub / super synchronous frequency band, and the phase range is kept in-50°-50°, meeting the requirement of impedance reshaping target 1. In the adaptive virtual impedance controller, with the increase of the virtual impedance coefficient k sa = -s 2 as an adaptive impedance, the original virtual impedance controller can adaptively increase its gain with the increase of frequency.

[0026] In the further embodiment, the amplitude-frequency characteristic of the adaptive virtual impedance controller linearly increases with the increase of frequency in the sub / super synchronous frequency band, and the phase range is kept in-50°-50°, meeting the requirement of impedance reshaping target 1. In the adaptive virtual impedance controller, with the increase of the virtual impedance coefficient k f , the amplitude-frequency characteristic of the adaptive virtual impedance controller increases, and the phase-frequency characteristic is not affected.

[0027] In the further embodiment, in step 5, the grid-side dq-axis modulation signals m sd , m sq are generated through current decoupling control according to the stator port current i Gd , i Gq , comprising:

[0028] The grid-side dq-axis modulation signals m Gd , m Gq are calculated in the following manner:

[0029]

[0030] In the formula, K and K d represent the proportional and integral coefficients of the d-axis and q-axis current controllers respectively, and K d represents the decoupling gain coefficient.

[0031] In the further embodiment, in step 6, the transfer function G m (s) of the adaptive active damping controller is expressed as:

[0032]

[0033] In the formula, G p (s) is a multi-order lead-lag correction controller, which is a phase compensation controller as a phase correction element, and is expressed as:

[0034]

[0035] In the formula, K p represents a compensation gain, and ω h , ω lω n represents the upper and lower limits of the modulation angle frequency, and n represents the order of the controller;

[0036] G p (s) The expression of the adjustable maximum phase angle is:

[0037]

[0038] Wherein, G f (s) represents a second-order Chebyshev low-pass filter, a noise suppression controller as a noise isolation link, and the expression is:

[0039]

[0040] In the formula, K c represents a gain coefficient, s pi represents a filter pole, i=1,2, s i =s i +jω i ; both satisfy the following expression:

[0041]

[0042] In the formula, ω cut represents the filter cutoff angular frequency, and ε represents the fluctuation coefficient.

[0043] Therefore, the present application aims at the problem that the existing wide-frequency oscillation suppression strategy of the double-fed wind turbine is difficult to effectively act in the case of grid operation condition change, and the suppression effect is also weakened when the wind turbine capacity and control parameters deviate, and proposes a double-fed wind turbine wide-frequency oscillation suppression method based on adaptive impedance remodeling, which remodels the DFIG impedance in the sub / super synchronous frequency band and the medium frequency band by designing RSC adaptive virtual impedance control and GSC adaptive active damping control.

[0044] In combination with the above embodiment of the double-fed wind turbine wide-frequency oscillation suppression method based on adaptive impedance remodeling, by analyzing the impedance characteristic curve of the DFIG in the wide frequency band, considering the main factors affecting the impedance characteristics of the double-fed wind turbine in each frequency band, the resonance suppression method for the sub / super synchronous frequency band selects to modify the RSC current control link, adds virtual impedance control, and adaptively improves the low-pass filter, so that the response characteristics in the passband can change with the frequency, and the change of the grid operation condition is considered; the resonance suppression method for the medium frequency band selects to add active damping control in the GSC current control link, considers the phase correction link and the noise isolation link, and considers the parameter deviation of the controlled object. Through the frequency band control, the negative damping characteristics of the DFIG in the sub / super synchronous frequency band and the medium frequency band are significantly improved as a whole.

[0045] From the above technical solutions of the present application, compared with the prior art, the present application has the following advantages:

[0046] 1. The RSC virtual impedance control considers the complex and changeable grid operating conditions, avoids affecting the medium and high frequency band by adaptively improving the Butterworth low-pass filter, makes the response characteristics in the passband adaptively increase with the frequency, has the characteristics of adapting to the change of oscillation frequency, and meanwhile, with the increase of the virtual impedance coefficient, the amplitude-frequency characteristics of the adaptive virtual impedance control also increase, and the phase-frequency characteristics are not affected;

[0047] 2. The GSC active damping control considers the parameter deviation and uncertainty of the controlled object, and the control effect is only related to the amplitude-frequency characteristics of the controller itself and K PWM , and therefore the impedance remodeling effect of the GSC is not affected by the parameters of the converter itself, and the phase compensation can be adaptively performed when the parameters of the converter change, and the characteristics of different units can be adapted.

[0048] It should be understood that all combinations of the aforementioned concepts and additional concepts described in greater detail below can be seen as part of the subject matter of the present disclosure as long as such concepts are not mutually inconsistent. In addition, all combinations of the claimed subject matter are considered as part of the subject matter of the present disclosure.

[0049] The foregoing and other aspects, embodiments and features of the present teachings can be understood and appreciated more fully by referring to the following description in conjunction with the accompanying drawings. Other aspects, embodiments and features of the present teachings will be apparent from the description that follows and from the claims. BRIEF DESCRIPTION OF DRAWINGS

[0050] The accompanying drawings are not intended to be drawn to scale. In the drawings, each identical, or nearly identical, component that is illustrated in various figures is represented with a like numeral. For purposes of clarity, not every component is called out in every drawing. Embodiments of various aspects of the present teachings will now be described, by way of example, with reference to the drawings, in which:

[0051] Figure 1 A schematic diagram of a grid-side topology of a doubly-fed wind turbine according to an example of the present application.

[0052] Figure 2 A schematic diagram of a machine-side topology of a doubly-fed wind turbine.

[0053] Figure 3 A schematic diagram of an RSC adaptive virtual impedance control structure according to an example of the present application.

[0054] Figure 4 A schematic diagram of an RSC adaptive virtual impedance controller amplitude-frequency characteristic according to an example of the present application.

[0055] Figure 5 Structure diagram of GSC adaptive active damping control according to the example of the present application.

[0056] Figure 6 Structure diagram of performance design of second-order Chebyshev low-pass filter in GSC adaptive active damping control according to the example of the present application.

[0057] Figure 7 Impedance characteristic comparison diagram of DFIG sub / super synchronous frequency band before and after introducing virtual impedance control according to the example of the present application.

[0058] Figure 8 Impedance characteristic comparison diagram of DFIG medium frequency band before and after introducing active damping control according to the example of the present application.

[0059] Figure 9 Inhibition effect diagram of sub / super synchronous oscillation caused by series compensation line according to the example of the present application, wherein (a), (b) and (c) respectively represent a-phase current, active power and electromagnetic torque waveform diagram of grid-connected point, FFT analysis result before remodeling and FFT analysis result after remodeling.

[0060] Figure 10 Inhibition effect diagram of sub / super synchronous oscillation caused by weak power grid according to the example of the present application, wherein (a), (b) and (c) respectively represent a-phase current, active power and electromagnetic torque waveform diagram of grid-connected point, FFT analysis result before remodeling and FFT analysis result after remodeling.

[0061] Figure 11 Inhibition effect diagram of medium frequency band oscillation caused by series compensation line grid connection according to the example of the present application, wherein (a), (b) and (c) respectively represent a-phase current, active power and electromagnetic torque waveform diagram of grid-connected point, FFT analysis result before remodeling and FFT analysis result after remodeling.

[0062] Figure 12 On the basis of the example of the present application, adaptive active damping control effect diagram when rated power of control object is different. Figure 11 DETAILED DESCRIPTION

[0063] In order to better understand the technical content of the present application, specific embodiments are described below with reference to the accompanying drawings.

[0064] ​Aspects of the present application are described in the disclosure with reference to the accompanying drawings, in which a number of illustrative embodiments are shown. The described embodiments of the disclosure are not meant to include all aspects of the present application. It should be understood that various concepts and embodiments introduced above and discussed in greater detail below can be implemented in any of numerous ways, as the disclosed concepts and embodiments are not limited to any one implementation. Additionally, some of the aspects of the present application can be used to advantage without combining with other aspects of the present application.

[0065] {Example 1}

[0066] Appendix Figure 1 and Appendix Figure 2 are the grid-side topology and machine-side topology of the doubly-fed wind turbine respectively, which includes the acquisition of the phase-locked angle θ PLL , the coordinate transformation of voltage and current, and the voltage control and current control links.

[0067] The adaptive impedance remodeling based wide frequency oscillation suppression method for doubly-fed wind turbine according to the embodiments of the present application comprises the following steps:

[0068] Step 1, the three-phase currents flowing into the stator port and the rotor winding are sampled by the current sensor and output, respectively, denoted as i sa,sb,sc and i ra,rb,rc ; the grid-side filter inductance current is sampled by the current sensor and output, denoted as i La,Lb,Lc ; the three-phase stator voltage is sampled by the voltage sensor and output, denoted as three-phase stator voltage v sa,sb,sc ;

[0069] Step 2, according to the phase-locked angle θ PLL , the three-phase currents i sa,sb,sc , i ra,rb,rc flowing into the stator port and the rotor winding, the current i sdq and i rdq in the dq rotating coordinate system, the grid-side filter inductance current i Ldq in the dq rotating coordinate system, and the three-phase stator voltage v sa,sb,sc and the output voltage v sdq in the dq rotating coordinate system are calculated by Park transformation;

[0070] Step 3, according to the dq-axis rotor output currents i rd , i rq and the d-axis stator output voltage v sd , the machine-side dq-axis modulation signals m Rd , m Rq are generated by current decoupling control and stator voltage feedforward control;

[0071] Step 4: Based on the dq axis rotor output current i rd i rq After frequency selection and phase correction by the adaptive virtual impedance controller, a modulation correction signal Δm is generated. d1 , △m q1 The signals are respectively superimposed on the dq-axis modulation signals m on the machine side. Rd m Rq In the process, the modified machine-side dq-axis modulation signal is obtained.

[0072] Step 5: Based on the current i flowing into the stator port along the dq axis sd i sq Through current decoupling control, a grid-side dq-axis modulation signal m is generated. Gd m Gq ;

[0073] Step 6: Based on the current i flowing into the stator port along the dq axis Ld i Lq After phase correction and noise isolation by the adaptive active damping controller, a modulation correction signal Δm is generated. d2 , △m q2 The signals are respectively superimposed on the dq-axis modulation signals m on the mesh side. Gd m Gq In the process, the modified mesh-side dq-axis modulation signal is obtained.

[0074] Step 7: Based on the phase-locked angle θ PLL The dq axis modulation signal m on the machine side Rd m Rq and the dq axis modulation signal m on the network side Gd m Gq After inverse Park transformation, the machine-side converter switching signals and grid-side converter switching signals in the three-phase stationary coordinate system are obtained. By changing the converter outputs on the machine side and grid side, impedance reshaping is performed on the DFIG sub / supersynchronous frequency band and intermediate frequency band, respectively.

[0075] Combined with appendix Figure 1 , 2 As shown, the attached Figure 1 The flow into the rotor winding and the attached Figure 2 The three-phase current flowing into the stator port is sampled by the current sensor and output, denoted as i. sa,sb,sc and i ra,rb,rc The current of the grid-side filter inductor is sampled by a current sensor and the output is recorded as i. La,Lb,Lc , will attach Figure 1 The three-phase voltage on the stator side is sampled by a voltage sensor and output as the three-phase stator voltage v. sa,sb,sc .

[0076] In step 2, the phase-locked angle θ PLL The q-axis component of the grid-side voltage is obtained through the phase-locked loop transfer function, and is expressed as:

[0077]

[0078] The phase-locked loop outputs an angle θ PLL The Park transformation matrix for the angular reference is:

[0079]

[0080] In the formula, k pp k pi These represent the proportional coefficient and integral coefficient of the phase-locked loop, respectively. q This represents the q-axis component of the grid-side voltage.

[0081] In a further embodiment, in step 3, based on the dq-axis rotor output current i rd i rq and d-axis stator output voltage v sd Through current decoupling control and stator voltage feedforward control, the machine-side dq-axis modulation signal m is generated. Rd m Rq ,include:

[0082] The machine-side dq-axis modulation signal m is calculated as follows: Rd m Rq :

[0083]

[0084] In the formula, These represent the proportional and integral coefficients K of the d-axis and q-axis of the machine-side current controller, respectively. rd K represents the decoupling gain coefficient. f This represents the voltage feedforward gain coefficient.

[0085]

[0086] In a further embodiment, in step 4, as Figure 3 As shown, the transfer function H of the aforementioned adaptive virtual impedance controller fa (s) is expressed as:

[0087]

[0088] In the formula, H f (s) is a fifth-order Butterworth low-pass filter. ω represents the normalized Laplace operator. c This represents the cutoff angular frequency, taken as 2π × 100 rad / s; kf represents a virtual impedance coefficient;

[0089] Wherein, the five-order Butterworth low-pass filter is selected as the virtual impedance controller, the influence on the medium and high frequency bands can be avoided, and sufficient gain size in the sub / super synchronous frequency band is ensured.

[0090] Since the impedance amplitude of the DFIG in the sub / super synchronous frequency band presents an overall increasing trend as the frequency increases, and the gain of the low-pass filter remains unchanged in the passband range, the low-pass filter needs to be improved, so that the response characteristics in the passband can be adaptively increased with the frequency. Therefore, in the embodiment of the application, the second-order differential element Z sa =-s 2 is constructed as an adaptive impedance, so that the original virtual impedance controller can adaptively increase its gain as the frequency increases.

[0091] For phase correction, the two main control targets of the virtual impedance control are respectively:

[0092] Target 1: control the impedance phase of the DFIG in-90°~90°, and weaken its negative resistance characteristics as much as possible, corresponding to the first and fourth quadrants in the polar coordinate system;

[0093] Target 2: make the impedance characteristics of the DFIG inductive as much as possible, corresponding to the first and second quadrants in the polar coordinate system, and the overlapping area of the two is the optimal adjustment target of Z DFIG .

[0094] Considering that the grid connection stability of the DFIG should be ensured first, in the embodiment of the application, target 1 is set as the primary target of impedance reshaping.

[0095] From the attached Figure 4 It can be seen that, compared with the traditional virtual impedance controller, the adaptive virtual impedance controller proposed in the application has a linear increase in the amplitude-frequency characteristics in the sub / super synchronous frequency band as the frequency increases, and the phase range is kept in-50°~50°, so it can meet the impedance reshaping target 1 and significantly improve the grid connection stability of the DFIG. At the same time, as the virtual impedance coefficient k f increases, the amplitude-frequency characteristics of the adaptive virtual impedance controller increase, and have no effect on the phase-frequency characteristics.

[0096] In a further embodiment, in step 5, according to the stator port current i sd , i sq , the grid-side dq-axis modulation signal m Gd , m Gq is generated through current decoupling control, including:

[0097] The grid-side dq-axis modulation signal mGd m Gq :

[0098]

[0099] In the formula, K represents the proportional and integral coefficients of the d-axis and q-axis of the grid-side current controller, respectively. d This represents the decoupling gain coefficient.

[0100] in,

[0101] In a further embodiment, in step 6, as Figure 5 As shown, the transfer function G of the aforementioned adaptive active damping controller m (s) is expressed as:

[0102]

[0103] Among them, G p (s) is a multi-order lead-lag compensator controller, which serves as the phase compensation controller for the phase correction stage. Its expression is as follows:

[0104]

[0105] In the formula, K p ω represents the compensation gain. h ω l This indicates the upper and lower limits of the adjustable angular frequency, and n represents the controller order;

[0106] G p The expression for the maximum adjustable phase angle (s) is:

[0107]

[0108] In embodiments of the present invention, the upper and lower limits of the adjustment angular frequency are set for the mid-frequency range as follows: ω l =2π×100rad / s, ω h = 2π × 1000 rad / s.

[0109] The higher the controller order, the larger the adjustable phase range. However, excessively high orders also increase the design complexity of the controller, hindering practical implementation. Therefore, in this embodiment, a controller order of n=2 is selected. It is 73°.

[0110] The transfer function design of the adaptive active damping controller in the above embodiments is as follows: Figure 6 As shown, G f(s) represents a second-order Chebyshev low-pass filter. Considering practical operability, it is used as a noise suppression controller in the noise isolation stage, and its expression is as follows:

[0111]

[0112] In the formula, K c s represents the gain coefficient. pi Denotes the filter poles, i = 1, 2, s i =s i +jω i Both must satisfy the following expression:

[0113]

[0114] In the formula, ω cut ε represents the filter cutoff angular frequency, and ε represents the ripple coefficient.

[0115] As an optional example, in step 6, for the noise isolation stage, the filter cutoff angular frequency ω cut The value is 2π×1000rad / s.

[0116] To further determine the value of ε, see Appendix Figure 5 This indicates the effect of the fluctuation coefficient on the actual cutoff frequency f. cut With passband ripple α max The influence of ε, where the actual cutoff frequency is taken as the frequency corresponding to the amplitude -3dB. It can be seen that with ε... 2 The increase of f cut It will continue to decrease, while α max This will continue to increase, therefore there is a balance point between the two. Therefore, in the embodiments of the present invention, ε is selected. 2 =0.24, at which point f cut =1070Hz, α max =1dB.

[0117] In a further embodiment, in step 7, the phase-locked loop outputs an angle θ. PLL The inverse Park transformation equation with the angle reference is as follows:

[0118] m a =m d cosθ PLL -m q sinθ PLL

[0119]

[0120] {Example 2}

[0121] Combining the methods of the above embodiments, since the line series compensation capacitor is an important cause of DFIG subsynchronous / supersynchronous oscillations, the impedance reshaping performance of DFIG in the scenario of grid connection with series-compensated lines is first analyzed. Based on the reshaped impedance model, with a series compensation degree of 40%, the impedance characteristics of DFIG subsynchronous / supersynchronous frequency bands before and after introducing virtual impedance control are shown in the attached figure. Figure 7 As shown, adaptive virtual impedance control can significantly improve the negative damping characteristics of DFIG in the sub / supersynchronous frequency band, enabling it to maintain a phase range of -90° to 90° in most frequency bands.

[0122] The parallel capacitor is one of the important reasons for the oscillation of DFIG in the mid-frequency range. The impedance reshaping effect of DFIG after parallel connection is shown in the attached figure. Figure 8 As shown, adaptive active damping control can improve the negative resistance and capacitive characteristics of the DFIG in the mid-frequency range, and reshape the Z-axis before resonating. DFIG With Z g Taking the intersection at 782Hz as an example, the phase difference at this intersection is 172°, and there is only an 8° phase margin, which poses a significant risk of instability.

[0123] After adaptive active damping control reshaping, its phase difference at 782Hz is reduced to 101°, the phase margin is increased to 81°, and the oscillation risk is greatly reduced.

[0124] We further utilized Matlab / Simulink to perform example simulations, combined with the attached... Figures 9-12 The technical solutions provided by the above-described embodiments of the invention will be further explained.

[0125] Scenario 1: DFIG connected to the grid via series compensation line

[0126] Figure 9 This demonstrates the effectiveness of adaptive virtual impedance control in suppressing subsynchronous / supersynchronous oscillations induced by series compensation lines. After the system is connected to a series compensation capacitor with a series compensation degree of 40% at 5 seconds, the grid-connected current, active power, and electromagnetic torque all exhibit oscillatory instability. At this time, the grid-connected current exhibits a 29Hz subsynchronous oscillation coupled with a 71Hz supersynchronous oscillation. After the adaptive virtual impedance control is applied at 6 seconds, the oscillations of various electrical quantities gradually disappear. At 7 seconds, the series compensation degree becomes 50%, and the system remains stable. This indicates that the frequency oscillation suppression method proposed in this invention can adapt well to different series compensation scenarios and has good adaptability.

[0127] Scenario 2: DFIG connection to a weak current grid

[0128] Figure 10 This indicates the effectiveness of adaptive virtual impedance control in suppressing subsynchronous / supersynchronous oscillations caused by weak power grids. (Previously, DFIG was connected to L...) g= 5.0mH inductive weak grid, when the system exists 25 / 75Hz sub / super synchronous oscillation, the oscillation gradually disappears in 0.2s after the virtual impedance control is put in at 3s, and the system is stable. L g changes to 5.5mH, at this time the system can still maintain stable operation, indicating that the adaptive virtual impedance control proposed in the application also has good adaptability in the weak grid scenario.

[0129] Scenario three: DFIG grid-connected through parallel compensation line

[0130] Figure 11 represents the suppression effect of adaptive active damping control on the intermediate frequency oscillation caused by the grid-connected through parallel compensation line. Before reshaping, the DFIG causes a 116Hz intermediate frequency oscillation due to the grid-connected through parallel compensation line (parallel compensation degree is 40%), and there is a 216Hz oscillation coupling component. After the adaptive active damping control is put in at 2s, the oscillation is suppressed in about 0.2s, and the system is stable. At 3s, the parallel compensation degree changes to 45%, it can be seen that at this time the intermediate frequency oscillation is not caused, and the adaptive active damping control proposed in the application has adaptability to the frequency shift of the intermediate frequency oscillation.

[0131] Scenario four: different control targets

[0132] Figure 12 represents the adaptive active damping control effect when the rated power of the control object is different. On the basis of scenario three, at the initial moment, the DFIG active power has an intermediate frequency oscillation due to the parallel compensation capacitor, at 2s the adaptive active damping controller is put in, and the oscillation phenomenon of the DFIG with three different rated powers is well suppressed. At 3s, the parallel compensation degree changes, at this time the impedance reshaping performance is consistent with the foregoing analysis, indicating that when the control object changes, the frequency oscillation suppression method proposed in the application does not need to be redesigned, and has good adaptability to the control target.

[0133] The contents described in detail in the specification belong to the prior knowledge known to those skilled in the art. Although the application has been disclosed as above with preferred embodiments, it is not intended to limit the application. Those skilled in the art without departing from the spirit and scope of the application can make various changes and modifications. Therefore, the protection scope of the application shall be subject to the scope defined by the claims.

Claims

1. A method for suppressing broadband oscillations in doubly-fed wind turbines based on adaptive impedance reshaping, characterized in that, include: Step 1: Sample the three-phase currents flowing into the stator terminals and the rotor windings using current sensors, and output them as i. sa,sb,sc and i ra,rb,rc The current of the grid-side filter inductor is sampled by a current sensor and the output is recorded as i. La,Lb,Lc The three-phase stator voltage is sampled by a voltage sensor and output, denoted as the three-phase stator voltage v. sa,sb,sc ; Step 2: Based on the phase-locked angle θ PLL The three-phase currents i flowing into the stator ports and the rotor windings are calculated using the Park transformation. sa,sb,sc i ra,rb,rc Current i in the dq rotating coordinate system sdq and i rdq The current i of the grid-side filter inductor in the dq rotating coordinate system Ldq and the three-phase stator voltage v sa,sb,sc Output voltage v in the dq rotating coordinate system sdq ; Step 3: Based on the dq axis rotor output current i rd i rq and d-axis stator output voltage v sd Through current decoupling control and stator voltage feedforward control, the machine-side dq-axis modulation signal m is generated. Rd m Rq ; Step 4: Based on the dq axis rotor output current i rd i rq After frequency selection and phase correction by the adaptive virtual impedance controller, a modulation correction signal Δm is generated. d1 Δm q1 The signals are respectively superimposed on the dq-axis modulation signals m on the machine side. Rd m Rq In the process, the modified machine-side dq-axis modulation signal is obtained. Step 5: Based on the current i flowing into the stator port along the dq axis sd i sq Through current decoupling control, a grid-side dq-axis modulation signal m is generated. Gd m Gq ; Step 6: Based on the current i flowing into the stator port along the dq axis Ld i Lq After phase correction and noise isolation by the adaptive active damping controller, a modulation correction signal Δm is generated. d2 Δm q2 The signals are respectively superimposed on the dq-axis modulation signals m on the mesh side. Gd m Gq In the process, the modified mesh-side dq-axis modulation signal is obtained. Step 7: Based on the phase-locked angle θ PLL The dq axis modulation signal m on the machine side Rd m Rq and the dq axis modulation signal m on the network side Gd m Gq After inverse Park transformation, the machine-side converter switching signals and grid-side converter switching signals in the three-phase stationary coordinate system are obtained. By changing the converter outputs on the machine side and grid side, impedance reshaping is performed on the DFIG sub / supersynchronous frequency band and intermediate frequency band, respectively.

2. The method for suppressing broadband oscillations of a doubly-fed wind turbine based on adaptive impedance reshaping according to claim 1, characterized in that, In step 2, the phase-locked angle θ PLL The q-axis component of the grid-side voltage is obtained through the phase-locked loop transfer function, and is expressed as: In the formula, k pp k pi These represent the proportional coefficient and integral coefficient of the phase-locked loop, respectively. q This represents the q-axis component of the grid-side voltage.

3. The method for suppressing broadband oscillations of doubly-fed wind turbines based on adaptive impedance reshaping according to claim 1, characterized in that, In step 3, based on the dq axis rotor output current i rd i rq and d-axis stator output voltage v sd Through current decoupling control and stator voltage feedforward control, the machine-side dq-axis modulation signal m is generated. Rd m Rq ,include: The machine-side dq-axis modulation signal m is calculated as follows: Rd m Rq : In the formula, These represent the proportional and integral coefficients K of the d-axis and q-axis of the machine-side current controller, respectively. rd K represents the decoupling gain coefficient. f This represents the voltage feedforward gain coefficient.

4. The method for suppressing broadband oscillations of a doubly-fed wind turbine based on adaptive impedance reshaping according to claim 1, characterized in that, In step 4, the transfer function H of the adaptive virtual impedance controller fa (s) is expressed as: In the formula, H f (s) is a fifth-order Butterworth low-pass filter. ω represents the normalized Laplace operator. c This represents the cutoff angular frequency, taken as 2π × 100 rad / s; k f Indicates the virtual impedance coefficient; In this design, a fifth-order Butterworth low-pass filter is selected as the virtual impedance controller, and a second-order differential element Z is constructed. sa =-s 2 As an adaptive impedance, it enables the original virtual impedance controller to adaptively increase its gain as the frequency increases.

5. The method for suppressing broadband oscillations of doubly-fed wind turbines based on adaptive impedance reshaping according to claim 4, characterized in that, The adaptive virtual impedance controller exhibits a linear increase in amplitude-frequency characteristics with increasing frequency in the sub / supersynchronous frequency band, while maintaining a phase range of -50° to 50°, thus meeting the requirements of impedance reshaping objective 1.

6. The method for suppressing broadband oscillations of a doubly-fed wind turbine based on adaptive impedance reshaping according to claim 4, characterized in that, In the adaptive virtual impedance controller, as the virtual impedance coefficient k... f As the amplitude increases, the amplitude-frequency characteristic of the adaptive virtual impedance controller increases accordingly, without affecting the phase-frequency characteristic.

7. The method for suppressing broadband oscillations of doubly-fed wind turbines based on adaptive impedance reshaping according to claim 1, characterized in that, In step 5, based on the current i flowing into the stator port along the dq axis... sd i sq Through current decoupling control, a grid-side dq-axis modulation signal m is generated. Gd m Gq ,include: The network-side dq-axis modulation signal m is calculated as follows: Gd m Gq : In the formula, K represents the proportional and integral coefficients of the d-axis and q-axis of the grid-side current controller, respectively. d This represents the decoupling gain coefficient.

8. The method for suppressing broadband oscillations of a doubly-fed wind turbine based on adaptive impedance reshaping according to claim 1, characterized in that, In step 6, the transfer function G of the adaptive active damping controller m (s) is expressed as: Among them, G p (s) is a multi-order lead-lag compensator controller, which serves as the phase compensation controller for the phase correction stage. Its expression is as follows: In the formula, K p ω represents the compensation gain. h ω l Indicates the upper and lower limits of the adjustable angular frequency, where n represents the controller order; ω c This represents the cutoff angular frequency, taken as 2π×100rad / s; G p The expression for the maximum adjustable phase angle (s) is: Among them, G f (s) represents a second-order Chebyshev low-pass filter, which acts as a noise suppression controller in the noise isolation stage. Its expression is: In the formula, K c s represents the gain coefficient. pi Denotes the filter poles, i = 1, 2, s i =σ i +jω i Both must satisfy the following expression: In the formula, ω cut ε represents the filter cutoff angular frequency, and ε represents the ripple coefficient.

9. The method for suppressing broadband oscillations of a doubly-fed wind turbine based on adaptive impedance reshaping according to claim 8, characterized in that, In step 6, for the phase correction stage, the upper and lower limits of the adjusted angular frequency are respectively set as: ω l =2π×100rad / s, ω h =2π×1000rad / s; the controller order n is set to 2.

10. The method for suppressing broadband oscillations of a doubly-fed wind turbine based on adaptive impedance reshaping according to claim 8, characterized in that, In step 6, for the noise isolation stage, the filter cutoff angular frequency ω cut The value is 2π × 1000 rad / s; ε 2 The value is 0.24.

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