Control method of Si IGBT and SiC MOSFET mixed frequency parallel system

Abstract

According to the control method of the Si IGBT and SiC MOSFET mixed frequency parallel system, when an open-circuit fault occurs in the SiC MOSFET high-frequency unit, the Si IGBT low-frequency unit works independently; under the light load working condition within the safe working range of the device, the SiC MOSFET high-frequency unit works independently; under the normal working condition, the Si IGBT low-frequency unit and the SiC MOSFET high-frequency unit share a common direct-current power supply and work in a coordinated mode, the Si IGBT low-frequency unit processes large power, the SiC MOSFET high-frequency unit compensates switch ripples generated by the Si IGBT low-frequency unit, specifically, the SiC MOSFET high-frequency unit adopts mQPR control and compensates third to twenty-third harmonic waves generated by an LFU, the Si IGBT low-frequency unit adopts SHEPWM control, the third to thirteenth harmonic waves generated by the Si IGBT low-frequency unit are reduced, and the switching ripple waves generated by the Si IGBT low-frequency unit are reduced. Therefore, the capacity requirement on the HFU is reduced.

Description

Technical Field

[0001] The present invention relates to the technical field of converter control, and in particular to a control method for a Si IGBT and SiC MOSFET mixed frequency parallel system. Background Art

[0002] In order to alleviate the contradiction between converter efficiency and power quality and make the output power quality meet the relevant grid-connected standards, existing research has proposed a variety of solutions, including: ① interleaved parallel technology; ② use of a new generation of wide bandgap (WBG) semiconductor components; ③ hybrid switch (Hybrid Switch, HyS) technology and mixed frequency parallel device technology based on the mixed application of traditional silicon (Si)-based insulated gate bipolar transistors (IGBTs) and new silicon carbide (SiC)-based metal-oxide semiconductor field effect transistors (MOSFETs).

[0003] Interleaved parallel technology is a traditional solution to increase the system equivalent switching frequency, reduce output harmonics, and meet the needs of high current output. With the increase in the number of parallel modules, the system composition and control will be greatly improved, and problems such as circulating current, current distribution, and system stability will become more prominent.

[0004] The switching loss of new WBG devices (such as SiC MOSFET) at high frequencies is much lower than that of traditional Si IGBT devices, making it possible to increase the switching frequency. However, at the same rated current, the market price of SiC MOSFET is much higher than that of SiIGBT, and at the same rated voltage, the rated current of SiC MOSFET is much smaller than that of Si IGBT.

[0005] The Si IGBT and SiC MOSFET mixed frequency parallel system is a hybrid switch formed by directly connecting high-current Si IGBT devices and low-current SiCMOSFET devices in parallel. It breaks through the switching frequency limit of Si IGBT and the rated current limit of SiC MOSFET and reduces the cost, providing a feasible idea for high-quality, high-efficiency and large-capacity power conversion. Existing research has applied the hybrid parallel idea to many types of converters such as bidirectional DC-DC converters, PFC, and single-phase inverters. In terms of system efficiency, compared with full SiC converters, various types of hybrid parallel converters have achieved basically the same efficiency and significantly reduced costs. In terms of key parameter design, the power distribution ratio of high / low frequency units and the selection of switching frequency are crucial to maximize the advantages of the mixed frequency parallel system. In terms of control strategy, the main control goal of the mixed frequency parallel system is to use high-frequency units to compensate for the low-frequency current ripple generated by low-frequency units, thereby increasing the frequency of the output current ripple and reducing the size and weight of the filter. However, at present, the mixing parallel technology is mostly used in simple topologies such as chopper circuits or single-phase inverter circuits. The mixing parallel of three-phase inverters is still rare, and the corresponding system design scheme is immature, with problems such as complex control methods and poor system output performance. Summary of the invention

[0006] The object of the present invention is to provide a control method for a Si IGBT and SiC MOSFET mixed frequency parallel system, which is used to control a mixed frequency parallel system composed of a Si IGBT three-phase inverter and a SiC MOSFET three-phase inverter. The control method can effectively take into account the efficiency and cost of the mixed frequency parallel system, maximize the advantages of power devices, and reduce the volume of passive filtering components while improving the dynamic performance and power density of the system, increase the control bandwidth of the system, reduce the control complexity of the converter, and improve the output performance of the mixed frequency system.

[0007] In order to solve the above technical problems, the present invention adopts the following technical methods: a control method for a Si IGBT and SiC MOSFET mixed frequency parallel system, when an open circuit fault occurs in the SiC MOSFET high frequency unit, the Si IGBT low frequency unit works alone, and under light load conditions within the safe working range of the device, the SiC MOSFET high frequency unit works alone;

[0008] Under other working conditions, the Si IGBT low-frequency unit and the SiC MOSFET high-frequency unit share a common DC power supply and work in coordination. The Si IGBT low-frequency unit handles high power, and the SiC MOSFET high-frequency unit compensates for the switching ripple generated by the Si IGBT low-frequency unit.

[0009] The Si IGBT low frequency unit and SiC MOSFET high frequency unit work in coordination, including:

[0010] Step S1: Adjust the active power output value P and reactive power output value Q of the mixed frequency parallel system to active power reference values ​​P and Q respectively. ref and reactive power reference value Q ref , and then derive the total output current reference value i of the system gref ;

[0011] Step S2, tracking the Si IGBT low frequency unit current sampling value i L With reference value i Lref The error between them generates the modulation signal u * L , the modulation signal u * L After SHEPWM processing, a switching signal for controlling the action of the Si IGBT low-frequency unit IGBT is obtained;

[0012] Step S3, using the mQPR controller to sample the current value i of the SiC MOSFET high frequency unit H Regulates to track the current reference value i Href , generate the modulation signal u * H , the modulation signal u * H After PWM processing, a switching signal is obtained to control the operation of the SiC MOSFET high-frequency unit MOSFET.

[0013] Further, in step S1, the active power output value P and the reactive power output value Q of the mixed frequency parallel system are adjusted to the active power reference value P respectively by using PQ control, or droop control, or VSG control. ref and reactive power reference value Q ref .

[0014] Furthermore, in step S2, the modulation signal u * L When performing SHEPWM processing:

[0015] First, calculate the Si IGBT low-frequency unit modulation signal amplitude M and modulation signal initial phase angle Δθ according to formula (1):

[0016]

[0017] In the formula, u * Ld 、u * Lq They are the d-axis and q-axis modulation signals of Si IGBT low-frequency unit respectively;

[0018] Then, based on the obtained M and Δθ, calculate the Si IGBT low-frequency unit a-phase modulation signal u * La , b-phase modulation signal u * Lb , c-phase modulation signal u * Lc , the calculation formula is as follows:

[0019]

[0020] Where, ω is the grid-connected voltage angular frequency;

[0021] Finally, according to the obtained a-phase modulation signal u * La , b-phase modulation signal u * Lb , c-phase modulation signal u * Lc , calculate the switching angles of the A-phase, B-phase, and C-phase bridge arms of the SiIGBT low-frequency unit to control the on and off of the two IGBTs in the A-phase, B-phase, and C-phase bridge arms respectively.

[0022] Furthermore, in step S2, the five switch angles α1, α2, α3, α4, and α5 of the first 1 / 4 cycle of the output phase voltage of the Si IGBT low-frequency unit bridge arm are selected as reference variables, and the Gaussian fitting method is used to fit the trajectory of the five switch angles changing with the modulation signal to obtain:

[0023]

[0024] Where: m is the Si IGBT low-frequency unit modulation signal; α k is the kth switching angle among the five switching angles;

[0025] When k = 1:

[0026] When n=1, a1=21.3, b1=-0.3641, c1=1.244;

[0027] When n=2, a2=0, b2=0.2745, c2=0.0003583;

[0028] When n=3, a3=3.937, b3=0.9583, c3=0.6273;

[0029] When k = 2:

[0030] When n=1, a1=3.193, b1=1.144, c1=0.2337;

[0031] When n=2, a2=8.636, b2=0.989, c2=0.5303;

[0032] When n=3, a3=20, b3=0.1601, c3=1.217;

[0033] When k = 3:

[0034] When n=1, a1=40.41, b1=-0.2299, c1=1.235;

[0035] When n=2, a2=11.7, b2=0.9276, c2=0.5762;

[0036] When n=3, a3=3.576, b3=1.141, c3=0.2539;

[0037] When k = 4:

[0038] When n=1, a1=5.435, b1=1.108, c1=0.1701;

[0039] When n=2, a2=16.44, b2=1.023, c2=0.4966;

[0040] When n=3, a3=40.48, b3=0.1932, c3=1.245;

[0041] When k = 5:

[0042] When n=1, a1=59.54, b1=-0.09161, c1=1.263;

[0043] When n=2, a2=17.91, b2=0.9769, c2=0.5182;

[0044] When n=3, a3=5.673, b3=1.105, c3=0.1828;

[0045] The a-phase modulation signal u is obtained by combining * La , b-phase modulation signal u * Lb , c-phase modulation signal u * Lc And formula (3) can calculate the switching angles of the A-phase, B-phase, and C-phase bridge arms of the Si IGBT low-frequency unit, thereby controlling the on and off of the two IGBTs of the A-phase, B-phase, and C-phase bridge arms respectively.

[0046] Furthermore, in step S3, the mQPR controller uses the following formulas (4)-(6) to calculate the SiC MOSFET high frequency unit a phase modulation signal u * Ha , b-phase modulation signal u* Hb , c-phase modulation signal u * Hc ;

[0047]

[0048] G mQPR (s)=H1(s)+H2(s)(5)

[0049]

[0050] In the formula, i Href(a) 、i Href(b) 、i Href(c) are the current reference values ​​of phase A, phase B and phase C of SiC MOSFET high frequency unit respectively; i H(a) 、i H(b) 、i H(c) They are the current sampling values ​​of phase A, phase B, and phase C of the SiC MOSFET high-frequency unit; K Hp K is the proportional coefficient of the fundamental wave controller; HR1 and K HRn are the resonance coefficients of the fundamental wave and the nth harmonic respectively; ω1 and ω n are the angular frequencies of the fundamental wave and the nth harmonic, ξ1 and ξ n are the correlation coefficients between the corresponding frequency cutoff frequency and the resonant frequency.

[0051] Furthermore, in step S3, the a-phase modulation signal u is processed using an SPWM or SVPWM strategy. * Ha , b-phase modulation signal u * Hb , c-phase modulation signal u * Hc , and obtain the switching signal that controls the action of the SiC MOSFET high-frequency unit MOSFET.

[0052] Preferably, in step S2, the Si IGBT low frequency unit d-axis modulation signal u * Ld and q-axis modulation signal u * Lq The generation process is as follows:

[0053] 1) Calculate the Si IGBT low-frequency unit current reference value i according to formula (7) Lref ;

[0054] i Lref =Ki gref (7)

[0055] Where K is the ratio of Si IGBT low-frequency unit power to full load power;

[0056] 2) According to formula (8), the current reference value i is obtained Lref Perform abc / dq changes;

[0057]

[0058] Where, ω is the grid-connected voltage angular frequency; i * Ld 、i * Lq are the d-axis and q-axis current reference values ​​of Si IGBT low-frequency unit respectively; i Lref (a) i Lref (b) i Lref (c) are the current reference values ​​of phase a, phase b, and phase c of Si IGBT low-frequency unit respectively;

[0059] 3) Calculate the d-axis modulation signal u of the Si IGBT low-frequency unit according to formula (9): * Ld and q-axis modulation signal u * Lq ;

[0060]

[0061] In the formula, K pc is the proportional coefficient of the current loop PI controller, K ic is the integral coefficient of the current loop PI controller; s is the differential operator; ω0 is the power frequency angular frequency, ω0=100π; L L It is the filter inductor of Si IGBT low frequency unit; Ld 、i Lq They are the d-axis and q-axis current sampling values ​​of the SiIGBT low-frequency unit; u Cd 、u Cq They are the system d-axis and q-axis capacitor voltage sampling values ​​respectively.

[0062] Preferably, in step S3, the SiC MOSFET high frequency unit current reference value i Href The following method is used to determine:

[0063] 1) To track the total output current reference value i of the system gref And meet the requirements of compensating the switching ripple generated by the Si IGBT low-frequency unit, determine the SiC MOSFET high-frequency unit current reference value i Href It is expressed as:

[0064] i Href =i gref -i L(10)

[0065] 2) Use the system discrete model to construct the predicted value Δi of Si IGBT low-frequency unit current change Lp The calculation formula is as follows:

[0066]

[0067] In the formula, x represents phase a, phase b, or phase c; Δi L is the current change of Si IGBT low frequency unit; k represents the time; S Lx is the main unit switching function, indicating the state of the Si IGBT low-frequency unit switch, S Lx ∈{0,1}; U dc is the common DC bus voltage; u Cx is the system capacitor voltage sampling value; u LxN is the voltage between the midpoint of the Si IGBT low-frequency unit bridge arm and the midpoint of the load; T SH is the switching cycle of the SiC MOSFET high frequency unit;

[0068] 3) Determine the Si IGBT low-frequency unit current i at time k+1 Lp The expression of is:

[0069] i Lp =i L (k+1)=i L (k)+Δi Lp (12)

[0070] 4) Combining equations (10), (11), and (12), we get equation (13) to calculate the SiC MOSFET high-frequency unit current reference value i Href ;

[0071] i Href =i gref -(i L +Δi Lp )(13).

[0072] The control method of the Si IGBT and SiC MOSFET mixed frequency parallel system provided by the present invention is used to control the mixed frequency parallel system composed of the SiIGBT three-phase inverter and the SiC MOSFET three-phase inverter. In the control method, the LFU and the HFU can work separately and coordinately. When working in coordination, the LFU processes high power at an extremely low switching frequency and uses the SHEPWM technology to reduce the harmonics generated by itself to generate the optimal output voltage waveform, reduce the current ripple, and improve the control performance. The HFU uses mQPR control to compensate for the switching ripple generated by the LFU to further improve the system output performance. In summary, the control method can not only effectively take into account the efficiency and cost of the mixed frequency parallel system, maximize the advantages of the power devices of the two, but also reduce the volume of passive filter components, increase the control bandwidth of the system, reduce the control complexity of the converter, and improve the output performance of the mixed frequency system while improving the dynamic performance and power density of the system. BRIEF DESCRIPTION OF THE DRAWINGS

[0073] Figure 1 It is a topological diagram of the mixed frequency parallel system of Si IGBT and SiC MOSFET in the present invention;

[0074] Figure 2 This is a closed-loop control architecture diagram of the Si IGBT and SiC MOSFET mixed frequency parallel system in the present invention;

[0075] Figure 3 Schematic diagram of the A-phase operating current waveform of the mixed frequency parallel system of Si IGBT and SiC MOSFET in the present invention ( Figure 3 In the figure, (a) is the A phase current waveform; (b) is the S phase current waveform in (a). La =1; (c) is the waveform of the amplified current when S in (a) La =0 when the amplified current waveform);

[0076] Figure 4 Schematic diagram of power path of Si IGBT and SiC MOSFET mixed frequency parallel system in the present invention;

[0077] Figure 5 A schematic diagram of harmonic elimination in a control method for a mixed frequency parallel system of Si IGBT and SiC MOSFET according to the present invention;

[0078] Figure 6 It is a schematic diagram of the SHEPWM principle in the control method of the Si IGBT and SiC MOSFET mixed frequency parallel system involved in the present invention;

[0079] Figure 7A schematic diagram of the trajectories of the five switching angles changing with the modulation signal in the control method of the Si IGBT and SiC MOSFET mixed frequency parallel system involved in the present invention;

[0080] Figure 8 This is a frequency response comparison diagram of commonly used controllers in the present invention;

[0081] Fig. 9 Schematic diagram of simulation waveform of mixing system in the embodiment of the present invention ( Fig. 9 In the figure, (a) is the output current waveform of the mixing system; (b) is the comparison diagram of the THD content of the output current of the mixing system). DETAILED DESCRIPTION

[0082] In order to facilitate the understanding of those skilled in the art, the present invention is further described below in conjunction with embodiments and drawings. The contents mentioned in the implementation modes are not intended to limit the present invention.

[0083] The Si IGBT and SiC MOSFET mixed frequency parallel system consists of a large-capacity three-phase Si IGBT low-frequency unit (LFU) and a small-capacity three-phase SiC MOSFET high-frequency unit (HFU) in parallel. The system topology is as follows: Figure 1 As shown in the figure, L L and L H are high and low frequency unit filter inductors, i L and i H is the current on the corresponding inductor, that is, the high and low frequency unit current; L g and C f Forming the grid-side LC filter, u C is the system capacitor voltage, i C is the system capacitance current; i g is the grid-connected current, u g is the grid voltage; the common DC bus voltage is U dc ; N' is the imaginary midpoint of the DC power supply, and N is the midpoint of the load.

[0084] The parallel LFU and HFU can operate independently or in coordination. Therefore, the mixed frequency parallel system provided by the present invention has three operation modes:

[0085] 1) LFU mode: When an open circuit failure occurs in the HFU, the LFU works alone to provide fault tolerance.

[0086] 2) HFU mode: Under light load conditions within the safe operating range of the device, the HFU can work alone to improve efficiency.

[0087] 3) Coordination mode: Under other working conditions, LFU and HFU share a common DC power supply and work in coordination. LFU processes high power at a very low switching frequency, and HFU compensates for the switching ripple generated by LFU to achieve a trade-off between efficiency and power quality.

[0088] The working principles of LFU mode and HFU mode are the same as those of traditional three-phase bridge converters. The working principle of coordination mode will be described in detail below.

[0089] Different from the traditional interleaved parallel converter, the hybrid frequency parallel system is composed of two converter units with different frequencies and different power handling capabilities. Therefore, the classic interleaved phase shift control is no longer applicable. To this end, the present invention proposes a new control strategy to achieve coordinated control of LFU and HFU. The working principle of the coordinated control is as follows:

[0090] The present invention introduces the switch function S Lx and S Hx (x=a,b,c,S Lx ,S Hx ∈{0,1}) to describe the low-frequency / high-frequency unit switching state. For example, if S La =1, then Figure 1 S La1 Shutdown, S La2 Open, and vice versa. Take phase A as an example to analyze, Figure 3 The A phase working current waveform is shown, and the working conditions of phases B and C are similar. Ignoring the capacitor current, the load current i ga The LFU output current i La and HFU output current i Ha Superposition composition:

[0091] i ga =i La +i Ha (14)

[0092] LFU output current i La By fundamental wave La(1) Harmonics La(h) The control goal of the inverter is to make the output current i ga With reference value i ref Therefore, the HFU output voltage i Ha Must meet:

[0093] i H =(i ref -i L(1) )-i L(h) (15)

[0094] The present invention introduces a basic power ratio K to describe the ratio of LFU current to total load current, i.e.La(1) =Ki ref , then (2) can be further expressed as:

[0095]

[0096] Figure 3 The blue and green lines show the La and i Ha Schematic waveform of Figure 3 From formula (16), it can be seen that HFU has two main functions: compensating LFU harmonics and processing part of the fundamental current. The comparison of LFU and HFU characteristics is shown in Table 1.

[0097] Table 1 Comparison of high and low frequency unit characteristics

[0098]

[0099]

[0100] The system power path is as follows Figure 4 As shown. Due to the use of extremely low switching frequency, the switching loss P of the low-frequency unit sw-L The total loss is almost only composed of the conduction loss of the power device and the diode P on-L At the same time, the extremely low switching frequency will generate more low-order harmonic currents, which will be compensated by the high-frequency unit. Therefore, the high- and low-frequency units have harmonic reactive power and harmonic currents of equal magnitude and opposite direction. Since the high-frequency unit has excellent switching characteristics and only processes very small power, the overall system efficiency will not be significantly reduced even at ultra-high switching frequencies. The high-frequency unit also processes a small amount of fundamental load power to accurately adjust the output current and improve control accuracy.

[0101] Regarding the steady state of the system, we take phase A as an example for analysis, and phases B and C are similar. Assume that the switching cycles of HFU and LFU are T SH and T SL In a T SH The load current variation ΔI ga is the LFU current change ΔI La and HFU current change ΔI Ha The sum is:

[0102] ΔI ga =ΔI La +ΔI Ha (17)

[0103] The following derivation of ΔI La and ΔI Ha The midpoint of the A-phase bridge arm of LFU is L aThe voltage between the imaginary midpoint N' of the DC power supply and the midpoint N of the load can be expressed as:

[0104]

[0105] In the formula, u NN’ The voltage between the load midpoint N and the DC power supply imaginary midpoint N' is. Similar to equations (18) and (19), the voltage between the midpoint of the bridge arm of phases B and C and N' and N can be obtained. Further, combined with (18) and (19), u NN’ It can be expressed as:

[0106]

[0107] make:

[0108]

[0109] Then (20) can be simplified as:

[0110]

[0111] Due to T SH Much lower than T SL , so in T SH In the LFU switch function S Lx (x=a,b,c) is considered as a constant (1 or 0), and then u LxN is also a constant. According to Kirchhoff’s voltage law, T SH The current change of the internal low-frequency unit is:

[0112]

[0113] Substituting (18), (19), and (22) into (23), we obtain:

[0114]

[0115] Similar to LFU, the midpoint of the A-phase bridge arm of HFU a The voltage between the imaginary midpoint N' of the DC power supply and the midpoint N of the load can be expressed as:

[0116]

[0117] T SH Internal HFU current i H Change ΔI H for:

[0118]

[0119] Where D SHa Represents S Ha2Substituting (22), (25), and (26) into (27), we can obtain:

[0120]

[0121] From (24) and (28), it can be seen that by modifying the HFU duty cycle D SH , that is, to adjust the load current to track the reference value and obtain a high-quality waveform. In order to ensure that each T SH The internal HFU current can fully compensate the LFU current ripple, ΔI L and ΔI H Must meet:

[0122] ΔI L +ΔI H =0(29)

[0123] Equation (29) shows that at each T SH The HFU current can fully compensate the LFU harmonics. This shows that the output current of the mixed frequency parallel system does not contain LFU harmonics, but only HFU harmonics. Therefore, the output current waveform quality is determined only by the HFU switching frequency and has nothing to do with the LFU switching frequency. This means that only a small part of the input power needs to be processed at high frequency, and the harmonics and waveform quality of the total current can obtain the same high switching frequency advantages as the full WBG converter without significantly increasing losses and costs. This feature provides a feasible idea for the trade-off between converter cost and performance.

[0124] Further, substituting (24) and (28) into (29), the theoretical duty cycle of HFU can be obtained as:

[0125]

[0126] Based on the above analysis, the focus of the present invention is how to suppress system harmonics. Specifically, system harmonics are suppressed by the following three methods: Figure 5As shown: ① HFU adopts multiple quasi proportional resonance (Multiple Quasi Proportional Resonances, mQPR) control to compensate for the 3rd to 23rd harmonics generated by LFU; ② LFU adopts specific harmonic elimination (Selective Harmonic Elimination, SHE) PWM control to reduce the 3rd to 13th harmonics generated by itself, so as to reduce the capacity demand for HFU; ③ The grid-side LC filter filters out the 23rd to 256th harmonics generated by LFU and HFU (this harmonic elimination method is a well-known technology in the industry, so it will not be repeated here). In a three-phase three-wire system under normal grid conditions, the harmonics of multiples of 3 in the output line voltage of the inverter bridge arm can be automatically offset, so the present invention only focuses on 6k±1 (k=1,2,3,...) harmonics (it should be noted that in a three-phase symmetrical system of a three-phase three-wire system, even harmonics and harmonics of integer multiples of three are not included).

[0127] Overall, the control method of the present invention consists of three parts: an instruction generation loop, an LFU current control loop and an HFU current control loop. Figure 2 :

[0128] Part 1: Instruction Generation Loop

[0129] There are many optional schemes for the command generation loop, such as PQ control, droop control, VSG control, etc. This embodiment takes PQ control as an example, and uses a proportional integral (PI) controller to adjust the active power output value P and reactive power output value Q of the mixed frequency parallel system to the active power reference value P respectively. ref and reactive power reference value Q ref , and then derive the total output current reference value i of the system gref .

[0130] Part 2: LFU Current Control Loop

[0131] The LFU current control is carried out in the dq coordinate system, and the PI controller is used to track the Si IGBT low-frequency unit current sampling value i L With reference value i Lref The error between them generates the modulation signal u * L , the modulation signal u * L After SHEPWM processing, the switching function S is obtained. L , control the IGBT action.

[0132] In this second part, SHEPWM control is used to reduce the 3rd to 13th harmonics generated by LHF itself, so as to reduce the capacity demand of HFU. SHEPWM technology is a PWM strategy with application prospects in high-power low switching frequency converters. This strategy uses the switching angle as a reference variable and selectively eliminates certain specific harmonics by calculating and reasonably selecting the switching moment position. SHEPWM technology can generate the optimal output voltage waveform at a lower switching frequency, thereby reducing current harmonics and improving control performance.

[0133] like Figure 6 The figure shows the voltage waveform of phase A output by the LFU bridge arm. The voltages of phases B and C lag behind phase A by 120° and 240° respectively.

[0134] from Figure 6 It can be seen that the voltage waveform of phase A output by the LFU bridge arm is an even function symmetrical about π / 2 and 3π / 2, and an odd function symmetrical about π. Due to the above symmetric relationship, the bridge arm switching angle only needs to be solved for the first 1 / 4 cycle, that is, α1, α2…α N There are N controllable switch angles in total. Since the phase voltage signal satisfies the Dirichlet condition, it can be expressed in Fourier series form:

[0135]

[0136] in:

[0137]

[0138] Wherein, ω′ is the fundamental angular frequency of the phase voltage signal, ω′=100π.

[0139] Due to the odd symmetry and 1 / 4 period symmetry of the phase voltage signal, the coefficients of the DC component, cosine component and even sine component in the Fourier decomposition are zero, that is:

[0140]

[0141] The odd-order sinusoidal component coefficients can be expressed as:

[0142]

[0143] Where n is the order of the fundamental wave and each harmonic, α k is the kth switching angle among the N switching angles in the interval [0,π / 2], and the size of each switching angle should satisfy the condition: 0°≤α1<α2…<α N ≤90°.

[0144] When there are N controllable switch angles, (N-1) harmonics can be eliminated through the SHEPWM strategy. To eliminate the 5th, 7th, 11th, and 13th harmonics, let there be 5 switch angles in 1 / 4 period (here, it is worth mentioning that by substituting the five switch angles into the Fourier function, the coefficients corresponding to the 5th, 7th, 11th, and 13th harmonics are calculated to be 0. Therefore, these harmonics can be suppressed), the following SHEPWM harmonic elimination equations can be written:

[0145]

[0146] In the formula, m is the modulation signal. By solving (35), the switching action time can be obtained. Figure 7 The obtained switching angle trajectory is shown as a function of the modulation signal m.

[0147] In the traditional implementation method of SHEPWM, it is usually necessary to store the obtained pulse switch angle value in a digital controller. In actual operation, the pulse angle corresponding to a certain modulation index m is found through an online table lookup to obtain the pulse waveform. This method requires the storage of a large number of switch angles, which is not easy to use flexibly, occupies a large amount of memory, and increases hardware costs. To this end, the present invention uses a Gaussian fitting method to fit the trajectory of the switch angle changing with the modulation signal. The general expression of n-dimensional Gaussian fitting is:

[0148]

[0149] Where: m is the Si IGBT low-frequency unit modulation signal; α k is the kth switching angle among the five switching angles.

[0150] Take n=3 (here n=3 is taken as an example. The value of n depends on the specific situation. The larger n is, the more accurate the fitting effect is; the smaller n is, the simpler the fitted function expression is), fit α1~α5, and the fitting curve is shown in Figure 7 , the fitting coefficients are shown in Table 2.

[0151] Table 2 Fitting parameters of the switching angle trajectory changing with the modulation signal

[0152] <![CDATA[α1(m)]]> <![CDATA[α2(m)]]> <![CDATA[α3(m)]]> <![CDATA[α4(m)]]> <![CDATA[α5(m)]]> <![CDATA[a1]]> 21.3 3.191 40.41 5.435 59.54 <![CDATA[a2]]> 0 8.636 11.7 16.44 17.91 <![CDATA[a3]]> 3.937 20 3.576 40.48 5.673 <![CDATA[b1]]> -0.3641 1.144 -0.2299 1.108 -0.09161 <![CDATA[b2]]> 0.2745 0.989 0.9276 1.023 0.9769 <![CDATA[b3]]> 0.9583 0.1601 1.141 0.1932 1.105 <![CDATA[c1]]> 1.244 0.2337 1.235 0.1701 1.263 <![CDATA[c2]]> 0.0003583 0.5303 0.5762 0.4966 0.5182 <![CDATA[c3]]> 0.6273 1.217 0.2539 1.245 0.1828

[0153] Specifically, the steps of the second part LFU current control loop are as follows:

[0154] 1) LFU handles most of the load power. Assuming the ratio of LFU power to full load power is K, the LFU current reference value i Lref for:

[0155] i Lref =Ki gref (7)

[0156] 2) According to formula (8), the current reference value i is obtained Lref Perform abc / dq changes;

[0157]

[0158] Where, ω is the grid-connected voltage angular frequency; i * Ld 、i * Lq are the d-axis and q-axis current reference values ​​of Si IGBT low-frequency unit respectively; i Lref (a) i Lref (b) i Lref (c) are the current reference values ​​of phase a, phase b, and phase c of Si IGBT low-frequency unit respectively;

[0159] 3) Calculate the d-axis modulation signal u of the Si IGBT low-frequency unit according to formula (9): * Ld and q-axis modulation signal u * L q ;

[0160]

[0161] In the formula, K pc is the proportional coefficient of the current loop PI controller, K ic is the integral coefficient of the current loop PI controller; s is the differential operator; ω0 is the power frequency angular frequency, ω0=100π; L L It is the filter inductor of Si IGBT low frequency unit; Ld 、i Lq They are the d-axis and q-axis current sampling values ​​of the SiIGBT low-frequency unit; u Cd 、u Cq They are the system d-axis and q-axis capacitor voltage sampling values ​​respectively.

[0162] 4) Calculate the modulation signal amplitude M and the modulation signal initial phase angle Δθ of the Si IGBT low-frequency unit according to formula (1):

[0163]

[0164] In the formula, u * Ld 、u * Lq They are the d-axis and q-axis modulation signals of Si IGBT low-frequency unit respectively;

[0165] Secondly, based on the obtained M and Δθ, calculate the Si IGBT low-frequency unit a-phase modulation signal u * La , b-phase modulation signal u* Lb , c-phase modulation signal u * Lc , the calculation formula is as follows:

[0166]

[0167] Where, ω is the grid-connected voltage angular frequency;

[0168] 5) Calculate the switching angle

[0169] The a-phase modulation signal u is obtained by combining * La And formula (3), calculate the switching angle of the A-phase bridge arm of the Si IGBT low-frequency unit, and control the on and off of the two IGBTs in the A-phase bridge arm.

[0170] Combined with the obtained b-phase modulation signal u * Lb And formula (3), calculate the switching angle of the B phase and bridge arm of the Si IGBT low-frequency unit, and control the on and off of the two IGBTs in the B phase bridge arm.

[0171] Combined with the obtained c-phase modulation signal u * Lc And formula (3), calculate the switching angle of the C-phase bridge arm of the Si IGBT low-frequency unit, and control the on and off of the two IGBTs in the C-phase bridge arm.

[0172] Part III: HFU Current Control Loop

[0173] The current sampling value i of the SiC MOSFET high frequency unit is measured using the mQPR controller H Regulates to track the current reference value i Href , generate the modulation signal u * H , the modulation signal u * H After PWM processing, a switching signal is obtained to control the operation of the SiC MOSFET high-frequency unit MOSFET.

[0174] In the third part, multiple quasi-proportional resonant mQPR control is used to compensate the 3rd to 23rd harmonics generated by LFU. Href is a complex AC signal with multiple frequencies superimposed. Therefore, the HFU current i is controlled by the mQPR controller in the abc coordinate system. H Regulate to track i Href The mQPR controller can make the specified frequency signal reach high gain, thereby realizing small error control. The mQPR consists of the fundamental wave controller H1 and the harmonic compensator H2, namely:

[0175] G mQPR (s)=H1(s)+H2(s)(5)

[0176] in:

[0177]

[0178] In the formula, K Hp K is the proportional coefficient of the fundamental wave controller; HR1 and K HRn are the resonance coefficients of the fundamental wave and the nth harmonic respectively; ω1 and ω n are the angular frequencies of the fundamental wave and the nth harmonic, ξ1 and ξ n are the correlation coefficients between the corresponding frequency cutoff frequency and the resonant frequency.

[0179] Take n = 5, 7, 11, 13, 17, 19, 23, Figure 8 The frequency domain characteristics of common controllers such as PI controller, PR controller, QPR controller and mQPR controller are shown. As can be seen from the figure, the mQPR controller has a large gain at the fundamental frequency and the selected harmonic frequency, so it can track complex signals with multiple frequencies superimposed, which is suitable for the scenario of the present invention.

[0180] Specifically, the steps of the third part HFU current control loop are as follows:

[0181] 1) To accurately track the total output current reference value i of the system gref And meet the requirements of compensating the switching ripple generated by the Si IGBT low-frequency unit, the SiC MOSFET high-frequency unit current reference value i Href It is expressed as:

[0182] i Href =i gref -i L (10)

[0183] In order to achieve accurate load current error tracking, the model predictive control idea is introduced to predict the LFU current ripple, that is, the sampling data at time k is used to predict the system state at time (k+1), so as to achieve control adjustment one cycle in advance. L In each high frequency switching cycle T SH can be regarded as a constant value, so the LFU current change prediction value Δi can be easily predicted using the system discrete model Lp As follows:

[0184]

[0185] In the formula, x represents phase a, phase b, or phase c; Δi Lis the current change of Si IGBT low frequency unit; k represents the time; S Lx is the main unit switching function, indicating the state of the Si IGBT low-frequency unit switch, S Lx ∈{0,1}; U dc is the common DC bus voltage; u Cx is the system capacitor voltage sampling value; u LxN is the voltage between the midpoint of the Si IGBT low-frequency unit bridge arm and the midpoint of the load; T SH is the switching cycle of the SiC MOSFET high frequency unit;

[0186] 3) Determine the Si IGBT low-frequency unit current i at time k+1 Lp The expression of is:

[0187] i Lp =i L (k+1)=i L (k)+Δi Lp (12)

[0188] 4) The predicted change value of the LFU current is incorporated into the HFU tracking error one cycle in advance, which not only achieves accurate tracking of the total current error, but also can compensate the LFU current ripple in real time. Combining equations (10), (11), and (12), we get equation (13) to calculate the SiC MOSFET high-frequency unit current reference value i Href ;

[0189] i Href =i gref -(i L +Δi Lp )(13).

[0190] 5) The mQPR controller uses formulas (4)-(6) to calculate the SiC MOSFET high-frequency unit a-phase modulation signal u * Ha , b-phase modulation signal u * Hb , c-phase modulation signal u * Hc ;

[0191]

[0192] In the formula, i Href(a) 、i Href(b) 、i Href(c) are the current reference values ​​of phase A, phase B and phase C of SiC MOSFET high frequency unit respectively; i H(a) 、i H(b) 、i H(c)They are the current sampling values ​​of phase A, phase B, and phase C of the SiC MOSFET high-frequency unit respectively.

[0193] 6) Use SPWM or SVPWM strategy to process the a-phase modulation signal u * Ha , b-phase modulation signal u * Hb , c-phase modulation signal u * Hc , and obtain the switching signal that controls the action of the SiC MOSFET high-frequency unit MOSFET.

[0194] To verify the effectiveness of the present invention, the control method of the present invention is used to simulate the mixing system. The simulation waveform is as follows: Fig. 9 The simulation parameters are shown in Table 3.

[0195] Initially, only the main unit operates independently. At t=0.5s, the high-frequency unit is connected, and the two units operate jointly. Before the auxiliary unit is connected, the total output current waveform of the grid-connected converter is distorted and the harmonic content is high. After the high-frequency unit is connected, the output current THD (abbreviation of total harmonic distortion in English, which means total harmonic distortion in Chinese, is used to reflect the quality of the output current waveform. After using the mixing system, THD is reduced, indicating that the waveform quality is improved) content is reduced from 6.02% to 2.30%. Preliminary simulation verification and comparative analysis show that the mixing system can achieve high efficiency and low cost effectively, maximize the advantages of power devices, and reduce the volume of passive filter components while improving the dynamic performance and power density of the system, increase the control bandwidth of the system, reduce the control complexity of the converter, and improve the output performance of the mixing system.

[0196] Table 3 Mixing system parameters

[0197] symbol meaning Numeric symbol meaning Numeric <![CDATA[V dc ]]> DC side voltage 1500V <![CDATA[L g ]]> Grid inductance 2mH <![CDATA[L f,L ]]> Low frequency unit filter 3.5mH <![CDATA[R g ]]> Grid resistance 0.5Ω <![CDATA[L f,H ]]> High frequency unit filter 1mH <![CDATA[C f ]]> Filter capacitor 15μF <![CDATA[f s,L ]]> Main unit switching frequency 1kHz <![CDATA[f s,H ]]> Auxiliary unit switching frequency 35kHz <![CDATA[i qref ]]> Q-axis rated current 0A <![CDATA[i gdref ]]> D-axis rated current 500A K Power allocation factor 0.75 <![CDATA[R d ]]> Damping resistor 5.5Ω

[0198] The above embodiments are preferred implementation schemes of the present invention. In addition, the present invention may also be implemented in other ways. Any obvious replacement without departing from the concept of the present technical solution is within the protection scope of the present invention.

[0199] In order to make it easier for ordinary technicians in the field to understand the improvements of the present invention over the prior art, some drawings and descriptions of the present invention have been simplified, and for the sake of clarity, some other elements are omitted in this application document. Ordinary technicians in the field should realize that these omitted elements may also constitute the content of the present invention.

Claims

1. A control method for a mixed frequency parallel system of Si IGBT and SiC MOSFET, wherein when an open circuit fault occurs in the high frequency unit of SiC MOSFET, the low frequency unit of Si IGBT works alone, and under light load conditions within the safe working range of the device, the high frequency unit of SiC MOSFET works alone, characterized in that: Under other working conditions, the Si IGBT low-frequency unit and the SiC MOSFET high-frequency unit share a common DC power supply and work in coordination. The Si IGBT low-frequency unit handles high power, and the SiC MOSFET high-frequency unit compensates for the switching ripple generated by the Si IGBT low-frequency unit. The Si IGBT low frequency unit and SiC MOSFET high frequency unit work in coordination, including: Step S1: Adjust the active power output value P and reactive power output value Q of the mixed frequency parallel system to active power reference values ​​P and Q respectively. ref and reactive power reference value Q ref , and then derive the total output current reference value i of the system gref ; Step S2, tracking the Si IGBT low frequency unit current sampling value i L With reference value i Lref The error between them generates the modulation signal u * L , the modulation signal u * L After SHEPWM processing, a switching signal for controlling the action of the Si IGBT low-frequency unit IGBT is obtained; Step S3, using the mQPR controller to sample the current value i of the SiC MOSFET high frequency unit H Regulates to track the current reference value i Href , generate the modulation signal u * H , the modulation signal u * H After PWM processing, a switching signal is obtained to control the operation of the SiC MOSFET high-frequency unit MOSFET.

2. The control method of the Si IGBT and SiC MOSFET mixed frequency parallel system according to claim 1, characterized in that: In step S1, the active power output value P and the reactive power output value Q of the mixed frequency parallel system are adjusted to the active power reference value P respectively by using PQ control, or droop control, or VSG control. ref and reactive power reference value Q ref .

3. The control method of the Si IGBT and SiC MOSFET mixed frequency parallel system according to claim 1 or 2, characterized in that: In step S2, the modulation signal u * L When performing SHEPWM processing: First, calculate the Si IGBT low-frequency unit modulation signal amplitude M and modulation signal initial phase angle Δθ according to formula (1): In the formula, u * Ld 、u * Lq They are the d-axis and q-axis modulation signals of Si IGBT low-frequency unit respectively; Then, based on the obtained M and Δθ, calculate the Si IGBT low-frequency unit a-phase modulation signal u * La , b-phase modulation signal u * Lb , c-phase modulation signal u * Lc , the calculation formula is as follows: Where, ω is the grid-connected voltage angular frequency; Finally, according to the obtained a-phase modulation signal u * La , b-phase modulation signal u * Lb , c-phase modulation signal u * Lc , calculate the switching angles of the A-phase, B-phase, and C-phase bridge arms of the Si IGBT low-frequency unit to control the on and off of the two IGBTs in the A-phase, B-phase, and C-phase bridge arms respectively.

4. The control method of the Si IGBT and SiC MOSFET mixed frequency parallel system according to claim 3, characterized in that: In step S2, the five switch angles α1, α2, α3, α4, and α5 of the first 1 / 4 cycle of the output phase voltage of the Si IGBT low-frequency unit bridge arm are selected as reference variables, and the Gaussian fitting method is used to fit the trajectory of the five switch angles changing with the modulation signal to obtain: Where: m is the Si IGBT low-frequency unit modulation signal; α k is the kth switching angle among the five switching angles; When k = 1: When n=1, a1=21.3, b1=-0.3641, c1=1.244; When n=2, a2=0, b2=0.2745, c2=0.0003583; When n=3, a3=3.937, b3=0.9583, c3=0.6273; When k = 2: When n=1, a1=3.193, b1=1.144, c1=0.2337; When n=2, a2=8.636, b2=0.989, c2=0.5303; When n=3, a3=20, b3=0.1601, c3=1.217; When k = 3: When n=1, a1=40.41, b1=-0.2299, c1=1.235; When n=2, a2=11.7, b2=0.9276, c2=0.5762; When n=3, a3=3.576, b3=1.141, c3=0.2539; When k = 4: When n=1, a1=5.435, b1=1.108, c1=0.1701; When n=2, a2=16.44, b2=1.023, c2=0.4966; When n=3, a3=40.48, b3=0.1932, c3=1.245; When k = 5: When n=1, a1=59.54, b1=-0.09161, c1=1.263; When n=2, a2=17.91, b2=0.9769, c2=0.5182; When n=3, a3=5.673, b3=1.105, c3=0.1828; The a-phase modulation signal u is obtained by combining * La , b-phase modulation signal u * Lb , c-phase modulation signal u * Lc And formula (3) can calculate the switching angles of the A-phase, B-phase, and C-phase bridge arms of the Si IGBT low-frequency unit, thereby controlling the on and off of the two IGBTs of the A-phase, B-phase, and C-phase bridge arms respectively.

5. The control method of the Si IGBT and SiC MOSFET mixed frequency parallel system according to claim 4, characterized in that: In step S3, the mQPR controller uses the following formulas (4)-(6) to calculate the SiC MOSFET high-frequency unit a-phase modulation signal u * Ha , b-phase modulation signal u * Hb , c-phase modulation signal u * Hc ; G mQPR (s)=H1(s)+H2(s) (5) In the formula, i Href(a) 、i Href(b) 、i Href(c) are the current reference values ​​of phase A, phase B and phase C of SiC MOSFET high frequency unit respectively; i H(a) 、i H(b) 、i H(c) They are the current sampling values ​​of phase A, phase B, and phase C of the SiC MOSFET high-frequency unit; K Hp K is the proportional coefficient of the fundamental wave controller; HR1 and K HRn are the resonance coefficients of the fundamental wave and the nth harmonic respectively; ω1 and ω n are the angular frequencies of the fundamental wave and the nth harmonic, ξ1 and ξ n are the correlation coefficients between the corresponding frequency cutoff frequency and the resonant frequency.

6. The control method of the Si IGBT and SiC MOSFET mixed frequency parallel system according to claim 2 or 5, characterized in that: In step S3, the a-phase modulation signal u is processed using the SPWM or SVPWM strategy. * Ha , b-phase modulation signal u * Hb , c-phase modulation signal u * Hc , and obtain the switching signal that controls the action of the SiC MOSFET high-frequency unit MOSFET.

7. The control method of the Si IGBT and SiC MOSFET mixed frequency parallel system according to claim 6, characterized in that: In step S2, the Si IGBT low frequency unit d-axis modulation signal u * Ld and q-axis modulation signal u * Lq The generation process is as follows: 1) Calculate the Si IGBT low-frequency unit current reference value i according to formula (7) Lref ; in Lref =To gref (7) Where K is the ratio of Si IGBT low-frequency unit power to full load power; 2) According to formula (8), the current reference value i is obtained Lref Perform abc / dq changes; Where, ω is the grid-connected voltage angular frequency; i * Ld 、i * Lq are the d-axis and q-axis current reference values ​​of Si IGBT low-frequency unit respectively; i Lref (a) i Lref (b) i Lref (c) are the current reference values ​​of phase a, phase b, and phase c of Si IGBT low-frequency unit respectively; 3) Calculate the d-axis modulation signal u of the Si IGBT low-frequency unit according to formula (9): * Ld and q-axis modulation signal u * L q ; In the formula, K pc is the proportional coefficient of the current loop PI controller, K ic is the integral coefficient of the current loop PI controller; s is the differential operator; ω0 is the power frequency angular frequency, ω0=100π; L L It is the filter inductor of Si IGBT low frequency unit; Ld 、i Lq They are the d-axis and q-axis current sampling values ​​of Si IGBT low-frequency unit respectively; u Cd 、u Cq They are the system d-axis and q-axis capacitor voltage sampling values ​​respectively.

8. The control method of the Si IGBT and SiC MOSFET mixed frequency parallel system according to claim 7, characterized in that: In step S3, the SiC MOSFET high frequency unit current reference value i Href The following method is used to determine: 1) To track the total output current reference value i of the system gref And meet the requirements of compensating the switching ripple generated by the Si IGBT low-frequency unit, determine the SiC MOSFET high-frequency unit current reference value i Href It is expressed as: i Href =i gref -i L (10) 2) Use the system discrete model to construct the predicted value Δi of Si IGBT low-frequency unit current change Lp The calculation formula is as follows: In the formula, x represents phase a, phase b, or phase c; Δi L is the change in the low-frequency unit current of Si IGBT; k represents the time; S Lx is the main unit switching function, indicating the state of the Si IGBT low-frequency unit switch, S Lx ∈{0,1}; U dc is the common DC bus voltage; u Cx is the system capacitor voltage sampling value; u LxN is the voltage between the midpoint of the Si IGBT low-frequency unit bridge arm and the midpoint of the load; T SH is the switching cycle of the SiCMOSFET high frequency unit; 3) Determine the Si IGBT low-frequency unit current i at time k+1 Lp The expression of is: i Lp =i L (k+1)=i L (k)+Δi Lp (12) 4) Combining equations (10), (11), and (12), we get equation (13) to calculate the SiC MOSFET high-frequency unit current reference value i Href ; i Href =i gref -(i L +Δi Lp ) (13)。

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