Three-phase four-leg inverter control method based on improved virtual oscillator

Abstract

The invention relates to the technical field of inverter control, in particular to a three-phase four-leg inverter control method based on an improved virtual oscillator. According to the method, a sequence-dividing control strategy is adopted, a positive sequence is controlled by a virtual oscillator control strategy improved based on voltage deviation compensation and frequency deviation compensation, a negative sequence is suppressed by virtual impedance combined with an MPR controller, and proportional resonance control is introduced to control a fourth bridge arm of the TPGL-VSI. The strategy can adjust the effective value and frequency of the voltage, so that the effective value and frequency can quickly recover to nominal values, thereby improving the stability of the system voltage. According to the method for controlling the fourth bridge arm in cooperation with the three-phase VOC, the compatibility of a VOC strategy in a three-phase four-wire system topological structure is improved.

Description

Technical Field

[0001] The present invention relates to the technical field of inverter control, and in particular to a voltage asymmetry suppression strategy for a three-phase four-bridge-arm inverter based on independent control of negative-sequence components and zero-sequence components, and an improved virtual oscillator control method based on secondary voltage and frequency deviation compensation. Background Art

[0002] Microgrid systems based on distributed renewable energy have gradually become an important means to improve energy utilization and supplement large power grids with their low pollution, high flexibility and plug-and-play characteristics. In low-voltage microgrids, three-phase asymmetric loads are common, which will cause the three-phase imbalance of the microgrid support voltage, and even affect the stable operation of the system in severe cases. In actual situations, a neutral line is often added to the three-phase circuit to keep the inverter at zero potential at the neutral point to achieve stable load operation. The three-phase four-wire inverter topologies developed from this include split capacitor type and three-phase four-leg type. The three-phase four-leg voltage source inverter (TPFL-VSI) is a topology structure obtained by adding a fourth bridge arm to the traditional three-phase three-wire voltage source inverter. By controlling the neutral point voltage, independent three-phase voltages can be obtained. This structure has the advantages of high voltage utilization and small inverter system size.

[0003] In recent years, a new inverter control strategy, virtual oscillator control (VOC), has been proposed and developed rapidly. VOC can achieve stable operation and phase synchronization without interconnection lines in both island and grid-connected operation modes of distributed generation systems. Studying the improved algorithm of VOC under complex working conditions such as unbalanced loads has become a hot research direction in this field. However, the virtual oscillator control VOC based on nonlinear Andronov-Hopf is mainly used in single-phase or three-phase three-wire power systems. For the TPFL-VSI topology, the VOC strategy is difficult to use directly, and the VOC strategy needs to be appropriately improved. Summary of the invention

[0004] The technical problem to be solved by the present invention is to provide a three-phase four-bridge-arm inverter control method based on an improved virtual oscillator to suppress voltage asymmetry and compensate for voltage and frequency deviations, so as to achieve safe and stable operation of the inverter system under unbalanced working conditions.

[0005] In order to solve the technical problem, the technical solution adopted by the present invention is: a three-phase four-bridge-arm inverter control method based on an improved virtual oscillator, comprising the following steps: S01), current sampling is performed on the three-phase four-bridge-arm inverter, and the three-phase current sampled is After Clark transformation, it enters the current positive and negative sequence component extraction module based on the second-order generalized integrator. The extracted current positive sequence component is controlled by a virtual oscillator to obtain a positive sequence reference voltage. The extracted current negative sequence component is passed through a virtual impedance module to generate a compensation voltage. The compensation voltage is superimposed on the positive sequence reference voltage and enters the voltage and current dual closed-loop controller after Clark inverse transformation to obtain the fundamental wave signal for controlling the first three bridge arms. ; S02), sampling the voltage of the three-phase four-bridge-arm inverter, and the sampled three-phase voltage Enter the zero-sequence voltage sequence control module to obtain the fundamental signal for controlling the fourth bridge arm ; S03), signal , Enter 3D-SVPWM to generate the modulation wave required to control the inverter switches.

[0006] Furthermore, the state equation of the virtual oscillator is: , in , represents the output of the virtual oscillator, , Respectively , The first derivative with respect to time, To adjust the state variable , The rate constant for convergence to the steady state, is the voltage proportionality coefficient, is the inverter nominal voltage RMS amplitude, Is and The column vector composed of is the Euclidean norm of the vector, is the fundamental frequency, is the resonant capacitor, , By formula Get, where is the phase angle, is the current proportionality coefficient, , The reference current The α-axis component and the β-axis component of , The positive sequence current components are The α-axis component and the β-axis component.

[0007] Furthermore, the voltage amplitude V and phase angle θ output by the virtual oscillator are passed through the voltage compensator and the frequency compensator to obtain the voltage deviation compensation value and frequency deviation compensation , voltage deviation compensation , Frequency deviation compensation Respectively with the active power setting value , reactive power setting value The difference is obtained by converting the matrix to obtain the reference current. ; Voltage deviation compensation , in , and are the voltage deviation compensation proportional coefficient and integral coefficient respectively, is the deviation between the reference voltage amplitude and the sampling voltage amplitude, is the inverter nominal voltage RMS amplitude, is the resonant capacitor, To adjust the state variable , The rate constant for convergence to the steady state, is the current proportionality coefficient, is the voltage proportionality coefficient; Frequency deviation compensation , , in and are the proportional and integral coefficients of the frequency compensator, is the reference apparent power, is the fundamental frequency, is the frequency sampling value, m is the gain coefficient, is the RMS amplitude of the inverter nominal voltage.

[0008] Further, m = 300, = 800.

[0009] Furthermore, the state equation of the virtual impedance module is: ; in, is the compensation voltage output by the virtual impedance module, is the fundamental compensation voltage, is the hth harmonic compensation voltage, , They are Axis and The fundamental component of the positive sequence current of the shaft, is the virtual negative impedance, is the virtual negative inductive reactance, h is the harmonic order, , They are Axis and The positive sequence current harmonic component of the shaft, is the fundamental frequency.

[0010] Furthermore, the voltage loop in the voltage-current dual closed-loop controller adopts an MPR controller. The MPR control adopts five resonant links in parallel to eliminate the resonant current at the resonant point and suppress the 3rd, 5th, 7th and 9th harmonic components.

[0011] Furthermore, the transfer function of the voltage and current dual closed-loop controller is: , in, is the transfer function of the multi-module proportional resonant controller, is the proportional controller transfer function, is the proportional coefficient of the multi-resonance controller, h is the harmonic order, is the cut-off frequency, is the integral coefficient at different harmonic orders, is the fundamental frequency, is the current loop proportionality coefficient.

[0012] Furthermore, the fundamental signal of the fourth bridge arm is controlled That is, the state equation of the zero-sequence voltage sequence control module is: , in, , is the zero axis scale factor, is the zero axis resonance coefficient, is the value of line filter inductance, is the fourth bridge arm current sampling signal, is the line impedance value, , , are the three-phase voltages, is the cut-off frequency, is the fundamental frequency.

[0013] Furthermore, The value is 0.4. The value is 100.

[0014] Furthermore, the current positive and negative sequence component extraction module includes a frequency-locked loop, a plurality of dual second-order generalized integrators, and a positive and negative separation control module equal to the number of dual second-order generalized integrators. The frequency-locked loop samples the output of the dual second-order generalized integrator to obtain the system angular frequency, and the system angular frequency is amplified according to the harmonic order and sent to the dual second-order generalized integrator. The positive and negative separation control modules are respectively connected after the dual second-order generalized integrator, and the inverter output current in the stationary coordinate system The phase difference at different frequencies is obtained by multiple biquad generalized integrators. The quadrature current signal and , and The positive-sequence and negative-sequence current components are obtained through the positive-negative separation control module.

[0015] Beneficial effects of the present invention: The present invention studies the three-phase voltage asymmetry and voltage deviation problems during the operation of the TPFL-VSI system under unbalanced load. A split-sequence control strategy is proposed, in which the positive sequence is controlled by a virtual oscillator control strategy based on voltage deviation compensation and frequency deviation compensation; the negative sequence is suppressed by a virtual impedance combined with an MPR controller; and proportional resonance control is introduced to control the fourth bridge arm of the TPGL-VSI. This strategy can adjust the effective value of the voltage and the frequency so that they can quickly return to the nominal value, thereby improving the stability of the system voltage. The method of the fourth bridge arm in conjunction with the three-phase VOC control proposed in the present invention improves the compatibility of the VOC strategy in the three-phase four-wire topology. BRIEF DESCRIPTION OF THE DRAWINGS

[0016] Figure 1 It is the overall principle block diagram of the present invention; Figure 2 It is the principle schematic diagram of the current positive and negative sequence component extraction module; Figure 3 It is a schematic diagram of the improved VOC principle based on voltage and frequency compensator; Figure 4 It is a schematic diagram of the voltage compensator principle when the line impedance is resistive; Figure 5 It is the principle diagram of the frequency compensator when the line impedance is resistive; Figure 6 It is the principle schematic diagram of the negative sequence virtual impedance module; Figure 7 It is a schematic diagram of the principle of a voltage-current dual closed-loop controller with an MPR controller; Figure 8 It is a schematic diagram of the main circuit topology of a three-phase four-bridge-arm voltage source inverter; Fig. 9 It is a principle schematic diagram of the zero-sequence voltage sequencing control module; Fig.10 Schematic diagram of the system frequency response curve when different values ​​of the gain coefficient m are selected; Fig.11 is the integration coefficient Schematic diagram of the system frequency response curve when different values ​​are selected. DETAILED DESCRIPTION

[0017] The present invention is further described below in conjunction with the accompanying drawings and specific embodiments. It should be understood that the specific implementation method described herein is only a preferred embodiment of the present invention, which is only used to explain the present invention and does not limit the protection scope of the present invention. All other embodiments obtained by ordinary technicians in this field without creative work are within the protection scope of the present invention.

[0018] Example 1 This embodiment discloses a three-phase four-bridge-arm inverter control method based on an improved virtual oscillator. This method is oriented to a three-phase four-bridge-arm voltage source inverter and adopts a sequence control strategy. The first three bridge arms of the inverter adopt a positive sequence VOC and negative sequence virtual impedance control strategy, and the fourth bridge arm is designed with a zero-sequence independent control loop, so as to suppress voltage imbalance and maintain voltage and frequency stability. Figure 1 As shown, the following steps are included: S01), current sampling is performed on the three-phase four-bridge-arm inverter, and the three-phase current sampled is After Clark transformation, the current positive and negative sequence component extraction module based on the second-order generalized integrator is entered to extract the current positive sequence component Positive sequence reference voltage obtained by virtual oscillator control , the extracted current negative sequence component Generate compensation voltage through virtual impedance module , compensation voltage With positive sequence reference voltage Superposition , after Clark inverse transformation, it enters the voltage and current double closed-loop controller to obtain the fundamental wave signal of the first three bridge arms ; S02), sampling the voltage of the three-phase four-bridge-arm inverter, and the sampled three-phase voltage Enter the zero-sequence voltage sequence control module to obtain the fundamental signal for controlling the fourth bridge arm ; S03), signal , Enter 3D-SVPWM to generate the modulation wave required to control the inverter switches.

[0019] Considering that the output voltage and current of the inverter can be decomposed into positive sequence, negative sequence and zero sequence in the case of three-phase imbalance, in order to improve the accuracy of the sequence control, it is necessary to obtain an accurate sequence signal. In this embodiment, a current positive and negative sequence component extraction module is used to extract the current positive and negative sequence components. The current positive and negative sequence component extraction module extracts the current positive and negative sequence components based on the positive and negative sequence separation structure of the double second-order generalized integrator, such as Figure 2 As shown, it includes a frequency lock loop (FLL), multiple double second-order generalized integrators (DSOGI) and positive and negative separation control modules (PNSC) equal to the number of double second-order generalized integrators. The frequency lock loop samples the output of the double second-order generalized integrator to obtain the system angular frequency. The system angular frequency is amplified according to the harmonic order and sent to the double second-order generalized integrator. The positive and negative separation control module is connected after the double second-order generalized integrator. The inverter output current in the stationary coordinate system The phase difference at different frequencies is obtained by multiple biquad generalized integrators. The quadrature current signal and , and The positive-sequence and negative-sequence current components are obtained through the positive-negative separation control module. is the SOGI gain coefficient, which is 0.707 and is input into the biquad generalized integrator; e is the error signal between the input and output of DSOGI, which is one of the outputs of the biquad generalized integrator; h Indicates the harmonic order. Select the harmonic order according to the influence of the harmonics on the system. .

[0020] In a three-phase three-wire inverter system, the virtual oscillator VOC can ensure that the voltage and frequency of each phase follow the droop relationship and achieve stable control. Therefore, it is feasible to use the virtual oscillator VOC to obtain the reference signal of the positive sequence voltage. In view of the problem of inaccurate compensation of voltage and frequency deviation under non-ideal line impedance, this embodiment improves the existing virtual oscillator VOC and designs a voltage and frequency compensator. By improving the droop relationship of VOC, the voltage and frequency regulation accuracy is improved.

[0021] like Figure 3 As shown, the improved VOC based on the voltage and frequency compensator in this embodiment is composed of an oscillating inductor and oscillating capacitor The resonant cavity is formed, and the fundamental frequency is , is the current flowing through the oscillating inductor, is the voltage across the oscillating capacitor, and the state variable of the circuit is and , and They are the current proportional coefficient, voltage proportional coefficient, and active power setting value. and reactive power setting value Get the reference current through the conversion matrix , the positive sequence current obtained by separation and reference current The input signal of the resonant circuit is obtained by subtracting and Since the influence of low-order odd harmonics on the system is dominant, and the influence of high-order harmonics on the system can be ignored, the current signal here is composed of the superposition of the positive sequence fundamental signal and the positive sequence low-order harmonic signals. The spatial state equation of the resonant circuit is: , in , represents the output of the virtual oscillator, , Respectively , The first derivative with respect to time, To adjust the state variable , The rate constant for convergence to the steady state, is the voltage proportionality coefficient, is the inverter nominal voltage RMS amplitude, Is and The column vector composed of is the Euclidean norm of the vector, is the fundamental frequency, , By formula get.

[0022] The output voltage signal obtained by the positive sequence current passing through VOC and As the reference value of the positive sequence voltage, .

[0023] and are nonlinear voltage source and nonlinear current source respectively, and the expressions are: , in, Voltage Amplitude and phase angle The dynamic equation is: , in is the voltage amplitude First-order derivative, is the phase angle The first derivative of is the active power sampling value, is the reactive power setting value, is the active power sampling value, It is the active power setting value.

[0024] The line impedance of the low-voltage power system is usually resistive. , and The droop relationship can be simplified to: , make and , when the line impedance is resistive, the VP droop relationship of VOC steady state is obtained as follows: .

[0025] When the line impedance is resistive, the VP at this time is nonlinear. Therefore, the relationship between active power and voltage is used to design the secondary voltage deviation compensation term, and the formula is as follows: , in, , and They are the voltage deviation compensation proportional coefficient and integral coefficient respectively. The structure of the voltage compensator is as follows: Figure 4 shown.

[0026] In the case of resistive line impedance, the frequency deviation compensation formula for reactive power correction is: In the formula, and are the proportional and integral coefficients of the frequency compensator, and VOC output voltage Axis and Axis component. The frequency compensator structure is as follows Figure 5 shown.

[0027] Proportional coefficient Defined as: , In the formula, m is the gain coefficient, is the system apparent power rating.

[0028] Taking the gain coefficient and integral coefficient design in the frequency response curve under resistive line impedance as an example, by observing Fig.10 and 11 , it can be seen that the selection of gain coefficient and integral coefficient requires a trade-off between frequency regulation speed and overshoot. In order to ensure that a good system response speed can be obtained within a reasonable parameter range, m = 300 is selected. = 800.

[0029] The positive and negative sequence separation structure includes the negative sequence current fundamental wave and low-order odd harmonic signals, and the positive sequence signal does not consider the influence of high-order harmonics. The detailed structure block diagram of the negative sequence virtual impedance module is as follows: Figure 6 As shown, is the virtual negative impedance, is the virtual negative inductive reactance. The voltage compensation amount generated after the introduction of negative sequence virtual impedance can be expressed as: , in, is the compensation voltage output by the virtual impedance module, is the fundamental compensation voltage, is the hth harmonic compensation voltage, , They are Axis and The fundamental component of the positive sequence current of the shaft, is the virtual negative impedance, is the virtual negative inductive reactance, h is the harmonic order, , They are Axis and The positive sequence current harmonic components of the shaft. , and are the oscillating inductance and oscillating capacitance respectively, and h is the harmonic order.

[0030] Considering the impedance of the low-voltage line is resistive and connected to a linear load, the negative-sequence harmonic component in the system has little effect on the voltage imbalance, so the fundamental negative-sequence component is mainly suppressed. .

[0031] In order to further suppress the ripple current at the resonant frequency, a multi-module PR (MPR) controller is added to the current closed loop according to the inner membrane principle. Figure 3 , Figure 6 The output voltage is superimposed to obtain the reference voltage , and input into the voltage and current double closed-loop control module. The voltage and current double closed-loop control block diagram is as follows Figure 7 As shown in Figure 1, the voltage and current dual-loop control structure is already very mature in this field and will not be described here. and current loop controller The transfer function is as follows: , In the formula, is the transfer function of the multi-module proportional resonant controller, is the proportional controller transfer function, is the cut-off frequency, h is the harmonic number, is the proportional coefficient of the multi-resonance controller, and its value is 0.5; is the integral coefficient under different harmonic orders, and its value is 75; is the current loop proportional coefficient, the value is 30; is the cut-off frequency, which mainly affects the bandwidth of the multi-resonance controller at the resonant frequency. , select .

[0032] Zero-sequence voltage is one of the reasons for three-phase imbalance. To solve this problem, this paper proposes a neutral point voltage control strategy. Figure 8 The TPFL-VSI main circuit topology shown in the figure can be obtained from Kirchhoff's voltage law: , For the same inverter, the filter inductance and line impedance can be considered equal, and the values ​​are as follows: , When the three-phase voltage is symmetrical, the voltage at the neutral point is 0, and we can get: .

[0033] in , , and are the output voltages of the four-leg inverter, , , and are the output currents of the four-leg inverter, , , and They are three-phase filter inductance and zero-sequence filter inductance respectively. , and are the load output voltage, is the voltage at point O, , , , are the line impedances of the four bridge arms respectively.

[0034] After transformation, the fourth bridge arm control voltage signal can be recorded as: , in, , is the zero axis scale factor, is the zero axis resonance coefficient, is the value of the line filter inductor, is the fourth bridge arm current sampling signal, is the line impedance value, , , are the three-phase voltages, is the cut-off frequency. Further, The value is 0.4. The value is 100.

[0035] The control signal of the fourth bridge arm can be obtained from the above two formulas. The principle block diagram of the zero-sequence voltage sequence control module is as follows: Fig. 9 shown.

[0036] Regarding the s in the above description, it represents the independent variable in the s domain. The transfer function adopts the expression in the s domain, so the independent variable is s, which is equivalent to t in the time domain. Those skilled in the art can clearly know the meaning of s based on the expression of the transfer function, so there is no explanation in the claims.

[0037] The above description is only the basic principle and preferred embodiments of the present invention. Improvements and substitutions made by those skilled in the art based on the present invention belong to the protection scope of the present invention.

Claims

1. A three-phase four-bridge-arm inverter control method based on an improved virtual oscillator, characterized in that: The following steps are involved: S01), current sampling is performed on the three-phase four-bridge-arm inverter, and the three-phase current sampled is After Clark transformation, it enters the current positive and negative sequence component extraction module based on the second-order generalized integrator. The extracted current positive sequence component is controlled by a virtual oscillator to obtain a positive sequence reference voltage. The extracted current negative sequence component is passed through a virtual impedance module to generate a compensation voltage. The compensation voltage is superimposed on the positive sequence reference voltage and enters the voltage and current dual closed-loop controller after Clark inverse transformation to obtain the fundamental wave signal for controlling the first three bridge arms. ; S02), sampling the voltage of the three-phase four-bridge-arm inverter, and the sampled three-phase voltage Enter the zero-sequence voltage sequence control module to obtain the fundamental signal for controlling the fourth bridge arm ; S03), signal , Enter 3D-SVPWM to generate the modulation wave required to control the inverter switches.

2. The three-phase four-bridge-arm inverter control method based on the improved virtual oscillator according to claim 1 is characterized in that: The state equation of the virtual oscillator is: ， in , represents the output of the virtual oscillator, , Respectively , The first derivative with respect to time, To adjust the state variable , The rate constant for convergence to the steady state, is the voltage proportionality coefficient, is the inverter nominal voltage RMS amplitude, Is and The column vector composed of is the Euclidean norm of the vector, is the fundamental frequency, is the resonant capacitor, , By formula Get, where is the phase angle, is the current proportionality coefficient, , The reference current The α-axis component and the β-axis component of , The positive sequence current components are The α-axis component and the β-axis component.

3. The three-phase four-bridge-arm inverter control method based on the improved virtual oscillator according to claim 2 is characterized in that: The voltage amplitude V and phase angle θ output by the virtual oscillator are passed through the voltage compensator and frequency compensator to obtain the voltage deviation compensation value and frequency deviation compensation , voltage deviation compensation , Frequency deviation compensation Respectively with the active power setting value , reactive power setting value The difference is obtained by converting the matrix to obtain the reference current. ; Voltage deviation compensation , in , and are the voltage deviation compensation proportional coefficient and integral coefficient respectively, is the deviation between the reference voltage amplitude and the sampling voltage amplitude, is the inverter nominal voltage RMS amplitude, is the resonant capacitor, To adjust the state variable , The rate constant for convergence to the steady state, is the current proportionality coefficient, is the voltage proportionality coefficient; Frequency deviation compensation , ， in and are the proportional and integral coefficients of the frequency compensator, is the reference apparent power, is the fundamental frequency, is the frequency sampling value, m is the gain coefficient, is the inverter nominal voltage RMS amplitude, is the resonant capacitor.

4. The three-phase four-bridge-arm inverter control method based on the improved virtual oscillator according to claim 3 is characterized in that: m = 300， = 800。 5. The three-phase four-bridge-arm inverter control method based on the improved virtual oscillator according to claim 1 is characterized in that: The state equation of the virtual impedance module is: ； in, is the compensation voltage output by the virtual impedance module, is the fundamental compensation voltage, is the hth harmonic compensation voltage, , They are Axis and The fundamental component of the positive sequence current of the shaft, is the virtual negative impedance, is the virtual negative inductive reactance, h is the harmonic order, , They are Axis and The positive sequence current harmonic component of the shaft, is the fundamental frequency.

6. The three-phase four-bridge-arm inverter control method based on the improved virtual oscillator according to claim 1 is characterized in that: The voltage loop in the voltage-current dual closed-loop controller adopts an MPR controller. The MPR control uses five resonant links in parallel to eliminate the resonant current at the resonance point and suppress the 3rd, 5th, 7th, and 9th harmonic components.

7. The three-phase four-bridge-arm inverter control method based on the improved virtual oscillator according to claim 6 is characterized in that: The transfer function of the voltage and current dual closed-loop controller is: ， in, is the transfer function of the multi-module proportional resonant controller, is the proportional controller transfer function, is the proportional coefficient of the multi-resonance controller, h is the harmonic order, is the cut-off frequency, is the integral coefficient at different harmonic orders, is the fundamental frequency, is the current loop proportionality coefficient.

8. The three-phase four-bridge-arm inverter control method based on the improved virtual oscillator according to claim 1, characterized in that: Control the fundamental signal of the fourth bridge arm That is, the state equation of the zero-sequence voltage sequence control module is: ， in , is the zero axis scale factor, is the zero axis resonance coefficient, is the value of the line filter inductor, is the fourth bridge arm current sampling signal, is the line impedance value, , , are the three-phase voltages, is the cut-off frequency, is the fundamental frequency.

9. The three-phase four-bridge-arm inverter control method based on the improved virtual oscillator according to claim 8, characterized in that: The value is 0.

4. The value is 100.

10. The three-phase four-bridge-arm inverter control method based on improved virtual oscillator according to claim 1, characterized in that: The current positive and negative sequence component extraction module includes a frequency-locked loop, multiple dual second-order generalized integrators, and positive and negative separation control modules equal to the number of dual second-order generalized integrators. The frequency-locked loop samples the output of the dual second-order generalized integrator to obtain the system angular frequency, which is amplified according to the harmonic order and sent to the dual second-order generalized integrator. The positive and negative separation control modules are respectively connected after the dual second-order generalized integrator. The inverter output current in the stationary coordinate system The phase difference at different frequencies is obtained by multiple biquad generalized integrators. The quadrature current signal and , and The positive-sequence and negative-sequence current components are obtained through the positive-negative separation control module.

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