Radiation-receiving type low-profile dual-beam array antenna and folding decoupling design method

Abstract

The invention discloses a radiation-receiving type low-profile dual-beam array antenna and a folding decoupling design method. The low-profile dual-beam array antenna is composed of a transmission metasurface, a polarization conversion metasurface and a horn feed source. The transmission metasurface is formed by periodically extending and arranging m * m radiation-receiving units in a plane at equal intervals, and efficient transmission and spin decoupling focusing phase regulation and control of electromagnetic waves of a specific frequency band are achieved. The radiation-receiving unit is composed of three layers of metal and two layers of dielectric plates. The polarization conversion metasurface is formed by periodically extending and arranging n * n polarization conversion units in a plane at equal intervals, and efficient reflection, polarization conversion and wave beam non-mirror image deflection phase regulation and control of electromagnetic waves of a specific frequency band are achieved. The polarization conversion unit is composed of two layers of metal and a layer of dielectric plate. The low-profile dual-beam array antenna provided by the invention can generate two independently regulated orthogonal circularly polarized waves through spin decoupling under the excitation of a circularly polarized wave feed source, and has the remarkable advantages of high aperture efficiency, low profile and the like.

Description

Technical Field

[0001] The present invention relates to the technical field of array antenna design, and in particular to a radiation-receiving low-profile dual-beam array antenna and a folding decoupling design method. Background Art

[0002] In recent years, in order to realize highly integrated transmission array antennas, folded transmission array antennas have become a research hotspot. The folded transmission array antenna is implemented based on the path tracing principle. It introduces a top metasurface with polarization selectivity, a bottom metasurface with reflectivity and polarization conversion, and a feed located at the center of the bottom metasurface. Since the electromagnetic waves emitted by the feed will be reflected twice by the top and bottom metasurfaces, its profile can be reduced to 1 / 3 of the focal length. In order to further improve the integration and radiation efficiency, the reflected electromagnetic wave front modulation technology is introduced into the metasurface, which can achieve a low-profile effect of multiple reflections through one reflection. However, the traditional folded transmission array antenna can only realize single linear polarization or circular polarization modulation within one band, which limits its engineering application. Therefore, the simultaneous independent regulation of left-handed circularly polarized waves and right-handed circularly polarized waves is of great significance to improving the performance of communication systems and has important engineering application value.

[0003] The radiation-receiving structure is a metamaterial based on artificial design. It can transmit electromagnetic waves in a specific frequency band while ensuring efficient reflection of electromagnetic waves in the frequency bands on the left and right sides of the frequency band. In addition, the introduction of spin decoupling theory into the radiation-receiving structure can break the conjugation of circularly polarized waves and realize two independently controlled circularly polarized beams; the traditional spin decoupling transmission array antenna has a high profile and is difficult to integrate. At the same time, the traditional folded transmission array antenna cannot achieve independent control of two orthogonal circularly polarized waves in one frequency band. Summary of the invention

[0004] The present invention combines the spin decoupling method with a non-mirror reflection folded transmission array for the first time. By adjusting the reflection phase of the polarization conversion metasurface, a single fold can achieve the low-profile effect of a secondary and tertiary fold, so as to reduce the energy loss caused by multiple reflections, thereby maintaining a low profile while achieving high efficiency. At the same time, the spin decoupling method is used in the transmission metasurface to realize dual beams, so that one folded transmission array can realize the functions of two traditional folded transmission arrays.

[0005] A folded transmission array antenna that simultaneously meets the requirements of low profile, dual beam and high radiation performance and its design method are disclosed.

[0006] The present invention discloses a radiation-receiving low-profile dual-beam array antenna, which comprises a three-part structure: a transmission metasurface, a polarization conversion metasurface and a feed horn;

[0007] The feed horn is placed at the center of the transmission metasurface, the aperture surface is flush with the surface of the lower receiving patch, and the polarization conversion metasurface is placed at a position H just below the transmission metasurface;

[0008] The transmission metasurface is composed of m×m radiation-receiving units with different rotation angles arranged in a plane at equal intervals and periodically extended to achieve efficient transmission of electromagnetic waves in a specific frequency band and spin decoupling focusing phase control;

[0009] The radiation-receiving unit is composed of three layers of metal structure and two layers of dielectric plates, with a period of P; the upper layer is a metal patch with a C-shaped groove; the middle layer is a metal floor with a through hole; the bottom layer is a C-shaped groove metal patch with two cut corners in the diagonal direction; the two layers of C-shaped groove metal patches are connected by metallized vias; in the two layers of dielectric plates, the first dielectric plate is located between the upper layer of C-shaped groove metal patch and the middle layer of metal floor with through holes, and the second dielectric plate is located between the metal floor with through holes and the bottom layer of C-shaped groove metal patch with two cut corners in the diagonal direction;

[0010] The polarization conversion metasurface is composed of n×n polarization conversion units with different opening angles, which are arranged in an equidistant and periodic manner in a plane, to achieve efficient reflection of electromagnetic waves in a specific frequency band, polarization conversion, and non-mirror deflection phase control of the beam; the upper layer of the polarization conversion unit is an I-shaped metal ring, and the bottom layer is a metal backplane, which is separated by a dielectric plate with a period of P1.

[0011] Furthermore, the radiation-receiving unit is composed of three layers of metal structure and two layers of dielectric plates; the upper structure of the unit is a metal patch with a C-shaped groove, the outer diameter of the metal patch with the C-shaped groove is R1, the inner diameter is R3, and the radius of the circular patch inside is R4; the middle layer is a metal base plate with a through hole diameter of D1; between the middle layer and the upper structure is a dielectric plate with a thickness of h1; the bottom layer is a metal patch with a C-shaped groove with two cut corners in a diagonal direction, the two cut angles are a, the outer diameter of the metal ring after cutting is R2, and the diameter of the metallized through hole is D2; between the bottom patch and the middle layer is a dielectric plate with a thickness of h1.

[0012] Furthermore, the radiation-receiving unit rotates the receiving patch and the radiation patch from 0°-360° respectively to obtain 360° phase coverage. The receiving / transmitting patch rotates counterclockwise by an angle of α / β:

[0013]

[0014] in, represents the incident electric field after the receiving patch is rotated by angle α, Represents the new coordinate system after the receiving patch is rotated by angle α, represents the new coordinate system after the radiation patch rotates by an angle β, E0 represents the electric field intensity of the incident wave, kz represents the spatial phase, represents the transmission electric field after the radiation patch rotates by an angle γ, T y-RCP It represents the transmission coefficient of right-hand circularly polarized wave converted into y-polarized wave;

[0015] The phase distribution of left-hand circularly polarized wave and right-hand circularly polarized wave components is obtained as follows:

[0016]

[0017] Among them, Φ LCP represents the phase distribution of the left-hand circularly polarized wave component, Φ RCP represents the phase distribution of the right-hand circularly polarized wave component;

[0018] When the phases of the bottom receiving patch and the top radiating patch of the transmissive metasurface meet the spin decoupling phase condition, two independently controlled orthogonal circularly polarized beams can be realized:

[0019]

[0020] in, represents the phase of the receiving patch after rotation of α, It represents the phase of the radiation patch after being rotated by β. The spin-decoupled spatial phase arrangement of the transmission metasurface is performed based on the phases of the receiving patch and the radiation patch.

[0021] Furthermore, the upper layer of the polarization conversion unit is an I-shaped metal ring, the thickness of the middle layer dielectric plate is h2, and the bottom layer is a metal backplate used to prevent the transmission of electromagnetic waves; the unit period is P1, the outer diameter of the I-shaped metal ring structure is R5, the inner diameter is R6, the opening angle of the I-shaped metal ring is γ1, the width of the middle metal strip is w, and the overall rotation angle of the I-shaped metal ring is γ2.

[0022] Furthermore, the polarization conversion unit can achieve 360° phase control within 11-15 GHz by changing the opening angle γ1 and the rotation angle β2 of the metal ring; when the left-handed circularly polarized wave is incident vertically, the compensation phase Φ(x, y0) required at different positions of the metasurface is:

[0023]

[0024] Φ(x,y) represents the compensation phase required to achieve electromagnetic wave focusing, (x,y) represents the relative position of the metasurface unit, k0=2π / λ0 is the free space wave vector, λ0 is the wavelength at the operating frequency f0, is the constant reference phase, OM is the initial focal length, and OM′ is the focal length after the electromagnetic wave path is changed.

[0025] A folding decoupling design method for a radiating-receiving low-profile dual-beam array antenna is also provided, and the folding decoupling design method for a radiating-receiving low-profile dual-beam array antenna comprises the following steps:

[0026] Step 1, designing a radiation-receiving unit of the radiation-receiving low-profile dual-beam array antenna;

[0027] Step 2, designing a polarization conversion unit of the radiating-receiving low-profile dual-beam array antenna;

[0028] Step 3, using spin decoupling phase to arrange the spatial phase of the transmission metasurface;

[0029] Step 4, using a phase compensation method to arrange the spatial phase of the polarization conversion metasurface;

[0030] Step 5: Based on the designed transmission metasurface and polarization conversion metasurface, construct the final radiating-receiving low-profile dual-beam array antenna.

[0031] Furthermore, in step 1, the radiation-receiving unit is composed of three layers of metal structure and two layers of dielectric plate; the upper structure of the unit is a metal patch with a C-shaped groove, the outer diameter of the metal patch with the C-shaped groove is R1, the inner diameter is R3, and the radius of the circular patch inside is R4; the middle layer is a metal bottom plate with a through hole diameter of D1; between the middle layer and the upper structure is a dielectric plate with a thickness of h1; the bottom layer is a metal patch with a C-shaped groove with two cut corners in a diagonal direction, the two cut angles are a, the outer diameter of the metal ring after the cut corners is R2, and the diameter of the metalized through hole is D2; between the bottom patch and the middle layer is a dielectric plate with a thickness of h1;

[0032] In step 2, the upper layer of the polarization conversion unit is an I-shaped metal ring, the thickness of the middle layer dielectric plate is h2, and the bottom layer is a metal backplane to prevent the transmission of electromagnetic waves; the unit period is P1, the outer diameter of the I-shaped metal ring structure is R5, the inner diameter is R6, the opening angle of the I-shaped metal ring is γ1, the width of the middle metal strip is w, and the overall rotation angle of the I-shaped metal structure is γ2.

[0033] Furthermore, in step 3, the receiving patch and the radiating patch of the radiating-receiving unit are rotated from 0° to 360° respectively to obtain 360° phase coverage. When the receiving / transmitting patch is rotated counterclockwise by an angle α / β, there is:

[0034]

[0035] in, represents the incident electric field after the receiving patch is rotated by angle α, Represents the new coordinate system after the receiving patch is rotated by angle α, represents the new coordinate system after the radiation patch rotates by an angle β, E0 represents the electric field intensity of the incident wave, kz represents the spatial phase, represents the transmission electric field after the radiation patch rotates by an angle β, T y-RCP It represents the transmission coefficient of right-hand circularly polarized wave converted into y-polarized wave;

[0036] The phase distribution of left-hand circularly polarized wave and right-hand circularly polarized wave components is obtained as follows:

[0037]

[0038] Among them, Φ LCP represents the phase distribution of the left-hand circularly polarized wave component, Φ RCP represents the phase distribution of the right-hand circularly polarized wave component;

[0039] When the phase of the bottom receiving patch and the phase of the top radiating patch of the transmissive metasurface meet the spin decoupling phase condition, two independently controlled orthogonal circularly polarized beams can be realized:

[0040]

[0041] in, represents the phase of the receiving patch after rotation of α, It represents the phase of the radiation patch after being rotated by β. The spin-decoupled spatial phase arrangement of the transmission metasurface is performed based on the phases of the receiving patch and the radiation patch.

[0042] Furthermore, in step 4, the polarization conversion unit can fully reflect the incident left-hand circularly polarized wave into a right-hand circularly polarized wave, and changing the opening angle γ1 and the rotation angle γ2 of the metal ring can achieve 360° phase control within 11-15 GHz. When the left-hand circularly polarized wave is vertically incident, the compensation phase Φ(x, y) required at different positions of the polarization conversion metasurface is:

[0043]

[0044] Φ(x,y) represents the compensation phase required to achieve electromagnetic wave focusing, (x,y) represents the relative position of the metasurface unit, k0=2π / λ0 is the free space wave vector, λ0 is the wavelength at the operating frequency f0, is the constant reference phase, OM is the initial focal length, and OM′ is the focal length after the electromagnetic wave path is changed.

[0045] The above formula is used to arrange the polarization conversion metasurface in spatial phase compensation.

[0046] Furthermore, in step 5, the transmission metasurface is placed above the polarization conversion metasurface, and the spacing is set to H; the circularly polarized horn feed is placed at the center of the transmission metasurface, and the horn aperture surface is flush with the receiving patch surface of the bottom layer of the transmission metasurface;

[0047] When H is equal to F / 3 or F / 4, F is the focal length after the electromagnetic wave path is changed. The left-hand circularly polarized wave emitted by the circularly polarized horn feed source is reflected as a right-hand circularly polarized wave by the polarization conversion metasurface, and then received by the transmission metasurface and generates independently controllable left-hand circularly polarized and right-hand circularly polarized beams radiating along the +z direction through spin decoupling.

[0048] The beneficial effects achieved by the present invention are:

[0049] Compared with the traditional folded transmission array antenna, the present invention can simultaneously achieve low profile, high aperture efficiency and independent control of dual circular polarization beams;

[0050] The present invention proposes an effective design method, which can design a low-profile dual-beam array antenna with a lower profile according to the design process as required;

[0051] The present invention has a very considerable antenna gain effect. When the overall cross-section of the antenna is 1 / 3 of that of the traditional transmission array, the measured peak gains of the left-hand circular polarization and the right-hand circular polarization beams are 24.53 dBic and 24.12 dBic respectively, and the corresponding aperture efficiencies are 37.12% and 33.78% respectively; at the same time, when the overall cross-section of the antenna is 1 / 4 of that of the traditional transmission array, the measured peak gains of the left-hand circular polarization and the right-hand circular polarization beams are 22.56 dBic and 22.42 dBic respectively, and the corresponding aperture efficiencies are 23.6% and 22.8% respectively. BRIEF DESCRIPTION OF THE DRAWINGS

[0052] Figure 1 Schematic diagram of a low-profile dual-beam array antenna.

[0053] Figure 2 The topological structure of the transmission metasurface radiation-receiving unit and the radiation patch and receiving patch.

[0054] Figure 3 Side view of the current distribution of the receiving-transmitting structure under the action of different electromagnetic waves.

[0055] (a) Left-hand circularly polarized wave; (b) right-hand circularly polarized wave.

[0056] Figure 4 Numerical characteristics simulated for the radiating-receiving unit.

[0057] (a) Cross-circular polarization transmission amplitude and phase when the receiving patch is rotated by angle α;

[0058] (b) Co-circular polarization transmission amplitude and phase when the receiving patch is rotated by angle α;

[0059] (c) Cross-circular polarization transmission amplitude and phase when the radiating patch is rotated by angle β;

[0060] (d) Co-circularly polarized transmission amplitude and phase when the radiating patch is rotated by angle β.

[0061] Figure 5 are the transmission amplitude and phase of the radiation-receiving unit under the incidence of right-hand circularly polarized waves at different angles.

[0062] (a) Cross circular polarization; (b) Co circular polarization.

[0063] Figure 6 Design and simulation results for polarization-converting metasurfaces.

[0064] (a) Top view of the polarization conversion metasurface and the cross-circular polarization reflection amplitude of the left-hand circularly polarized wave at different incident angles when γ1 and γ2 are 20° and 45° respectively.

[0065] (b) Cross-circularly polarized reflection amplitude and phase at 13 GHz when γ2 is 45° and −45°, and γ1 varies from 7° to 132°.

[0066] Figure 7 The theoretical synthesized phase distribution of the transmission metasurface at the center frequency of 13 GHz.

[0067] (a) Phase distribution of left-hand circularly polarized beam at (10°, 0°); (b) Phase distribution of right-hand circularly polarized beam at (-25°, 0°);

[0068] (c) Phase distribution of the receiving patch; (d) Phase distribution of the transmitting patch.

[0069] Figure 8 Theoretical synthesized phase distribution of the polarization-conversion metasurface at a center frequency of 13 GHz.

[0070] (a) Phase distribution of H1=F / 3; (b) Phase distribution of H2=F / 4.

[0071] Fig. 9 3D simulated radiation pattern of a left-hand circularly polarized feed excited at 13 GHz.

[0072] (a) H1 = F / 3; (b) H2 = F / 4.

[0073] Fig.10 (a) Antenna model manufactured for processing; (b) experimental setup.

[0074] Fig.11 The simulated and measured radiation patterns of the array antenna with H1 = F / 3 at 13 GHz. (a) Left-hand circularly polarized and (b) right-hand circularly polarized beams on the xz plane; (c) yz L Left-hand circularly polarized beam on a plane; (d)yz R Right-hand circularly polarized beam in a plane.

[0075] Fig.12 The gain and axial ratio curves of the simulated and experimental H1=F / 3 array antenna. (a) Left-hand circularly polarized beam; (b) Right-hand circularly polarized beam.

[0076] Fig.13 The simulated and measured radiation patterns of the array antenna with H2 = F / 4 at 13 GHz. (a) Left-hand circularly polarized and (b) right-hand circularly polarized beams on the xz plane; (c) yz L Left-hand circularly polarized beam on a plane; (d)yz R Right-hand circularly polarized beam in a plane.

[0077] Fig.14 The simulated and experimental gain and axial ratio curves of the H2=F / 4 array antenna.

[0078] (a) Left-hand circularly polarized beam; (b) right-hand circularly polarized beam. DETAILED DESCRIPTION

[0079] The present invention will be further described below in conjunction with specific embodiments, and the advantages and features of the present invention will become clearer as the description proceeds. However, these embodiments are exemplary only and do not constitute any limitation to the scope of the present invention. It should be understood by those skilled in the art that the details and forms of the technical solution of the present invention may be modified or replaced without departing from the spirit and scope of the present invention, but these modifications and replacements all fall within the scope of protection of the present invention.

[0080] like Figure 1 As shown, the low-profile dual-beam array antenna provided by the present invention is composed of a transmission metasurface, a polarization conversion metasurface and a horn feed. The polarization conversion metasurface is placed at H directly below the transmission metasurface, the horn feed is placed at the center of the transmission metasurface, and the aperture surface is parallel to its underlying receiving patch. Under the excitation of the horn feed, the polarization conversion metasurface performs a non-mirror reflection of the electromagnetic wave, which is then received by the transmission metasurface and transmitted and converted into two independently controlled orthogonal circularly polarized waves through the spin decoupling theory. This design achieves the effect of folding the traditional folded transmission array antenna three times through one reflection, greatly reducing the energy loss in the reflection process and improving the overall radiation efficiency and integration of the antenna.

[0081] The transmission metasurface is composed of m×m radiation-receiving units with different rotation angles arranged in a plane at equal intervals and periodically extended to achieve efficient transmission of electromagnetic waves in a specific frequency band and spin decoupling focusing phase regulation; the radiation-receiving unit is composed of three layers of metal structure and two layers of dielectric plates, with a period of P; the upper layer is a metal patch with C-shaped grooves; the middle layer is a metal floor with through holes; the bottom layer is a C-shaped groove metal patch with two cut corners along the diagonal direction; the two layers of C-shaped groove metal patches are connected by metal vias; in the two layers of dielectric plates, the first dielectric plate is located on the upper layer Between the C-shaped slot metal patch and the metal floor with through holes in the middle layer, the second dielectric plate is located between the metal floor with through holes and the C-shaped slot metal patch with two cut corners in the diagonal direction of the bottom layer; the polarization conversion metasurface is composed of n×n polarization conversion units with different opening angles arranged in a plane at equal intervals and periodically extended to achieve efficient reflection of electromagnetic waves in a specific frequency band, polarization conversion and non-mirror deflection phase control of beams; the upper layer of the polarization conversion unit is an I-shaped metal patch, and the bottom layer is a metal backplane, and the two are separated by a dielectric plate with a period of P1;

[0082] The structural parameters of the two units are as follows: in the first part, the outer diameter of the metal patch with a C-shaped groove is R1, the inner diameter is R3, and the radius of the circular patch inside is R4; the diameter of the through hole of the metal floor with a through hole is D1; ​​the metal patch with a C-shaped groove with two cut corners in the diagonal direction has a cut angle a, and the outer diameter of the metal ring after cutting is R2; the diameter of the metalized via is D2; the thickness of the first and second dielectric plates is h1; in the second part, the outer diameter of the I-shaped metal ring structure is R5, the inner diameter is R6, the opening angle of the I-shaped metal ring is γ1, the width of the middle metal strip is w, and the overall rotation angle of the I-shaped metal structure is γ2; the thickness of the dielectric plate is h2, and the distance between the polarization conversion metasurface and the transmission metasurface is H.

[0083] The structural parameters of the low-profile dual-beam array antenna according to the design process and method are as follows: P = 10mm, H1 = 36mm, H2 = 27mm, R1 = 3.2mm, R2 = 2.6mm, R3 = 2.3mm, R4 = 1mm, a = 90°, D1 = 1mm, D2 = 0.4mm, P1 = 6mm, R5 = 2.5mm, R6 = 1.6mm, w = 0.9mm, γ1 = 20°, γ2 = 45°, h1 = 2mm, h2 = 2.5mm; the metal material parameters are copper with a thickness of 0.017mm and a conductivity of 5.8×10 7 S / m; the dielectric plate is made of F4B, the dielectric constant is 2.65, and the electrical tangent loss is 0.001.

[0084] According to the requirements of the low-profile dual-beam array antenna, the present invention optimizes the design of the two unit structures and the final antenna structure. The specific steps are as follows:

[0085] Step 1, designing a radiation-receiving unit;

[0086] Compared with the traditional multi-layer stacked dielectric structure, the radiation-receiving unit can broaden the working bandwidth and flexibly control electromagnetic waves. Based on this, the present invention designs a radiation-receiving unit, which has the functions of efficient transmission and spin decoupling focusing phase control of the chirality channel in the antenna working frequency band, and can perform polarization conversion through the upper and lower layers of metal patches.

[0087] The structural unit consists of three layers of metal structure and two layers of dielectric plates, with a period of P. The upper structure of the unit is a metal patch with a C-shaped groove. The outer diameter of the metal patch with a C-shaped groove is R1, the inner diameter is R3, and the radius of the circular patch inside is R4. The middle layer is a metal bottom plate with a through hole diameter of D1. Between the middle layer and the upper structure is a dielectric plate with a thickness of h1. The bottom layer is a metal patch with a C-shaped groove with two cut corners in the diagonal direction. The two cut angles are a, the outer diameter of the metal ring after the cut corners is R2, and the diameter of the metalized through hole is D2. Between the bottom structure and the middle layer is a dielectric plate with a thickness of h1. Rotating the lower layer receiving patch and the upper layer radiating patch from 0°-360° respectively can obtain 360° phase coverage. The scanning parameter function of CST software is used to simulate and optimize the unit structure parameters to determine the optimal structural parameters. The optimized structural parameters are: P=10mm, R1=3.2mm, R2=2.6mm, R3=2.3mm, R4=1mm, a=90°, D1=1mm, D2=0.4mm.

[0088] In order to study the electromagnetic characteristics of the radiation-receiving unit, the present invention uses CST software to perform a series of simulations on the unit. During the unit simulation process, the x and y directions are set as periodic boundary conditions, the z direction is set as an open boundary condition, and the unit is excited by a wave port. Figure 3 The current distribution diagram of the unit is given when the circularly polarized electromagnetic wave is incident along the +z direction. The current caused by the incident left-handed circularly polarized wave is mainly concentrated on the receiving patch, and there is no current on the radiating patch. This phenomenon shows that most of the left-handed circularly polarized wave is reflected by the receiving patch and the metal floor. On the other hand, the current caused by the incident right-handed circularly polarized wave can reach the radiating patch through the metallized via. Therefore, the unit works in the transmission mode when the right-handed circularly polarized wave is incident. Figure 4 The transmission coefficient and phase of the unit when the right-hand circularly polarized wave is incident along the +z direction are given. Here, the present invention uses the cross circular polarization transmission coefficient T LR and the co-circular polarization transmission coefficient T RR The left-hand circularly polarized wave and right-hand circularly polarized wave components of the transmission field are studied independently. Since the linearly polarized transmission wave is evenly divided into left-hand circularly polarized waves and right-hand circularly polarized waves, the maximum transmission amplitude is close to -3dB. Figure 4As shown in (a) and (b), when the rotation angle α of the receiving patch changes from 0° to 360°, the transmission amplitude is close to -3dB, and the phase can achieve 360° coverage in the range of 11.5 to 14.5GHz. Similarly, changing the rotation angle β of the radiating patch will also produce similar results, such as Figure 4 (c) and (d). Since the feed source emits electromagnetic waves at different angles, the effect of oblique incidence at these angles on the transmittance of the transmissive metasurface should be considered. Figure 5 As can be seen from (a) and (b), the transmission amplitude of the right-hand circularly polarized wave can still achieve a high transmittance at different incident angles. The phase error is acceptable within the working frequency band, so the designed radiation-receiving unit is angle-insensitive.

[0089] Step 2, designing a polarization conversion unit;

[0090] In the previous step, a radiation-receiving structural unit was designed, which can efficiently transmit electromagnetic waves at 11.5-14.5 GHz. In order to build a low-profile dual-beam array antenna, the present invention designs an I-shaped polarization conversion structure. The upper layer of the polarization conversion unit is an I-shaped metal patch, the thickness of the middle layer dielectric plate is h2, and the bottom layer is a metal backplane to prevent the transmission of electromagnetic waves. The unit period is P1, the outer diameter of the I-shaped metal ring structure is R5, the inner diameter is R6, the opening angle of the I-shaped metal ring is γ1, the width of the middle metal strip is w, and the overall rotation angle of the I-shaped metal structure is γ2. Using the full-wave simulation software CSTMICROWAVE STUDIO (2020) for simulation calculation, the results show that the polarization conversion unit can fully reflect the incident left-handed circularly polarized wave into a right-handed circularly polarized wave, and changing the metal ring opening angle γ1 and the rotation angle γ2 can achieve 360° phase control within 11-15 GHz.

[0091] The structural parameters optimized by the full-wave simulation software CST 2020 are: P1 = 6mm, R5 = 2.5mm, R6 = 1.6mm, w = 0.9mm, γ1 = 20°, γ2 = 45°, h2 = 2.5mm. Finally, the unit characteristics are simulated using CST 2020. Here, R RL It represents the cross-circular polarization reflection amplitude when left-hand circularly polarized wave is incident. Figure 6 As shown in (a), in the frequency range of 11.5 to 14.5 GHz, the left-hand circularly polarized wave is efficiently converted into the right-hand circularly polarized wave, and the reflection amplitude remains stable at different incident angles. Figure 6 (b) shows that at 13 GHz, fixing γ2 = 45° and changing the opening angle γ1 from 7° to 132° can achieve 180° phase coverage; changing γ2 from 45° to -45° and adjusting γ1 from 7° to 132° can achieve an additional 180° reflection phase coverage.

[0092] Step 3, using spin decoupling phase to arrange the spatial phase of the transmission metasurface;

[0093] Combination Figure 2 , the present invention derives the spin decoupling theory accordingly.

[0094] Assuming that the receiving patch without rotation is illuminated by a right-hand circularly polarized plane wave propagating in the +z direction, the incident electric field can be written as follows:

[0095]

[0096] Among them, E0 is the electric field intensity of the incident wave, and kz represents the spatial phase.

[0097] After transmission, the right-hand circularly polarized wave is converted into a y-polarized wave, and the transmitted electric field can be expressed as:

[0098]

[0099] Among them, T y-RCP is the transmission coefficient of right-hand circularly polarized wave converted into y-polarized wave.

[0100] When the receiving / transmitting patch rotates counterclockwise by an angle α / β, the coordinate system will change from Transfer to The new coordinate system can be written as:

[0101]

[0102] According to equation (3), the incident electric field and the transmitted electric field can be written as:

[0103]

[0104] in, represents the incident electric field after the receiving patch rotates by an angle α, E0 represents the electric field strength of the incident wave, represents the transmission electric field after the radiation patch rotates by an angle β, T y-RCP It represents the transmission coefficient of right-hand circularly polarized wave converted into y-polarized wave;

[0105] The phase distribution of left-hand circularly polarized wave and right-hand circularly polarized wave components is obtained as follows:

[0106]

[0107] Among them, Φ LCP represents the phase distribution of the left-hand circularly polarized wave component, Φ RCPrepresents the phase distribution of the right-hand circularly polarized wave component; therefore, when the phase of the bottom receiving patch and the phase of the top radiating patch of the transmission metasurface satisfy the spin decoupling phase condition shown in equation (7), two independently controlled orthogonal circularly polarized beams can be realized.

[0108]

[0109] in represents the phase of the receiving patch after rotation of α, represents the phase of the radiating patch after rotation of β,

[0110] According to formula (7), the required phase at different positions of the transmission metasurface is calculated. The present invention arranges the corresponding radiation-receiving units at the corresponding positions, and finally designs an 18×18 transmission metasurface. The transmission metasurface can perform spin decoupling when a right-hand circularly polarized wave is incident, thereby generating two independently controlled orthogonal circularly polarized waves. The phase calculation of left-hand circularly polarized and right-hand circularly polarized waves is as shown in formula (8):

[0111]

[0112] Where k0 is the wave vector in free space, F is the focal length, the F / D ratio is set to 0.6, the aperture D is 180 mm, and the focal length F is 108 mm. Here, is the left-hand circular polarization / right-hand circular polarization beam direction, where θ is the elevation angle, is the azimuth angle, where the left-hand circular polarization and right-hand circular polarization beam directions are selected as (10°, 0°) and (-25°, 0°), respectively. According to the calculated phase distribution at different positions, Figure 7 (a) and (b) give the required phase distributions for two separate left-hand circularly polarized and right-hand circularly polarized beams. In addition, substituting equation (8) into equation (7) yields the required phase distributions for the receiving and radiating layers, as Figure 7 As shown in (c) and (d).

[0113] Step 4, using a phase compensation method to arrange the spatial phase of the polarization conversion metasurface;

[0114] When the left-hand circularly polarized wave is incident vertically, the compensation phase φ(x, y) required at different positions of the polarization conversion metasurface is calculated according to formula (9):

[0115]

[0116] Among them, Φ(x,y) represents the compensation phase required to achieve electromagnetic wave focusing, (x,y) represents the relative position of the metasurface unit, k0=2π / λ0 is the free space wave vector, λ0 is the wavelength at the operating frequency f0, is the constant reference phase, OM is the initial focal length, and OM′ is the focal length after the electromagnetic wave path is changed. Figure 1 It can be seen that OM′=F, OM=2H, and H is the distance between the transmission metasurface and the polarization conversion metasurface.

[0117] According to formula (9), the required compensation phase at different positions is calculated. The present invention arranges metasurface units with corresponding reflection functions at corresponding positions, and finally designs a 30×30 polarization conversion metasurface. The polarization conversion metasurface can perform phase compensation when a left-handed circularly polarized wave is incident, thereby realizing a non-mirror reflection function, and can also realize the change of the rotation direction of the circularly polarized wave.

[0118] Step 5, design of low-profile dual-beam array antenna.

[0119] With the designed transmission metasurface and polarization conversion metasurface, the ultimate radiation-receiving low-profile dual-beam array antenna can be constructed. Specifically as follows: Place the transmission metasurface above the polarization conversion metasurface, and the interval is set to H. The size of H depends on the phase compensation distance of the polarization conversion metasurface. Here, the present invention sets H1=F / 3 and H2=F / 4. A circularly polarized horn feed with a diameter of 20 mm is placed at the center of the transmission metasurface, and the horn aperture surface is flush with the receiving patch surface of the bottom layer of the transmission metasurface. The left-handed circularly polarized wave emitted by the circularly polarized horn feed is reflected as a right-handed circularly polarized wave by the polarization conversion metasurface, and then received by the transmission metasurface and generated independently controllable left-handed circularly polarized and right-handed circularly polarized beams radiated along the +z direction through spin decoupling.

[0120] Finally, the present invention designs two low-profile dual-beam array antennas to verify this design method and performs full-wave simulation in CST2020. Fig. 9 As shown, in the case of H1=F / 3 or H2=F / 4, both left-hand circular polarization and right-hand circular polarization beams can be generated in the set direction.

[0121] In order to further verify the correctness of the low-profile dual-beam array antenna design method, the present invention uses PCB technology to process and manufacture the designed physical samples based on the designed simulation model, such as Fig.10 As shown in (a), four 50 mm high nylon pillars are used to ensure the distance H between the transmission metasurface and the polarization conversion metasurface. When actually testing the antenna characteristics of the processed sample, the present invention is carried out in a microwave darkroom, which can effectively avoid the interference of the surrounding complex electromagnetic environment, such as Fig.10 (b) The antenna sample is placed at the center of a cylindrical foam that can rotate around its central axis, while a standard gain receiving horn operating at 8-18 GHz is placed on another cylindrical foam 10 m away from the sample, and a laser level is used to ensure that the centers of the two are in the same straight line.

[0122] Fig.11 The radiation pattern of H1=F / 3 is given, and it can be seen that the directions of the left-hand circular polarization and right-hand circular polarization beams are completely consistent with the previously designed directions. In the measurement, the present invention obtains that the sidelobe levels of the left-hand circular polarization and right-hand circular polarization beams are -15dB and -14.8dB respectively. Fig.12 The axial ratio and gain curves of the designed antenna are given. It can be seen from the figure that in the operating frequency band of 11-15GHz, the axial ratios of the left-hand circular polarization and right-hand circular polarization beams are both less than 3dB. At 13GHz, the measured peak gains of the left-hand circular polarization and right-hand circular polarization beams are 24.53dBic and 24.12dBic respectively. Using the aperture efficiency formula Calculations show that the corresponding aperture efficiencies are 37.12% and 33.78% respectively. Meanwhile, the 3dB gain bandwidths of left-hand circularly polarized and right-hand circularly polarized beams are 23.1% and 23.1% respectively. Fig.13 The radiation pattern of H2 = F / 4 is given, and it can be seen that the directions of the left-hand circularly polarized and right-hand circularly polarized beams are exactly the same as the previously designed directions. The measured sidelobe levels of the left-hand circularly polarized and right-hand circularly polarized beams are -15.5dB and -15.1dB respectively. Fig.14 (a) and (b) show the simulated and measured performance of the gain and axial ratio curves achieved in the frequency range of 11-15 GHz. At 13 GHz, the measured peak gains of the left-hand circularly polarized and right-hand circularly polarized beams are 22.56 dBic and 22.42 dBic, respectively, and the corresponding aperture efficiencies are 23.6% and 22.8%, respectively. At the same time, the measurement results show that the 3 dB gain bandwidths of the left-hand circularly polarized and right-hand circularly polarized beams are 23.1% and 23%, respectively. In the entire operating frequency band, the axial ratios of the left-hand circularly polarized and right-hand circularly polarized beams are both lower than 3 dB. The slight difference between the above simulation and measurement results is mainly caused by the distance error between the transmission metasurface and the polarization conversion metasurface when using nylon struts to install the transmission metasurface. The alignment error of the feed horn and the sample in the measurement is also one of the reasons.

[0123] The above are only specific steps of the present invention and do not constitute any limitation to the protection scope of the present invention; any technical solutions formed by equivalent transformation or equivalent replacement shall fall within the protection scope of the present invention; some parts not elaborated in detail in the present invention belong to the known technologies of those skilled in the art.

Claims

1. A radiating-receiving low-profile dual-beam array antenna, characterized in that: The radiating-receiving low-profile dual-beam array antenna includes three structures: a transmission metasurface, a polarization conversion metasurface, and a feed horn; The feed horn is placed at the center of the transmission metasurface, the aperture surface is flush with the surface of the lower receiving patch, and the polarization conversion metasurface is placed at a position H just below the transmission metasurface; The transmission metasurface is composed of m×m radiation-receiving units with different rotation angles arranged in a plane at equal intervals and periodically extended to achieve efficient transmission of electromagnetic waves in a specific frequency band and spin decoupling focusing phase control; The radiation-receiving unit is composed of three layers of metal structure and two layers of dielectric plates, with a period of P; the upper layer is a metal patch with a C-shaped groove; The middle layer is a metal floor with through holes; the bottom layer is a C-shaped groove metal patch with two cut corners along the diagonal direction; Two layers of C-shaped slot metal patches are connected through metallized vias; in the two layers of dielectric boards, the first dielectric board is located between the upper C-shaped slot metal patch and the middle metal floor with through holes, and the second dielectric board is located between the metal floor with through holes and the C-shaped slot metal patch with two cut corners in the diagonal direction of the bottom layer; The polarization conversion metasurface is composed of n×n polarization conversion units with different opening angles, which are arranged in an equidistant and periodic manner in a plane, to achieve efficient reflection of electromagnetic waves in a specific frequency band, polarization conversion, and non-mirror deflection phase control of the beam; the upper layer of the polarization conversion unit is an I-shaped metal ring, and the bottom layer is a metal backplane, which is separated by a dielectric plate with a period of P1.

2. The radiating-receiving low-profile dual-beam array antenna according to claim 1, characterized in that: The radiation-receiving unit is composed of three layers of metal structure and two layers of dielectric plates; the upper structure of the unit is a metal patch with a C-shaped groove, the outer diameter of the metal patch with the C-shaped groove is R1, the inner diameter is R3, and the radius of the circular patch inside is R4; the middle layer is a metal base plate with a through hole diameter of D1; between the middle layer and the upper structure is a dielectric plate with a thickness of h1; the bottom layer is a metal patch with a C-shaped groove with two cut corners in a diagonal direction, the two cut angles are a, the outer diameter of the metal ring after cutting is R2, and the diameter of the metallized through hole is D2; between the bottom patch and the middle layer is a dielectric plate with a thickness of h1.

3. The radiating-receiving low-profile dual-beam array antenna according to claim 1, characterized in that: The radiation-receiving unit rotates the receiving patch and the radiation patch from 0°-360° respectively to obtain 360° phase coverage. The receiving / transmitting patch rotates counterclockwise by an angle of α / β: in, represents the incident electric field after the receiving patch is rotated by angle α, Represents the new coordinate system after the receiving patch is rotated by angle α, represents the new coordinate system after the radiation patch is rotated by an angle β, E0 represents the electric field intensity of the incident wave, kz represents the spatial phase, represents the transmission electric field after the radiation patch rotates by an angle β, T y-RCP It represents the transmission coefficient of right-hand circularly polarized wave converted into y-polarized wave; The phase distribution of left-hand circularly polarized wave and right-hand circularly polarized wave components is obtained as follows: Among them, Φ LCP represents the phase distribution of the left-hand circularly polarized wave component, Φ RCP represents the phase distribution of the right-hand circularly polarized wave component; When the phase of the bottom receiving patch and the phase of the top radiating patch of the transmissive metasurface meet the spin decoupling phase condition, two independently controlled orthogonal circularly polarized beams can be realized: in, represents the phase of the receiving patch after rotation of α, It represents the phase of the radiation patch after being rotated by β. The spin-decoupled spatial phase arrangement of the transmission metasurface is performed based on the phases of the receiving patch and the radiation patch.

4. The radiating-receiving low-profile dual-beam array antenna according to claim 1, characterized in that: The upper layer of the polarization conversion unit is an I-shaped metal ring, the thickness of the middle layer dielectric plate is h2, and the bottom layer is a metal backplane used to prevent the transmission of electromagnetic waves; the unit period is P1, the outer diameter of the I-shaped metal ring structure is R5, the inner diameter is R6, the opening angle of the I-shaped metal ring is γ1, the width of the middle metal strip is w, and the overall rotation angle of the I-shaped metal ring is γ2.

5. The radiating-receiving low-profile dual-beam array antenna according to claim 1, characterized in that: The polarization conversion unit can achieve 360° phase control within 11-15 GHz by changing the opening angle γ1 and the rotation angle γ2 of the metal ring; when the left-handed circularly polarized wave is incident vertically, the compensation phase Φ(x, y) required at different positions of the metasurface is: Φ(x,y) represents the compensation phase required to achieve electromagnetic wave focusing, (x,y) represents the relative position of the metasurface unit, k0=2π / λ0 is the free space wave vector, λ0 is the wavelength at the operating frequency f0, is the constant reference phase, OM is the initial focal length, and OM′ is the focal length after the electromagnetic wave path is changed.

6. A folding decoupling design method for a radiating-receiving low-profile dual-beam array antenna, characterized in that: The folding decoupling design method of the radiating-receiving low-profile dual-beam array antenna comprises the following steps: Step 1, designing a radiation-receiving unit of the radiation-receiving low-profile dual-beam array antenna; Step 2, designing a polarization conversion unit of the radiating-receiving low-profile dual-beam array antenna; Step 3, using spin decoupling phase to arrange the spatial phase of the transmission metasurface; Step 4, using a phase compensation method to arrange the spatial phase of the polarization conversion metasurface; Step 5: Based on the designed transmission metasurface and polarization conversion metasurface, construct the final radiating-receiving low-profile dual-beam array antenna.

7. The radiating-receiving low-profile dual-beam array antenna and folded decoupling design method according to claim 6, characterized in that: In step 1, the radiation-receiving unit is composed of three layers of metal structure and two layers of dielectric plate; the upper structure of the unit is a metal patch with a C-shaped groove, the outer diameter of the metal patch with the C-shaped groove is R1, the inner diameter is R3, and the radius of the circular patch inside is R4; the middle layer is a metal bottom plate with a through hole diameter of D1; between the middle layer and the upper structure is a dielectric plate with a thickness of h1; the bottom layer is a metal patch with a C-shaped groove with two cut corners in a diagonal direction, the two cut angles are a, the outer diameter of the metal ring after the cut corners is R2, and the diameter of the metalized through hole is D2; between the bottom patch and the middle layer is a dielectric plate with a thickness of h1; In step 2, the upper layer of the polarization conversion unit is an I-shaped metal ring, the thickness of the middle layer dielectric plate is h2, and the bottom layer is a metal backplane to prevent the transmission of electromagnetic waves; the unit period is P1, the outer diameter of the I-shaped metal ring structure is R5, the inner diameter is R6, the opening angle of the I-shaped metal ring is γ1, the width of the middle metal strip is w, and the overall rotation angle of the I-shaped metal structure is γ2.

8. The folded decoupling design method of the radiating-receiving low-profile dual-beam array antenna according to claim 7, characterized in that: In step 3, the receiving patch and the radiating patch of the radiating-receiving unit are rotated from 0° to 360° respectively to obtain 360° phase coverage. When the receiving / transmitting patch is rotated counterclockwise by an angle α / β, there is: in, represents the incident electric field after the receiving patch is rotated by angle α, Represents the new coordinate system after the receiving patch is rotated by angle α, represents the new coordinate system after the radiation patch rotates by an angle β, E0 represents the electric field intensity of the incident wave, kz represents the spatial phase, represents the transmission electric field after the radiation patch rotates by an angle β, T y-RCP It represents the transmission coefficient of right-hand circularly polarized wave converted into y-polarized wave; The phase distribution of left-hand circularly polarized wave and right-hand circularly polarized wave components is obtained as follows: Among them, Φ LCP represents the phase distribution of the left-hand circularly polarized wave component, Φ RCP represents the phase distribution of the right-hand circularly polarized wave component; When the phases of the bottom receiving patch and the top radiating patch of the transmissive metasurface meet the spin decoupling phase condition, two independently controlled orthogonal circularly polarized beams can be realized: in, represents the phase of the receiving patch after rotation of α, It represents the phase of the radiation patch after being rotated by β. The spin-decoupled spatial phase arrangement of the transmission metasurface is performed based on the phases of the receiving patch and the radiation patch.

9. The folded decoupling design method of the radiating-receiving low-profile dual-beam array antenna according to claim 7, characterized in that: In step 4, the polarization conversion unit can fully reflect the incident left-hand circularly polarized wave into a right-hand circularly polarized wave, and changing the opening angle γ1 and the rotation angle γ2 of the metal ring can achieve 360° phase control within 11-15 GHz. When the left-hand circularly polarized wave is vertically incident, the compensation phase Φ(x, y) required at different positions of the polarization conversion metasurface is: Φ(x,y) represents the compensation phase required to achieve electromagnetic wave focusing, (x,y) represents the relative position of the metasurface unit, k0=2π / λ0 is the free space wave vector, λ0 is the wavelength at the operating frequency f0, is the constant reference phase, OM is the initial focal length, and OM′ is the focal length after the electromagnetic wave path is changed. The above formula is used to arrange the polarization conversion metasurface in spatial phase compensation.

10. The folded decoupling design method of the radiating-receiving low-profile dual-beam array antenna according to claim 6, characterized in that: In step 5, the transmission metasurface is placed above the polarization conversion metasurface, and the spacing is set to H; the circularly polarized horn feed is placed at the center of the transmission metasurface, and the horn aperture surface is flush with the receiving patch surface of the bottom layer of the transmission metasurface; When H is equal to F / 3 or F / 4, F is the focal length after the electromagnetic wave path is changed. The left-hand circularly polarized wave emitted by the circularly polarized horn feed source is reflected as a right-hand circularly polarized wave by the polarization conversion metasurface, and then received by the transmission metasurface and generates independently controllable left-hand circularly polarized beam and right-hand circularly polarized beam radiated along the +z direction through spin decoupling.

Images