Frequency division multiplexing RFID tag signal parallel reception method based on SIC architecture

Abstract

This invention discloses a parallel reception method for frequency division multiplexing (FDM) RFID tag signals based on the SiC architecture. First, a FDM UHF RFID system is constructed. Tags use M values ​​of 4, 8, 12, and 16 to modulate their respective Miller codes using subcarriers. Then, based on the Wiener-Khinchin theorem, the power spectral density of the Miller subcarrier modulation sequence is derived. The interference level generated by tags in other channels within different frequency channels is quantitatively analyzed. Power estimation is performed using FFT. Then, using the known interference level, the amplitude and RF phase of each tag are estimated. The SiR of each tag is calculated and sorted. Demodulation is performed sequentially according to the SIR value until the last tag signal is demodulated. This method enables parallel reception of FDM RFID tags, allowing a reader to simultaneously store multiple tags within a single time slot. This effectively improves the identification efficiency and throughput of the UHF RFID system, enhancing its ability to adapt to large-scale IoT applications requiring rapid access to massive numbers of tags.

Description

Technical Field

[0001] This invention belongs to the field of radio frequency identification technology, specifically relating to a parallel reception method for frequency division multiplexing RFID tag signals based on the SIC architecture. Background Technology

[0002] Radio Frequency Identification (RFID) technology is an automatic identification technology that involves contactless data communication between a reader and a passive tag. It is widely used in logistics management, inventory control, intelligent transportation systems, and other scenarios, serving as a fundamental support for building the Internet of Things (IoT). With the continuous increase in RFID devices and applications, higher demands are being placed on the terminal capacity and coverage of RFID systems.

[0003] When multiple tags are simultaneously within the reader's recognition range, tag collisions are likely to occur, leading to decreased system recognition efficiency and data loss. Current anti-collision algorithms mainly include tree-structured algorithms and the Aloha algorithm, both based on time-division multiplexing. The QT algorithm is a typical tree-structured algorithm with a throughput of approximately 0.348. The dynamic Aloha algorithm has a maximum throughput of only 0.368. However, as the communication distance and coverage of readers continue to increase, the number of connected tags also increases. But the signal bandwidth returned by the tags is limited, making it difficult to improve the overall uplink throughput within the time slot returned by a single tag. Therefore, anti-collision algorithms based on time-division multiplexing have reached a bottleneck in improving throughput.

[0004] Currently, UHF RFID systems generally use the ISO / IEC 18000-6C protocol, which stipulates that a single reader can only communicate with the tag using one frequency channel via time-division multiplexing. However, the ISO / IEC 18000-6C protocol allows tags to use Miller-coded subcarrier modulation, with different tags selecting different M values ​​to achieve modulation on different subcarrier frequencies, thus enabling uplink frequency division multiplexing on the tag side.

[0005] Because of the tag's backscatter communication mechanism, the reflected signal cannot be filtered and shaped to suppress sidelobe / harmonic interference, leading to interference between different channels. Therefore, solving the interference introduced by multiple tags simultaneously responding to reader commands using frequency division multiplexing technology is one of the key issues in improving the throughput of RFID systems. Summary of the Invention

[0006] To address the aforementioned technical problems, this invention provides a parallel reception method for frequency division multiplexing RFID tag signals based on the SIC (Successive Interference Cancellation) architecture. It uses the Wiener-Khinchin theorem to derive the power spectral density of the Miller subcarrier modulation sequence, quantitatively analyzes the interference levels generated by tags in other channels within different frequency channels, and introduces the SIC mechanism into the frequency division multiplexing UHF RFID system to reduce sidelobe and harmonic interference of the tag signal.

[0007] The technical solution adopted in this invention is: a parallel reception method for frequency division multiplexing RFID tag signals based on the SIC architecture, the specific steps of which are as follows:

[0008] S1. Construct a frequency division multiplexing UHF RFID system, including: 4 RFID tags and 1 reader / writer;

[0009] The reader activates four RFID tags by sending a 920-925MHz tag excitation signal through its transmitting antenna. After receiving the inventory command from the reader, the four RFID tags simultaneously return a 920-925MHz signal to the reader's receiving antenna.

[0010] The reader / writer includes: a baseband signal processing module and an RF front-end. The baseband signal processing module includes: a downsampling module, a bandpass filter (BLF), an FFT module, a power estimation module, an amplitude estimation module, a SIR calculation and sorting module, an RF phase compensation module, a Costas loop, a decoding module, and a signal reconstruction module. The RF front-end includes: a quadrature tuner and an ADC.

[0011] S2. The four RFID tags respectively use M values ​​of 4, 8, 12, and 16 to perform subcarrier modulation on their respective Miller codes to form Miller subcarrier modulation sequences.

[0012] After the reader sends the inventory command, the four RFID tags communicate with the reader to send the Miller subcarrier modulation sequence; the bandwidth of each frequency channel is the first zero-point bandwidth of the RFID tag signal, which is 4 times the data rate of the tag baseband signal.

[0013] S3. After the reader receives the signals returned by four RFID tags at the same time, it first performs down-conversion and ADC sampling through the radio frequency front-end to convert the analog signal into a digital signal.

[0014] S4. Estimate the power of the total signal received by the reader using FFT;

[0015] First, estimate the signal power in each frequency band. Using the power spectral density of Miller subcarrier modulation sequences with M values ​​of 4, 8, 12, and 16 derived from the Wiener-Khinchin theorem, the power of each Miller subcarrier modulation sequence in different frequency bands is obtained through numerical integration. Then, the interference level H generated by tags in other channels within different frequency channels is obtained. Finally, an estimate of the actual first null bandwidth power of each tag is obtained. Complete power estimation.

[0016] Here, H is a 4×4 matrix, and the element h in the i-th row and j-th column is... ij This represents the ratio of the power of the RFID tag located in the j-th frequency channel in the i-th frequency channel to the first null-point bandwidth power of the tag in the j-th frequency channel, and

[0017] S5. Based on step S4, perform amplitude estimation and radio frequency phase estimation according to Parseval's theorem;

[0018] S6. Based on step S4, after completing the power estimation, calculate the SIR of each tag and sort the SIRs.

[0019] S7. Based on step S6, first perform SIC demodulation on the tag with the largest SIR.

[0020] First, the signal passes through an FIR bandpass filter to remove out-of-band interference. Then, it passes through an RF phase compensation module to cancel out the influence of RF phase. The signal is then input into a Costas loop to obtain the tag's subcarrier signal and Miller code. The signal is then input into a decoding module to make a decision on the Miller code and obtain the tag's original data. Finally, the signal is input into a signal reconstruction module to obtain the reconstructed tag signal, which is the reconstructed Miller subcarrier modulation sequence of the tag.

[0021] Costas generates a sinusoidal signal with the same frequency and phase as the original Miller subcarrier in an internal numerically controlled oscillator, and outputs a Miller-encoded waveform through an internal low-pass filter. The Miller-encoded waveform is then polarized to obtain the Miller code.

[0022] S8. Based on step S7, repeat the demodulation process until the last tag signal is demodulated, thus completing the parallel reception of RFID tags.

[0023] Based on the reconstructed tag signal obtained by SIC demodulation of the tag with the largest SIR in step S7, the reconstructed signal is divided into I and Q paths using the estimated radio frequency phase. The reconstructed tag signal is subtracted from both I and Q paths. Then, the tag signal with the second largest SIR is demodulated. The demodulation process is repeated until the last tag signal is demodulated, thus completing the final reception process, i.e., completing the parallel reception of RFID tags.

[0024] Furthermore, the M value used for the four RFID tags is not limited to 4, 8, 12, or 16, and can be expanded according to actual needs.

[0025] Furthermore, step S3 is specifically as follows:

[0026] The down-converted tag signal X received by the reader's receiving antenna RF The expression for (t) is as follows:

[0027]

[0028] Where t represents time, T i (t)=M i (t)Square i (2πf i t) represents the Miller subcarrier modulation sequence of tag i, where tags i = 1, 2, 3, 4 represent tags with M values ​​of 4, 8, 12, and 16, respectively. i (t) represents the baseband Miller code, Square i (2πf i t) represents a square wave signal, a i , t i , and f i These represent the amplitude, radio frequency phase, transmission delay, and subcarrier frequency of tag i, respectively. N(t) represents additive white Gaussian noise.

[0029] Then X RF The expression for the received complex signal X(n) after ADC sampling is as follows:

[0030]

[0031] Where n represents a discrete time series, f s Let X(n) represent the ADC sampling frequency; and since the reader receiver uses quadrature demodulation, the I and Q components of X(n) are rewritten as follows:

[0032]

[0033] Among them, two real factors and This includes the effects of radio frequency phase.

[0034] Furthermore, step S4 is specifically as follows:

[0035] The I and Q digital signals sampled by the ADC in step S3 are used for power estimation via FFT. First, the power spectral density (PSD) of the Miller subcarrier modulation sequences with M values ​​of 4, 8, 12, and 16, derived from the Wiener-Khinchin theorem, is used. The PSD estimation expression is as follows:

[0036]

[0037] Among them, D I (n), D Q (n) represent the estimated power spectral densities of the I and Q paths, respectively. denoted as Fourier transform, and N represents the length of the Fourier transform.

[0038] In the FFT module, the reader calculates the power estimates for the four frequency channels using FFT.

[0039]

[0040] in,[·] T This indicates a transpose operation. The power estimate includes the first null bandwidth signal of the tag corresponding to each channel and the sidelobe or harmonic interference of the tags of the other three channels in that frequency channel.

[0041] For path I, the power estimate entering the i-th channel is obtained by numerically integrating the power spectral density estimated by the FFT in the i-th channel. set up This represents the estimated actual power of the i-th tag in the first zero-point bandwidth. Then... and The following relationship exists:

[0042]

[0043] in, This represents an estimate of the actual power of the tag within the first zero-point bandwidth across the four frequency channels of the I-path. H represents the linear expression of tag interference in different channels, where the element h in the i-th row and j-th column is... ij This represents the ratio of the power of the tag located in the j-th frequency channel in the i-th frequency channel to the first null-point bandwidth power of the tag in the j-th frequency channel; and This represents the power estimate across the four frequency channels of path I.

[0044] Similarly, the power estimation for the i-th channel in the Q-path can be obtained. Estimation of the actual power of the i-th tag in the first zero-point bandwidth

[0045] In summary, the bandwidth power estimates of the first zero point for each tag distributed across the I and Q channels are calculated. Power estimates for each frequency channel in both I and Q paths are obtained.

[0046] Furthermore, step S5 is specifically as follows:

[0047] The first zero-point bandwidth power of the tag accounts for 0.77% of the total Miller subcarrier modulated signal power. According to Parseval's theorem, the time-domain power and the frequency-domain power are equal, therefore the amplitude value estimated by path I is... Similarly, the amplitude value of the Q-path estimation can be obtained.

[0048] Then, the estimated amplitudes from both the I and Q channels are used to estimate the RF phase. The calculation method is as follows

[0049] Furthermore, step S6 is specifically as follows:

[0050] After power estimation is completed, the SIR of each sub-channel tag i is estimated using the SIR calculation and sorting module. The calculation expression for the SIR of tag i is as follows:

[0051]

[0052] After the reader calculates the SIR of each tag, based on the power estimate of each channel obtained in step S5, it determines whether the power value in each channel is lower than the preset threshold. If it is lower, the channel is considered to have no signal and will not be considered in the subsequent SIC demodulation process. Then, the SIR of the remaining channels is sorted and demodulated in descending order of SIR. During demodulation, the reader first selects the tag with the largest SIR for SIC demodulation.

[0053] Furthermore, the reader performs SIC demodulation on the current channel, reconstructing the tag signal. If the reader detects a bit error in the tag signal, when proceeding to the next-level channel for SIC demodulation, there are two scenarios: subtracting the reconstructed signal from the remaining total signal or not subtracting the reconstructed signal. The reader compares whether bit errors occur in the SIC demodulation of the next channel under these two scenarios. If no bit error occurs under either scenario, the current tag signal continues to be reconstructed for subsequent SIC demodulation. If bit errors occur under both scenarios, SIC demodulation stops.

[0054] Furthermore, in step S7, the demodulation of the tag is specifically performed as follows:

[0055] For each sub-channel label i, a bandpass filter (BPF) is used to filter out all out-of-band spectral components, followed by RF phase compensation. The resulting digital signal X at the Costas loop input is then... i The expression for (n) is as follows:

[0056]

[0057] The signal input to the Costas ring is the signal after RF phase compensation, BPF i This indicates that the signal passes through a digital FIR bandpass filter.

[0058] The Costas ring includes a phase detector, a loop filter, and a digitally controlled oscillator. The Costas ring performs demodulation, carrier phase locking, and carrier frequency multiplication of the BPSK signal through two loops. The Costas ring's function is to first demodulate the tag subcarrier modulation sequence, removing the subchannel carrier f... i The signal is filtered using a low-pass filter (LPF) to remove 2f. i The spectrum is then analyzed, and bit determination is performed according to the zero-threshold decoding rule.

[0059] The phase detector employs a multiplicative phase detector cascaded with an FIR low-pass filter structure. The sign of the I-channel phase detector result multiplied by the Q-channel phase detector result is used as the phase detection result input to the loop filter. The frequency control word of the numerically controlled oscillator (NCO) is based on the initial control word with the loop filter output added. The NCO generates primary carrier sine and cosine carrier signals through the frequency control word and generates a tertiary carrier cosine signal through the triple frequency control word. The primary carrier sine and cosine signals are used as inputs to the phase detector multiplication terminal, and the tertiary carrier cosine signal is used as the module output for harmonic interference suppression.

[0060] Then reconstruct the reader to receive input X. i The tag signal at (n) is used in the loop to generate the sine wave sin[2πf] to be synchronized. i (n / f s -t i The sine wave has the same frequency and phase as the subcarrier of the Miller subcarrier modulation sequence. Therefore, the reconstructed Miller code M can be obtained by taking the polarity of the low-pass filter of the Costas ring. i (n / f s -t i ), sin[2πf i (n / f s -t i After determining the polarity, the reconstructed subcarrier signal Square can be obtained. i [2πf i (n / fs -t i )).

[0061] If the reconstructed signal is in square wave form, then a method for signal reconstruction using a series of odd harmonics is proposed, namely, the square wave signal Square(2πf) s The Fourier expansion of t is as follows:

[0062]

[0063] The reconstructed Miller subcarrier is obtained by fitting a square wave with a finite number of odd-order sinusoidal harmonics, as shown in the following expression:

[0064]

[0065] in, and This represents the delay and frequency of the subcarrier obtained through synchronization with the Costas ring.

[0066] Finally, the sinusoidal signal output from the Costas ring is harmonicized by 3, 5, and 7, and then superimposed with each odd harmonic component to fit the Miller subcarrier signal. Multiplying the Miller subcarrier signal with the Miller code yields the reconstructed tag signal. The reconstructed signal expression for tag i is as follows:

[0067]

[0068] The beneficial effects of this invention are as follows: The method of this invention first constructs a frequency division multiplexing UHF RFID system. Tags use M values ​​of 4, 8, 12, and 16 to modulate their respective Miller codes with subcarriers, achieving different reverse link frequencies (BLF), thereby realizing frequency division multiplexing of tag signals. Then, based on the Wiener-Khinchin theorem, the power spectral density of the Miller subcarrier modulation sequences with M values ​​of 4, 8, 12, and 16 is derived. The interference level generated by tags in other channels within different frequency channels is quantitatively analyzed. The power of the superimposed signal in each frequency band is estimated by FFT. Then, using the known interference level, the amplitude and radio frequency phase of each tag are estimated. During reception, the reader uses FFT to calculate the signal-to-interference ratio (SIR) of each signal and sorts them. The tag with the largest SIR is demodulated first. During demodulation, the subcarrier and Miller code of the tag are reconstructed using a Costas ring, and the reconstructed signal is subtracted from the total signal. Then, the tag signal with the second largest SIR is demodulated. The demodulation process is repeated until the last tag signal is demodulated. The method of this invention enables parallel reception of frequency division multiplexing RFID tags, allowing the reader to store multiple tags simultaneously within a single time slot. This effectively improves the identification efficiency and throughput of the UHF RFID system, enhancing its ability to adapt to application scenarios requiring rapid access to a massive number of tags in large-scale Internet of Things (IoT) applications. Attached Figure Description

[0069] Figure 1 This is a flowchart of a parallel reception method for frequency division multiplexing RFID tag signals based on the SIC architecture according to the present invention.

[0070] Figure 2 This is a schematic diagram of the frequency division multiplexing UHF RFID system structure in an embodiment of the present invention.

[0071] Figure 3 This is a block diagram of a parallel receiving frequency division multiplexing tag signal reader / writer based on the SIC architecture in an embodiment of the present invention.

[0072] Figure 4 This is a schematic diagram of the power spectral density (PSD) of the Miller subcarrier modulation sequence with a reverse link frequency (BLF) of 80kHz when the M value is 4 in an embodiment of the present invention.

[0073] Figure 5 This is a flowchart of the frequency domain channel collision solution in an embodiment of the present invention.

[0074] Figure 6 This is a schematic diagram of the radio frequency phase compensation module, Costas ring, and signal reconstruction module in the parallel receiving block diagram of the reader in this embodiment of the invention. Detailed Implementation

[0075] The method of the present invention will be further described below with reference to the accompanying drawings and embodiments.

[0076] like Figure 1 The flowchart shown below illustrates a parallel reception method for frequency division multiplexing RFID tag signals based on the SIC architecture of the present invention. The specific steps are as follows:

[0077] S1, such as Figure 2 As shown, a frequency division multiplexing UHF RFID system is constructed, including: 4 RFID tags and 1 reader / writer;

[0078] The reader activates four RFID tags by sending a 920-925MHz tag excitation signal through its transmitting antenna. After receiving the inventory command from the reader, the four RFID tags simultaneously return a 920-925MHz signal to the reader's receiving antenna.

[0079] like Figure 3 As shown, the reader / writer includes: a baseband signal processing module and an RF front-end. The baseband signal processing module includes: a downsampling module, a bandpass filter (BLF), an FFT module, a power estimation module, an amplitude estimation module, a SIR calculation and sorting module, an RF phase compensation module, a Costas loop, a decoding module, and a signal reconstruction module. The RF front-end includes: a quadrature tuner and an ADC.

[0080] S2. The four RFID tags respectively use M values ​​of 4, 8, 12, and 16 to perform subcarrier modulation on their respective Miller codes to form Miller subcarrier modulation sequences.

[0081] After the reader sends the inventory command, the four RFID tags communicate with the reader to send the Miller subcarrier modulation sequence; the bandwidth of each frequency channel is the first zero-point bandwidth of the RFID tag signal, which is 4 times the data rate of the tag baseband signal.

[0082] S3. After the reader receives the signals returned by four RFID tags at the same time, it first performs down-conversion and ADC sampling through the radio frequency front-end to convert the analog signal into a digital signal.

[0083] S4. Estimate the power of the total signal received by the reader using FFT;

[0084] First, estimate the signal power in each frequency band. Using the power spectral density of Miller subcarrier modulation sequences with M values ​​of 4, 8, 12, and 16 derived from the Wiener-Khinchin theorem, the power of each Miller subcarrier modulation sequence in different frequency bands is obtained by numerical integration. Then, the interference level H generated by tags in other channels within different frequency channels is obtained. Finally, the estimated value of the actual first zero-point bandwidth power of each tag is obtained. Complete power estimation.

[0085] Here, H is a 4×4 matrix, and the element h in the i-th row and j-th column is... ij This represents the ratio of the power of the RFID tag located in the j-th frequency channel in the i-th frequency channel to the first null-point bandwidth power of the tag in the j-th frequency channel, and

[0086] S5. Based on step S4, perform amplitude estimation and radio frequency phase estimation according to Parseval's theorem;

[0087] S6. Based on step S4, after completing the power estimation, calculate the SIR of each tag and sort the SIRs.

[0088] S7. Based on step S6, first perform SIC demodulation on the tag with the largest SIR.

[0089] First, the signal passes through an FIR bandpass filter to remove out-of-band interference. Then, it passes through an RF phase compensation module to cancel out the influence of RF phase. The signal is then input into a Costas loop to obtain the tag's subcarrier signal and Miller code. The signal is then input into a decoding module to make a decision on the Miller code and obtain the tag's original data. Finally, the signal is input into a signal reconstruction module to obtain the reconstructed tag signal, which is the reconstructed Miller subcarrier modulation sequence of the tag.

[0090] Costas generates a sinusoidal signal with the same frequency and phase as the original Miller subcarrier in an internal numerically controlled oscillator, and outputs a Miller-encoded waveform through an internal low-pass filter. The Miller-encoded waveform is then polarized to obtain the Miller code.

[0091] S8. Based on step S7, repeat the demodulation process until the last tag signal is demodulated, thus completing the parallel reception of RFID tags.

[0092] Based on the reconstructed tag signal obtained by SIC demodulation of the tag with the largest SIR in step S7, the reconstructed signal is divided into I and Q paths using the estimated radio frequency phase. The reconstructed tag signal is subtracted from both I and Q paths. Then, the tag signal with the second largest SIR is demodulated. The demodulation process is repeated until the last tag signal is demodulated, thus completing the final reception process, i.e., completing the parallel reception of RFID tags.

[0093] In this embodiment, the M value used for the four RFID tags is not limited to 4, 8, 12, and 16, but can be expanded according to the actual situation, as shown in Table 1.

[0094] To facilitate parallel reception of data returned by frequency division multiplexing (FDM) UHF RFID tags by the reader / writer, this embodiment divides the channels based on different M values ​​and BLF values ​​at the same data rate. The ISO / IEC 18000-6C standard specifies that the tag encoding BLF should be in the range of 40–640 kHz. Therefore, except for the data rate of 110 kbps, the BLF of all other channels falls within this range. Furthermore, interference between adjacent channels should be minimized during channel division. The channel division results are shown in Table 1.

[0095] Table 1

[0096]

[0097] In Table 1, the data rate unit is kbps, and the BLF unit is kHz.

[0098] As shown in Table 1, the number of channels available for a tag within a time slot can be 2, 3, 4, 5, or 8. Specifically, when the data rate is 5kbps, 10kbps, or 20kbps, there are 8 channels available for the tag to choose from within a time slot. When there is only one channel in a time slot, the tag complies with the ISO / IEC 18000-6C standard protocol. Although harmonic interference exists on some channels, the time required to suppress harmonic interference is less than the time required to re-identify the tag. Furthermore, when the harmonic interference signal is much smaller than the useful signal, the interfered tag can be correctly identified even without harmonic interference. Therefore, considering all factors, channels with harmonic interference can be included in the channel allocation process.

[0099] In this embodiment, step S3 is specifically as follows:

[0100] The down-converted tag signal X received by the reader's receiving antenna RF The expression for (t) is as follows:

[0101]

[0102] Where t represents time, Ti (t)=M i (t)Square i (2πf i t) represents the Miller subcarrier modulation sequence of tag i, where tags i = 1, 2, 3, 4 represent tags with M values ​​of 4, 8, 12, and 16, respectively. i (t) represents the baseband Miller code, Square i (2πf i t) represents a square wave signal, a i , , t i , and f i These represent the amplitude, radio frequency phase, transmission delay, and subcarrier frequency of tag i, respectively. N(t) represents additive white Gaussian noise.

[0103] Then X RF The expression for the received complex signal X(n) after ADC sampling is as follows:

[0104]

[0105] Where n represents a discrete time series, f s Let X(n) represent the ADC sampling frequency; and since the reader receiver uses quadrature demodulation, the I and Q components of X(n) are rewritten as follows:

[0106]

[0107] Among them, two real factors and This includes the effects of radio frequency phase.

[0108] In this embodiment, step S4 is specifically as follows:

[0109] The I and Q digital signals sampled by the ADC in step S3 are used for power estimation via FFT. First, the power spectral density (PSD) of the Miller subcarrier modulation sequences with M values ​​of 4, 8, 12, and 16, derived using the Wiener-Khinchin theorem, is used. Under the assumption of a stationary signal, FFT provides a good approximation of the received signal PSD (using a rectangular window). The PSD estimation calculation expression is as follows:

[0110]

[0111] Among them, D I (n), D Q (n) represent the estimated power spectral densities of the I and Q paths, respectively. denoted as Fourier transform, and N represents the length of the Fourier transform.

[0112] Because of the tag backscatter communication mechanism, it is impossible to filter and shape the reflected signal to suppress sidelobe / harmonic interference, so its reflected and scattered signal has a wide bandwidth. Figure 4 The power spectral density of the Miller subcarrier modulation sequence with an M value of 4 and a reverse link frequency of 80 kHz. Figure 4 Tag distribution in the sidelobes or harmonic components of other frequency channels can cause interference. The Miller subcarrier modulation sequence belongs to a first-order Markov random process, and the power spectral densities D4, D8, and D of Miller subcarrier modulation sequences with M values ​​of 4, 8, 12, and 16 can be derived based on the Wiener-Khinchin theorem. 12 D 16 The interference level generated by tags in other channels within different frequency channels was quantitatively analyzed, specifically the interference on D4, D8, and D... 12 D 16 The power of each Miller subcarrier modulation sequence in different frequency bands was obtained by numerical integration, as shown in Table 2.

[0113] Table 2

[0114] Channel Tag 1 Tag 2 Tag 3 Tag 4 Sum 1 -12.3dB -17.6dB -20.6dB -10.7dB 2 -24.6dB, -13.4dB, -18.8dB -12.1dB 3 -9.8dB -20.0dB -14.0dB -8.1dB 4 -31.0dB -37.8dB -18.4dB -18.1dB

[0115] In the FFT module, the reader calculates the power estimates for the four frequency channels using FFT.

[0116]

[0117] in,[·] T This indicates a transpose operation. The power estimate includes the first null bandwidth signal of the tag corresponding to each channel and the sidelobe or harmonic interference of the tags of the other three channels in that frequency channel.

[0118] For path I, the power estimate entering the i-th channel is obtained by numerically integrating the power spectral density estimated by the FFT in the i-th channel. set up This represents the estimated actual power of the i-th tag in the first zero-point bandwidth. Then... and The following relationship exists:

[0119]

[0120] in, This represents an estimate of the actual power of the tag within the first zero-point bandwidth across the four frequency channels of the I-path. H represents the linear expression of tag interference in different channels (obtained by converting the values ​​in Table 1 into a linear expression), where the element h in the i-th row and j-th column is... ijThis represents the ratio of the power of the tag located in the j-th frequency channel in the i-th frequency channel to the first null-point bandwidth power of the tag in the j-th frequency channel; and This represents the power estimate across the four frequency channels of path I.

[0121] Similarly, the power estimation for the i-th channel in the Q-path can be obtained. The estimate of the actual power of the i-th label in the first zero-point bandwidth

[0122] In summary, the bandwidth power estimates of the first zero point for each tag distributed across the I and Q channels are calculated. Power estimates for each frequency channel in both I and Q paths are obtained.

[0123] In this embodiment, step S5 is specifically as follows:

[0124] The first zero-point bandwidth power of the tag accounts for 0.77% of the total Miller subcarrier modulated signal power. According to Parseval's theorem, the time-domain power and the frequency-domain power are equal, therefore the amplitude value estimated by path I is... Similarly, the amplitude value of the Q-path estimation can be obtained.

[0125] Then, the estimated amplitudes from both the I and Q channels are used to estimate the RF phase. The calculation method is as follows

[0126] In this embodiment, step S6 is specifically as follows:

[0127] The interference of tags in different channels, as shown in Table 2, makes the demodulation of frequency division multiplexing tag signals very challenging. Therefore, this embodiment proposes using the SIC mechanism at the reader receiver for parallel demodulation of multi-tag signals. Figure 3 This is a block diagram of the SIC mechanism in the reader receiver. After power estimation, the SIR of each sub-channel tag i is estimated through the SIR calculation and sorting module. The calculation expression for the SIR of tag i is as follows:

[0128]

[0129] In a frequency division multiplexing (FDM) UHF RFID system, tags randomly select different time slots and M values ​​to communicate with the reader. If multiple tags select the same M value within the same time slot, frequency domain collisions will occur. This embodiment proposes a frequency domain channel collision solution, the flowchart of which is shown below. Figure 5As shown. After calculating the SIR of each tag, the reader determines whether the power value in each channel is lower than the preset threshold based on the power estimate of each channel obtained in step S5. If it is lower, the channel is considered to have no signal and will not be considered in the subsequent SIC demodulation process. Then, the SIRs of the remaining channels are sorted and demodulated in descending order of SIR. During demodulation, the reader first selects the tag with the largest SIR for SIC demodulation.

[0130] In this embodiment, since the reader cannot calculate the bit error rate of each tag when receiving tag signals, the reader adopts the following mechanism during reception:

[0131] The reader demodulates the current channel using SIC, reconstructing the tag signal. If the reader detects a bit error in the tag signal (CRC check fails), it proceeds to the next channel for SIC demodulation. There are two scenarios: subtracting the reconstructed signal from the remaining total signal or not subtracting it. The reader compares whether bit errors occur in the next channel's SIC demodulation under these two scenarios. If no bit error occurs in either scenario, the current tag signal continues to be reconstructed for subsequent SIC demodulation. If bit errors occur in both scenarios, SIC demodulation stops.

[0132] In this embodiment, the demodulation of the tag in step S7 is as follows:

[0133] The structure of tag Miller subcarrier modulation sequence signal demodulation and tag signal reconstruction is as follows: Figure 6 As shown. For each sub-channel label i, a bandpass filter (BPF) is used to filter out all out-of-band spectral components, followed by RF phase compensation (RF phase compensation is to counteract the effect of RF phase imbalance on the I and Q channels). The digital signal X at the Costas loop input is then... i The expression for (n) is as follows:

[0134]

[0135] The signal input to the Costas ring is the signal after RF phase compensation, BPF i This indicates that the signal passes through a digital FIR bandpass filter. Considering that the output of the low-pass filter is a baseband Miller-coded waveform, and a strictly linear phase characteristic is required within the passband, an FIR low-pass filter is used, whose passband bandwidth is the first zero-point bandwidth of the tag.

[0136] Costas rings are widely used for tracking carrier-suppressed signals with low signal-to-noise ratios. A Costas ring includes a phase detector, a loop filter, and a digitally controlled oscillator. The Costas ring performs demodulation, carrier phase locking, and carrier frequency multiplication of the BPSK signal through two loops, as described below. Figure 6 As shown. Furthermore, the Costas ring's function is to first demodulate the tag subcarrier modulation sequence, removing the subchannel carrier f first. i The signal is filtered using a low-pass filter (LPF) to remove 2f. i The spectrum is then analyzed, and bit determination is performed according to the zero-threshold decoding rule.

[0137] To further reduce overhead, the phase detector does not use a traditional multiplier, but instead employs a signed decision phase detection algorithm, namely a multiplicative phase detector connected in series with an FIR low-pass filter structure. The polarity of the baseband signal output from the in-phase branch controls whether the low-pass filter output of the quadrature branch flips. The sign of the I-channel phase detector result multiplied by the Q-channel phase detector result is used as the phase detection result input to the loop filter, which is a second-order active proportional loop filter. The frequency control word of the numerically controlled oscillator (NCO) is based on the initial control word plus the loop filter output. The NCO is implemented using a Direct Digital Synthesis (DDS) signal generator method, generating primary carrier sine and cosine carrier signals through the frequency control word, and generating a tertiary carrier cosine signal through the triple frequency control word. The primary carrier sine and cosine signals are used as inputs to the phase detector multiplier, and the tertiary carrier cosine signal is used as the module output for harmonic interference suppression.

[0138] Then reconstruct the reader to receive input X. i The tag signal at (n) is used in the loop to generate the sine wave sin[2πf] to be synchronized. i (n / f s -t i The sine wave has the same frequency and phase as the subcarrier of the Miller subcarrier modulation sequence. Therefore, the reconstructed Miller code M can be obtained by taking the polarity of the low-pass filter of the Costas ring. i (n / f s -t i ), sin[2πf i (n / f s -t i After determining the polarity, the reconstructed subcarrier signal Square can be obtained. i [2πf i (n / f s -t i )).

[0139] Because the reconstructed signal is in square wave form, it deviates significantly from the original signal at the frequency hopping edges. After interference suppression, spikes appear in the remaining signal. Therefore, to suppress interference, it is not advisable to reconstruct the original signal into square wave form. This embodiment proposes a novel method for signal reconstruction using a series of odd harmonics, namely, a square wave signal Square(2πf s The Fourier expansion of t is as follows:

[0140]

[0141] As shown in equation (8), a square wave signal can be represented as the sum of an infinite number of odd-order sine waves, with amplitudes of 1, 1 / 3, 1 / 5, 1 / 7, etc., and frequencies that are odd multiples of the fundamental frequency. Since a sinusoidal modulated signal can be represented as a spectral shift of a baseband signal, the Miller subcarrier modulation sequence is equivalent to the sum of an infinite number of odd-order sine waves modulated by the baseband Miller code. Therefore, a square wave can be fitted with a finite number of odd-order sine harmonics to obtain the reconstructed Miller subcarrier, as expressed below:

[0142]

[0143] in, and This represents the time delay and frequency of the subcarrier obtained through synchronization with the Costas ring. The sinusoidal signal output from the Costas ring is multiplied by 3, 5, and 7. This frequency multiplication is achieved by multiplying the frequency control word of the input numerically controlled oscillator by 3, 5, and 7, respectively. The sinusoidal signal is then superimposed with each odd harmonic component to fit the Miller subcarrier signal. Multiplying the Miller subcarrier signal by the Miller code yields the reconstructed tag signal. Therefore, the reconstructed signal expression for tag i is as follows:

[0144]

[0145] In summary, the method of the present invention enables parallel reception of frequency division multiplexing RFID tags, allowing the reader to simultaneously store multiple tags within a single time slot. This effectively improves the identification efficiency and throughput of the UHF RFID system, enhancing its ability to adapt to application scenarios requiring rapid access to a massive number of tags in large-scale Internet of Things (IoT) applications.

[0146] Those skilled in the art will recognize that the embodiments described herein are intended to help the reader understand the principles of the invention, and should be understood that the scope of protection of the invention is not limited to such specific statements and embodiments. Those skilled in the art can make various other specific modifications and combinations based on the technical teachings disclosed in this invention without departing from the spirit of the invention, and these modifications and combinations are still within the scope of protection of this invention.

Claims

1. A method for parallel reception of frequency division multiplexing RFID tag signals based on SIC architecture, the specific steps of which are as follows: S1. Construct a frequency division multiplexing UHF RFID system, including: Four RFID tags and one reader / writer; The reader activates four RFID tags by sending a 920-925MHz tag excitation signal through its transmitting antenna. After receiving the inventory command from the reader, the four RFID tags simultaneously return a 920-925MHz signal to the reader's receiving antenna. The reader / writer includes: a baseband signal processing module and an RF front-end; the baseband signal processing module includes: a downsampling module, a bandpass filter (BLF), an FFT module, a power estimation module, an amplitude estimation module, an SIR calculation and sorting module, an RF phase compensation module, a Costas loop, a decoding module, and a signal reconstruction module; the RF front-end includes: a quadrature tuner and an ADC; S2. The four RFID tags respectively use M values ​​of 4, 8, 12, and 16 to perform subcarrier modulation on their respective Miller codes to form Miller subcarrier modulation sequences. After the reader sends the inventory command, the four RFID tags communicate with the reader to send the Miller subcarrier modulation sequence; the bandwidth of each frequency channel is the first zero-point bandwidth of the RFID tag signal, which is 4 times the data rate of the tag baseband signal. S3. After the reader receives the signals returned by four RFID tags at the same time, it first performs down-conversion and ADC sampling through the radio frequency front-end to convert the analog signal into a digital signal. S4. Estimate the power of the total signal received by the reader using FFT; First, estimate the signal power in each frequency band. Using the power spectral density of Miller subcarrier modulation sequences with M values ​​of 4, 8, 12, and 16 derived from the Wiener-Khinchin theorem, the power of each Miller subcarrier modulation sequence in different frequency bands is obtained through numerical integration. Then, the interference level H generated by tags in other channels within different frequency channels is obtained. Finally, an estimate of the actual first null bandwidth power of each tag is obtained. Complete power estimation; Here, H is a 4×4 matrix, and the element h in the i-th row and j-th column is... ij This represents the ratio of the power of the RFID tag located in the j-th frequency channel in the i-th frequency channel to the first null-point bandwidth power of the tag in the j-th frequency channel, and S5. Based on step S4, perform amplitude estimation and radio frequency phase estimation according to Parseval's theorem; S6. Based on step S4, after completing the power estimation, calculate the SIR of each tag and sort the SIRs. S7. Based on step S6, first perform SIC demodulation on the tag with the largest SIR. First, the signal passes through an FIR bandpass filter to remove out-of-band interference. Then, it passes through an RF phase compensation module to cancel out the influence of RF phase. The signal is then input into a Costas loop to obtain the tag's subcarrier signal and Miller code. The signal is then input into a decoding module to make a decision on the Miller code and obtain the tag's original data. Finally, the signal is input into a signal reconstruction module to obtain the reconstructed tag signal, which is the reconstructed Miller subcarrier modulation sequence of the tag. Costas generates a sinusoidal signal with the same frequency and phase as the original Miller subcarrier in an internal numerically controlled oscillator, and outputs a Miller-encoded waveform through an internal low-pass filter. The Miller-encoded waveform is then polarized to obtain the Miller code. S8. Based on step S7, repeat the demodulation process until the last tag signal is demodulated, thus completing the parallel reception of RFID tags. Based on the reconstructed tag signal obtained by SIC demodulation of the tag with the largest SIR in step S7, the reconstructed signal is divided into I and Q paths using the estimated radio frequency phase. The reconstructed tag signal is subtracted from both I and Q paths. Then, the tag signal with the second largest SIR is demodulated. The demodulation process is repeated until the last tag signal is demodulated, thus completing the final reception process, i.e., completing the parallel reception of RFID tags. Step S3 is as follows: The down-converted tag signal X received by the reader's receiving antenna RF The expression for (t) is as follows: Where t represents time, T i (t)=M i (t)Square i (2πf i t) represents the Miller subcarrier modulation sequence of tag i, where tags i = 1, 2, 3, 4 represent tags with M values ​​of 4, 8, 12, and 16, respectively. i (t) represents the baseband Miller code, Square i (2πf i t) represents a square wave signal, a i , t i , and f i These represent the amplitude, radio frequency phase, transmission delay, and subcarrier frequency of tag i, respectively; N(t) represents additive white Gaussian noise. Then X RF The expression for the received complex signal X(n) after ADC sampling is as follows: Where n represents a discrete time series, f s Let X(n) represent the ADC sampling frequency; and since the reader receiver uses quadrature demodulation, the I and Q components of X(n) are rewritten as follows: Among them, two real factors and Including the effects of radio frequency phase; Step S4 is as follows: The I and Q digital signals sampled by the ADC in step S3 are used for power estimation via FFT. First, the power spectral density (PSD) of the Miller subcarrier modulation sequences with M values ​​of 4, 8, 12, and 16, derived from the Wiener-Khinchin theorem, is used. The PSD estimation expression is as follows: Among them, D I (n), D Q (n) represent the estimated power spectral densities of the I and Q paths, respectively. This represents the Fourier transform, where N represents the length of the Fourier transform. In the FFT module, the reader calculates the power estimates for the four frequency channels using FFT. in,[·] T This indicates a transpose operation, and the power estimate includes the first null bandwidth signal of the tag corresponding to each channel and the sidelobe or harmonic interference of the tags of the other three channels in that frequency channel; For path I, the power estimate entering the i-th channel is obtained by numerically integrating the power spectral density estimated by the FFT in the i-th channel. set up Let represent the estimated actual power of the i-th tag in the first zero-point bandwidth; then and The following relationship exists: in, This represents an estimate of the actual power of the tag within the first zero-point bandwidth across the four frequency channels of the I-path. H represents the linear expression of tag interference in different channels, where the element h in the i-th row and j-th column is... ij This represents the ratio of the power of the tag located in the j-th frequency channel in the i-th frequency channel to the first null-point bandwidth power of the tag in the j-th frequency channel; and This represents the power estimate across the four frequency channels of path I; Similarly, the power estimation for the i-th channel in the Q-path can be obtained. Estimation of the actual power of the i-th tag in the first zero-point bandwidth In summary, the bandwidth power estimates of the first zero point for each tag distributed across the I and Q channels are calculated. Power estimates for each frequency channel in both I and Q paths are obtained. Step S5 is as follows: The first zero-point bandwidth power of the tag accounts for 0.77% of the total Miller subcarrier modulated signal power. According to Parseval's theorem, the time-domain power and the frequency-domain power are equal, therefore the amplitude value estimated by path I is... Similarly, the amplitude value of the Q-path estimation can be obtained. Then, the estimated amplitudes from both the I and Q channels are used to estimate the RF phase. The calculation method is as follows Step S6 is as follows: After power estimation is completed, the SIR of each sub-channel tag i is estimated using the SIR calculation and sorting module. The calculation expression for the SIR of tag i is as follows: After the reader calculates the SIR of each tag, based on the power estimate of each channel obtained in step S5, it determines whether the power value in each channel is lower than the preset threshold. If it is lower, the channel is considered to have no signal and will not be considered in the subsequent SIC demodulation process. Then, the SIR of the remaining channels is sorted and demodulated in descending order of SIR. During demodulation, the reader first selects the tag with the largest SIR for SIC demodulation. In step S7, the demodulation of the tag is performed as follows: For each sub-channel label i, a bandpass filter (BPF) is used to filter out all out-of-band spectral components, followed by RF phase compensation. The resulting digital signal X at the Costas loop input is then... i The expression for (n) is as follows: The signal input to the Costas ring is the signal after RF phase compensation, BPF i This indicates that the signal passes through a digital FIR bandpass filter; The Costas ring includes a phase detector, a loop filter, and a digitally controlled oscillator. The Costas ring performs demodulation, carrier phase locking, and carrier frequency multiplication of the BPSK signal through two loops. The Costas ring's function is to first demodulate the tag subcarrier modulation sequence, removing the subchannel carrier f... i The signal is filtered using a low-pass filter (LPF) to remove 2f. i The spectrum is then analyzed, and bit determination is performed according to the zero-threshold decoding rule. The phase detector employs a multiplicative phase detector cascaded with an FIR low-pass filter structure. The sign of the I-channel phase detector result multiplied by the Q-channel phase detector result is used as the phase detection result input to the loop filter. The frequency control word of the numerically controlled oscillator (NCO) is based on the initial control word with the loop filter output added. The NCO generates primary carrier sine and cosine carrier signals through the frequency control word and generates a tertiary carrier cosine signal through the triple frequency control word. The primary carrier sine and cosine signals are used as inputs to the phase detector multiplication terminal, and the tertiary carrier cosine signal is used as the module output for harmonic interference suppression. Then reconstruct the reader to receive input X. i The tag signal at (n) is used in the loop to generate the sine wave sin[2πf] to be synchronized. i (n / f s -t i The sine wave has the same frequency and phase as the subcarrier of the Miller subcarrier modulation sequence. Therefore, the reconstructed Miller code M can be obtained by taking the polarity of the low-pass filter of the Costas ring. i (n / f s -t i ), sin[2πf i (n / f s -t i After determining the polarity, the reconstructed subcarrier signal Square can be obtained. i [2πf i (n / f s -t i )]; If the reconstructed signal is in square wave form, then a method for signal reconstruction using a series of odd harmonics is proposed, namely, the square wave signal Square(2πf) s The Fourier expansion of t is as follows: The reconstructed Miller subcarrier is obtained by fitting a square wave with a finite number of odd-order sinusoidal harmonics, as shown in the following expression: in, and This represents the delay and frequency of the subcarrier obtained through synchronization with the Costas ring. Finally, the sinusoidal signal output from the Costas ring is harmonicized by 3, 5, and 7, and then superimposed with each odd harmonic component to fit the Miller subcarrier signal. Multiplying the Miller subcarrier signal with the Miller code yields the reconstructed tag signal. The reconstructed signal expression for tag i is as follows:

2. The method for parallel reception of frequency division multiplexing RFID tag signals based on SIC architecture according to claim 1, characterized in that, The reader demodulates the current channel using SIC, reconstructing the tag signal. If the reader detects a bit error in the tag signal, it proceeds to the next channel for SIC demodulation, with two options: subtracting the reconstructed signal from the remaining total signal or not subtracting it. The reader then compares whether bit errors occur in the next channel's SIC demodulation under these two conditions. If no bit error occurs under either condition, the reader continues to reconstruct the current tag signal for subsequent SIC demodulation. If bit errors occur under both conditions, SIC demodulation is stopped.

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